Resonance type power electronic transformer based on current clamping and operation control method thereof

Abstract

The invention discloses a resonant power electronic transformer based on current clamping and an operation control method thereof, the input voltage of the resonant power electronic transformer is Ui, the output voltage of the resonant power electronic transformer is Uo, and the topology comprises an inversion unit, a transformer, a resonant capacitor Cr, a rectification unit and an output inductor Lo; according to an instruction of an external controller, the inversion unit periodically inputs positive, zero and negative levels to the primary side of the transformer, the output inductor is utilized to apply a clamping effect on the resonant current, and the resonant capacitor is utilized to realize zero-current turn-off of the inversion unit. The resonant current is approximate to a trapezoidal wave, the peak value and the effective value of the current are obviously reduced under the same power, and the device has the advantages of small current stress and small conduction loss; zero-current turn-off of all switching tubes of the inverter unit is realized, and the advantage of low switching loss is achieved; the problems of high current stress and high conduction loss caused by sine wave resonance current of an existing resonance type power electronic transformer are solved.

Description

Technical Field

[0001] The present invention belongs to the field of power electronic transformers, and relates to a resonant power electronic transformer, specifically to a resonant power electronic transformer based on current clamping and its operation control method. Background Art

[0002] Industrial parks are important platforms and powerful engines for promoting national economic growth, industrial agglomeration and innovation. According to data from the National Development and Reform Commission, industrial parks contribute more than 30% to the national economy, but also account for 31% of the national carbon dioxide emissions. Under the strategic goal of "carbon peak and carbon neutrality", increasing the proportion of renewable energy used in industrial parks is an important way to achieve the national carbon emission reduction goal. Among them, photovoltaic power generation has become the first choice for renewable new energy access in industrial parks due to its low cost, environmental friendliness, and no terrain restrictions.

[0003] Traditional new energy AC collection and transmission technologies face many problems such as large transmission line losses, multiple power conversion links, low efficiency, and poor stability. Their capacity and scale are limited, making it difficult to meet the electricity demand of large industrial parks. In contrast, the photovoltaic power generation scheme of medium-voltage DC collection and transmission has significant advantages. The collection and step-up transformer is the core equipment of this photovoltaic DC collection system. To improve the system operation efficiency and enhance the photovoltaic power generation benefit, a resonant power electronic transformer scheme is usually adopted.

[0004] CN118264120A proposes a power electronic transformer scheme with LC series resonance. By making the inductance and capacitance resonate, the resonant current shows a periodic sine wave, thus realizing the soft switching of the primary side switching tubes and reducing the switching loss. The sine wave resonant current of this scheme increases the peak value and effective value of the current flowing through the switching tubes, increases the device current stress and conduction loss, and reduces the operation efficiency of the power electronic transformer. CN116316529A proposes a three-active-bridge power electronic transformer scheme. Through the coordinated control of the primary and secondary side switching tubes, trapezoidal wave resonant current operation can be achieved, reducing the amplitude and effective value of the resonant current. The primary side switching tubes of this scheme can only achieve zero-voltage turn-on, which is not suitable for the medium-voltage DC collection scenario widely using IGBT devices. The voltage withstand level of the secondary side switching tubes also limits the step-up output voltage level, and multiple module outputs in series are required to complete the step-up output.

[0005] In summary, to further improve the operation efficiency of the photovoltaic medium-voltage DC collection and transmission system, a resonant power electronic transformer with both soft switching characteristics and low resonant current amplitude and effective value is still needed. Summary of the Invention

[0006] To solve the problems of high current stress and high conduction loss caused by the sinusoidal resonant current in the current resonant power electronic transformer, the present invention provides a resonant power electronic transformer based on current clamping and its operation control method. According to the instructions of an external controller, the inverter unit of the present invention periodically inputs three levels of positive, zero, and negative to the primary side of the transformer, uses the output inductor to clamp the resonant current, and uses the resonant capacitor to achieve zero-current turn-off of the inverter unit. The resonant current of the present invention is approximately trapezoidal, and the current peak value and effective value are significantly reduced under the same power, having the advantages of small device current stress and small conduction loss; realizing zero-current turn-off of all switching tubes of the inverter unit and having the advantage of small switching loss.

[0007] The object of the present invention is achieved through the following technical solutions:

[0008] A resonant power electronic transformer based on current clamping, with an input voltage of U i , and an output voltage of U o , the topology includes an inverter unit, a transformer (with a turns ratio of 1:n and a leakage inductance of L δ ), a resonant capacitor C r , a rectifier unit, and an output inductor L o ;

[0009] The inverter unit includes a positive DC-side port, a negative DC-side port, an AC-side port 1, and an AC-side port 2;

[0010] The rectifier unit includes a positive DC-side port, a negative DC-side port, an AC-side port 1, and an AC-side port 2;

[0011] The positive pole of the input voltage U i is connected to the positive DC-side port of the inverter unit, and the negative pole of the input voltage U i is connected to the negative DC-side port of the inverter unit;

[0012] One end of the primary winding of the transformer is connected to the AC-side port 1 of the inverter unit, the other end of the primary winding of the transformer is connected to the AC-side port 2 of the inverter unit, one end of the secondary winding of the transformer is connected to one pole of the resonant capacitor C r , the other pole of the resonant capacitor C r is connected to the AC-side port 1 of the rectifier unit, and the other end of the secondary winding of the transformer is connected to the AC-side port 2 of the rectifier unit;

[0013] The positive DC-side port of the rectifier unit is connected to one end of the output inductor L o , the other end of the output inductor L o is connected to the positive pole of the output voltage U o , and the negative DC-side port of the rectifier unit is connected to the negative pole of the output voltage U o ;

[0014] The inverter unit is a full-bridge inverter topology, a type-I three-level half-bridge inverter topology, a type-T three-level half-bridge inverter topology, a flying-capacitor type three-level half-bridge inverter topology, a type-I three-level full-bridge inverter topology, a type-T three-level full-bridge inverter topology or a flying-capacitor type three-level full-bridge inverter topology;

[0015] The rectifier unit is a full-bridge uncontrolled rectifier topology.

[0016] An operating control method for the above-mentioned resonant type power electronic transformer topology based on current clamping. Taking the full-bridge inverter topology of the inverter unit as an example, it includes the following steps:

[0017] Step 1: The first switch tube S1 and the second switch tube S2 conduct complementarily and both conduct for half of the switching control period; the third switch tube S3 and the fourth switch tube S4 conduct complementarily and both conduct for half of the switching control period; the fourth switch tube S4 conducts one zero-level time earlier than the first switch tube S1. Let the zero-level time be T z ; Let a switching control period be T s , and the time is t0 ≤ t < t8, where t0 ≤ t < t4 is the positive half-cycle and t4 ≤ t < t8 is the negative half-cycle;

[0018] Step 2: Let t0 be the initial moment and the starting point of a switching control period. At the initial moment t0, the first switch tube S1 and the fourth switch tube S4 conduct;

[0019] Step 3: Enter the first switching mode: When t0 ≤ t < t1, at the moment t0, the second switch tube S2 is turned off and the first switch tube S1 is turned on. The voltage u p of the primary winding of the transformer is U i , the first diode D1 and the fourth diode D4 conduct under positive voltage, the second diode D2 and the third diode D3 turn off under negative voltage, the output inductor L o and the leakage inductance L δ of the transformer are in series, and the secondary current i s of the transformer is approximately constant and equal to the output current I o ;

[0020] Step 4: Enter the second switching mode: When t1 ≤ t < t2, at the moment t1, the voltage u r of the resonant capacitor C Cr reaches the amplitude of nU i , the voltage polarity between the AC port 1 and the AC port 2 of the rectifier unit is reversed, the first diode D1 and the fourth diode D4 commutate to the second diode D2 and the third diode D3, and the resonant capacitor C r and the leakage inductance L δ of the transformer start to resonate in series, and the primary current ip Decrease according to the sine function law;

[0021] Step Five: Enter the third switching mode: when t2 ≤ t < t3, at the moment of t2, the primary side current i of the transformer p Decreases to 0, meanwhile, the fourth switching tube S4 turns off with zero current, the third switching tube S3 turns on, and the voltage u of the primary side winding of the transformer p Is 0, the first diode D1 and the fourth diode D4 continue to commutate to the second diode D2 and the third diode D3, and the resonant capacitor C r And the leakage inductance L of the transformer δ Continue to be in series resonance, and the primary side current i of the transformer p Continues to decrease according to the cosine function law;

[0022] Step Six: Enter the fourth switching mode: when t3 ≤ t < t4, at the moment of t3, the secondary side current i of the transformer s Decreases to -I o , the first diode D1 and the fourth diode D4 are turned off under negative voltage, the second diode D2 and the third diode D3 are turned on under positive voltage, and the output inductor L o And the leakage inductance L of the transformer δ Are in series, and the secondary side current i of the transformer s Is approximately constant and equal to the output current -I o ;

[0023] Step Seven: Enter the fifth switching mode: when t4 ≤ t < t5, at the moment of t4, the first switching tube S1 turns off and the second switching tube S2 turns on, and the voltage u of the primary side winding of the transformer p Is -U i , the first diode D1 and the fourth diode D4 are turned off under negative voltage, the second diode D2 and the third diode D3 are turned on under positive voltage, and the output inductor L o And the leakage inductance L of the transformer δ Are in series, and the secondary side current i of the transformer s Is approximately constant and equal to the output current -I o ;

[0024] Step Eight: Enter the sixth switching mode: when t5 ≤ t < t6, at the moment of t5, the voltage u r Of the resonant capacitor C Cr Reaches -nU in amplitude, the voltage polarity between port 1 and port 2 of the rectification unit is reversed, the second diode D2 and the third diode D3 commutate to the first diode D1 and the fourth diode D4, and the resonant capacitor C i And the leakage inductance L of the transformer r Start to be in series resonance, and the primary side current i of the transformer δ Increases according to the sine function law; p

[0025] Step Nine: Enter the seventh switching mode: when t6 ≤ t < t7, the primary current i of the transformer at time t6 p increases to 0. At the same time, the third switch S3 turns off with zero current, and the fourth switch S4 turns on. The voltage u of the primary winding of the transformer p is 0. The second diode D2 and the third diode D3 continue to commutate to the first diode D1 and the fourth diode D4. The resonant capacitor C r and the leakage inductance L of the transformer δ continue to resonate in series, and the primary current i of the transformer p continues to increase according to the cosine function law;

[0026] Step Ten: Enter the eighth switching mode: when t7 ≤ t < t8, the secondary current i of the transformer at time t7 s increases to I o . The first diode D1 and the fourth diode D4 conduct under positive voltage, and the second diode D2 and the third diode D3 turn off under negative voltage. The output inductor L o and the leakage inductance L of the transformer δ are in series. The secondary current i of the transformer s is approximately constant and equal to the output current I o .

[0027] In the present invention, the inverter unit is a full-bridge inverter topology, an I-type three-level half-bridge inverter topology, a T-type three-level half-bridge inverter topology, a flying-capacitor type three-level half-bridge inverter topology, an I-type three-level full-bridge inverter topology, a T-type three-level full-bridge inverter topology or a flying-capacitor type three-level full-bridge inverter topology; the rectifier unit is a full-bridge uncontrolled rectifier topology.

[0028] In the present invention, the resonant angular frequency ω r has a value range of:

[0029]

[0030] where U i_min is the lowest input voltage, I o_max is the maximum output current, and T sN is the rated switching control period;

[0031] In the present invention, the resonant capacitor C r has a value of:

[0032]

[0033] In the present invention, the output inductor L o should make the output current I o present as a DC current with a small ripple. Preferably, the value range of the output inductor L o is:

[0034]

[0035] Among them, ε i is the ripple rate of the output current.

[0036] In the present invention, the switching control period T s takes the value of:

[0037]

[0038] The zero-level time is T z takes the value of:

[0039]

[0040] The external controller can adjust the switching control period T s and the zero-level time T z to realize the adjustment of the input voltage U i or the output voltage U o .

[0041] Compared with the prior art, the present invention has the following advantages:

[0042] 1. The resonant current of the present invention is approximately trapezoidal. Under the same power, the current peak value and the effective value are significantly reduced. It has the advantages of small device current stress, small conduction loss, low device cost, and high power conversion efficiency;

[0043] 2. The present invention realizes zero-current turn-off of all switching tubes in the inverter unit, has the advantage of small switching loss, and high power conversion efficiency;

[0044] 3. The present invention uses the output inductor to clamp the resonant current, and can still maintain low current stress and effective value operation even when the input and output voltages are extremely mismatched, and can operate in a wide input and output voltage range;

[0045] 4. The present invention can adjust the input voltage U s or the output voltage U z by adjusting the switching control period T i and the zero-level time T o to realize the MPPT of the photovoltaic array or the wind turbine and the adaptive adjustment of the load;

[0046] 5. The rectifier unit of the present invention is composed of diodes. In the boost scenario, the withstand voltage level can be increased by diode series connection, which has the advantage of low cost;

[0047] 6. The inverter unit of the present invention adopts a full-bridge inverter topology, a three-level half-bridge inverter topology or a three-level full-bridge inverter topology, which can flexibly adapt to scenarios with different voltages and powers, and solves the problems of high current stress and high conduction loss caused by the sinusoidal resonant current of the existing resonant power electronic transformer. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] Figure 1 is a schematic topology diagram of a resonant power electronic transformer based on current clamping;

[0049] Figure 2 is a schematic diagram of typical driving and voltage-current waveforms of a resonant power electronic transformer based on current clamping;

[0050] Figure 3 is a schematic diagram of the current path of a resonant power electronic transformer based on current clamping operating in the first to fifth switching modes;

[0051] Figure 4 is a schematic diagram of the simulation waveforms of the primary current of the transformer and the resonant capacitor voltage under the voltage regulation state of a resonant power electronic transformer based on current clamping using a full-bridge inverter unit;

[0052] Figure 5 is a schematic diagram of the simulation waveforms of the primary current of the transformer and the resonant capacitor voltage under the non-voltage regulation state of a resonant power electronic transformer based on current clamping using a full-bridge inverter unit;

[0053] Figure 6 is a derivative schematic diagram of the topology structure proposed in Embodiment 2;

[0054] Figure 7 is a derivative schematic diagram of the topology structure proposed in Embodiment 3;

[0055] Figure 8 is a derivative schematic diagram of the topology structure proposed in Embodiment 4;

[0056] Figure 9 is a derivative schematic diagram of the topology structure proposed in Embodiment 5;

[0057] Figure 10 is a derivative schematic diagram of the topology structure proposed in Embodiment 6;

[0058] Figure 11 is a derivative schematic diagram of the topology structure proposed in Embodiment 7. DETAILED DESCRIPTION OF THE EMBODIMENTS

[0059] The technical solutions of the present invention will be further described below in conjunction with the accompanying drawings, but are not limited thereto. Any modifications or equivalent replacements of the technical solutions of the present invention without departing from the spirit and scope of the technical solutions of the present invention shall be covered within the protection scope of the present invention.

[0060] Embodiment 1:

[0061] This embodiment provides a resonant power electronic transformer based on current clamping, as Figure 1 shown. The input voltage of the resonant power electronic transformer is U i , and the output voltage is U o . The topology includes an inverter unit, a transformer (with a turns ratio of 1:n and a leakage inductance of L δ ), a resonant capacitor C r , a rectifier unit, and an output inductor L o , where:

[0062] The inverter unit is a full-bridge inverter topology, including an input capacitor C i , a first switching tube S1, a second switching tube S2, a third switching tube S3, a fourth switching tube S4, and ports 1 to 4. One end of port 1 of the inverter unit is simultaneously connected to the positive electrode of the input capacitor C i , the collector of the first switching tube S1, and the collector of the third switching tube S3. Port 2 of the inverter unit is simultaneously connected to the negative electrode of the input capacitor C i , the emitter of the second switching tube S2, and the emitter of the fourth switching tube S4. Port 3 of the inverter unit is simultaneously connected to the emitter of the first switching tube S1 and the collector of the second switching tube S2. Port 4 of the inverter unit is simultaneously connected to the emitter of the third switching tube S3 and the collector of the fourth switching tube S4;

[0063] The rectifier unit includes a first diode D1, a second diode D2, a third diode D3, a fourth diode D4, and ports 1 to 4. Port 1 of the rectifier unit is simultaneously connected to the positive electrode of the first diode D1 and the negative electrode of the second diode D2. Port 2 of the rectifier unit is simultaneously connected to the positive electrode of the third diode D3 and the negative electrode of the fourth diode D4. Port 3 of the rectifier unit is simultaneously connected to the negative electrode of the first diode D1 and the negative electrode of the third diode D3. Port 4 of the rectifier unit is simultaneously connected to the positive electrode of the second diode D2 and the positive electrode of the fourth diode D4;

[0064] The positive electrode of the input voltage U i is connected to port 1 of the inverter unit, and the negative electrode of the input voltage U i is connected to port 2 of the inverter unit;

[0065] One end of the primary side winding of the transformer is connected to port 3 of the inverter unit, and the other end of the primary side winding of the transformer is connected to port 4 of the inverter unit. One end of the secondary side winding of the transformer is connected to one pole of the resonant capacitor C r , and the other pole of the resonant capacitor C r is connected to port 1 of the rectifier unit. The other end of the secondary side winding of the transformer is connected to port 2 of the rectifier unit;

[0066] Port 3 of the rectification unit is connected to one end of the output inductor L o and the other end of the output inductor L o is connected to the positive pole of the output voltage U o and port 4 of the rectification unit is connected to the negative pole of the output voltage U o .

[0067] Preferably, the inverter unit is a full-bridge inverter topology, a three-level half-bridge inverter topology or a three-level full-bridge inverter topology, and the three-level topology is a type-I three-level topology, a T-type three-level topology or a flying-capacitor three-level topology.

[0068] This embodiment also provides an operation control method for a resonant power electronic transformer topology based on current clamping, as Figure 2 and Figure 3 shown. The method includes the following steps:

[0069] Step 1: The first switch tube S1 and the second switch tube S2 conduct complementarily and both conduct for half of the switching control period; the third switch tube S3 and the fourth switch tube S4 conduct complementarily and both conduct for half of the switching control period; the fourth switch tube S4 conducts one zero-level time earlier than the first switch tube S1. Let the zero-level time be T z ; let one switching control period be T s , and the time is t0 ≤ t < t8, where t0 ≤ t < t4 is the positive half-cycle and t4 ≤ t < t8 is the negative half-cycle;

[0070] Step 2: Let t0 be the initial moment and the starting point of one switching control period. At the initial moment t0, the first switch tube S1 and the fourth switch tube S4 conduct;

[0071] Step 3: Enter the first switching mode: when t0 ≤ t < t1, at the moment t0, the second switch tube S2 is turned off and the first switch tube S1 is turned on. The voltage u p of the primary side winding of the transformer is U i , the first diode D1 and the fourth diode D4 conduct under positive voltage, the second diode D2 and the third diode D3 turn off under negative voltage, the output inductor L o and the leakage inductance L δ of the transformer are in series, and the current i s of the secondary side of the transformer is approximately constant and equal to the output current I o ;

[0072] Step 4: Enter the second switching mode: when t1 ≤ t < t2, at the moment t1, the voltage u r of the resonant capacitor C Cr reaches the amplitude of nU i, the voltage polarity between port 1 and port 2 of the rectifier unit is reversed, and the first diode D1 and the fourth diode D4 commutate to the second diode D2 and the third diode D3, and the resonant capacitor C r and the leakage inductance L of the transformer δ start to resonate in series, and the primary current i of the transformer p decreases according to the sine function law;

[0073] Step Five: Enter the third switching mode: when t2 ≤ t < t3, at time t2, the primary current i of the transformer p decreases to 0, and at the same time, the fourth switching tube S4 turns off with zero current and the third switching tube S3 turns on. The voltage u of the primary winding of the transformer p is 0, and the first diode D1 and the fourth diode D4 continue to commutate to the second diode D2 and the third diode D3, and the resonant capacitor C r and the leakage inductance L of the transformer δ continue to resonate in series, and the primary current i of the transformer p continues to decrease according to the cosine function law;

[0074] Step Six: Enter the fourth switching mode: when t3 ≤ t < t4, at time t3, the secondary current i of the transformer s decreases to -I o , the first diode D1 and the fourth diode D4 are turned off under negative voltage, and the second diode D2 and the third diode D3 are turned on under positive voltage. The output inductor L o and the leakage inductance L of the transformer δ are in series, and the secondary current i of the transformer s is approximately constant and equal to the output current -I o ;

[0075] Step Seven: Enter the fifth switching mode: when t4 ≤ t < t5, at time t4, the first switching tube S1 is turned off and the second switching tube S2 is turned on. The voltage u of the primary winding of the transformer p is -U i , the first diode D1 and the fourth diode D4 are turned off under negative voltage, and the second diode D2 and the third diode D3 are turned on under positive voltage. The output inductor L o and the leakage inductance L of the transformer δ are in series, and the secondary current i of the transformer s is approximately constant and equal to the output current -I o ;

[0076] Step Eight: Enter the sixth switching mode: when t5 ≤ t < t6, at time t5, the voltage u of the resonant capacitor C r reaches the amplitude of -nU Cr i ​, the voltage polarity between port 1 and port 2 of the rectification unit is reversed, and the second diode D2 and the third diode D3 commutate to the first diode D1 and the fourth diode D4, and the resonant capacitor C r and the leakage inductance L of the transformer δ start to resonate in series, and the primary side current i of the transformer p increases according to the sine function law;

[0077] Step Nine: Enter the seventh switching mode: when t6 ≤ t < t7, at the moment t6, the primary side current i of the transformer p increases to 0, at the same time the third switching tube S3 turns off with zero current and the fourth switching tube S4 turns on, and the voltage u of the primary side winding of the transformer p is 0, and the second diode D2 and the third diode D3 continue to commutate to the first diode D1 and the fourth diode D4, and the resonant capacitor C r and the leakage inductance L of the transformer δ continue to resonate in series, and the primary side current i of the transformer p continues to increase according to the cosine function law;

[0078] Step Ten: Enter the eighth switching mode: when t7 ≤ t < t8, at the moment t7, the secondary side current i of the transformer s increases to I o , the first diode D1 and the fourth diode D4 conduct under positive voltage, and the second diode D2 and the third diode D3 turn off under negative voltage. The output inductor L o and the leakage inductance L of the transformer δ are in series, and the secondary side current i of the transformer s is approximately constant and equal to the output current I o ;

[0079] Preferably, the value range of the resonant angular frequency ω r is:

[0080]

[0081] where U i_min is the lowest input voltage, and I o_max is the maximum output current.

[0082] Preferably, the value of the switching control period T s is:

[0083]

[0084] Preferably, the value of the zero-level time T z is:

[0085]

[0086] The external controller adjusts the switching control period T s and the zero-level time T z to achieve the adjustment of the input voltage U i or the output voltage U o , as shown in Figure 4 and Figure 5 .

[0087] Embodiment 2:

[0088] As shown in Figure 6 , using the type-I three-level half-bridge as the inverter unit, a resonant power electronic transformer based on current clamping can still be constructed, and the operation control method remains unchanged.

[0089] Embodiment 3:

[0090] As shown in Figure 7 , using the type-I three-level full-bridge as the inverter unit, a resonant power electronic transformer based on current clamping can still be constructed, and the operation control method remains unchanged.

[0091] Embodiment 4:

[0092] As shown in Figure 8 , using the T-type three-level half-bridge as the inverter unit, a resonant power electronic transformer based on current clamping can still be constructed, and the operation control method remains unchanged.

[0093] Embodiment 5:

[0094] As shown in Figure 9 , using the T-type three-level full-bridge as the inverter unit, a resonant power electronic transformer based on current clamping can still be constructed, and the operation control method remains unchanged.

[0095] Embodiment 6:

[0096] As shown in Figure 10 , using the flying-capacitor type three-level half-bridge as the inverter unit, a resonant power electronic transformer based on current clamping can still be constructed, and the operation control method remains unchanged.

[0097] Embodiment 7:

[0098] As shown in Figure 11 , using the flying-capacitor type three-level full-bridge as the inverter unit, a resonant power electronic transformer based on current clamping can still be constructed, and the operation control method remains unchanged.

Claims

1. A resonant power electronic transformer based on current clamping, characterized in that The input voltage of the resonant power electronic transformer is U i , and the output voltage is U o . The topology includes an inverter unit, a transformer, a resonant capacitor C r , a rectifier unit, and an output inductor L o ; The turns ratio of the transformer is 1:n, and the leakage inductance is L δ ; The inverter unit is a full-bridge inverter topology, including an input capacitor C i , a first switching transistor S1, a second switching transistor S2, a third switching transistor S3, a fourth switching transistor S4, and ports 1 to 4. One end of port 1 of the inverter unit is simultaneously connected to the positive electrode of the input capacitor C i , the collector of the first switching transistor S1, and the collector of the third switching transistor S3. Port 2 of the inverter unit is simultaneously connected to the negative electrode of the input capacitor C i , the emitter of the second switching transistor S2, and the emitter of the fourth switching transistor S4. Port 3 of the inverter unit is simultaneously connected to the emitter of the first switching transistor S1 and the collector of the second switching transistor S2. Port 4 of the inverter unit is simultaneously connected to the emitter of the third switching transistor S3 and the collector of the fourth switching transistor S4; The rectifier unit is a full-bridge uncontrolled rectifier topology, including a first diode D1, a second diode D2, a third diode D3, a fourth diode D4, and ports 1 to 4. Port 1 of the rectifier unit is connected to the positive electrode of the first diode D1 and the negative electrode of the second diode D2 at the same time. Port 2 of the rectifier unit is connected to the positive electrode of the third diode D3 and the negative electrode of the fourth diode D4 at the same time. Port 3 of the rectifier unit is connected to the negative electrode of the first diode D1 and the negative electrode of the third diode D3 at the same time. Port 4 of the rectifier unit is connected to the positive electrode of the second diode D2 and the positive electrode of the fourth diode D4 at the same time; The positive electrode of the input voltage U i is connected to port 1 of the inverter unit, and the negative electrode of the input voltage U i is connected to port 2 of the inverter unit; One end of the primary winding of the transformer is connected to port 3 of the inverter unit, and the other end of the primary winding of the transformer is connected to port 4 of the inverter unit. One end of the secondary winding of the transformer is connected to one pole of the resonant capacitor C r , and the other pole of the resonant capacitor C r is connected to port 1 of the rectifier unit, and the other end of the secondary winding of the transformer is connected to port 2 of the rectifier unit; Port 3 of the rectification unit is connected to one end of output inductor L o and the other end of output inductor L o is connected to the positive pole of output voltage U o Port 4 of the rectification unit is connected to the negative pole of output voltage U o .

2. The resonant power electronic transformer based on current clamping according to claim 1, wherein The inverter unit is replaced with a type-I three-level half-bridge inverter topology, a type-T three-level half-bridge inverter topology, a flying-capacitor type three-level half-bridge inverter topology, a type-I three-level full-bridge inverter topology, a type-T three-level full-bridge inverter topology, or a flying-capacitor type three-level full-bridge inverter topology.

3. The resonant power electronic transformer based on current clamping according to claim 1, characterized in that The resonant capacitor C r has a value of: Among them, ω r is the resonant angular frequency.

4. The resonant power electronic transformer based on current clamping according to claim 3, wherein The resonant angular frequency ω r has a value range of: Among them, U i_min is the lowest input voltage, I o_max is the maximum output current, and T sN is the rated switching control period.

5. The resonant power electronic transformer based on current clamping according to claim 1, wherein The output inductor L o has a value range of: Among them, ε i is the ripple rate of the output current.

6. A method for operating and controlling a current-clamped resonant power electronic transformer topology according to any one of claims 1-5, characterized in that The method includes the following steps: Step 1: The first switching transistor S1 and the second switching transistor S2 conduct complementarily and both conduct for half of the switching control period; the third switching transistor S3 and the fourth switching transistor S4 conduct complementarily and both conduct for half of the switching control period; the fourth switching transistor S4 conducts one zero-level time earlier than the first switching transistor S1, and let the zero-level time be T z ; Let a switching control period be T s , the time is t0 ≤ t < t8, where t0 ≤ t < t4 is the positive half-cycle, and t4 ≤ t < t8 is the negative half-cycle; Step 2: Let t0 be the initial moment, and it is the starting point of a switching control period. At the initial moment t0, the first switch tube S1 and the fourth switch tube S4 are turned on; Step 3: Enter the first switching mode: when t0 ≤ t < t1, at the moment t0, the second switching transistor S2 is turned off and the first switching transistor S1 is turned on, and the voltage u of the primary winding of the transformer p is U i . The first diode D1 and the fourth diode D4 conduct under positive voltage, and the second diode D2 and the third diode D3 turn off under negative voltage. The output inductor L o and the leakage inductance L of the transformer δ are connected in series. The current i of the secondary side of the transformer s is approximately constant and equal to the output current I o . Step 4: Enter the second switching mode: when t1 ≤ t < t2, at the moment of t1, the voltage u of the resonant capacitor C r reaches the amplitude of nU Cr , the voltage polarity between the AC port 1 and the AC port 2 of the rectifying unit is reversed, and the first diode D1 and the fourth diode D4 commutate to the second diode D2 and the third diode D3. The resonant capacitor C i and the leakage inductance L of the transformer r start series resonance, and the primary side current i of the transformer δ decreases according to the sine function law; p ​ Step Five: Enter the third switching mode: when t2 ≤ t < t3, at the moment of t2, the primary side current i of the transformer p decreases to 0. At the same time, the fourth switching transistor S4 turns off with zero current, and the third switching transistor S3 turns on. The voltage u of the primary side winding of the transformer p is 0. The first diode D1 and the fourth diode D4 continue to commutate to the second diode D2 and the third diode D3. The resonant capacitor C r and the leakage inductance L of the transformer δ continue to be in series resonance. The primary side current i of the transformer p continues to decrease according to the cosine function law; Step 6: Enter the fourth switching mode: when t3 ≤ t < t4, at time t3, the current i on the secondary side of the transformer s decreases to -I o , the first diode D1 and the fourth diode D4 are turned off under negative voltage, the second diode D2 and the third diode D3 are turned on under positive voltage, and the output inductor L o and the leakage inductance L of the transformer δ are in series, and the current i on the secondary side of the transformer s is approximately constant and equal to the output current -I o ; Step 7: Enter the fifth switching mode: when t4 ≤ t < t5, the first switching transistor S1 is turned off and the second switching transistor S2 is turned on at the moment of t4, and the voltage u of the primary side winding of the transformer p is -U i . The first diode D1 and the fourth diode D4 are turned off under negative voltage, and the second diode D2 and the third diode D3 are turned on under positive voltage. The output inductor L o and the leakage inductance L of the transformer δ are connected in series. The current i of the secondary side of the transformer s is approximately constant and equal to the output current -I o ; Step 8: Enter the sixth switching mode: when t5 ≤ t < t6, at the moment of t5, the resonant capacitor C r voltage u Cr amplitude reaches -nU i , the voltage polarity between port 1 and port 2 of the rectification unit is reversed, the second diode D2 and the third diode D3 commutate to the first diode D1 and the fourth diode D4, and the resonant capacitor C r and the leakage inductance L of the transformer δ start series resonance, and the primary side current i of the transformer p increases according to the sine function law; Step Nine: Enter the seventh switching mode: when t6 ≤ t < t7, the primary side current i of the transformer at the moment of t6 p increases to 0. Meanwhile, the third switching transistor S3 turns off with zero current, and the fourth switching transistor S4 turns on. The voltage u of the primary side winding of the transformer p is 0. The second diode D2 and the third diode D3 continue to commutate to the first diode D1 and the fourth diode D4. The resonant capacitor C r and the leakage inductance L of the transformer δ continue to resonate in series. The primary side current i of the transformer p continues to increase according to the cosine function law; Step Ten: Enter the eighth switching mode: when t7 ≤ t < t8, the secondary-side current i of the transformer at time t7 s increases to I o . The first diode D1 and the fourth diode D4 conduct under positive voltage, and the second diode D2 and the third diode D3 turn off under negative voltage. The output inductor L o and the leakage inductance L of the transformer δ are in series. The secondary-side current i of the transformer s is approximately constant and equal to the output current I o .

7. The operating control method of the resonant power electronic transformer topology based on current clamping according to claim 6, characterized in that The switching control period T s has a value of:

8. The operating control method of the resonant power electronic transformer topology based on current clamping according to claim 6, characterized in that The zero-level time is T z The value of

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