Motor control device and power conversion system

Abstract

The invention provides a motor control device capable of controlling a current phase angle based on a change characteristic quantity of inductance to reduce loss. A motor control device according to the present invention is connected to a power converter that performs power conversion from DC power to three-phase AC power to drive a motor with the three-phase AC power, and controls power conversion of the power converter, the motor control device comprising: an inductance change detection unit that detects a change in inductance of the motor; an inductance change characteristic amount calculation unit that calculates an inductance change characteristic amount indicating a change in the inductance of the motor in a first phase interval including a phase in which the absolute value of an arbitrary phase of the three-phase alternating current becomes the maximum; and a current phase angle control unit that controls a current phase angle on the basis of the inductance change characteristic quantity.

Description

Technical Field

[0001] The present invention relates to a motor control device and a power conversion system. Background Art

[0002] The motor control device determines a three-phase AC voltage command that satisfies a torque command or the like and reduces losses (copper loss and iron loss) in a three-phase synchronous motor, and controls an inverter according to the three-phase AC voltage command.

[0003] As a technique for suppressing such losses, there is Patent Document 1. In Patent Document 1, it is described in claim 1 that "the carrier frequency adjustment unit adjusts the phase difference between the voltage command and the carrier in such a manner as to reduce the eddy current loss generated in the magnet of the rotor of the AC motor in accordance with the d-axis current supplied to the AC motor and the rotational speed of the AC motor", in paragraph 0061 that "the eddy current loss We is expressed by the proportional relationship shown in the following formula (11)", in paragraph 0063 that "formula (11) can be replaced and expressed as the proportional relationship shown in the following formula (12)", and in paragraph 0072 that "the fixed triangular wave phase determination unit 1633 determines the value of the carrier phase difference Δθcarr based on the d-axis current and sum calculated by the d-axis current and calculation unit 1632. Here, the value of the carrier phase difference Δθcarr is determined in such a manner that the value of the d-axis current and sum becomes minimum".

[0004] Then, in paragraph 0076 of Patent Document 1, it is described that "in the voltage phase error calculation unit 163, as described above, the carrier phase difference Δθcarr is determined, and the voltage phase error Δθv is calculated. Thus, the voltage phase error Δθv can be determined in such a manner that the d-axis current and sum become minimum in accordance with the d-axis current Id and the motor rotational speed ωr. As a result, the phase difference between the voltage command for the inverter 3 and the carrier used in pulse width modulation can be changed in such a manner as to reduce the eddy current loss generated in the magnet of the rotor of the motor 2, and the carrier frequency fc can be set. As a result, the rise of the magnet temperature Tmag can be suppressed, and irreversible demagnetization can be prevented".

[0005] Prior Art Documents

[0006] Patent Documents

[0007] Patent Document 1: Japanese Unexamined Patent Application Publication No. 2022-18168 Summary of the Invention

[0008] Problems to be Solved by the Invention

[0009] However, iron losses such as eddy current losses depend on the magnetic flux generated in the coil, and this magnetic flux can be reduced by increasing the d-axis current. Therefore, there are also cases where iron losses can be more effectively suppressed even when the d-axis current is not at its minimum. Consequently, there is a problem of having to appropriately select the d-axis current in order to minimize losses.

[0010] In addition, since iron losses change due to inductance, there is a problem of having to detect changes in inductance and control the current accordingly in order to suppress losses.

[0011] Accordingly, an object of the present invention is to provide a motor control device and a power conversion system that can control the current phase angle based on a change characteristic amount of inductance to reduce losses.

[0012] Technical means for solving the problem

[0013] To solve the above problems, a motor control device of the present invention is connected to a power converter, for example, and controls the power conversion of the power converter. The power converter performs the power conversion from DC power to three-phase AC power to drive a motor with the three-phase AC power. The motor control device includes: an inductance change detection unit that calculates an inductance change characteristic amount representing a change in the inductance of the motor in a first phase interval where the absolute value of an alternating current including any phase of the three-phase alternating current becomes maximum; and a current phase angle control unit that controls the current phase angle based on the inductance change characteristic amount.

[0014] In addition, a power conversion system of the present invention includes the motor control device and the power converter, for example.

[0015] Effects of the invention

[0016] According to the present invention, it is possible to provide a motor control device and a power conversion system that can control the current phase angle based on a change characteristic amount of inductance to reduce losses. Description of the drawings

[0017] Figure 1 is an overall structural diagram of a motor drive system having a motor control device.

[0018] Figure 2 is a block diagram showing the functional structure of the motor control device according to the embodiment.

[0019] Figure 3 is a diagram for explaining the sampling period of the average value of the current.

[0020] Figure 4 is a diagram for explaining the sampling period of the average value of the current.

[0021] Figure 5It is a diagram showing an example of the relationship between the current I flowing in a coil and the magnetic flux Φ generated in the coil.

[0022] Figure 6 It is a diagram for explaining the sampling period of the current used in the calculation of the characteristic quantity of inductance change.

[0023] Figure 7 It is a diagram for explaining the sampling period of the current used in the calculation of the characteristic quantity of inductance change.

[0024] Figure 8 It is a diagram for explaining the relationship between the carrier frequency and the current ripple.

[0025] Figure 9 It is a diagram for explaining the relationship between the three-phase alternating current and the carrier frequency.

[0026] Figure 10 It is a diagram for explaining the relationship between the loss and the current phase angle. Detailed implementation mode

[0027] Hereinafter, the present invention will be described in detail with reference to the accompanying drawings. In addition, the present invention is not limited to the embodiments described below. These embodiments are merely examples, and the present invention can be implemented in various modified and improved forms based on the knowledge of those skilled in the art. In addition, in each of the accompanying drawings used in the following description, the same reference numerals are attached to common devices and equipment, and the description of some devices, equipment, and operations that have been described may be omitted.

[0028] Figure 1 It is an overall structural diagram of a motor drive system having a motor control device. Figure 1 In this embodiment, the motor drive system 100 of the present embodiment includes a motor control device 1, a motor 2, an inverter (power converter) 3, a rotational position detector 4, a high-voltage battery 5, a current detection unit 7, and a rotational position sensor 8.

[0029] For the motor control device 1, the rotational position θr of the motor 2 is input from the rotational position detector 4. In addition, Iu, Iv, and Iw representing the three-phase alternating current flowing in the motor 2 are input from the current detection unit 7, and a torque command T* is input from an upper-level control device (not shown). Based on these input information, the motor control device 1 generates a gate signal for controlling the drive of the motor 2 and outputs it to the inverter 3. Thereby, the operation of the inverter 3 is controlled, and the drive of the motor 2 is controlled. In addition, the details of the motor control device 1 will be described later.

[0030] The inverter 3 includes an inverter circuit 31, a PWM signal drive circuit 32, and a smoothing capacitor 33. The PWM signal drive circuit 32 generates a PWM signal for controlling each switching element included in the inverter circuit 31 based on a gate signal input from the motor control device 1, and outputs it to the inverter circuit 31. The inverter circuit 31 has switching elements corresponding to the upper arms and lower arms of the U-phase, V-phase, and W-phase respectively. By controlling these switching elements according to the PWM signals input from the PWM signal drive circuit 32, the DC power supplied from the high-voltage battery 5 is converted into AC power and output to the motor 2. The smoothing capacitor 33 smoothes the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

[0031] The high-voltage battery 5 is a DC voltage source of the motor drive system 100, and outputs a power supply voltage Hvdc to the inverter 3. The power supply voltage Hvdc of the high-voltage battery 5 is converted by the inverter circuit 31 and the PWM signal drive circuit 32 of the inverter 3 into a pulsed three-phase AC voltage with variable voltage and variable frequency, and is applied to the motor 2 as an inter-phase voltage. Thus, based on the DC power of the high-voltage battery 5, AC power is supplied from the inverter 3 to the motor 2. In addition, the power supply voltage Hvdc of the high-voltage battery 5 varies according to its state of charge.

[0032] The motor 2 is a three-phase motor driven to rotate by the AC power supplied from the inverter 3, and has a stator and a rotor. In this embodiment, an example of using a permanent magnet synchronous motor as the motor 2 is described, but it is not limited thereto. When AC power input from the inverter 3 is applied to the three-phase coils Lu, Lv, and Lw provided in the stator, three-phase AC currents Iu, Iv, and Iw flow through the motor 2, and magnetic fluxes are generated in the respective coils. An attractive / repulsive force is generated between the magnetic fluxes of the respective coils and the magnetic flux of the permanent magnet disposed in the rotor, thereby generating a torque in the rotor and driving the motor 2 to rotate.

[0033] In the motor 2, a rotation position sensor 8 for detecting the rotation position θr of the rotor is installed. The rotation position detector 4 calculates the rotation position θr based on the input signal of the rotation position sensor 8. The calculation result of the rotation position θr obtained by the rotation position detector 4 is input to the motor control device 1 and is used in the phase control of the AC power in which the motor control device 1 generates a pulsed gate signal corresponding to the phase of the induced voltage of the motor 2.

[0034] Here, as the rotational position sensor 8, a synchro resolver composed of an iron core and windings is more suitable, but it is also acceptable to use a sensor such as a magnetoresistive element using GMR (Giant Magneto Resistive effect) or a Hall element. As long as the magnetic pole position of the rotor can be measured, any sensor can be used as the rotational position sensor 8. Additionally, the rotational position detector 4 may not use the input signal from the rotational position sensor 8, but instead use the three-phase alternating current Iu, Iv, Iw flowing in the motor 2 and the three-phase alternating voltage Vu, Vv, Vw applied to the motor 2 from the inverter 3 to estimate the rotational position θr.

[0035] A current detection unit 7 is arranged in the current path between the inverter 3 and the motor 2. The current detection unit 7 detects the three-phase alternating current Iu, Iv, Iw (U-phase alternating current Iu, V-phase alternating current Iv, and W-phase alternating current Iw) flowing through the motor 2. As the current detection unit 7, for example, a Hall current sensor or the like can be used. The detection results of the three-phase alternating current Iu, Iv, Iw obtained by the current detection unit 7 are input to the motor control device 1 and are used for the generation of the gate signal performed by the motor control device 1. Additionally, Figure 1 An example in which the current detection unit 7 is composed of three current detectors is shown, but two current detectors can also be employed, and for the alternating current of the remaining one phase, it can be calculated based on the fact that the sum of the three-phase alternating current Iu, Iv, Iw is zero. Additionally, a pulsed direct current flowing from the high-voltage battery 5 into the inverter 3 can be detected using a shunt resistor inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase alternating current Iu, Iv, Iw can be obtained based on this direct current and the three-phase alternating voltage Vu, Vv, Vw applied to the motor 2 from the inverter 3.

[0036] Next, the details of the motor control device 1 will be described. Figure 2 It is a block diagram showing the functional structure of the motor control device 1 according to the embodiment.

[0037] The motor control device 1 has functional modules such as a current command generation unit 10, a speed calculation unit 11, a current conversion unit 12, a current control unit 13, a carrier frequency adjustment unit 14, a carrier generation unit 15, a phase calculation unit 16, a three-phase voltage conversion unit 18, a gate signal generation unit 19, an inductance change detection unit 21, a current phase angle control unit 22, and a carrier frequency switching unit 23. Here, the current command generation unit 10, the current conversion unit 12, the current control unit 13, and the three-phase voltage conversion unit 18 function as a three-phase AC voltage command generation unit. The structure of the three-phase AC voltage command generation unit is not limited to this, and other functions can also be included. The motor control device 1 is constituted by, for example, a microcomputer, and these functional modules can be realized by executing a prescribed program in the microcomputer. Alternatively, a part or all of these functional modules can also be realized using a hardware circuit such as a logic IC or an FPGA (Field Programmable Gate Array).

[0038] The speed calculation unit 11 calculates a motor rotation speed ωr indicating the rotation speed (rotational speed) of the motor 2 based on the change over time of the rotation position θr of the motor 2. In addition, the motor rotation speed ωr can be a value expressed in either angular velocity (rad / s) or rotational speed (rpm). In addition, these values can also be used interchangeably.

[0039] The carrier frequency adjustment unit 14 determines a carrier frequency fc indicating the frequency of the carrier used to generate the gate signal based on the rotation speed ωr obtained by the speed calculation unit 11. For example, the carrier frequency fc is determined such that the synchronous PWM carrier number Nc indicating the number of carriers corresponding to one cycle of the voltage waveform in synchronous PWM control becomes a prescribed integer. Here, the synchronous PWM carrier number Nc can be set, for example, to a number that satisfies the conditional expression Nc = 3×(2×n - 1) among multiples of 3. In this conditional expression, n represents an arbitrary natural number, and for example, n = 1 (Nc = 3), n = 2 (Nc = 9), n = 3 (Nc = 15), etc. can be selected. In addition, the synchronous PWM carrier number Nc can also be changed corresponding to the rotation speed ωr. Then, the carrier frequency adjustment unit 14 determines fc based on the rotation speed ωr and the synchronous PWM carrier number Nc.

[0040] In addition, the carrier frequency adjustment unit 14 adjusts the carrier frequency fc in accordance with the switching flag FlgFrqSw from the carrier frequency switching unit 23 described later. For example, an example will be described in which the carrier frequency adjustment unit 14 uses a coefficient K to make the carrier frequency fc = K·ωr·Nc / 2π and changes K in accordance with the value of the switching flag FlgFrqSw, but it is not limited thereto. If the coefficient K when the switching flag FlgFrqSw = 1 is larger than the coefficient K when the switching flag FlgFrqSw = 0, it is possible to achieve switching of the carrier frequency such that the carrier is a high frequency when the switching flag FlgFrqSw = 1 and the carrier is a low frequency when the switching flag FlgFrqSw = 0. In addition, the method of switching the carrier frequency fc is not limited thereto. In addition, the carrier frequency adjustment unit 14 may use the rotational position θr of the motor 2 as an input and be able to switch the carrier frequency only when the rotational position θr of the motor 2 is within a specific range. Thus, the carrier frequency switching unit 23 performs calculations only when the rotational position θr of the motor 2 is within a specific range, and the processing load on the motor control device 1 can be reduced.

[0041] Based on the carrier frequency fc determined by the carrier frequency adjustment unit 14, the carrier signal generation unit 15 generates carrier signals Sc for the three-phase AC voltage commands Vu*, Vv*, and Vw*, respectively.

[0042] Based on the rotational position θr, the rotational speed ωr, and the carrier frequency fc, the phase calculation unit 16 calculates the estimated rotational position θe of the motor 2 using the following equations (1) to (3).

[0043] θe = θr + φv…(1)

[0044] φv = ωr·1.5Tc…(2)

[0045] Tc = 1 / fc…(3)

[0046] Here, φv represents the arithmetic delay compensation value of the voltage phase, and Tc represents the carrier period. The arithmetic delay compensation value φv is a value for compensating for the arithmetic delay of 1.5 control periods during the period from obtaining the rotational position θr from the rotational position detector 4 until the motor control device 1 outputs a gate signal to the inverter 3.

[0047] Figure 3 and 4 are diagrams illustrating the sampling period of the average value of the current. Figure 3 and 4 In, the vertical axis is the current and the horizontal axis is the time, showing an example of the current waveform of one phase of the three-phase AC current. The actual current generates current ripple and vibrates at a frequency close to the carrier frequency fc. For ease of explanation, the following will be described assuming that the frequency of the current ripple is equal to the carrier frequency fc, but it is not limited thereto. As shown in Figure 3As shown, the average value of the current is obtained at each sampling period Ts determined based on the carrier frequency fc, for example, Ts = 1 / fc. Additionally, when the carrier frequency fc is high, the average value of the current can also be obtained every integer multiple of 1 / fc, for example, every Ts = 2 / fc as shown in Figure 4 When the sampling period becomes longer, the number of times the average value is obtained decreases, so the processing load on the motor control device 1 can be reduced.

[0048] Figure 5 FIG. is an example showing the relationship between the current I flowing in the coil and the magnetic flux Φ generated in the coil. The horizontal axis is the current I, and the vertical axis is the magnetic flux Φ. When the current I is small, the current I and the magnetic flux Φ are linearly related, and when the current I is large, the current I and the magnetic flux Φ are non-linearly related. Hereinafter, the portion of the curve showing the relationship between the current I and the magnetic flux Φ where the current I and the magnetic flux Φ are linearly related will be referred to as the linear portion, and the portion where the current I and the magnetic flux Φ are non-linearly related will be referred to as the non-linear portion.

[0049] Since the inductance L of the coil is defined by Φ = LI, when calculating the inductance L at the point P1 on the linear portion and the point P2 on the non-linear portion respectively, the inductance L at the point P1 is the slope L0 of the straight line connecting the origin and P1, and the inductance L at the point P2 is the slope L2 of the straight line connecting the origin and P2. Thus, at any point on the linear portion, the inductance L takes a fixed value L0, but as shown by the inductance L2 in Figure 5 at any point on the non-linear portion, the inductance L takes a value different from L0 and varies according to the point. That is, when the current I and the magnetic flux Φ are linearly related, the inductance L takes a fixed value, and when the current increases and the current I and the magnetic flux Φ become non-linearly related, the inductance L changes. This is because magnetic saturation occurs when the current increases. Additionally, the difference between the inductance L at the point on the non-linear portion and the inductance L0 at the linear portion increases as the current increases. In order to detect such a change in inductance with respect to L0, it is preferable to directly calculate the inductance L based on the magnetic flux Φ and the current I, but there is a problem that it is difficult to obtain the magnetic flux Φ. Thus, in this embodiment, Figure 5 the tangent of the curve shown in, that is, dΦ / dI, is defined as the inductance change characteristic quantity ΔL (= dΦ / dI), and the inductance change with respect to L0 is detected using the inductance change characteristic quantity ΔL.

[0050] The inductance change characteristic quantity ΔL at the point P1 on the linear part is consistent with the inductance L0 at the linear part. Similarly, at any point on the linear part, the inductance change characteristic quantity ΔL is L0. Thus, the inductance change characteristic quantity ΔL at any point on the linear part is ΔL = L0, which is fixed. On the other hand, the inductance change characteristic quantity ΔL at any point on the non-linear part such as the point P2 takes a value different from the inductance change characteristic quantity ΔL = L0 at the linear part and gradually approaches 0 as the current increases. Such a change characteristic of the inductance change characteristic quantity ΔL, that is, being fixed at L0 on the linear part and deviating from the inductance change characteristic quantity L0 at the linear part as the current increases on the non-linear part, is the same as the change characteristic of the inductance L. Therefore, by monitoring the inductance change characteristic quantity ΔL, the change of the inductance can be detected. In addition, as will be described later, different from the inductance, the inductance change characteristic quantity ΔL can be easily calculated without using the magnetic flux Φ. Thus, in this embodiment, the inductance change detection unit 21 calculates the inductance change characteristic quantity ΔL to detect the inductance change.

[0051] For the inductance change detection unit 21, as Figure 2 shown, the three-phase alternating current Iu, Iv, and Iw, and the previous values Vu*_z, Vv*_z, and Vw*_z of the three-phase alternating voltage command are input. Then, the inductance change detection unit 21 calculates the inductance change characteristic quantity ΔL using the formula V = ΔL×(dI / dt) for the phase with the largest absolute value of the current among the three-phase alternating currents Iu, Iv, and Iw. In addition, regarding V = ΔL×(dI / dt), it can be derived by using dΦ / dI = ΔL for the formula V = dΦ / dt that holds between the voltage V applied to the coil and the magnetic flux Φ generated in the coil.

[0052] Figure 6 and 7 are diagrams illustrating the sampling period of the current used in the calculation of the inductance change characteristic quantity. Figure 6 and 7 In, the vertical axis is the current and the horizontal axis is the time, showing an example of the current waveform of one phase of the three-phase alternating current. The inductance change detection unit 21 is as Figure 6As shown, dI / dt is obtained using the minimum value I_min and the maximum value I_max of the current represented by square dots in the actual current. Therefore, the inductance change detection unit 21 acquires the current values at the timing when the current ripple becomes the minimum value and the maximum value. For example, in the case where the average value (dots) of the current is acquired every sampling period Ts as described above, the inductance change detection unit 21 acquires the minimum value of the current ripple at the time (4N + 1)×Ts / 4 (N is an integer), and acquires the maximum value of the current ripple at the time (4N + 3)×Ts / 4. In this case, the time interval between the minimum value I_min and the maximum value I_max of the current ripple acquired within the sampling period Ts of the average value of the current is approximately Ts / 2. Thus, the inductance change detection unit 21 calculates dI / dt as (I_max - I_min) / (Ts / 2). Then, the inductance change detection unit 21 substitutes the calculated dI / dt and the previous value of the AC voltage command of the phase in which dI / dt is calculated into V = ΔL×(dI / dt) to calculate the inductance change characteristic amount ΔL, and outputs the inductance change characteristic amount ΔL to the current phase angle control unit 22.

[0053] In addition, the inductance change characteristic amount ΔL is calculated for the phase with the largest absolute value of the current among the phase currents Iu, Iv, and Iw as described above, but it is not limited thereto. For example, the inductance change characteristic amount ΔL can be calculated for each phase regardless of the magnitudes of the absolute values of the phase currents Iu, Iv, and Iw, and the inductance change characteristic amount ΔL that deviates the most from L0 among them can be output to the current phase angle control unit 22.

[0054] When the carrier frequency fc is high, the inductance change detection unit 21 can Figure 7 as shown, alternately acquire the minimum value I_min of the current ripple and the maximum value I_max of the current ripple every 1 / fc. In this case, the inductance change detection unit 21 regards the acquired minimum value I_min (blackened square dots) of the current ripple as the minimum value (dashed square dots) of the current ripple in the next cycle to calculate dI / dt.

[0055] The current phase angle control unit 22 is as Figure 2As shown, the current phase angle θcrr corresponding to the inductance change characteristic amount ΔL calculated by the inductance change detection unit 21 is determined. Here, the current phase angle θcrr is the leading phase angle of the current vector determined by the three-phase alternating current Iu, Iv, and Iw with respect to the q-axis. The current phase angle control unit 22 preferably obtains and stores in advance the current phase angle θcrr with the minimum loss for each inductance change characteristic amount ΔL. For example, the current phase angle θcrr with the minimum loss is obtained in advance for each inductance change characteristic amount ΔL through experiments or analyses, etc. In addition, when the characteristics of the motor, etc. are clear, the current phase angle θcrr with the minimum loss can also be calculated and obtained in advance for each inductance change characteristic amount ΔL based on mathematical formulas. The current phase angle control unit 22 refers to the current phase angle θcrr of each inductance change characteristic amount ΔL obtained in advance and outputs the current phase angle θcrr corresponding to the input inductance change characteristic amount ΔL.

[0056] The carrier frequency switching unit 23 determines the switching of the carrier frequency based on the current phase angle θcrr and the estimated rotational position θe of the motor 2. Therefore, the direction of the current vector is obtained based on the estimated rotational position θe and the current phase angle θcrr of the motor 2, and the phase angles of the U-phase current, the V-phase current, and the W-phase current are obtained. The carrier frequency switching unit 23 outputs a switching flag FlgFrqSw as described later based on the phase angles of the U-phase current, the V-phase current, and the W-phase current. For example, for two carrier frequencies, a high frequency and a low frequency, when the carrier frequency is set to the high frequency, the switching flag FlgFrqSw is set to 1 and output, and when the carrier frequency is set to the low frequency, the switching flag FlgFrqSw is set to 0 and output. The details of the operation of the carrier frequency switching unit 23 are described later.

[0057] The current command generation unit 10 calculates the d-axis current command Id* and the q-axis current command Iq* based on the input torque command T*, the power supply voltage Hvdc, and the current phase angle θcrr. Here, for example, using a preset current command map and a mathematical formula representing the relationship between the d-axis current Id, the q-axis current Iq, and the motor torque, etc., the d-axis current command Id* and the q-axis current command Iq* corresponding to the torque command T*, the power supply voltage Hvdc, and the current phase angle θcrr are obtained. In addition, after obtaining the d-axis current command Id* and the q-axis current command Iq* that satisfy the requirements of the torque command T* and the power supply voltage Hvdc, the d-axis current command Id* and the q-axis current command Iq* can also be corrected using the current phase angle θcrr to obtain the final d-axis current command Id* and the q-axis current command Iq*.

[0058] The current conversion unit 12 performs dq conversion on the three-phase AC currents Iu, Iv, and Iw detected by the current detection unit 7 based on the rotational position θr obtained by the rotational position detector 4, and calculates the d-axis current value Id and the q-axis current value Iq.

[0059] Based on the deviation between the d-axis current command Id* and the q-axis current command Iq* output from the current command generation unit 10 and the d-axis current value Id and the q-axis current value Iq output from the current conversion unit 12, the current control unit 13 calculates the d-axis voltage command Vd* and the q-axis voltage command Vq* corresponding to the torque command T* in such a way that these values are made to match respectively. Here, for example, by means of a control method such as PI control, the d-axis voltage command Vd* corresponding to the deviation between the d-axis current command Id* and the d-axis current value Id and the q-axis voltage command Vq* corresponding to the deviation between the q-axis current command Iq* and the q-axis current value Iq are calculated.

[0060] The three-phase voltage conversion unit 18 calculates and outputs three-phase AC voltage commands Vu*, Vv*, and Vw* (U-phase voltage command Vu*, V-phase voltage command Vv*, and W-phase voltage command Vw*) based on the d-axis current command Id* and the q-axis current command Iq* output from the current command generation unit 10 and the estimated rotational position θe calculated by the phase calculation unit 16.

[0061] The gate signal generation unit 19 performs pulse width modulation on the three-phase AC voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18 using the carrier signal Sc output from the carrier generation unit 15, and generates gate signals for controlling the operation of the inverter 3. Specifically, based on the comparison result between the three-phase AC voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18 and the carrier signal Sc output from the carrier generation unit 15, pulsed voltages are generated for each of the U-phase, V-phase, and W-phase. Then, based on the generated pulsed voltages, pulsed gate signals corresponding to the switching elements of each phase of the inverter 3 are generated. At this time, the logical inverses of the gate signals Gup, Gvp, and Gwp of the upper arms of each phase are respectively obtained to generate the gate signals Gun, Gvn, and Gwn of the lower arms. The gate signals generated by the gate signal generation unit 19 are output from the motor control device 1 to the PWM signal drive circuit 32 of the inverter 3, and are converted into PWM signals by the PWM signal drive circuit 32. Thereby, the on / off control of each switching element of the inverter circuit 31 is performed to adjust the output voltage of the inverter 3.

[0062] Figure 8 It is a diagram showing the relationship between the carrier frequency and the current ripple. The vertical axis represents current and the horizontal axis represents time. A portion within the time width T1 is extracted from the current waveform of one of the three-phase AC currents. Additionally, Figure 8 the carrier frequency in (b) is higher than Figure 8The carrier frequency of (a) is high. Figure 8 In (a), the current ripple includes two cycles within the time width T1. On the other hand, Figure 8 In (b), the current ripple includes four cycles within the same time width T1. Regarding the current ripple included within the same time width T1, Figure 8 more of (b) has a higher carrier frequency. Therefore, regarding the current difference ΔI and the time width ΔT used to calculate dI / dt, Figure 8 those of (b) with a higher carrier frequency are smaller. When ΔI and ΔT decrease, the calculation accuracy of dI / dt also improves. Therefore, Figure 8 (b) with a higher carrier frequency can calculate dI / dt with better accuracy. Thus, the accuracy of the inductance change characteristic quantity ΔL calculated using dI / dt also improves. Then, the motor control device of the present embodiment increases the carrier frequency in the phase interval where the inductance changes, that is, in the phase interval including the phase where the absolute value of the alternating current of any phase of the three-phase alternating current is the largest. Thereby, the inductance change can be detected with good accuracy.

[0063] Figure 9 is a diagram illustrating the relationship between the three-phase alternating current and the carrier frequency. Figure 9 is a diagram in which the gate signal is superimposed on the current waveform of the three-phase alternating current, and the horizontal axis is the phase. Additionally, Figure 9 in PR1 is the first phase interval, PR2 is the second phase interval, and the line width of the gate signal indicates the high or low carrier frequency. The lower the carrier frequency, the thicker the line width of the gate signal. The carrier frequency adjustment unit 14, as Figure 9 shown, increases the frequency of the carrier in the first phase interval PR1 including the phase where the absolute value of the alternating current of any phase of the three-phase alternating current becomes the largest. Within the range of one cycle (2π) of the phase of the three-phase alternating current, there are three phases where any one of Iu, Iv, and Iw is the largest, and there are three phases where any one of Iu, Iv, and Iw is the smallest. Therefore, the first phase interval PR1 is preset to include six phases where any one of Iu, Iv, and Iw is the largest or the smallest. Figure 9 shows an example where the six regions near the six phases where any one of Iu, Iv, and Iw is the largest or the smallest, that is, the six regions where the line width of the gate signal is thinner, are used as the first phase region PR1, but it is not limited thereto. The carrier frequency switching unit 23 sets the switching flag FlgFrqSw to 1 when the phase of the three-phase alternating current calculated based on the current phase angle θcrr and the estimated rotational position θe of the motor 2 is within the first phase interval PR1. Then, the carrier frequency adjustment unit 14 sets the frequency of the carrier to a high frequency when the switching flag FlgFrqSw is 1. In this way, by increasing the carrier frequency in the phase interval where the inductance changes, the inductance change can be detected with good accuracy.

[0064] In addition, in a second phase interval PR2 where the absolute value of an alternating current including any phase of a three-phase alternating current becomes 0 as shown, the carrier frequency adjustment unit 14 lowers the carrier frequency. Within one cycle (2π) of the phases of the three-phase alternating current, there are six phases where any one of Iu, Iv, and Iw is 0. Therefore, the second phase interval PR2 is preset to include the six phases where any one of Iu, Iv, and Iw is 0. Figure 9 In, an example is shown in which a region of six ranges with a relatively thick line width of the gate signal, that is, the vicinity of the six phases where any one of Iu, Iv, or Iw is 0, is used as the second phase interval PR2, but it is not limited thereto. When the phase of the three-phase alternating current calculated based on the current phase angle θcrr and the estimated rotational position θe of the motor 2 is within the second phase interval PR2, the carrier frequency switching unit 23 sets the switching flag FlgFrqSw to 0. Then, when the switching flag FlgFrqSw is 0, the carrier frequency adjustment unit 14 sets the carrier frequency to a low frequency. In this way, by lowering the carrier frequency in a phase interval where inductance changes are less likely to occur, switching losses can be suppressed. Figure 9 is a diagram for explaining the relationship between loss and the current phase angle. As

[0065] Figure 10 shown, the current phase angle with the minimum loss deviates from the current phase angle with the minimum current. In the present invention, instead of controlling the current phase angle so that the current becomes minimum, the change characteristic amount of the inductance is calculated based on the measured current value, and the current phase angle is controlled based on the calculated change characteristic amount of the inductance, whereby the loss can be minimized. Figure 10 shown, the current phase angle with the minimum loss deviates from the current phase angle with the minimum current. In the present invention, instead of controlling the current phase angle so that the current becomes minimum, the change characteristic amount of the inductance is calculated based on the measured current value, and the current phase angle is controlled based on the calculated change characteristic amount of the inductance, whereby the loss can be minimized.

[0066] Description of Reference Numerals

[0067] 1... Motor control device, 2... Motor, 3... Inverter, 4... Rotational position detector, 5... High-voltage battery, 7... Current detection unit, 8... Rotational position sensor, 10... Current command generation unit, 11... Speed calculation unit, 12... Current conversion unit, 13... Current control unit, 14... Carrier frequency adjustment unit, 15... Carrier generation unit, 16... Phase calculation unit, 18... Three-phase voltage conversion unit, 19... Gate signal generation unit, 21... Inductance change detection unit, 22... Current phase angle control unit, 23... Carrier frequency switching unit, 31... Inverter circuit, 32... PWM signal drive circuit, 33... Smoothing capacitor, 34... Voltage detection unit, 100... Motor drive system.

Claims

1. A motor control device is connected to a power converter and controls the power conversion of the power converter, wherein the power converter performs the power conversion from DC power to three-phase AC power to drive a motor with the three-phase AC power, and the motor control device is characterized by comprising: an inductance change detection unit that calculates an inductance change characteristic quantity representing a change in the inductance of the motor in a first phase interval where the absolute value of the AC current in any phase including the three-phase AC current becomes maximum; and a current phase angle control unit that controls a current phase angle based on the inductance change characteristic quantity.

2. The motor control device according to claim 1, wherein: it further has a three-phase AC voltage command generation unit that generates a three-phase AC voltage command, and the inductance change detection unit calculates the inductance change characteristic quantity based on the three-phase AC voltage command and the three-phase AC current of the motor.

3. The motor control device according to claim 2, wherein: the inductance change detection unit calculates the inductance change characteristic quantity based on the time derivative of the three-phase AC current and the three-phase AC voltage command.

4. The motor control device according to claim 2, characterized in that, It comprises: a gate signal generation unit that performs pulse width modulation on the three-phase AC voltage command to generate a gate signal for controlling the power conversion of the power converter; and a carrier frequency adjustment unit that adjusts the frequency of a carrier used in the pulse width modulation based on the current phase angle and the rotational position of the motor, wherein the carrier frequency adjustment unit increases the frequency of the carrier in the first phase interval compared to the frequency of the carrier in a second phase interval where the absolute value of the AC current in any phase including the three-phase AC current becomes zero.

5. A power conversion system, characterized by: comprising the motor control device according to any one of claims 1 to 4 and the power converter.

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