Design method and system of wideband active emi filter based on phasor analysis

Abstract

This invention belongs to the field of electromagnetic interference filter design technology, and discloses a broadband active EMI filter design method and system based on phasor analysis, including: modeling the transfer function and impedance of each component of the AEF in the target frequency band using initial active parameters; calculating the phase difference θ between the interference current generated by the common-mode interference source and the compensation current output by the AEF. Δ And calculate the phase difference θ between the current on the grid side and the phase difference θ between them. c Adjust the initial active parameters to reduce θ in the frequency band where the interference amplitude exceeds the limit. Δ The absolute value of the difference from 180° is Δθ Δ or reduce θ c The absolute value of the difference from 90° is Δθ c The common-mode EMI spectrum of the AEF (Active Filter) after inserting a common-mode interference source is measured based on the adjusted active parameters. When the common-mode EMI spectrum meets EMC standards within a preset margin, the corresponding active parameters are the target design parameters of the AEF. This invention can improve the interference suppression performance of active EMI filters over a wide frequency band.

Description

TECHNICAL FIELD

[0001] The present application belongs to the technical field of electromagnetic interference filter design, and more particularly, to a wideband active EMI filter design method and system based on phasor analysis. BACKGROUND

[0002] Electromagnetic interference (EMI) is an unavoidable problem in the operation of power electronic devices, because the pulse width modulation (PWM) method determines that the power switch will be in a high-speed on-off state, resulting in a large voltage jump (dv / dt) and current jump (di / dt), forming an EMI source containing rich high-frequency components. Among them, common mode (CM) EMI interference will cause excessive ground leakage current and aggravate radiated EMI, ultimately leading to abnormal device operating state, so CM EMI is the key to suppression in the design process of power electronic devices. For CM EMI suppression, the traditional strategy relies on inserting passive EMI filters mainly composed of inductors and capacitors in the interference propagation path to achieve interference blocking or shunting, while the high-power density development trend of power electronic equipment puts forward more urgent demands for the miniaturization and light weight of EMI filters.

[0003] In order to reduce the volume and weight of EMI, active suppression methods represented by active EMI filters (AEF) have attracted widespread attention from academia and industry. AEF cancels EMI current by generating a signal opposite to the interference, breaking the dependence on passive components, and its volume and weight are much smaller than passive EMI filters composed of inductors and capacitors at the same power level. At the same time, AEF has higher control freedom, which means that the insertion loss of AEF can be precisely controlled in different frequency bands, so that the interference spectrum can be reduced below the electromagnetic compatibility (EMC) standard line in a wide frequency range. In order to optimize the interference suppression performance of AEF, topology optimization, parameter fine design and impedance matching are generally used, among which: topology optimization refers to the use of a multi-stage cascaded structure for the internal amplification circuit of AEF, thereby improving the effective bandwidth and loop gain of AEF; parameter fine design refers to calculating the required insertion loss of AEF according to the interference spectrum and standard limit, thereby precisely designing the parameters to make the actual insertion loss approximate the required value in the full frequency band; impedance matching refers to adjusting the impedance on the interference source side and the impedance on the grid side to ensure that the AEF output current is shunted to the grid side to the greatest extent, thereby realizing effective cancellation of interference on the grid side.

[0004] The AEF designed based on the existing method can theoretically output a compensation signal opposite to the interference, but the non-ideal characteristics of each link of the actual circuit will cause a phase difference between the compensation signal and the interference signal, resulting in that the interference compensation effect is seriously weakened. It is urgent to improve the AEF design method considering the phase relationship. SUMMARY

[0005] In view of the above defects or improvement needs of the prior art, the present application provides a wideband active EMI filter design method and system based on phasor analysis, which aims to improve the interference suppression performance of the active EMI filter in a wide frequency band.

[0006] To achieve the above-mentioned purpose, the present application provides a wideband active EMI filter design method based on phasor analysis, comprising:

[0007] determining initial active parameters of an active EMI filter AEF;

[0008] modeling transfer functions of each link of the AEF in a target frequency band and common-mode interference source impedance Z CM and grid-side impedance using the initial active parameters;

[0009] based on the transfer functions, the common-mode interference source impedance Z CM and the grid-side impedance, calculating a phase difference θ Δ between an interference current generated by the common-mode interference source and a compensation current output by the AEF, and calculating a phase difference θ c between a current of the grid side and the compensation current output by the AEF;

[0010] adjusting the initial active parameters to reduce an absolute value Δθ Δ of the difference between the phase difference θ Δ and 180°, or to reduce an absolute value Δθ c of the difference between the phase difference θ c and 90° in a frequency band where the interference amplitude exceeds the limit, measuring a common-mode EMI spectrum of the AEF after the common-mode interference source is inserted based on the adjusted active parameters, and when the common-mode EMI spectrum in the target frequency band meets the EMC standard within a preset margin, the corresponding active parameters are the target design parameters of the AEF.

[0011] Further, the links of the AEF include a sampling link, a signal amplification link and a signal compensation link; the initial active parameters include a turns ratio n of a current transformer CT used in the sampling link, a ratio R f of a feedback resistor R and an input resistor R1 of an inverting amplifier used in the signal amplification link.f / R1, the compensation resistor R in the signal compensation circuit comp With compensation capacitor C comp .

[0012] Furthermore, the initial active parameters are adjusted to reduce the phase difference θ in the frequency band where the interference amplitude exceeds the limit. Δ The absolute value of the difference from 180° is Δθ Δ Or reduce the phase difference θ c The absolute value of the difference from 90° is Δθ c ,include:

[0013] Calculate the low corner frequency f of the CT transfer function in the frequency domain. L-CT and high transition frequency f H-CT ; Frequency f within the target frequency band <f L-CT When, or f>f H-CT At that time, reduce the primary-to-secondary turns ratio n of the CT to reduce Δθ. Δ ;f L-CT ≤f≤f H-CT When, determine f and θ c The turning frequency f across 90° c-θc The relationship between f <f c-θc Reduce the primary-to-secondary turns ratio n of the CT to reduce Δθ. c If f≥f c-θc Then increase the primary-to-secondary turn ratio n of CT to reduce Δθ. c ;

[0014] Calculate the corner frequency f of the closed-loop gain of the inverting amplifier. c-op ;f≥f c-op When R is reduced f / R1, to reduce Δθ Δ ;f <f c-op When, determine f and f c-θc The relationship between f <f c-θc Reduce R f / R1, to reduce Δθ c If f≥f c-θc Then increase R f / R1, to reduce Δθ c ;

[0015] Calculate the compensation resistor R comp With the compensation capacitor C comp The corner frequency f of the series equivalent impedance c-comp ;f <f c-comp When, increase the compensation capacitor C comp To reduce Δθ Δ ;f≥f c-comp When, determine f and fc-θc The relationship between f <f c-θc Increase the compensation resistor R comp The value of Δθ is used to reduce the value of Δθ. c If f≥f c-θc Then reduce the compensation resistor R comp The value of Δθ is used to reduce the value of Δθ. c .

[0016] Furthermore, the low transition frequency f L-CT and high transition frequency f H-CT The calculation method is as follows:

[0017]

[0018] Among them, L m1 R is the equivalent inductance of CT. c The equivalent resistance, R, is used to characterize the loss. sense For sensing resistance; C pp The parasitic capacitance of the primary edge of the CT;

[0019] The transition frequency f c-op With R f The following conditions must be met between / R1:

[0020]

[0021] Among them, G op f1 and f2 are the open-loop gain of the operational amplifier in the inverting amplifier, and f1 and f2 are the two open-loop corner frequencies of the operational amplifier, respectively.

[0022] The transition frequency f c-comp The calculation method is as follows:

[0023] Furthermore, the phase difference θ Δ and the phase difference θ c The calculation method is as follows:

[0024]

[0025] Where A is Amplitude, The compensation current representing the output of AEF With interference current The quotient, whose phase is θ Δ G CT G is the transfer function of CT. A Z represents the transfer function of an inverting amplifier; || denotes the parallel connection symbol; Z LISN Z is the impedance on the grid side. L Z represents the equivalent impedance of an external magnetic ring, and is an empirical value.out The output impedance of the push-pull amplifier used for the signal amplification link.

[0026] Further, the transfer function G CT of the inverting amplifier is calculated as follows: A

[0027]

[0028] Where s is the Laplace operator, R c is the equivalent resistance representing the loss, R sense is the induced resistance; C pp is the parasitic capacitance of the primary side of the CT, L m is the excitation inductance; Z o is the output impedance of the operational amplifier in the inverting amplifier, Z in The input impedance of the push-pull amplifier used for the signal amplification link.

[0029] The application also provides a wide-frequency active EMI filter obtained by the wide-frequency active EMI filter design method based on phasor analysis.

[0030] The application also provides a wide-frequency active EMI filter design system based on phasor analysis, comprising a computer readable storage medium and a processor.

[0031] The computer readable storage medium is used to store executable instructions.

[0032] The processor is used to read the executable instructions stored in the computer readable storage medium to execute the wide-frequency active EMI filter design method based on phasor analysis.

[0033] The application also provides a computer readable storage medium, which stores a computer program, and the program is executed by the processor to realize the wide-frequency active EMI filter design method based on phasor analysis.

[0034] The application also provides a computer program product, which comprises a computer program, and when the computer program is executed on a computer, the computer program makes the computer execute the wide-frequency active EMI filter design method based on phasor analysis.

[0035] Overall, the above technical solutions conceived by the application can achieve the following beneficial effects:

[0036] (1) The application takes into account the phase difference θ between the interference current and the compensation current output by the AEF due to the non-ideal characteristics of the internal circuit of the AEF.​Δ Typically, the angle deviates by 180°; at this point, the AEF outputs a compensation current. Neutralizing interference current The component with a 90° phase difference not only fails to contribute to interference compensation but also amplifies the interference. Therefore, the compensation current output by AEF... After a phase shift occurs, the interference compensation performance of AEF is severely affected. Further analysis reveals that the factors determining the effectiveness of interference compensation include not only... and The phase difference θ between Δ There is also AEF output compensation current. Amplitude I c Size; when θ Δ Fixed, I c When changes occur, the current on the grid side and The phase difference θ between c When the angle is 90°, the current amplitude I on the grid side r It is possible to achieve the minimum value. Based on the above analysis, the optimization objective of the constructed AEF is to reduce... and Phase difference θ Δ The absolute value of the difference from 180° is Δθ Δ ,as well as and The phase difference θ between c The absolute value of the difference from 90° is Δθ c By adjusting the active parameters of AEF, Δθ can be reduced in the frequency band where the interference amplitude exceeds the limit. Δ or Δθ c This invention ensures that the AEF (Active Element Filter) corresponding to the adjusted active parameters, after insertion into the target power electronic device, meets EMC standards within the target frequency band for the CM EMI spectrum. The resulting AEF parameters are then the required active EMI filter parameters, and the active EMI filter based on these AEF parameters also meets the requirements. This invention solves the problem of traditional common-mode EMI filter design methods in power electronic devices neglecting the phase relationship between the compensation current and the interference current, thereby fully optimizing the filter's interference suppression performance over a wide frequency band.

[0037] (2) Furthermore, this invention simplifies the complex broadband optimization problem into the calculation of several cutoff frequencies, and establishes a mapping relationship between the key parameters of AEF and its interference suppression performance in each frequency band, which has important guiding significance for the broadband performance optimization of AEF. The method of this invention is highly feasible in actual engineering practice. In addition, the proposed design method requires no additional software or hardware and has strong portability.

[0038] In summary, the application considers that due to the non-ideal characteristics of the internal links of the AEF, the compensation current and the interference current are not necessarily inversely related, but an additional phase shift is introduced, and the characteristics change with the frequency, which seriously affects the interference suppression performance of the AEF in a wide frequency range (10 kHz to 10 MHz). In order to solve this problem, the application proposes an AEF optimization design method based on the phase relationship between the compensation current and the interference current, which breaks through the limitation of the traditional AEF design scheme that only considers the insertion loss as a single evaluation factor and does not consider the phase information. Based on the cutoff frequency at which the phase response of each link turns over, the influence mechanism of each key parameter of the AEF on its attenuation degree is revealed, which can help the AEF to achieve interference suppression performance optimization in the full frequency range. BRIEF DESCRIPTION OF DRAWINGS

[0039] Figure 1 A schematic diagram of the common-mode path of the inverter in the embodiment of the application;

[0040] Figure 2 A schematic diagram of the principle of the AEF in the embodiment of the application;

[0041] Figure 3 An equivalent circuit diagram of the common-mode path after the AEF is adopted in the embodiment of the application;

[0042] Figure 4 A circuit diagram of the internal structure of the AEF in the embodiment of the application;

[0043] Figure 5 A comparison diagram of the frequency response modeling results of the key links and the network analyzer measurement results in the embodiment of the application; wherein, Figure 5 (a) in the figure is the frequency response of the current transformer CT; Figure 5 (b) in the figure is the frequency response of the signal amplification link; Figure 5 (c) in the figure is the output impedance of the operational amplifier and the push-pull amplifier;

[0044] Figure 6 A simplified equivalent circuit diagram of the common-mode path containing the AEF in the embodiment of the application;

[0045] Figure 7 A phase relationship between the compensation current, the common-mode interference current and the compensated current of the AEF in the embodiment of the application; wherein, Figure 7 (a) in the figure is the case without introducing phase shift, i.e. θ Δ = 180°; Figure 7 (b) in the figure is the case of introducing phase shift, i.e. θ Δ ≠ 180°;

[0046] Figure 8Time-domain waveforms of the AEF compensation current, common-mode interference current and compensated current in the embodiment of the present application; wherein, Figure 8 (a) is the case when θ Δ = 180°, the interference cancellation effect is significant; Figure 8 (b) is the case when θ Δ ≠ 180°, the interference cancellation effect is greatly weakened;

[0047] Figure 9 Time-domain waveforms of the AEF compensation current, common-mode interference current and compensated current in the embodiment of the present application; wherein, Δ ≠ 180°, the current phase relationship is shown in the following figure: Figure 9 (a) is the case when θ Δ is unchanged, and the compensation current amplitude changes; Figure 9 (b) is the case when the compensation current amplitude is unchanged, and θ Δ changes;

[0048] Figure 10 Time-domain waveforms of the AEF compensation current, common-mode interference current and compensated current in the embodiment of the present application; wherein, Δ and θ c are calculated values in the frequency range of 10 kHz to 10 MHz;

[0049] Figure 11 Frequency response schematic diagram of the AEF key link in the embodiment of the present application; wherein, Figure 11 (a) is the frequency response of CT; Figure 11 (b) is the frequency response of the open-loop gain and closed-loop gain of the inverting amplifier; Figure 11 (c) is the impedance of the compensation link;

[0050] Figure 12 Amplitude-frequency characteristics of CT, inverting amplifier and impedance characteristics of the compensation link in the embodiment of the present application;

[0051] Figure 13 Flow chart of the AEF design method based on phasor analysis in the embodiment of the present application;

[0052] Figure 14 Schematic diagram of the AEF design method based on phasor analysis in the embodiment of the present application;

[0053] Figure 15 Two EMI measured frequency spectrums that do not meet the standard in the embodiment of the present application;

[0054] Figure 16 EMI measured frequency spectrum that meets the standard after optimizing the AEF parameters in the embodiment of the present application, and the comparison with the frequency spectrum of the non-standard working condition. DETAILED DESCRIPTION

[0055] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the invention. Furthermore, the technical features involved in the various embodiments of this invention described below can be combined with each other as long as they do not conflict with each other.

[0056] Example 1

[0057] In this embodiment of the invention, the power electronic device selects an inverter and analyzes the common-mode path of the inverter.

[0058] The common-mode interference path of a typical inverter is as follows: Figure 1 As shown, the high-speed switching of the power switch causes a large voltage jump dv / dt at the midpoint of the bridge arm, and this jump is transmitted through the parasitic capacitance C between the power module and the load to the heat sink. p Induced common-mode current i CM As shown in equation (1). Then, i CM The signal returns to the line impedance stabilization network (LISN) via the device's protective ground, and finally to the DC bus (grid). The LISN's function is to isolate interference introduced by the DC power supply and simulate the grid-side impedance, providing a reference for measuring common-mode interference of the target device (power electronic device).

[0059]

[0060] To compensate for common-mode EMI current, the AEF (Automatic External Filter) needs to sample the EMI current in the interference loop, generate a corresponding compensation current through signal processing circuitry, and finally inject it back into the interference loop to achieve EMI current compensation. The principle is as follows: Figure 2 As shown in the diagram. In the sampling stage, current-sampling AEFs typically use current transformers (CTs) to acquire common-mode current and convert it into a voltage signal. For the signal amplification stage, an inverting amplifier is typically used for signal inversion and voltage amplification, and a push-pull amplifier for power amplification. The compensation stage is usually an RC branch used to isolate the internal circuitry of the AEF from the DC power loop, while simultaneously injecting the amplified compensation signal (the output current of the signal amplification stage) into the interference loop, i.e., into the power grid, to cancel common-mode electromagnetic interference. Since the AEF constructs a loop from the converter to the PE, the compensation capacitor should be considered a Y capacitor, and its value should not exceed the leakage current standard's limit on the Y capacitor value. Figure 3 The common-mode loop circuit after inserting AEF is shown.

[0061] Figure 4The equivalent circuit diagrams of the CT, inverting amplifier, and push-pull amplifier are shown. First, the CT converts the common-mode current i... CM Converted into the input voltage V of the signal amplification stage in , where C pp With C sp These are the parasitic capacitances of the primary and secondary sides of the CT, C sp =n 2 C pp L m For the magnetizing inductance, R c The equivalent resistance characterizing the loss is the transfer function G of the CT. CT It can be represented by equation (2), I CM Let be the output current of the CT, which is also the input current of the signal amplifier; n be the turns ratio of the primary and secondary sides of the CT; s be the Laplace operator; and R be the input current of the signal amplifier. sense This is the sensing resistor. Subsequently, the inverting amplifier circuit uses an operational amplifier to convert the input voltage V... in Invert and amplify to V out After considering the non-ideal characteristics of the operational amplifier, the transfer function G of the inverting amplifier circuit is... A This can be expressed as equation (3). Where, V out R is the output voltage of the inverting amplifier circuit. f R1 and R2 are the feedback resistor and input resistor, respectively; G op The open-loop gain of the operational amplifier can be found in the technical datasheet. o Z is the output impedance of the operational amplifier. in This is the input impedance of the push-pull amplifier. The main function of a push-pull amplifier is to provide a constant voltage (V) while maintaining the input impedance of the amplifier. AEF ≈V op To construct a smaller output impedance Z out Its expression is shown in equation (5), thereby improving the load-carrying capacity of AEF. Wherein, R E Z is the emitter resistor of the transistor. ZoCBC Representing Z o Base resistance R B With base-collector capacitance C BC The parallel impedance, Z BE Represents the base-emitter resistance R BE With capacitor C BE Parallel impedance, r o β is the transistor output impedance, and β is the transistor gain. Figure 5 The modeling results of the above three stages are presented and compared with the measured results. It can be seen that the model established by the embodiment of the present invention has high accuracy in the 10kHz-10MHz frequency band of interest (target frequency band). At the same time, it can be seen that the output impedance Z of the push-pull amplifier is... out Less than the op-amp impedance Z o.

[0062]

[0063] In this embodiment of the invention, phasor analysis is performed on the AEF compensation current and interference current. After establishing equivalent models of each component of the AEF, it can be... Figure 3 Simplified to Figure 6 The circuit shown; where V CM As a common-mode interference voltage source, Z CM Both are common-mode interference source impedances, obtained through measurement; Z comp For compensation resistor R comp With compensation capacitor C comp The series equivalent impedance, which serves as the transfer function of the compensation element; Z out The output impedance of the push-pull amplifier circuit is calculated using equation (4); V AEF Z is the output voltage of AEF; LISN Z represents the equivalent impedance of the LISN; in practical applications, it represents the impedance on the grid side, obtained through measurement. L This is the equivalent impedance (empirical value) of an external magnetic ring, used to provide a high-impedance loop at high frequencies to compensate for the insufficient performance of the AEF at high frequencies. According to the principle of AEF, the interference current... (that is, the common-mode current i mentioned above) CM The sampled current is used as the input to AEF, and then AEF outputs the corresponding compensation current. Ultimately, the compensated current on the LISN side is achieved. Its amplitude is the smallest. Theoretically, and The phase difference θ between Δ Determined solely by the inverting amplifier, i.e., θ Δ =180°, therefore All components are used for The offset, such as Figure 7 As shown in (a) and (5). However, due to the additional phase shift introduced by the non-ideal characteristics of the internal circuitry of the AEF, θ Δ It usually deviates by 180°, such as Figure 7 (b) is shown in equation (6); at this time, the AEF outputs the corresponding compensation current. It can be decomposed into two orthogonal components and This component serves to compensate for interference; its current direction (phase) is opposite to the interference current. On the contrary, it can therefore play a role in interference compensation; and Phase and interference current A 90° phase difference not only fails to contribute to interference compensation but also amplifies the interference. Therefore, the compensation current corresponding to the AEF output... After a phase shift occurs, the interference compensation performance of AEF will be severely affected. Figure 8 The time-domain waveforms further illustrate the consequences of the phase shift: Figure 8 (a) in the diagram shows θ Δ When the angle is 180°, the compensation current and the interference current are approximately equal in magnitude but opposite in phase. Therefore, the current after compensation (the current on the LISN side) The amplitude is very small, and the AEF suppression performance is excellent; Figure 8 (b) shows θ Δ When the angle deviates by 180°, the current amplitudes before and after compensation are roughly the same, and AEF provides almost no interference compensation. Therefore, it is crucial to optimize the interference compensation capability of AEF by performing phasor analysis on the AEF compensation current and interference current.

[0064]

[0065] As can be seen from equation (6), the effect of interference compensation is determined by... Amplitude I r Not only sizes and The phase difference θ between Δ There is also the compensation current corresponding to the AEF output. Amplitude I c . Figure 9 The effects of these two factors will be analyzed separately. When θ Δ Fixed, I c When changes occur (I) c byI c1 Gradually increase to I c3 ),like Figure 9 As shown in (a) in the diagram; obviously, when and The phase difference θ between c When it is 90°, I r To reach the minimum value. When I c Fixed, θ Δ When changes occur, such as Figure 9 As shown in (b) in the figure, when θ Δ When I equals 180° r Obtain the minimum value. (By...) Figure 6 The circuit shown can be calculated using equation (7) based on Kirchhoff's circuit laws. Δ ;in, The compensation current representing the output of AEF With interference current The quotient, whose phase is θΔ ;|| represents the parallel circuit symbol. Meanwhile, based on the sine theorem, θ can be calculated. c As shown in equation (8). Figure 10 Showing θ Δ With θ c Calculated values ​​within the 10kHz-10MHz frequency band; where, f c-θc Represents θ c The frequency of crossing a 90° turn means that in f c-θc Previously, due to θ c If the angle is less than 90°, I needs to be reduced. c Make θ c Approaching 90°; while at f c-θc After that, due to θ c If the angle is less than 90°, I needs to be increased. c Make θ c Approximately 90°. To simplify the expression, equation (9) defines Δθ. Δ With Δθ c , which are the absolute values ​​of the differences between the two and 180° and 90°, respectively; therefore, according to phasor analysis, the optimization objective of AEF is to minimize Δθ as much as possible. Δ or Δθ c (equivalent to the current on the LISN side) (Minimizing the amplitude), the AEF parameters designed based on this optimization objective are the required active EMI filter parameters. In this embodiment of the invention, f c-θc You can directly use θ c Read the value, for example, based on Figure 10 As shown, f c-θc Represents θ c The frequency of the 90° turn can be used to read f at this time. c-θc =10 6 Hz; or, according to equation (8), let θ c =90°, at which point the calculated frequency is f. c-θc .

[0066]

[0067] Furthermore, in this embodiment of the invention, an analysis is conducted based on the internal electrical characteristics of the AEF to illustrate the specific design method for AEF parameters. This is done to minimize Δθ. Δ With Δθ c Therefore, a thorough analysis of the frequency characteristics of the internal components of AEF is required.

[0068] For the sampling stage, based on equation (2), equation (10) uses two corner frequencies f. L-CT with f H-CT The CT transfer function is divided into three parts in the frequency domain, where fL-CT and respectively the low and high cut-off frequencies of the CT transfer function in the frequency domain, L m1 is the equivalent inductance of the CT, As Figure 11 shown in (a) and f L-CT is the frequency at which the CT transfer function crosses the 90° phase shift, H-CT can be calculated by equation (11). When the frequency f < f L-CT within the target frequency band, the CT exhibits inductive behavior, thus the CT introduces a leading phase shift, i.e. contributes to θ Δ , θ c and the gain G CT-lowfre at this time (G L-CT of the CT when f < f L-CT ) increases with frequency; when f H-CT < f < f CT-ideal , the CT exhibits resistive behavior, at this time the gain G H-CT does not change with frequency and also does not introduce phase shift. When f > f CT-highfre , the CT exhibits capacitive behavior, at this time a lagging phase shift is introduced and the gain G L-CT at this time decreases with frequency. Ideally, when designing the AEF, one would like the resistive region of the CT to be as wide as possible, i.e. f H-CT as small as possible while f L-CT as large as possible; however, as can be seen from equation (10) and equation (11), the width of the resistive region is in conflict with the gain. For example, decreasing the CT primary-to-secondary turns ratio n, can decrease f H-CT and increase f CT-ideal , i.e. widen the width of the resistive region, however will result in a decrease of G Δ . In other words, the change of n has different effects on Δθ c and Δθ L-CT at different frequency bands, and needs to be balanced according to the actual spectrum distribution of the interference. Specifically, when the CT introduces phase shift, i.e. when the frequency f < f H-CT within the target frequency band, or f > f Δ , increasing the CT primary-to-secondary turns ratio n, will increase Δθ Δ , and vice versa, decreasing the CT primary-to-secondary turns ratio n, will decrease Δθ L-CT . When the CT does not introduce phase shift, i.e. when f H-CT , since n is increased, the gain at this time will also increase, at this time one needs to judge the relationship between f and the turning frequency f c at which θ c-θc crosses 90°, if f < f c-θc , the gain increase will result in an increase of Δθ c , i.e. f < f c-θcIncreasing the primary-to-secondary turns ratio n of the CT will increase Δθ. c Conversely, reducing the primary-to-secondary turns ratio n of the CT will reduce Δθ. c ;f≥f c-θc When the gain increases, Δθ will increase. c Decrease, i.e., f≥f c-θc Increasing the primary-to-secondary turns ratio n of the CT will decrease Δθ. c Conversely, reducing the primary-to-secondary turns ratio n of the CT will increase Δθ. c The ratio of the number of turns on the primary and secondary sides of the CT to Δθ Δ or Δθ c The specific impacts are shown in Table 1 below.

[0069]

[0070] For an inverting amplifier, equation (12) shows the expression for its gain, where G op f1 and f2 are the open-loop gain and two open-loop corner frequencies of the operational amplifier, respectively. The open-loop gain and corner frequencies can be obtained from the datasheet. c-op This is the cutoff frequency corresponding to the closed-loop gain of the inverting amplifier. Therefore, the frequency response of the inverting amplifier can be derived from... Figure 11 (b) in the text indicates that G op G is the open-loop gain of the operational amplifier. close The closed-loop gain of the operational amplifier is related to the feedback resistor R. f The ratio of the input resistance R1 to the input resistance R1 is equal. When f <f c-op At this time, the inverting amplifier has a fixed closed-loop gain R. f / R1, and the inverting amplifier does not introduce additional phase shift; when f≥f c-op At this time, the gain of the inverting amplifier is equal to the open-loop gain G. op At this point, the gain decreases with increasing frequency and introduces an additional phase shift. The intersection of the open-loop gain and the horizontal axis is called the gain-bandwidth product (GBW), which is determined by the properties of the operational amplifier itself. Therefore, when designing an AEF, it is desirable to use an operational amplifier with the highest possible GBW. In summary, once the operational amplifier is determined, the parameter that determines the frequency response of the inverting amplifier is R. f / R1:R f The larger / R1 is, the higher the closed-loop gain, but f c-op The smaller the value, the narrower the effective bandwidth; R f The smaller / R1 is, the lower the closed-loop gain, but f c-op The higher the value, the wider the effective bandwidth. In summary, R... fThe effect of / R1 on the inverting amplifier is similar to the effect of n on the CT; the effective bandwidth and gain are contradictory and need to be adjusted in conjunction with the actual interference characteristics. Specifically, when the inverting amplifier introduces a phase shift, i.e., f≥f c-op When, increase R f / R1 will increase Δθ Δ Conversely, reducing R f / R1 will decrease Δθ Δ When the inverting amplifier does not introduce a phase shift, i.e., f <f c-op At that time, R f As / R1 increases, the gain will also increase. At this point, it is necessary to determine the relationship between f and θ. c The turning frequency f across 90° c-θc The relationship between f <f c-θc When the gain increases, Δθ will increase. c Increase, i.e., f <f c-θc When, increase R f / R1 will increase Δθ c Conversely, reducing R f / R1 will decrease Δθ c ;f≥f c-θc When the gain increases, Δθ will increase. c Decrease, i.e., f≥f c-θc When, increase R f / R1 will reduce Δθ c Conversely, reducing R f / R1 will increase Δθ c R f / R1 for Δθ Δ or Δθ c The specific impacts are shown in Table 1 below.

[0071]

[0072] For the output compensation stage, equation (13) represents its corner frequency f. c-comp Obviously, for an RC series branch, in the low-frequency range, i.e., f <f c-comp At this frequency, the overall capacitance is capacitive, therefore the compensation capacitor introduces a 90° phase shift hysteresis, and the impedance decreases as the frequency increases; in the high-frequency range, i.e., f ≥ f c-comp When the capacitance impedance is negligible, the overall behavior is resistive, thus no phase shift is introduced, and the impedance value is equal to the compensation resistor R. comp ,like Figure 11 As shown in (c) in the diagram. Ideally, we would like f to... c-comp The smaller the better, i.e., the smaller the compensation resistor R. comp With compensation capacitor C compthe product of R and C is as large as possible; however, limited by the leakage current, the compensation capacitor has a maximum value, and R comp The increase of R c-comp may be equivalent to the increase of the internal resistance of AEF, which will lead to the decrease of the compensation current in the frequency band after f comp Therefore, R comp and C c-comp have a contradictory effect on the gain and effective bandwidth of the compensation link. Specifically, when the compensation capacitor introduces phase shift, i.e. f < f comp , increasing the compensation capacitor C Δ will reduce Δθ comp , and vice versa, reducing the compensation capacitor C Δ will increase Δθ c-comp . When the compensation capacitor does not introduce phase shift, i.e. f ≥ f comp , the increase of the compensation resistance R c will lead to the decrease of the gain, at this time, it is necessary to determine the relationship between f and θ c-θc , f < f c-θc , increasing the value of the compensation resistance R comp will reduce Δθ c , and vice versa, reducing the value of the compensation resistance R comp will increase Δθ c ; f ≥ f c-θc , increasing the value of the compensation resistance R comp will increase Δθ c , and vice versa, reducing the value of the compensation resistance R comp will reduce Δθ c . The effects of R comp and C comp on Δθ Δ or Δθ c are shown in Table 1 below.

[0073]

[0074] Table 1 Influence of key parameters on Δθ Δ or Δθ c

[0075]

[0076] In Table 1, ↑: positively correlated with parameter change; ↓: negatively correlated with parameter change; --: no direct relationship.

[0077] Figure 12 The frequency response and impedance of the three links of sampling, amplification and compensation are combined. It can be seen that the phase shift inside the AEF is affected by the three links, and the effect can be characterized by the cut-off frequencies. For example, when f c-op ​>f>f H-CT When the phase shift is introduced, only the CT is introduced, and the inverting amplifier and the compensation link are in the resistive working area, and the phase shift is not introduced. When the corresponding key parameters of each link change, the gain and the phase shift θ Δ of each frequency band will change accordingly, wherein, based on the value of θ c crossing 90°, f c-θc is judged, so as to represent the influence of the change of the gain on Δθ c . In addition, these cut-off frequencies will also change with the change of the parameters, so it is necessary to recalculate after adjusting the parameters each time.

[0078] As Figure 13 shown, the AEF design flow based on the phasor analysis in the embodiment of the application. First, the initial active parameters of the AEF are determined based on the experience values, including: the CT primary and secondary turns ratio n in the sampling circuit, the compensation resistance R comp and the compensation capacitance C comp in the signal compensation circuit, the ratio R f / R1 of the feedback resistance R f and the input resistance R1 of the inverting amplifier in the signal amplification circuit.

[0079] Based on the determined initial active parameters, the transfer function of the internal links of the AEF in the target frequency band is modeled, and the transformer impedance Z CM and the grid-side impedance are measured. The related calculation modes are shown in formula (2)-(4).

[0080] Subsequently, the CMEMI spectrum after the AEF is inserted in the target power electronic device is measured, if the EMC standard is met, the initial active parameters are directly taken as the parameter results of the AEF; if the standard is not met, the phase difference θ Δ between f and f is calculated, and the phase difference θ c between f and f is calculated, the related calculation modes are shown in formula (7) and formula (8); and each cut-off frequency f c-comp , f L-CT , f c-θc , f H-CT , f c-op is calculated.

[0081] According to each cut-off frequency, the target frequency band is divided into multiple subintervals; for the subinterval frequency band with the exceeding interference amplitude, the rule in table 1 is referred to, and each active parameter is adjusted to reduce Δθ Δ or Δθ cThen the interference spectrum is measured again, and this process is repeated until the measured CM EMI spectrum in the target frequency band under the active parameters of the current AEF meets the standard limit.

[0082] Figure 15 Two out-of-limit interference conditions and the corresponding parameter influence laws are shown, and Table 2 shows the respective cutoff frequencies of the two conditions. In Figure 15 (a) of FIG. 1, the interference is out of limit in the I-e frequency band (i.e., not up to standard, not meeting the EMC standard); since the interference in the I-d frequency band does not have sufficient margin from the standard limit (the margin of the interference from the standard limit is an empirical value of 6db), this frequency band should also be optimized. According to the parameter influence law, R f / R1 and R comp may be reduced to reduce Δθ c ; even if this method will cause an increase in Δθ c in the I-a, I-b and I-c frequency bands, the interference in these three frequency bands has sufficient margin, so a certain degree of AEF performance degradation is acceptable. In Figure 15 (b) of FIG. 1, the interference is out of limit in the II-a frequency band, and an effective method is to reduce Δθ comp in this frequency band by increasing C Δ ; in addition, R comp may be reduced to reduce Δθ c in the II-e frequency band, thereby improving the margin in this frequency band. Figure 16 The conditions before and after the parameters are optimized using the proposed AEF design method are shown. After the optimization design, the interference in the two originally out-of-limit frequency bands is lower than the standard limit, and the highest degree of attenuation is improved by 10dB; Table 3 compares the key parameters of the AEF corresponding to these conditions.

[0083] Table 2: Cutoff frequencies of two conditions

[0084]

[0085] Table 3: Key parameter values of AEF

[0086]

[0087] In summary, the AEF design method based on phasor analysis considers the phase relationship between the compensation current and the interference current, reveals the influence mechanism of the key parameters in different frequency bands on the interference suppression performance of the AEF, and has high reference value for the optimization design of the parameters in a wide frequency band.

[0088] In the embodiment of the present application, a kind of AEF design method based on compensation current and interference current phasor analysis is provided, and the dominant factor and influence mechanism of the attenuation degree of each frequency band are identified.Firstly, the frequency response and impedance of each link of common-mode interference loop are obtained by combining the transfer function model and the way of network analyzer measurement, including the impedance characteristics of inverter, line impedance stabilizing network and the transfer function of AEF internal link.Subsequently, based on the equivalent circuit of common-mode interference loop and the characteristics of AEF control loop, the phase relationship between AEF output compensation current and interference current Can be calculated.The optimization of interference suppression performance of AEF can ultimately be attributed to the adjustment of the phase difference between current, and there is a mapping relationship between phase angle and active control parameters inside AEF, so the current phase angle can be regulated based on this relationship by optimizing AEF parameters, and then the interference suppression performance of AEF in different frequency bands is targeted to improve.

[0089] Embodiment 2

[0090] The embodiment of the present application provides a wide-frequency active EMI filter obtained by the wide-frequency active EMI filter design method based on phasor analysis in the above-mentioned embodiment 1.

[0091] The related technical solutions are the same as above, and will not be repeated here.

[0092] Embodiment 3

[0093] The embodiment of the present application provides a wide-frequency active EMI filter design system based on phasor analysis, which comprises a memory and a processor, the memory stores a computer program, and the processor realizes the steps of the wide-frequency active EMI filter design method based on phasor analysis in the above-mentioned embodiment 1 when executing the computer program.

[0094] The related technical solutions are the same as above, and will not be repeated here.

[0095] Embodiment 4

[0096] The embodiment of the present application provides a computer readable storage medium, which stores a computer program, and the computer program is executed by a processor to realize the steps of the wide-frequency active EMI filter design method based on phasor analysis in the above-mentioned embodiment 1.

[0097] Specifically, the memory can include a high-speed random access memory, and can also include a non-volatile memory, such as a hard disk, a memory, a plug-in hard disk, a smart media card (SMC), a secure digital (SD) card, a flash card, at least one magnetic disk storage device, a flash memory device, or other volatile solid-state storage device.

[0098] The related technical solutions are the same as above, and will not be repeated here.

[0099] Embodiment 5

[0100] The embodiment of the present application provides a computer program product, comprising a computer program, when the computer program runs on a computer, so that the computer executes the steps of the wideband active EMI filter design method based on the phasor analysis in the above-mentioned embodiment 1.

[0101] The related technical solutions are the same as above, and will not be repeated here.

[0102] Those skilled in the art can easily understand that the above description is only the preferred embodiment of the present application, and is not used to limit the present application, and any modification, equivalent replacement and improvement made within the spirit and principle of the present application should be included in the protection scope of the present application.

Claims

1. A broadband active EMI filter design method based on phasor analysis, characterized in that, include: Determine the initial active parameters of the active EMI filter AEF; Using the initial active parameters, the transfer function and common-mode interference source impedance of each component of the AEF in the target frequency band are analyzed. Z CM Modeling the grid-side impedance; Based on the aforementioned transfer function and common-mode interference source impedance Z CM Calculate the interference current generated by the common-mode interference source based on the grid-side impedance. Compensation current with AEF output phase difference between θ Δ And calculate the current on the grid side. Compensation current with AEF output phase difference between θ c ; The initial active parameters are adjusted to reduce the phase difference in the frequency band where the interference amplitude exceeds the limit. θ Δ The absolute value of the difference from 180° Δθ Δ Or reduce the phase difference θ c The absolute value of the difference from 90° Δθ c The common-mode EMI spectrum of the AEF after inserting a common-mode interference source is measured based on the adjusted active parameters. When the common-mode EMI spectrum meets the EMC standard within the preset margin in the target frequency band, the corresponding active parameters are the target design parameters of the AEF. The AEF (Automatic Enhancement Function) includes a sampling stage, a signal amplification stage, and a signal compensation stage; the initial active parameters include: the turns ratio of the primary and secondary sides of the current transformer (CT) used in the sampling stage. n The feedback resistor of the inverting amplifier used in the signal amplification stage R f and input resistance R The ratio of 1 R f / R 1. Compensation resistor in the signal compensation stage R comp With compensation capacitor C comp ; The initial active parameters are adjusted to reduce the phase difference in the frequency band where the interference amplitude exceeds the limit. θ Δ The absolute value of the difference from 180° Δθ Δ Or reduce the phase difference θ c The absolute value of the difference from 90° Δθ c ,include: Calculate the low corner frequency of the CT transfer function in the frequency domain. f L-CT and high turning frequency f H-CT ; Frequency within the target frequency band f < f L-CT Sometimes, or f > f H-CT At the same time, reduce the primary and secondary turn ratio of CT. n In order to reduce Δθ Δ ; f L-CT ≤ f ≤ f H-CT At that time, make a judgment f and θ c Frequency of turning 90° f c-θc The relationship between them, if f < f c-θc Reduce the primary and secondary turn ratio of CT n In order to reduce Δθ c ;like f ≥ f c-θc This increases the primary-to-secondary turn ratio of the CT. n In order to reduce Δθ c ; Calculate the corner frequency of the closed-loop gain of the inverting amplifier. f c-op ; f ≥ f c-op At that time, reduce R f / R 1. To reduce Δθ Δ ; f < f c-op At that time, make a judgment f and f c-θc The relationship between them, if f < f c-θc ,reduce R f / R 1. To reduce Δθ c ;like f ≥ f c-θc Then increase R f / R 1. To reduce Δθ c ; Calculate the compensation resistor R comp With the compensation capacitor C comp The corner frequency of the series equivalent impedance f c-comp ; f < f c-comp When, increase the compensation capacitor C comp In order to reduce Δθ Δ ; f ≥ f c-comp At that time, make a judgment f and f c-θc The relationship between them, if f < f c-θc Increase the compensation resistor R comp The value of , to reduce Δθ c ;like f ≥ f c-θc Then reduce the compensation resistor. R comp The value of , to reduce Δθ c .

2. The broadband active EMI filter design method based on phasor analysis according to claim 1, characterized in that, The low transition frequency f L-CT and high turning frequency f H-CT The calculation method is as follows: in, The equivalent inductance of a CT. R c The equivalent resistance is used to characterize the loss. For sensing resistance; C pp The parasitic capacitance of the primary edge of the CT; The transition frequency f c-op and R f / R The following conditions must be met between 1 and 2: in, G op This refers to the open-loop gain of the operational amplifier in the inverting amplifier. f 1 and f 2 represents the two open-loop corner frequencies of the operational amplifier; The transition frequency f c-comp The calculation method is as follows: .

3. The broadband active EMI filter design method based on phasor analysis according to claim 1 or 2, characterized in that, The phase difference θ Δ and the phase difference θ c The calculation method is as follows: in, A For Amplitude, The compensation current representing the output of AEF With interference current The quotient, its phase is θ Δ ; The transfer function of CT. This is the transfer function of the inverting amplifier; Indicates the parallel connection symbol; Z LISN The impedance on the power grid side. Z L The equivalent impedance of an external magnetic ring is an empirical value. Z out The output impedance of the push-pull amplifier used in the signal amplification stage; For the compensation resistor R comp With the compensation capacitor C comp The series equivalent impedance.

4. The broadband active EMI filter design method based on phasor analysis according to claim 3, characterized in that, CT transfer function and the transfer function of the inverting amplifier The calculation method is as follows: in, s For the Laplace operator, R c The equivalent resistance is used to characterize the loss. For sensing resistance; C pp The parasitic capacitance of the primary edge of the CT. L m For magnetizing inductance; Z o The output impedance of the operational amplifier in the inverting amplifier is... The input impedance is the push-pull amplifier used in the signal amplification stage.

5. A broadband active EMI filter obtained by the broadband active EMI filter design method based on phasor analysis as described in any one of claims 1-4.

6. A broadband active EMI filter design system based on phasor analysis, characterized in that, Includes computer-readable storage media and processors; The computer-readable storage medium is used to store executable instructions; The processor is used to read executable instructions stored in the computer-readable storage medium and execute the broadband active EMI filter design method based on phasor analysis as described in any one of claims 1-4.

7. A computer-readable storage medium having a computer program stored thereon, characterized in that, When the program is executed by the processor, it implements the broadband active EMI filter design method based on phasor analysis as described in any one of claims 1-4.

8. A computer program product, characterized in that, Includes a computer program that, when run on a computer, causes the computer to perform the broadband active EMI filter design method based on phasor analysis as described in any one of claims 1-4.

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