Motor control device

Abstract

Provided is a motor control device capable of performing stable motor driving by suppressing an estimation error of an interlinking magnetic flux in a magnetic flux observer system. The present invention is provided with: a current detection element, a PWM signal generation unit and a current detection unit which are capable of executing first, second, and third output modes so that the current detection unit can detect a two-phase current at a two-point timing within a carrier period, and also with: a magnetic flux estimation unit which estimates an interlinking magnetic flux of an armature winding; a signal switching output unit that estimates the rotating magnetic field angle and speed of the motor on the basis of the interlinked magnetic flux, and outputs a switching command such that a first output scheme is executed when the modulation factor of the motor applied voltage is less than a first threshold value, a second output scheme is executed when the modulation factor is equal to or greater than the first threshold value and less than a second threshold value, and a third output scheme is executed when the modulation factor is equal to or greater than the second threshold value; and an angle correction unit that generates an angle calculated on the basis of the speed estimated in the last control cycle when the motor current cannot be detected in one cycle of the electrical angle.

Description

Technical Field

[0001] An embodiment of the present invention relates to a control device that controls a motor via an inverter circuit by performing PWM control on a plurality of switching elements connected in a three-phase bridge. Background Art

[0002] When detecting the currents in each of the U, V, and W phases for motor control, some technologies use a single shunt resistor inserted into the DC section of the inverter circuit for current detection. To detect the currents in all three phases in this manner, it is necessary to generate a three-phase PWM (Pulse Width Modulation) signal so that the currents in two or more phases can be detected within one cycle of the PWM carrier.

[0003] To this end, Patent Document 1 (Japanese Patent No. 5178799) proposes a technique that, by shifting the phase of the PWM signal within one cycle, enables the detection of currents in two or more phases without increasing noise, even in a region where the modulation rate of the motor applied voltage is low. Meanwhile, a flux observer is proposed, for example, in Non-Patent Document 1, as a method for estimating motor speed and angle based on estimated magnetic flux. In this flux observer method, the α-axis component Ψα and the β-axis component Ψβ of the interlinked magnetic flux of the motor winding are estimated based on, for example, the two-phase currents Iα and Iβ, the two-phase voltages Vα and Vβ, and the motor winding resistance R obtained from the current sensor. Furthermore, the motor's rotating magnetic field angle, the rotor phase angle, and the generated torque T are estimated.

[0004] In addition, the applicant proposed the following technology in Japanese Patent Application No. 2022-130097: when a flux observer method that infers the phase angle and speed of the rotating magnetic field of the motor based on the inferred interlinkage flux is combined with a shunt output method, the inference error of the interlinkage flux that may occur in the low-speed area from startup is suppressed.

[0005] In the flux observer described in non-patent document 1 (Inoue et al., "Expanding the Transformation Domain of Direct Torque Control in PMSM: A Method for Estimating the Magnetic Flux," Proceedings of the 2013 National Conference of the Institute of Electrical Engineers of Japan, IEEE, March 1, 2013, 5-095), it is envisioned that a current sensor such as a CT and a three-shunt output method be used to detect the current used to estimate the magnetic flux. However, in household appliances, a single-shunt output method is often used to reduce inverter costs. If a single-shunt output method is used, if the modulation rate of the voltage applied to the motor is low at low speeds from startup, the current may not be detected, resulting in errors in the estimation of the magnetic flux. In particular, when using the α-axis and β-axis currents that change sinusoidally in the calculation of flux estimation, if the previous value is used when the current is not detected, there is a problem that the estimation error of the interlinkage flux becomes large.

[0006] Furthermore, the aforementioned application provides a motor control device that combines a flux observer method, which estimates the phase angle and speed of the motor's rotating magnetic field based on estimated interlinkage flux, with a single-shunt output method. By shifting the phase of the PWM signal within one cycle using a three-phase modulation method, the device is able to suppress interlinkage flux estimation errors that may occur in the low-speed range from startup, thereby achieving stable motor drive. However, shifting the phase of the PWM signal using the three-phase modulation method in the medium-speed range causes a sharp decrease in the current detection rate, resulting in a larger interlinkage flux estimation error. Summary of the Invention

[0007] Provided is a motor control device that, in a flux observer method, can suppress an estimation error of interlinkage flux throughout the entire motor driving range in a high-speed region from startup, thereby stably driving the motor.

[0008] A motor control device according to an embodiment controls switching of a plurality of switching elements connected in a three-phase bridge according to a PWM signal, thereby driving a motor via an inverter circuit that converts direct current into three-phase alternating current, and includes:

[0009] a current detection element connected to the DC side of the inverter circuit and generating a signal corresponding to the current value;

[0010] a PWM signal generating unit that determines a rotor position based on at least a phase current of the motor and generates a PWM signal so as to follow the rotor position; and

[0011] a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal;

[0012] The PWM signal generating unit can execute a first output mode, a second output mode, and a third output mode so that the current detecting unit can detect the current of the two phases at two timings within a carrier cycle of the PWM signal.

[0013] The first output method is a method of outputting a three-phase phase-shifted PWM signal to cause the current detection unit to detect the current at a fixed timing.

[0014] The second output method is a method of outputting a two-phase phase-shifted PWM signal to cause the current detection unit to detect the current at a fixed timing.

[0015] The third output method is a method of outputting a three-phase or two-phase symmetrical PWM signal to enable the current detection unit to detect the current at a fixed or variable timing.

[0016] The motor control device further comprises:

[0017] a magnetic flux estimation unit for estimating interlinkage magnetic flux of an armature winding of the motor based on a phase current and an output voltage command of the motor;

[0018] a signal switching output unit that estimates the rotating magnetic field angle and speed of the motor based on the interlinked magnetic flux and outputs a switching instruction so that the PWM signal generating unit performs the first output mode when the modulation rate of the voltage applied to the motor is less than a first threshold value, performs the second output mode when the modulation rate is greater than or equal to the first threshold value and less than a second threshold value, and performs the third output mode when the modulation rate is greater than or equal to the second threshold value; and

[0019] When the motor current cannot be detected within one cycle of the electrical angle, the angle correction unit uses the speed estimated in the previous control cycle and generates an angle calculated based on the speed estimated in the previous control cycle. BRIEF DESCRIPTION OF THE DRAWINGS

[0020] Figure 1 This is a functional block diagram showing the configuration of a motor control device according to the first embodiment.

[0021] Figure 2 This is a diagram showing a vector control block using a flux observer.

[0022] Figure 3 This is a functional block diagram showing the detailed structure of the position estimation control unit.

[0023] Figure 4 This is a functional block diagram showing the configuration of an integrator used in the magnetic flux estimation unit.

[0024] Figure 5This is a diagram showing two-phase PWM signals obtained by shifting their phases.

[0025] Figure 6 This is a flowchart showing the angle correction process in a shunt detection method.

[0026] Figure 7 This is a diagram showing examples of current detection rates according to each current detection method.

[0027] Figure 8 This is a flowchart showing a process of switching the current detection method according to the level of the modulation rate.

[0028] Figure 9 This is a diagram showing an example of switching the current detection method.

[0029] Figure 10 This is a functional block diagram showing the configuration of a motor control device according to the second embodiment.

[0030] Figure 11 This is a functional block diagram showing the configuration of an integrator used in a magnetic flux estimation unit according to the third embodiment.

[0031] Figure 12 This is a diagram showing the relationship between the actual angle and the estimated angle in the low speed range in the first embodiment.

[0032] Figure 13 This is a diagram showing the relationship between the actual angle and the estimated angle in the low speed range in the third embodiment.

[0033] Figure 14 The fourth embodiment is a flowchart showing a process when the motor is started.

[0034] Figure 15 It is a diagram showing the waveform of each signal.

[0035] Figure 16 It will Figure 15 A diagram showing an enlarged portion of a portion.

[0036] Figure 17 This figure shows the principle of correcting the phase θ of the rotating magnetic field in accordance with the fifth embodiment.

[0037] Figure 18 This is a functional block diagram showing the configuration of a motor control device according to the sixth embodiment.

[0038] Figure 19 This is a flowchart showing a process of switching the carrier frequency and current detection method of PWM control according to the level of the modulation rate. DETAILED DESCRIPTION

[0039] Therefore, a motor control device is provided that can suppress an estimation error of interlinkage flux in the entire motor driving range in a high-speed region from startup in a flux observer method, thereby stably driving the motor.

[0040] A motor control device according to an embodiment controls switching of a plurality of switching elements connected in a three-phase bridge according to a PWM signal, thereby driving a motor via an inverter circuit that converts direct current into three-phase alternating current, and includes:

[0041] a current detection element connected to the DC side of the inverter circuit and generating a signal corresponding to the current value;

[0042] a PWM signal generating unit that determines a rotor position based on at least a phase current of the motor and generates a PWM signal so as to follow the rotor position; and

[0043] a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal;

[0044] The PWM signal generating unit can execute a first output mode, a second output mode, and a third output mode so that the current detecting unit can detect the current of the two phases at two timings within a carrier cycle of the PWM signal.

[0045] The first output method is a method of outputting a three-phase phase-shifted PWM signal to cause the current detection unit to detect the current at a fixed timing.

[0046] The second output method is a method of outputting a two-phase phase-shifted PWM signal to cause the current detection unit to detect the current at a fixed timing.

[0047] The third output method is a method of outputting a three-phase or two-phase symmetrical PWM signal to enable the current detection unit to detect the current at a fixed or variable timing.

[0048] The motor control device further comprises:

[0049] a magnetic flux estimation unit for estimating interlinkage magnetic flux of an armature winding of the motor based on a phase current and an output voltage command of the motor;

[0050] a signal switching output unit that estimates the rotating magnetic field angle and speed of the motor based on the interlinked magnetic flux and outputs a switching instruction so that the PWM signal generating unit performs the first output mode when the modulation rate of the voltage applied to the motor is less than a first threshold value, performs the second output mode when the modulation rate is greater than or equal to the first threshold value and less than a second threshold value, and performs the third output mode when the modulation rate is greater than or equal to the second threshold value; and

[0051] When the motor current cannot be detected within one cycle of the electrical angle, the angle correction unit uses the speed estimated in the previous control cycle and generates an angle calculated based on the speed estimated in the previous control cycle.

[0052] A "symmetrical PWM signal" refers to a signal in which the direction of increase or decrease of the pulse width of the PWM signal is the same for each phase, based on any phase of the carrier cycle. A "phase-shifted PWM signal" refers to a signal in which the direction of increase or decrease of the pulse width of the PWM signal is different for each phase, based on any phase of the carrier cycle.

[0053] (First embodiment)

[0054] Figure 1 This is a functional block diagram showing the configuration of the motor control device of this embodiment, which is in the patent document 1. Figure 1 is a diagram with several functional blocks added. The DC power supply unit 1 is represented by the symbol of a DC power supply, and when generating DC power from a commercial AC power supply, it includes a rectifier circuit, a smoothing capacitor, and the like. An inverter circuit 3 is connected to the DC power supply unit 1 via a positive bus 2a and a negative bus 2b, and a shunt resistor 4 serving as a current detection element is inserted on the negative bus 2b side. The inverter circuit 3 is constructed by bridging three phases of, for example, N-channel power MOSFETs 5 (U+, V+, W+, U-, V-, W-), serving as switching elements. The output terminals of each phase are connected to the windings of each phase of, for example, a brushless DC motor.

[0055] The terminal voltage of the shunt resistor 4 is detected by the current detection unit 7. Based on this terminal voltage and the three-phase PWM signals output to the inverter circuit 3, the current detection unit 7 detects the currents Iu, Iv, and Iw of the U, V, and W phases, respectively. The currents detected by the current detection unit 7 are supplied to the duty cycle (DUTY) generation unit 8, which performs A / D conversion and reads the currents. The currents are then calculated based on, for example, the control conditions of the motor 6. The resulting duty cycles U_DUTY, V_DUTY, and W_DUTY used to generate the PWM signals for each phase are determined.

[0056] For example, when vector control is performed, if a rotational speed command ωref for motor 6 is given to DUTY generator 8 from a microcomputer or other computer that sets control conditions, a torque current command Iqref is generated based on the difference between this command and the estimated actual rotational speed of motor 6. Once motor 6's rotor position θ is determined based on each of motor 6's phase currents Iu, Iv, and Iw, a vector control operation using this rotor position θ is performed to calculate torque current Iq and excitation current Id. A PI control operation, for example, is performed on the difference between torque current command Iqref and torque current Iq to generate voltage command Vq. Similar processing is performed on the excitation current Id side to generate voltage command Vd. The voltage commands Vq and Vd are then converted into three-phase voltages Vu, Vv, and Vw using the rotor position θ. Duty ratios U, V, and W_DUTY for each phase are then determined based on these three-phase voltages Vu, Vv, and Vw.

[0057] The duty cycles U, V, and W_DUTY of each phase are assigned to the PWM signal generator 9 and compared with the carrier wave level to generate a three-phase PWM signal. Furthermore, a signal for the lower arm side is generated by inverting the three-phase PWM signal. After adding dead time as required, these signals are output to the drive circuit 10. In accordance with the assigned PWM signal, the drive circuit 10 outputs a gate signal to each gate of the six power MOSFETs 5 (U+, V+, W+, U-, V-, and W-) that comprise the inverter circuit 3. Furthermore, a potential boosted to the required level is output to the upper arm side.

[0058] The DC voltage detection unit 11 detects the voltage of the DC power supply 1 and outputs the detection result to the motor-applied voltage modulation rate calculation unit 12. The motor-applied voltage modulation rate calculation unit 12 calculates the modulation rate of the voltage applied to the motor 6 via the inverter circuit 3 based on the duty cycle information input from the duty generation unit 8. The calculated modulation rate is output to the PWM output mode selection unit 13. The PWM output mode selection unit 13, which serves as a signal switching output unit, outputs a switching signal that switches the output mode of the PWM signal from the PWM signal generation unit 9 according to the input modulation rate.

[0059] Figure 2 Represents a vector control block using a flux observer. Figure 2 In the AB / α / β conversion unit 21 of the DUTY generator 8, the currents of each phase detected by the current detection unit 7 are converted into the α-axis component and β-axis component of the motor current, Iα and Iβ. The currents Iα and Iβ obtained by this conversion are supplied to the position estimation control unit 23. The position estimation control unit 23 estimates the magnetic flux based on the currents Iα and Iβ and the voltage commands Vα and Vβ input from the dq / α / β conversion unit 26 described later.

[0060] The position estimation control unit 23 includes a magnetic flux estimation unit 23a and a speed position estimation unit 23b. The magnetic flux estimation unit 23a estimates the α-axis and β-axis components of the interlinkage magnetic flux according to the following equations (1) and (2). and Mutual inductance is used for L. Self-inductance, d-axis inductance Ld, or q-axis inductance Lq can also be used instead.

[0061]

[0062] As the voltages of the α-axis and β-axis components used in the calculations of the equations (1) and (2), the previously calculated values may be used.

[0063] The speed and position estimation unit 23b first estimates the magnetic flux and The phase θ and torque T of the rotating magnetic field with respect to the α-axis are estimated according to the following equations (3) and (4), respectively.

[0064]

[0065] The integrators on the right side of each of equations (1) and (2) use an incomplete integration method based on an LPF (Low Pass Filter) with a cutoff angular frequency ωc as shown in the transfer function of equation (5), as shown in Figure 4 The figure shows the integration of the magnetic flux. “s” is the differentiation operator.

[0066] G(S)=1 / (s+ωc)……(5)

[0067] When the frequency of the magnetic flux is sufficiently greater than the cutoff angular frequency ωc, a good estimation result can be obtained. ω is estimated by differentiating θ estimated using equation (3). As an LPF, in addition to conventional LPFs, IIR (Infinite Impulse Response) filters or FIR (Finite Impulse Response) filters can also be used.

[0068] Figure 3 1 is a functional block diagram showing the internal structure of the position estimation control unit 23 in more detail corresponding to the above-mentioned calculation. Figure 3 The integrator 29 of the magnetic flux estimation unit 23a shown is actually as follows Figure 4As shown, it is composed of a combination of an integrator 29a and a low-pass filter, namely LPF29b, and adopts a so-called incomplete integration method. The output signal of the integrator 29a contains a bias. The bias component is extracted by filtering the output signal by LPF29b, and the bias component is eliminated by subtracting the bias component by a subtractor at the subsequent stage. In addition, the LPF29b and the subtractor at the subsequent stage can also be composed of an HPF (High Pass Filter). For speed control, information on the rotor speed of the motor 6 is required. When a flux observer is used in the configuration of vector control, the fact that the rotating magnetic field speed of the motor 6 is stably consistent with the rotor speed is utilized.

[0069] Refer again Figure 2 The rotational speed command ωref for motor 6 is provided by a higher-level control device, such as a microcomputer, which sets control conditions. Speed control unit 24 generates torque current command Iqref based on the difference between the rotational speed command ωref and the rotational speed ω estimated by position estimation unit 23. αβ / dq conversion unit 22 calculates torque current Iq and excitation current Id from currents Iα and Iβ using vector control calculations using rotor position θ.

[0070] The current control unit 25 performs a PI control calculation, for example, on the difference between the torque current command Iqref and the torque current Iq, generating a voltage command Vq. The excitation current Id is similarly processed to generate a voltage command Vd. The space vector generation unit 27 uses the rotor position θ to convert the voltage commands Vq and Vd into three-phase voltages Vu, Vv, and Vw. Based on the three-phase voltages Vu, Vv, and Vw, the duty ratios U_DUTY, V_DUTY, and W_DUTY used to generate the PWM signals for each phase are determined.

[0071] The duty ratios U, V, and W_DUTY of each phase are given to the PWM forming unit 28, and a two-phase or three-phase PWM signal is generated by comparing the level with the carrier. In addition, a signal on the lower arm side is generated by inverting the two-phase or three-phase PWM signal, and after adding a dead time as needed, it is output to the drive circuit 10. Regarding the method in which the PWM forming unit 28 generates a three-phase PWM signal obtained by shifting the phase, for example, the method of the fourth embodiment disclosed in Patent Document 1 is used. In the method in which the PWM forming unit 28 generates a two-phase PWM signal obtained by shifting the phase, for example Figure 5 As shown, regarding the duty ratio during two-phase modulation, an inverse triangle wave is used as the comparison carrier of the U phase, an inverse sawtooth wave is used as the comparison carrier of the V phase, and a sawtooth wave is used as the comparison carrier of the W phase.

[0072] Figure 1The motor applied voltage modulation rate calculation unit 12 shown calculates the modulation rate of the motor applied voltage per carrier cycle as shown in Formula (6) based on Vα and Vβ calculated by the duty generation unit 8 .

[0073] (Modulation rate) = 100 × Vdc / (√3 × √(Vq2 + Vd2)) ... (6)

[0074] This calculation result is output to the PWM output mode selection unit 13. Based on this information, the PWM output mode selection unit 13 outputs a signal for switching the PWM output signal to the PWM signal generation unit 9. Furthermore, the PWM signal generation unit 9 outputs a current detection timing signal to the current detection unit 7. Alternatively, the modulation rate of the motor applied voltage may be replaced by simply using the motor rotation speed or the like.

[0075] Currents Iα and Iβ vary sinusoidally over time. Therefore, if the phase currents of motor 6 cannot be detected using the single-shunt output method, using the currents Iα and Iβ estimated during the previous control cycle to estimate magnetic flux may result in reduced estimation accuracy. Furthermore, since the angle varies in a sawtooth pattern, using the angle estimated during the previous control cycle when the phase currents of motor 6 cannot be detected can result in errors in vector control system calculations. Furthermore, in this embodiment, the control period is equal to the carrier period.

[0076] Figure 6 This flowchart shows the angle correction process in a single-shunt output method using a flux observer. The currents and voltages of the α-axis and β-axis are calculated (S1), and flux observer control is performed (S2). If the phase current can be detected (S3: Success), normal control is performed, and the estimated angle θ, load torque T, and speed ω are calculated in sequence (S4-S6). On the other hand, if the phase current cannot be detected (S3: Failure), the previously estimated speed ω is used (S7), and the angle θ calculated by integrating this speed ω is used (S8). This process is performed in the flux estimation unit 23a, which also serves as the angle correction unit.

[0077] Figure 7An example of the current detection rate in each current detection method is shown. The greater the speed and load torque of the motor, the closer the modulation rate of the voltage applied to the motor is to 100%. The PWM output method selection unit 13 switches the output method of the PWM signal generation unit 9 outputting the PWM signal according to the height of the modulation rate, and switches the detection method of the current detection unit 7 detecting the phase current. When the modulation rate is in a relatively low area, the PWM signal generation unit 9 generates a 3-phase PWM signal obtained by offsetting the output phase of the PWM signal pulse of each phase in a manner different from that of the conventional patent document 1. This is used as the first output method. In addition, when the modulation rate is in the middle area of 50% to 70%, as shown in FIG. Figure 5 As shown in FIG, a PWM signal is generated by shifting the output phase of the 2-phase PWM signal pulse. This is used as the second output method. In addition, when the modulation rate is in a high region, such as the one in Patent Document 1, Figure 7 As shown in FIG. 1 , a switching command is output so as to generate a PWM signal with two or three phases, which is a pulse signal symmetrical with respect to the midpoint of the PWM cycle. This is referred to as the third output mode.

[0078] Hereinafter, the method in which the current detection unit 7 detects the two-phase current at a fixed timing, corresponding to the first output method described above, is referred to as the first detection method. Furthermore, the method in which the current detection unit 7 detects the two-phase current at a fixed timing, corresponding to the second output method, is referred to as the second detection method. Furthermore, the method in which the current detection unit 7 detects the two-phase current at a fixed or variable timing, corresponding to the third output method, is referred to as the third detection method. When driving the motor 6 from startup to the high-speed range, the PWM output method selection unit 13 switches between the first detection method, the second detection method, and the third detection method in this order, depending on the modulation rate.

[0079] Conversely, when the speed of the motor 6 is shifted from the high speed range to the low speed range, the third detection method, the second detection method, and the first detection method are switched in this order. In this case, by adding hysteresis to the threshold value for switching the current detection method, it is possible to suppress frequent switching of the current detection method even when the speed of the motor 6 increases or decreases.

[0080] Figure 8This flowchart shows the current detection method switching process. After the motor 6 starts, the process branches to Case 1 (S9), selecting either the first or second detection method. The current detection method switching threshold is compared with the modulation factor (S10). If the modulation factor is less than 50%, for example, the first detection method is selected (S11), and the process branches to Case 1 again (S12). If the modulation factor is 50% or higher, the second detection method is selected (S13), transitioning to Case 2 (S14), and the process branches to Case 2, selecting the first to third detection methods. A modulation factor of 50% corresponds to the first threshold.

[0081] For example, if the modulation rate is less than 45% (S15), the first detection mode is selected (S16), and the process returns to Case 1 (S17). If the modulation rate is 45% or higher (S18), the current detection mode switching threshold is again compared with the modulation rate. For example, if the modulation rate is less than 60% (S18), the second detection mode is continued (S19). If the modulation rate is 60% or higher (S18), the third detection mode is selected (S21), and the process transitions to Case 3 (S22). A modulation rate of 60% corresponds to the second threshold. After transitioning to Case 3, if the modulation rate is less than 55%, for example (S23), the second detection mode is selected (S24), and the process transitions to Case 2 (S25). If the modulation rate is 55% or higher (S23), the third detection mode is selected (S26), and the process transitions to Case 3 again (S27). A modulation rate of 55% corresponds to the third threshold. The threshold value can also be appropriately adjusted to take into account current ripple, A / D conversion time, and other factors.

[0082] Figure 9 This example shows how the motor 6 is driven by switching the single-shunt detection method using a flux observer to the third detection method, the second detection method, and the first detection method. When the drive frequency is increased after starting the motor 6, the detection methods are switched in the order of the first, second, and third detection methods. When the drive frequency is lowered after switching to the third detection method, the detection methods are switched in the order of the third, second, and first detection methods.

[0083] As described above, according to this embodiment, the PWM signal generator 9 determines the rotor position based on at least the phase current of the motor 6 and generates a PWM signal to track the rotor position. The current detector 7 detects the phase current of the motor 6 based on the signal generated by the shunt resistor 4 and the PWM signal. The PWM signal generator 9 outputs the PWM signal using the first to third output modes, enabling the current detector 7 to detect the current of the two phases at two points within the carrier cycle according to the first to third detection modes. In the first output mode, the duty cycle of one of the three phases is increased or decreased in both directions, either in the delayed or advanced direction, based on an arbitrary phase of the carrier cycle. The duty cycle of another phase is increased or decreased in one direction, either in the delayed or advanced direction, and the duty cycle of the remaining phase is increased or decreased in the opposite direction.

[0084] The magnetic flux estimation unit 23a estimates the interlinked magnetic flux of the armature winding of the motor 6 based on the phase current and output voltage command of the motor 6, and estimates the rotating magnetic field angle and speed of the motor 6 based on this interlinked magnetic flux. The PWM output mode selection unit 13 outputs a switching command to cause the PWM signal generation unit 9 to execute the first output mode when the modulation rate of the motor applied voltage is in the low range, the second output mode when it is in the intermediate range, and the third output mode when it is in the high range. In conjunction with this, the current detection unit 7 switches to the first through third detection modes. If the magnetic flux estimation unit 23a cannot detect the motor current, it uses the previously estimated speed and generates an angle calculated based on the previously estimated speed.

[0085] Here, the condition under which motor current cannot be detected is when the duration of the PWM signal used for current detection within one electrical angle cycle is shorter than the current detection time, for example, 5 to 10 μsec, which is sufficient to account for current ripple and A / D conversion time. Therefore, even if a PWM signal is generated to detect two-phase currents at two points, this does not necessarily mean a 100% detection rate. The actual detection rate is generally within the range of 70% to 100%. Furthermore, the detection rate decreases sharply when the modulation factor exceeds approximately 50% to 60%, which can lead to unstable motor drive depending on the application.

[0086] Therefore, by employing the second detection method in the modulation rate range exceeding approximately 50% to 60%, the current detection rate can be improved compared to the first detection method. With this configuration, even when the single-shunt detection method is employed, the three-phase currents Iu, Iv, and Iw can be detected at a high current detection rate from a low to a high modulation rate of the motor applied voltage, enabling magnetic flux estimation based on the α-axis and β-axis currents and the voltage command vector. Furthermore, even when the phase currents of motor 6 cannot be detected, using the previously estimated speed value prevents degradation in position estimation accuracy.

[0087] (Second embodiment)

[0088] Hereinafter, the same reference numerals will be assigned to the same parts as in the first embodiment, and descriptions thereof will be omitted. The following describes the different parts. In the first embodiment, the angle θ and speed ω of the motor 6 are estimated based on the estimated magnetic flux, and vector control is applied. In the second embodiment, flux observer control is applied to direct torque control.

[0089] like Figure 10 As shown, in direct torque control using a flux observer, a UVW / αβ conversion unit 31, a torque calculation unit 32 as a direct torque control execution unit, a binary level output unit 33, and a switching table 34 are used instead of the αβ / dq conversion unit 22, the speed estimation unit 24 to the space vector formation unit 27. The target torque command Tref and the target flux command Tref are input from the upper control device. ωref is substituted for the rotation speed command. Then, referring to the switching table 34, a three-phase PWM signal is generated. Direct torque control is a known technique, so a detailed description is omitted. Furthermore, as a motor, in addition to a permanent magnet motor, a synchronous reluctance motor or an induction motor can also be applied.

[0090] (Third embodiment)

[0091] In the first embodiment, the integrators on the right side of equations (1) and (2) are respectively used with an incomplete integration method. In the second embodiment, for the integration of the magnetic flux in the same situation, the following is used: Figure 11 And the second-order generalized integral form of the transfer function shown in equation (7).

[0092] G(S)=kω' / (s2+Kω'S+ω'2)......(7)

[0093] ω' is the natural angular frequency in the second-order filter, and k is the coefficient that determines the attenuation.

[0094] When estimating the magnetic flux based on the incomplete integration method, the magnetic flux can be estimated well when the magnetic flux frequency is sufficiently large compared to ωc. However, due to the use of LPF29b, Figure 12 The accuracy is reduced in the low speed area indicated by the double-headed arrows. Figure 13 In the second-order generalized integral method shown above, the frequency characteristics in the low-speed range are improved, thereby expanding the operable range.

[0095] As described above, according to the third embodiment, when the modulation rate of the motor applied voltage is in a relatively low region, the output phase of the PWM signal pulses of each phase is offset in the manner of Patent Document 1 to generate a three-phase PWM signal and perform sensorless operation. By integrating the magnetic flux using a second-order generalized integration method, the estimation accuracy of the motor magnetic flux can be maintained even in a lower speed region, thereby enabling sensorless control.

[0096] (Fourth embodiment)

[0097] According to the third embodiment, the accuracy of estimating the motor flux can be maintained even at low speeds. However, when the motor 6 is started, the motor flux cannot be estimated with sufficient accuracy. Therefore, it is considered to perform forced commutation at startup by applying a d-axis current at an angle corresponding to the command speed, and after increasing the speed of the motor 6, switch to sensorless operation. However, if the load torque of the motor 6 is too large, imbalance may occur. If a sufficient d-axis current is applied during forced commutation, the load at startup can be coped with, but the power at startup becomes larger.

[0098] Therefore, in the fourth embodiment, when the motor 6 is started, forced commutation is implemented, applying a d-axis current in accordance with the estimated angle of the motor magnetic flux. Furthermore, a torque current command Iqref is also applied based on the difference between the commanded rotational speed and the estimated rotational speed of the motor 6. As a result, during forced commutation at startup, a d-axis current is applied when the motor load is light, and a q-axis current is applied when the motor load increases. Thus, even during forced commutation at startup, the motor current is varied in accordance with the load, enabling the motor 6 to be started without wasting power.

[0099] exist Figure 14In the control sequence for startup shown, positioning control is first performed (S31). Here, the excitation current command Idref is set to a predetermined value, the torque current command Iqref is set to zero, and the angle is set to the target angle. Next, "forced commutation control 1" is performed (S32). The current commands Idref and Iqref, as well as the angle, are set to the same as in step S31. The speed command ωref is then increased. If the speed command ωref exceeds speed threshold 1 in step S33 (S33: Yes), the process transitions to "forced commutation control 2" (S34).

[0100] In "Forced Commutation Control 2," the excitation current command Idref is set to, for example, approximately half the value specified in "Forced Commutation Control 1." The torque current command Iqref uses the result of speed control during vector control, and the angle is set to a value estimated by the flux observer. If the speed command ωref exceeds speed threshold 2 (Yes) in step S35, the system transitions to sensorless control (S36). Here, the excitation current command Idref is set to zero. The same determination as in step S35 is then made (S37). If the speed command ωref exceeds speed threshold 2, the startup process ends; if it is below speed threshold 2, the system returns to step S34.

[0101] Figure 15 This is the motor operating waveform when forced commutation is performed according to the estimated motor magnetic flux angle. Figure 16 Will Figure 15 A portion of the load application interval, i.e., the portion enclosed by a rectangle, is shown enlarged. As the load increases during forced commutation, the torque current component Iq also increases. Therefore, it can be seen that even during forced commutation, the motor output torque can be varied in accordance with the load torque.

[0102] As described above, according to the fourth embodiment, when starting the motor 6, forced commutation is performed to apply a d-axis current in accordance with the estimated motor magnetic flux angle, and a torque current command Iqref is also applied based on the difference between the estimated rotational speed of the motor 6 and the command rotational speed. This allows the output torque of the motor 6 to be varied in accordance with the load torque even during forced commutation. Consequently, the motor 6 can be started without wasting power.

[0103] (Fifth embodiment)

[0104] In the first and second embodiments, the phase θ of the rotating magnetic field is estimated using equation (3). As the motor's load torque increases, the error between the phase θ of the rotating magnetic field and the phase of the magnet increases. Therefore, the phase θ of the rotating magnetic field is corrected using equation (9) using the torque angle δ calculated using equation (8) based on the q-axis inductance Lq, d-axis inductance Ld, q-axis current Iq, d-axis current Id, and magnet flux Φ. Figure 17 This is a diagram showing the principle of correction.

[0105] δ=tan -1 {(Lq×Iq) / (Ld×Id+Φ)}......(8)

[0106] θ=θs-δ……(9)

[0107] As described above, according to the fifth embodiment, when the motor 6 is started, the position estimation error between the phase θ of the rotating magnetic field and the phase of the magnet, which is generated when the load torque of the motor 6 increases, can be reduced. As a result, the motor 6 can be stably driven even when the load torque of the motor 6 increases.

[0108] (Sixth embodiment)

[0109] When the output phase of the PWM signal pulses of each phase is shifted as in the above-mentioned embodiment, if the carrier frequency is within the human audible range, such as 4 kHz, noise may become a problem. On the other hand, if the carrier frequency is increased for the entire range of motor drive, there is a concern that the switching loss of the inverter circuit 3 will increase, resulting in a decrease in overall efficiency. Therefore, in the sixth embodiment, the PWM output mode selection unit 13 switches the current detection unit 7 between the first and second current detection modes and the third current detection mode in accordance with the height of the modulation rate. Figure 18 The PWM frequency changing unit 41 shown changes the carrier frequency.

[0110] like Figure 19 As shown, if the modulation rate is less than 50% in step S10, the carrier frequency is set to, for example, 8k to 16kHz or more (S27), and the first detection method (S11) is executed. In addition, if the modulation rate is less than 55% in step S23, the carrier frequency is also set to, for example, 8k to 16kHz or more (S30), and the second detection method (S24) is executed.

[0111] On the other hand, if the modulation factor is 60% or higher in step S18, the carrier frequency is set low, such as 4 kHz (S29), and the third detection method is executed (S21). With this detection method, the noise caused by the carrier frequency is smaller than when the output phase is shifted. Furthermore, either the current detection method or the carrier frequency change can be performed first. Furthermore, the carrier frequency change can be performed in stages or all at once. Furthermore, the modulation factor of the motor applied voltage can be simply replaced by the motor rotation speed, for example.

[0112] (Other embodiments)

[0113] As a method of widening the pulse width of each phase PWM signal, the first to third embodiments of Patent Document 1 may also be applied.

[0114] Magnetic Flux and The inference of can also be calculated as in formula (10) and formula (11).

[0115]

[0116] The control period does not necessarily have to be consistent with the carrier period, and may be set to be twice or more of the carrier period, or 1 / 2 of the carrier period.

[0117] While several embodiments of the present invention have been described above, these embodiments are provided as examples and are not intended to limit the scope of the invention. These novel embodiments may be implemented in various other forms and may be omitted, replaced, or modified without departing from the spirit of the invention. These embodiments and their variations are intended to be included within the scope and spirit of the invention and within the scope of the invention set forth in the claims and their equivalents.

Claims

1. A motor control device that controls the switching of a plurality of switching elements connected in a three-phase bridge according to a PWM signal, thereby driving a motor via an inverter circuit that converts direct current into three-phase alternating current, wherein: have: a current detection element connected to the DC side of the inverter circuit and generating a signal corresponding to the current value; a PWM signal generating unit that determines a rotor position based on at least a phase current of the motor and generates a PWM signal so as to follow the rotor position; as well as a current detection unit that detects a phase current of the motor based on a signal generated in the current detection element and the PWM signal; The PWM signal generating unit can execute a first output mode, a second output mode, and a third output mode so that the current detecting unit can detect the current of the two phases at two timings within a carrier cycle of the PWM signal. The first output method is a method of outputting a three-phase phase-shifted PWM signal to cause the current detection unit to detect the current at a fixed timing. The second output method is a method of outputting a two-phase phase-shifted PWM signal to cause the current detection unit to detect the current at a fixed timing. The third output method is a method of outputting a three-phase or two-phase symmetrical PWM signal to enable the current detection unit to detect the current at a fixed or variable timing. The motor control device further comprises: a magnetic flux estimation unit for estimating interlinkage magnetic flux of an armature winding of the motor based on a phase current and an output voltage command of the motor; a signal switching output unit that estimates the rotating magnetic field angle and speed of the motor based on the interlinked magnetic flux and outputs a switching instruction so that the PWM signal generating unit performs the first output mode when the modulation rate of the voltage applied to the motor is less than a first threshold value, performs the second output mode when the modulation rate is greater than or equal to the first threshold value and less than a second threshold value, and performs the third output mode when the modulation rate is greater than or equal to the second threshold value; as well as The angle corrector uses the speed estimated in the previous control cycle when the current detector cannot detect the phase current in one cycle of the electrical angle, and generates an angle calculated based on the speed estimated in the previous control cycle.

2. The motor control device according to claim 1, wherein: After outputting the switching instruction to execute the third output mode, the signal switching output unit outputs the switching instruction to execute the second output mode if the modulation rate is lower than a third threshold value set to be greater than or equal to the first threshold value and less than the second threshold value.

3. The motor control device according to claim 1, wherein: In the first output mode, the PWM signal generating unit Based on an arbitrary phase of the carrier cycle, the pulse width is increased or decreased in both the delay side and the advance side to generate one phase of the three-phase PWM signal. Based on an arbitrary phase of the carrier cycle, the pulse width is increased or decreased in one direction of the delay side and the advance side to generate the other phase of the three-phase PWM signal. The remaining one phase of the three-phase PWM signal is generated by increasing or decreasing the pulse width in the direction opposite to the above direction with reference to an arbitrary phase of the carrier cycle.

4. The motor control device according to claim 1, wherein: The flux inference unit infers the linkage flux by performing time integration based on the motor current value obtained by converting the three-phase AC motor current into a two-phase AC motor current, the output voltage command obtained by converting the DC component into a two-phase AC component, and the winding resistance value of the motor.

5. The motor control device according to claim 4, wherein: The magnetic flux estimation unit estimates the interlinkage magnetic flux by integrating the output voltage command converted into the two phases, the two-phase AC current of the motor, and a value calculated based on the winding resistance value of the motor over time using a double integrator.

6. The motor control device according to any one of claims 1 to 5, wherein: The magnetic flux estimation unit uses the torque angle calculated based on the q-axis inductance Lq, the d-axis inductance Ld, the q-axis current Iq, the d-axis current Id, and the magnet magnetic flux Φ to correct the position estimation error caused by the increase or decrease in load torque at the motor angle calculated based on the estimated magnetic flux.

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