Synchronous machine control device

Abstract

The purpose of the present invention is to provide a synchronous machine control device that suppresses overcurrent or torque fluctuation when switching from a three-phase open or short circuit mode to a torque mode (PWM mode). In order to solve the problem, this synchronous machine control device controls a power converter that supplies power to a synchronous machine, and is provided with: a first magnetic flux command calculation unit that calculates a first magnetic flux command value on the basis of a current command value of the synchronous machine; a magnetic flux estimation unit that estimates a magnetic flux value of the synchronous machine on the basis of a current detection value of the synchronous machine; and a second magnetic flux command calculation unit that calculates a second magnetic flux command value such that the first magnetic flux command value and the magnetic flux value coincide using an integration controller, the d-axis side of the integration controller being initialized with a d-axis magnetic flux value and the q-axis side being initialized with a q-axis magnetic flux value when the power converter is switched from the three-phase short circuit to the PWM control.

Description

Technical Field

[0001] The present invention relates to a synchronous machine control device for driving a synchronous machine such as a synchronous motor. Background Art

[0002] In order to reduce the size of synchronous motors, the motors are being developed to have higher performance, such as higher rotation speeds and higher magnetic flux densities. This trend is particularly pronounced in electric vehicles, such as electric cars, where the weight of the motor affects power consumption.

[0003] Patent Document 1 discloses a control device that can achieve higher performance for electric motors. In the control device described in Patent Document 1, a first magnetic flux command value is calculated based on a current command value for the synchronous machine, and the magnetic flux value of the synchronous machine is estimated based on a current detection value of the synchronous machine. A voltage command value for the power converter is generated so that the first magnetic flux command value and the magnetic flux value match. Prior art literature Patent Literature

[0004] Patent Document 1: Japanese Patent Application Laid-Open No. 2021-151003 Summary of the Invention Problems to be solved by the invention

[0005] In the control device described in Patent Document 1, the PI controller that calculates the second d-axis magnetic flux command value calculates the difference between the first d-axis magnetic flux command value and the estimated d-axis magnetic flux value using an adder-subtractor. Furthermore, the difference is integrated by an integrator, and the integral value is multiplied by an integral gain (KI). The difference multiplied by a proportional gain 97 and the integral multiplied by the integral gain are added together by an adder 99 to calculate the second d-axis magnetic flux command value. A second q-axis magnetic flux command value is also calculated in a similar manner.

[0006] At this time, when the power conversion device is switched from the three-phase open-circuit or short-circuit mode to the torque mode (PWM mode), if the initial value of the integrator is incorrect, a difference will occur between the output voltage of the power conversion device and the voltage of the synchronous machine when the PWM mode is switched, which may cause overcurrent or torque fluctuations.

[0007] The present invention has been made in view of the above problems, and an object of the present invention is to provide a synchronous machine control device that suppresses overcurrent and torque fluctuation when switching from a three-phase open or short-circuit mode to a torque mode (PWM mode). Technical means to solve the problem

[0008] In order to solve the above-mentioned problems, the synchronous machine control device of the present invention is a synchronous machine control device that controls a power converter that supplies power to the synchronous machine, and comprises: a first flux command operation unit that calculates a first flux command value based on a current command value of the synchronous machine; a flux estimation unit that estimates the flux value of the synchronous machine based on a current detection value of the synchronous machine; and a second flux command operation unit that uses an integral controller to calculate the second flux command value so that the first flux command value and the flux value are consistent. When the power converter switches from three-phase short-circuit control to PWM control, the d-axis side of the integral controller is initialized with the d-axis flux value, and the q-axis side is initialized with the q-axis flux value. Effects of the Invention

[0009] According to the present invention, it is possible to suppress overcurrent and torque fluctuation when switching from a three-phase open-circuit or short-circuit mode to a torque mode (PWM mode).

[0010] Other problems, structures, and effects than those described above will become clear from the following description of the embodiments. BRIEF DESCRIPTION OF THE DRAWINGS

[0011] Figure 1 This is a block diagram showing the functional configuration of the synchronous machine control device according to the embodiment. Figure 2 3 is a block diagram showing the functional configuration of the PI controller in the second dq-axis magnetic flux command calculation unit 25 . Figure 3 The configuration of the voltage vector calculation unit 19 based on the inverse model represented by equation (1) is shown. Figure 4 This is an example showing the relationship between magnetic flux and current. Figure 5 This is a block diagram showing the configuration of an integrator according to the embodiment. Figure 6 This is a block diagram showing the configuration of an integrator according to a modified example of the embodiment. Figure 7 This is a diagram showing a current waveform when switching from a three-phase short-circuit mode to a PWM mode in conventional control. Figure 8 This is a diagram showing a current waveform when switching from a three-phase short-circuit mode to a PWM mode in the control according to the embodiment. Figure 9 This is a diagram showing a current waveform when switching from a three-phase open mode to a PWM mode in conventional control. Figure 10 This is a diagram showing a current waveform when switching from the three-phase open mode to the PWM mode in the control according to the embodiment. DETAILED DESCRIPTION

[0012] Hereinafter, embodiments of the present invention will be described using the accompanying drawings. In each of the drawings, the same reference numerals represent the same components or components having similar functions.

[0013] In the following description, a synchronous machine to be controlled is a permanent magnet synchronous motor (hereinafter referred to as “PMSM” (abbreviation for Permanent Magnet Synchronous Motor)).

[0014] [Overall composition] Figure 1 This is a block diagram showing the functional configuration of the synchronous machine control device of this embodiment. In addition, in this embodiment, a computer system such as a microcomputer executes a predetermined program as Figure 1 The synchronous machine control device shown functions.

[0015] exist Figure 1 In the power converter 2, DC power from a DC voltage source 9 (e.g., a battery) is converted into AC power and output to the PMSM 1. This AC power drives the PMSM 1. Power converter 2 includes an inverter main circuit composed of semiconductor switching elements. Gate signals control the on / off operation of the semiconductor switching elements, converting DC power into AC power. For example, IGBTs (Insulated Gate Bipolar Transistors) are used as semiconductor switching elements.

[0016] The phase current detector 3 detects the three-phase motor current flowing from the power converter 2 to the PMSM 1, that is, the U-phase current I u , V phase current I v and W phase current I w , respectively as the U phase current detection value I uc , V phase current detection value I vc And W phase current detection value I wc In addition, a Hall CT (Current Transformer) or the like is used as the phase current detector 3 .

[0017] The magnetic pole position detector 4 detects the magnetic pole position of the PMSM 1 and outputs magnetic pole position information θ * As the magnetic pole position detector 4, a resolver or the like is applied.

[0018] The frequency calculation unit 5 calculates the magnetic pole position information θ outputted from the magnetic pole position detector 4. * , through time differential operation and other operations and output speed information ω1 * .

[0019] The coordinate conversion unit 7 uses the magnetic pole position information θ* , the phase current detector output I uc , I vc , I wc Converted to the dq axis current detection value I in the rotating coordinate system dc , I qc , output I dc , I qc .

[0020] The dq axis magnetic flux estimation unit 23 calculates the dq axis current detection value I from the coordinate conversion unit 7. dc , I qc , refer to the lookup table (table data) and estimate the dq axis magnetic flux estimated value φ dc 、φ qc The lookup table (table data) referred to by the dq-axis magnetic flux estimation unit 23 is a table indicating I dc , I qc and φ dc 、φ qc The corresponding table data is stored in a storage device (not shown) provided in the synchronous machine control device of this embodiment. In addition, a predetermined function (approximate expression, etc.) may be used instead of the lookup table.

[0021] The first dq axis magnetic flux command calculation unit 21 generates a dq axis magnetic flux command value I according to the dq axis current command value I supplied from the host control device or the like. dc * , I qc * , refer to the lookup table (table data), calculate and output the first dq axis flux command value φ d * 、φ q * The lookup table (table data) referred to by the first dq axis magnetic flux command operation unit 21 is a table representing I dc * , I qc * and φ d * 、φ q * The corresponding table data is stored in a storage device (not shown) provided in the synchronous machine control device of this embodiment. In addition, a predetermined function (approximate expression, etc.) may be used instead of the lookup table.

[0022] The second dq axis magnetic flux command calculation unit 25 calculates and outputs the second dq axis magnetic flux command value φ by a proportional integral (PI) controller. d ** 、φ q ** , so that the first dq-axis flux command value φ d * 、φq * and the estimated dq-axis magnetic flux value φ dc 、φ qc consistent.

[0023] [PI controller] Figure 2 3 is a block diagram showing the functional configuration of the PI controller in the second dq-axis magnetic flux command calculation unit 25 .

[0024] like Figure 2 As shown in the figure above, when calculating the second d-axis magnetic flux command value φ d ** In the PI controller, the first d-axis magnetic flux command value φ is calculated by the adder-subtractor 81. d * and the estimated d-axis magnetic flux value φ dc The difference (φ d * -φ dc ), multiply the differential operation value by the proportional gain 87 (K P ). In addition, the difference operation value is integrated by the integrator 83, and the integrated value is multiplied by the integral gain 85 (K I The adder 89 adds the difference calculated by multiplying the proportional gain 87 and the integral value multiplied by the integral gain 85 to calculate the second d-axis magnetic flux command value φ. d ** .

[0025] like Figure 2 As shown in the figure below, when calculating the second q-axis magnetic flux command value φ q ** In the PI controller, the first q-axis magnetic flux command value φ is calculated by the adder-subtractor 91. q * and the estimated q-axis magnetic flux value φ qc The difference (φ q * -φ qc ), multiply the differential operation value by the proportional gain 97 (K P ). In addition, the difference operation value is integrated by the integrator 93, and the integrated value is multiplied by the integral gain 95 (K I The difference calculated by the proportional gain 97 and the integral value multiplied by the integral gain 95 are added by the adder 99 to calculate the second d-axis magnetic flux command value φ. q ** .

[0026] [Voltage vector calculation] Figure 1 The voltage vector calculation unit 19 shown generates a voltage command value using an inverse model of the motor model.

[0027] In the inverse model of the motor model, the d-axis magnetic flux and q-axis magnetic flux of the motor are set to φ d and φ q , the d-axis voltage and q-axis voltage of the motor are set to V d and V q , the motor speed is set to ω1, and then it is expressed by the voltage equation as shown in equation (1).

[0028] [Formula 1] In this embodiment, the inverse model represented by equation (1) is applied, and V d and V d Let d-axis voltage command value V d * and q-axis voltage command value V q * , φ d and φ q The second d-axis magnetic flux command value φ is respectively d ** and the second q-axis magnetic flux command value φ q ** , set ω1 as the speed information ω1 * .

[0029] As will be described later, in equation (1), magnetic saturation of the motor is taken into consideration.

[0030] Figure 3 The configuration of the voltage vector calculation unit 19 based on the inverse model represented by equation (1) is shown. d 、L q , K e They are the winding resistance, d-axis inductance, q-axis inductance, and magnet flux in PMSM1.

[0031] like Figure 3 As shown, the differentiator 45 calculates φ d ** In addition, adder-subtractor 44 calculates φ d ** With K e The difference (φ d ** -K e ), multiply the difference calculation value by R / L d (46) The differential operation value performed by the differentiator 45 is multiplied by R / L as a gain. d The difference calculated value of (46) is added by adder 47. In addition, ω1 is multiplied by multiplier 48. * With φ q **Furthermore, the adder-subtractor 49 calculates the difference between the addition value of the adder 47 and the multiplication value of the multiplier 48, thereby generating V d * .

[0032] In addition, if Figure 3 As shown, the differentiator 35 calculates φ q ** In addition, φ q ** Multiply by R / L q (36) The differential operation value of the differentiator 35 is multiplied by R / L q (36) after φ q ** The sum is added by adder 37. In addition, ω1 is multiplied by multiplier 38. * With φ d ** Furthermore, adder 39 adds the added value of adder 37 and the multiplied value of multiplier 38 to generate V q * .

[0033] In this manner, the voltage vector calculation unit can be configured based on the voltage equation representing the inverse model of the motor model.

[0034] Figure 1 The coordinate conversion unit 11 shown in FIG. 1 uses the magnetic pole position information θ detected by the magnetic pole position detector 4 to convert the magnetic pole position information θ into a coordinate. * The dq-axis voltage command value V for the power converter 2 outputted from the voltage vector calculation unit 19 is d * 、V q * Perform coordinate conversion to generate and output the three-phase voltage command value V for the power converter 2. u * 、V v * 、V w * .

[0035] The DC voltage detector 6 detects the voltage of the DC voltage source 9 and outputs DC voltage information V dc .

[0036] The PWM controller 12 receives the three-phase voltage command value V from the voltage vector calculation unit 19. u * 、V v * 、V w * , and receives DC voltage information V from DC voltage detector 6 dcBased on these, the gate signal provided to the power converter 2 is generated and output by pulse width modulation. The PWM controller 12 generates the gate signal by pulse width modulation of the three-phase voltage command value as a modulation wave using, for example, a triangular wave as a carrier signal.

[0037] [Method for generating voltage command reception] Hereinafter, a description will be given of a means for generating a voltage command value in consideration of magnetic saturation of the PMSM 1 , which is used in the voltage vector calculation unit 19 of the present embodiment.

[0038] First, the current (dq axis current I d , I q ) as the state quantity, if magnetic saturation is considered, the voltage equation is as shown in formula (2).

[0039] [Formula 2] Here, L dh 、L qh 、L dqh 、L qdh Indicates dynamic inductance, L d 、L q 、L dq 、L qd Indicates the static inductance. Use Figure 4 These inductors are described below.

[0040] Figure 4 An example of the relationship between magnetic flux and current is shown. The vertical axis represents magnetic flux, and the horizontal axis represents current. The solid line in the relationship diagram between magnetic flux and current represents an example of the relationship between magnetic flux and current.

[0041] like Figure 4 As shown, due to the influence of magnetic saturation, the q-axis current (I q ) is larger, the q-axis magnetic flux (φ q ) increases more slowly. Therefore, as inductance, dynamic inductance and static inductance are defined as follows. Dynamic inductance L qh Is an action point (I q ,φ q ) is the slope (dφ) of the tangent line (dashed line in the figure) q / dt). In addition, the static inductance L q is the connected q-axis current (I q ) is 0 and the slope (φ) of the straight line (dashed line in the figure) of the action point q / I q ).

[0042] Although not shown in the figure, the relationship between the d-axis magnetic flux and the d-axis current, the dynamic inductance L dh , static inductance Ld and Figure 4 same.

[0043] In formula (2), the coefficient (matrix) in the current differential term (the second term on the right) is the dynamic inductance, and the coefficient (matrix) in the induced voltage term (the third term on the right) is the static inductance.

[0044] In addition, when magnetic saturation is significant, mutual interference occurs between the control axes, that is, between the d and q axes. This mutual interference is determined by the dynamic inductance L in the coefficient (matrix) of the current differential term. dqh 、L qdh , the static inductance L in the coefficient (matrix) of the induced voltage term dq 、L qd express.

[0045] Based on formula (2), that is, when the current is used as the state quantity and magnetic saturation is considered to control PMSM1, the 8 types of inductance (L dh 、L qh 、L dqh 、L qdh 、L d 、L q 、L dq 、L qd ). Therefore, in this case, the synchronous machine control device includes eight table data or functions (approximate expressions, etc.) indicating the correspondence between each inductance value and current value (d-axis current value and q-axis current value).

[0046] Furthermore, if the temperature dependence of these inductances is taken into consideration, the table data or mathematical expressions (approximate expressions, etc.) each become table data or functions of three variables: the d-axis current value, the q-axis current value, and the temperature.

[0047] In addition, since the magnet flux K in formula (2) e For q-axis current I q and temperature T, so the synchronous machine control device has a q and T are variables representing two variables and K e A table of data or function (approximate formula, etc.) of two variables that express the relationship between them.

[0048] In this manner, when the PMSM 1 is controlled by taking the magnetic saturation into consideration using the current as a state quantity, the synchronous machine control device includes a plurality of multivariable table data or multivariable functions.

[0049] Therefore, as described below, in this embodiment, by setting the magnetic flux as a state quantity as in the inverse model of the motor model represented by the above-mentioned formula (1), while taking magnetic saturation into consideration, the total number of table data or functions (approximate formulas, etc.) used in the synchronous machine control device is reduced (to 9 when the current is set as the state quantity as described above).

[0050] In the case of magnetic flux (dq axis magnetic flux φ d 、φ q ) as the state quantity, if magnetic saturation is considered, the voltage equation is as shown in formula (3).

[0051] [Formula 3] In many high-efficiency PMSMs used in automobiles, the winding resistance R is sufficiently small, so the influence of the first term of equation (3) on motor control is relatively small. Therefore, even if we approximate it as equation (1), and further reduce L d 、L q , K e Setting it to a constant value also has little effect on the motor control. Therefore, the voltage vector calculation unit 19 in this embodiment is based on the above equation (1) which takes the magnetic flux as the state quantity, and L in equation (1) is set to d 、L q , K e Set to a certain value, according to the dq axis magnetic flux command value (φ d ** 、φ q ** ) generates dq axis voltage command value (V d * 、V q * ).

[0052] In this case, the synchronous machine control device has a d-axis magnetic flux (φ d ) and q-axis magnetic flux (φ q ) respectively with the current (d-axis current I d , q-axis current I q ) of the corresponding relationship. Therefore, the synchronizer controller includes a total of two table data or functions.

[0053] By using magnetic flux as a state variable, the number of tables and functions used in motor control is reduced. This simplifies the control system while taking magnetic saturation into account, thereby reducing the computational load on the synchronous machine control device and shortening parameter identification time.

[0054] In addition, in this embodiment, the inverse model of the motor model is used to calculate the second dq-axis magnetic flux command value φ generated by the second dq-axis magnetic flux command calculation unit 25. d ** 、φ q ** , generate dq axis voltage command value V d * 、V q* Therefore, even in the high-speed region, the d-axis magnetic flux estimated value φ can be made dc and the estimated q-axis magnetic flux value φ qc and the second d-axis magnetic flux command value φ d ** and the second q-axis magnetic flux command value φ q ** Therefore, according to the synchronous machine control device of this embodiment, it is possible to control the high-speed rotation of the PMSM 1.

[0055] In addition, in this embodiment, the influence of the temperature dependence of the magnetic flux is mitigated by the PI controller or I controller provided in the second dq axis magnetic flux command calculation unit 25. Therefore, in the case of magnetic flux (φ d 、φ q ) may also be a table or function (e.g., an approximate expression) that does not include temperature as a variable but only uses current as a variable. This can reduce the computational load on the synchronous machine control device and shorten the parameter identification time.

[0056] In addition, by using the same table data or function in the first dq axis magnetic flux command calculation unit 21 and the dq axis magnetic flux estimation unit 23, the control I dc , I qc , so that it is respectively connected to I through the so-called magnetic flux d * , I q * In this case, a current control system is actually formed.

[0057] In addition, by using I in the input of the first dq axis magnetic flux command calculation unit 21 and the dq axis magnetic flux estimation unit 23 dc , I qc and I d * , I q * In this case, the first dq-axis magnetic flux command calculation unit 21 and the dq-axis magnetic flux estimation unit 23 use the dq-axis magnetic flux command value (φ d * 、φ q * ) and dq axis current command value (I d * , I q * ) and the table data or function showing the correspondence relationship between the dq axis magnetic flux estimation value (φ dc 、φ qc ) and dq axis current detection value (I dc , Iqc ) table data or function of the corresponding relationship.

[0058] Furthermore, since the synchronous machine control device of this embodiment substantially takes the dynamic inductance and static inductance of the motor into consideration, it is suitable for application to electric vehicles such as electric vehicles that use PMSMs that are greatly affected by magnetic saturation and require accurate torque response.

[0059] The above-mentioned lookup table, table data, and function (approximate expression), which are information indicating the correspondence relationship between magnetic flux and current in the PMSM 1 , can be set based on actual measurement, magnetic field analysis, or the like.

[0060] [Initialization of the integrator] Next, use Figures 5 to 8 Initialization of the integrator according to this embodiment will be described.

[0061] When switching power converter 2 from three-phase open-circuit mode or three-phase short-circuit mode to torque mode (PWM mode), it is necessary to set initial values ​​for integrators 83 and 93. If the initial values ​​are incorrect, a difference may occur between the output voltage of power converter 2 and the voltage of PMSM 1 when switching to PWM mode, potentially causing overcurrent or torque fluctuation.

[0062] Figure 7 This is a diagram showing the current waveform when switching from three-phase short circuit to PWM mode in conventional control. e The waveforms of the d-axis current, q-axis current, torque, and three-phase AC current when the d-axis integrator 83 and the q-axis integrator 93 are initialized to 0, and the gate mode switching signal are shown. Figure 7 The dotted line in represents the switching timing of the power converter 2 from the three-phase short-circuit mode to the PWM mode.

[0063] like Figure 7 As shown in the figure, the fluctuations in the d-axis and q-axis current values ​​increase after the switching time. This is due to a difference between the output voltage of power converter 2 and the voltage of the motor. Therefore, to prevent overcurrent, it is necessary to set appropriate initial values ​​for integrators 83 and 93 at the time of switching.

[0064] Figure 5 This is a block diagram of the integrator of this embodiment. Figure 5 As shown, the integrator 83 receives a gate mode switching signal when the driving state (gate mode) of the power converter 2 is switched by the upper controller (not shown). In this case, the d-axis magnetic flux estimated value φ is used for the integrator 83 on the d-axis side. dc Similarly, for the integrator 93 on the q-axis side, the q-axis magnetic flux estimated value φ is used. qcAlternatively, the estimated value φ of the d-axis magnetic flux can be used. dc and the estimated q-axis magnetic flux value φ qc The value after low-pass filtering is used as the initial value.

[0065] In the vector control using magnetic flux as the controlled variable as in this embodiment, in a steady state, the integral value on the d-axis side and the d-axis magnetic flux estimated value φ dc In addition, the integral value on the q-axis side is consistent with the q-axis magnetic flux estimated value φ qc Therefore, by using the d-axis magnetic flux estimated value φ dc The d-axis side integrator 83 is initialized and the q-axis magnetic flux estimated value φ is used. qc The integrator 93 on the q-axis side is initialized, and since the voltage of the PMSM 1 and the voltage of the power converter 2 coincide with each other, changes in the current during switching can be suppressed.

[0066] Specifically, when the power converter 2 is in a three-phase short-circuit state, the voltage becomes 0V, and a current that causes the voltage to become 0V flows through the PMSM 1. Therefore, when the power converter 2 switches from a three-phase short-circuit state to a PWM mode, it is sufficient to output a voltage that causes the current that flows during the three-phase short-circuit state to flow. That is, the integrator 83 on the d-axis side uses the d-axis magnetic flux estimated value φ dc Initialization is performed, and the integrator 93 on the q-axis side is initialized with the q-axis magnetic flux estimated value φ qc By performing initialization, the voltage of PMSM 1 and the voltage of power converter 2 become consistent with each other, and thus a change in current during switching can be suppressed.

[0067] Figure 8 This is a diagram showing the change in current when switching from three-phase short-circuit mode to PWM mode in the control of this embodiment. dc Initialize the d-axis side integrator 83 and use the q-axis magnetic flux estimated value φ qc The waveforms of the d-axis current, q-axis current, torque, and three-phase AC current when the integrator 93 on the q-axis side is initialized, as well as the switching signal of the gate mode. Figure 8 As shown, the control device of this embodiment suppresses changes in the d-axis current and the q-axis current after switching of the power converter 2. This can suppress the occurrence of overcurrent.

[0068] When the power converter 2 is in a three-phase open circuit, the current in the power converter 2 becomes 0 A (i d =i q =0A), only the induced voltage (V q =ωK e). Therefore, when the power converter 2 switches from the three-phase open circuit mode to the PWM mode, the power converter 2 can output a voltage that cancels the induced voltage of the PMSM 1. That is, the integrator 83 on the d-axis side uses the d-axis magnetic flux estimated value φ dc Initialization, the q-axis side integrator 93 uses the q-axis magnetic flux estimated value φ qc = 0 initialization. Thus, even when the three-phase short-circuit mode is switched to the PWM mode, the generation of overcurrent can be suppressed.

[0069] [Modification] Next, use Figure 6 Modifications of this embodiment will be described. Figure 6 This is a diagram showing an integrator according to a modified example of the present embodiment.

[0070] like Figure 6 As shown in the figure above, the initialization value of the d-axis side integrator 183 is configured to be different depending on the state before switching the drive mode of the power converter 2. As a result, the integrator 183 can be initialized with an appropriate value according to the state before switching the drive mode of the power converter 2.

[0071] For example, in Figure 6 In the configuration of , when the driving state of the power converter 2 before switching is three-phase open, the magnet flux K considering the rotor temperature is e Initialization is performed, and when the driving state of the power converter 2 before switching is a three-phase short circuit, the d-axis magnetic flux estimated value φ dc Initialization is performed. Since the d-axis magnetic flux estimated value φ dc Since the rotor temperature of PMSM1 changes, the magnet flux K is used with consideration of the rotor temperature. e , enabling more accurate control. The reason for considering rotor temperature only in three-phase open circuit mode is that in automobiles and other applications, three-phase open circuit mode is actively used during operation to reduce losses. Therefore, when switching between three-phase open circuit mode and PWM mode, it is desirable to minimize torque fluctuations to prevent deterioration in ride comfort. On the other hand, three-phase short circuit mode can be implemented under special conditions, such as when power converter 2 fails. Therefore, when switching between three-phase short circuit mode and PWM mode, it is sufficient to suppress overcurrent. This allows for efficient operation of PMSM 1.

[0072] Figure 9 This figure shows the current waveform when switching from three-phase open circuit to PWM mode in conventional control. e The d-axis current, q-axis current, torque, and time changes of the three-phase AC current when the d-axis integrator 83 and the q-axis integrator 93 are initialized to 0, as well as the gate mode switching signal. Figure 9 The dotted line in represents the switching timing of the power converter 2 from the three-phase open mode to the PWM mode.

[0073] like Figure 9 As shown in FIG, the fluctuation of the d-axis current value and the q-axis current value increases after the switching time. This is because the magnet flux of PMSM1 changes according to the rotor temperature, so the magnet flux K set as the initial value is different from the original value. e As a result, there is a possibility of torque fluctuations due to the difference between the output voltage of power converter 2 and the voltage of PMSM 1. Therefore, in order to prevent torque fluctuations, it is necessary to set appropriate initial values ​​for integrators 83 and 93 at the switching time.

[0074] Figure 10 This is a diagram showing the current waveform when switching from three-phase open circuit to PWM mode in the control of a modified example of this embodiment. e Initialize the d-axis side integrator 83 and use the q-axis magnetic flux estimated value φ qc (=0) The d-axis current, q-axis current, torque, and time variation of the three-phase AC current when the integrator 93 on the q-axis side is initialized, as well as the switching signal of the gate mode. Figure 10 As shown, the control device of this embodiment suppresses changes in the d-axis current and the q-axis current after switching of the power converter 2. This can suppress the occurrence of deterioration in ride comfort caused by a sudden torque fluctuation.

[0075] Alternatively, the dq-axis magnetic flux estimation unit 23 can estimate the d-axis and q-axis magnetic flux values ​​of the PMSM 1 based on a three-dimensional table that takes into account the rotor temperature of the PMSM 1. While this increases the motor adaptation effort, since the rotor temperature can be taken into account, it can more accurately prevent overcurrent and torque fluctuations caused by changes in the driving state of the power converter 2.

[0076] Furthermore, when the rotor temperature information cannot be obtained, the integrator may be initialized using the rotor temperature as a predetermined value. This prevents overcurrent and torque fluctuations caused by changes in the driving state of the power converter 2 even when the rotor temperature cannot be obtained.

[0077] The present embodiment described above is summarized as follows.

[0078] A synchronous machine control device for controlling a power converter that supplies power to a synchronous machine includes: a first flux command calculation unit that calculates a first flux command value based on a current command value of the synchronous machine; a flux estimation unit that estimates d-axis and q-axis magnetic flux values ​​of the synchronous machine based on a current detection value of the synchronous machine; a voltage calculation unit that generates a voltage command value for the power converter so that the magnetic flux value matches the first flux command value; and a second flux command calculation unit that uses an integrator to calculate second flux command values ​​for the d-axis side and the q-axis side, respectively, so that the first flux command value and the magnetic flux value match. The integrator is initialized to the d-axis magnetic flux value and the q-axis magnetic flux value when the drive state of the power converter switches. This prevents overcurrent and other issues caused by changes in the drive state of the power converter.

[0079] Furthermore, when the power converter's driving state switches, the integral controller can be configured to initialize different values ​​depending on the driving state before the switch. This allows the integrator to be initialized with appropriate values, thereby more accurately preventing torque fluctuations caused by changes in the power converter's driving state.

[0080] Furthermore, when the driving state before the switching is three-phase open, the d-axis side may be initialized with a value determined based on the temperature of the synchronous machine rotor. This can prevent degradation of ride comfort caused by abrupt torque fluctuations when the power conversion device switches from three-phase open to PWM mode.

[0081] Furthermore, it is also possible to configure that when the driving state before switching is three-phase short circuit, the d-axis side is initialized with the d-axis magnetic flux value, the q-axis side is initialized with the q-axis magnetic flux value, and when the driving state before switching is three-phase open circuit, the d-axis side is initialized with the magnet magnetic flux K taking into account the temperature of the rotor of the synchronous machine. e Initialization is performed so that the q-axis side is initialized with the q-axis magnetic flux value. This makes it possible to more accurately prevent overcurrent and the like caused by changes in the driving state of the power converter.

[0082] Furthermore, the magnetic flux estimation unit can be configured to estimate the d-axis and q-axis magnetic flux values ​​of the synchronous machine based on a three-dimensional table that takes into account the rotor temperature of the synchronous machine. This allows for more accurate prevention of overcurrent and torque fluctuations caused by changes in the power converter's driving state, since the rotor temperature can be taken into account.

[0083] Furthermore, if the rotor temperature information cannot be obtained, the integrator may be initialized using the rotor temperature as a predetermined value. This prevents overcurrent and torque fluctuations caused by changes in the power converter's driving state, even when the rotor temperature cannot be obtained.

[0084] Furthermore, the present invention is not limited to the above-described embodiments and includes various variations. For example, the above-described embodiments are described in detail to facilitate understanding of the present invention and are not necessarily limited to embodiments having all of the described configurations. Furthermore, portions of the configurations of each embodiment may be added, deleted, or replaced with other configurations.

[0085] For example, the synchronous machine to be controlled is not limited to the PMSM, and may be a synchronous reluctance motor, a permanent magnet synchronous generator, a winding-excitation synchronous motor, a winding-excitation synchronous generator, or the like.

[0086] In addition, the PMSM may be either a buried magnet type or a surface magnet type, and may be either an external-type or internal-type type.

[0087] Furthermore, the semiconductor switching element constituting the inverter main circuit is not limited to the IGBT, and may be a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) or the like.

[0088] Furthermore, the synchronous machine control device of each of the above-described embodiments can be used as a control device in various synchronous machine drive systems including a synchronous machine, a power converter for driving the synchronous machine, and a control device for controlling the power converter. Explanation of symbols

[0089] 1: PMSM, 2: Power converter, 3: Phase current detector, 4: Magnetic pole position detector, 5: Frequency calculation unit, 6: DC voltage detector, 7: Coordinate conversion unit, 10: Voltage vector addition unit, 11: Coordinate conversion unit, 15: dq-axis voltage command calculation unit, 19: Voltage vector calculation unit, 21: First dq-axis magnetic flux command calculation unit, 23: dq-axis magnetic flux estimation unit, 25: Second dq-axis magnetic flux command calculation unit, 35: Differentiator, 37: Adder, 38: Multiplier, 39: Adder, 44: Adder-subtractor, 45: Differentiator, 47: Adder, 48: Multiplier, 49: Adder-subtractor, 55: Multiplier, 57: Multiplier, 81: Adder-subtractor, 83: Integrator, 85: Integral gain, 87: Proportional gain, 89: Adder, 91: Adder-subtractor, 93: Integrator, 95: Integral gain, 97: Proportional gain, 98: Adder, 99: Adder, 183: Integrator, 193: Integrator.

Claims

1. A synchronous machine control device for controlling a power converter that supplies power to a synchronous machine, the synchronous machine control device comprising: a first magnetic flux command calculation unit for calculating a first magnetic flux command value based on a current command value of the synchronous machine; a magnetic flux estimating unit for estimating d-axis and q-axis magnetic flux values ​​of the synchronous machine based on a current detection value of the synchronous machine; a voltage calculation unit that generates a voltage command value for the power converter so that the first magnetic flux command value and the magnetic flux value coincide with each other; and a second flux command calculation unit for calculating second flux command values ​​on the d-axis side and the q-axis side using an integral controller so that the first flux command value and the flux value coincide with each other; When the driving state of the power converter is switched, the d-axis side of the integral controller is initialized with a d-axis magnetic flux value, and the q-axis side is initialized with a q-axis magnetic flux value.

2. The synchronous machine control device according to claim 1, wherein: When the driving state of the power converter is switched, the integral controller is initialized to a different value depending on the driving state before the switching.

3. The synchronous machine control device according to claim 2, wherein: When the driving state before the switching is three-phase open, the d-axis side is initialized with the magnet flux.

4. The synchronous machine control device according to claim 2, wherein: When the driving state before the switching is a three-phase short circuit, the d-axis side is initialized with the d-axis magnetic flux value, and the q-axis side is initialized with the q-axis magnetic flux value. When the driving state before the switching is three-phase open, the d-axis side is initialized with the magnet flux taking into account the rotor temperature of the synchronous machine, and the q-axis side is initialized with the q-axis flux value.

5. The synchronous machine control device according to claim 1, wherein: The magnetic flux estimating unit estimates d-axis and q-axis magnetic flux values ​​of the synchronous machine based on a three-dimensional table that takes into account a rotor temperature of the synchronous machine.

6. The synchronous machine control device according to claim 4 or 5, characterized in that: When the rotor temperature information cannot be acquired, the integral controller is initialized using the rotor temperature as a predetermined value.

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