Multi-mode control method and device for energy storage converter

Abstract

The invention relates to a multi-mode control method and device for an energy storage converter. The method comprises the following steps: aiming at the operation of an energy storage converter in a network following mode, establishing a second-order infinite impulse response IIR filter, and establishing a low-voltage ride-through control strategy based on conforming to repetitive control; for the operation of the energy storage converter in an off-grid mode, a three-phase four-bridge-arm topology is adopted to provide a zero-sequence path, and a preset proportional multi-resonance control strategy is established; establishing a parallel pre-synchronization control strategy aiming at the operation of the energy storage converter in a network construction mode; and realizing multi-mode control of the energy storage converter based on a low-voltage ride-through control strategy conforming to repetitive control, a preset proportional multi-resonance control strategy and a parallel pre-synchronization control strategy. Any resonant frequency and filtering radius can be set to suppress the harmonic peak, the system stability is ensured, meanwhile, compared with a traditional pre-synchronization method, pre-synchronization can be completed without additional control links by only utilizing the self frequency modulation characteristic, and control resources are greatly saved.

Description

Technical Field

[0001] The present disclosure relates to the field of energy storage converter control, and in particular to a multi-mode control method and device for an energy storage converter. Background Art

[0002] In the global transition to a low-carbon energy structure, power grid systems face major technical challenges brought about by the large-scale integration of renewable energy. These challenges include: the high penetration of intermittent distributed power sources such as wind and solar, leading to deteriorating power quality and low voltage ride-through during grid faults; distributed microgrids, as the primary means of absorbing new energy, face the problem of three-phase imbalance in load-measured AC voltage; and the increasing proportion of power electronic equipment, resulting in low inertia and weak damping characteristics in the grid, which impacts system stability. In this context, as a key device for the stable operation of renewable energy power systems, energy storage converters, with their multi-mode operation capabilities, play a key role in achieving bidirectional energy control, inertial support, and power quality management. Therefore, studying multi-mode control strategies for energy storage converters has both theoretical value and practical significance.

[0003] It should be noted that the information disclosed in the above background technology section is only used to enhance the understanding of the background of the present disclosure, and therefore may include information that does not constitute prior art known to ordinary technicians in the field. Summary of the Invention

[0004] The purpose of the present disclosure is to provide a multi-mode control method and device for an energy storage converter, thereby overcoming one or more problems caused by limitations and defects of related technologies, at least to a certain extent.

[0005] According to one aspect of the present disclosure, a multi-mode control method for an energy storage converter is provided, comprising:

[0006] For the energy storage converter operating in grid-following mode, a low voltage ride-through control strategy based on repetitive control is established by creating a second-order infinite impulse response (IIR) filter.

[0007] For the energy storage converter operating in off-grid mode, a three-phase four-leg topology is used to provide a zero-sequence path and establish a preset proportional multi-resonance control strategy.

[0008] A parallel pre-synchronization control strategy is established for the energy storage converter operating in grid-forming mode;

[0009] Multi-mode control of the energy storage converter is achieved based on the low voltage ride-through control strategy that complies with repetitive control in the grid-following mode, the preset proportional multi-resonance control strategy in the off-grid mode, and the parallel pre-synchronization control strategy in the grid-forming mode.

[0010] In one aspect of the present disclosure, a multi-mode control device for an energy storage converter is provided, comprising:

[0011] The grid-following strategy building module is used to establish a low voltage ride-through control strategy based on repetitive control by creating a second-order infinite impulse response (IIR) filter when the energy storage converter operates in grid-following mode.

[0012] An off-grid strategy building module is used to establish a preset proportional multi-resonance control strategy for the energy storage converter operating in off-grid mode, using a three-phase four-leg topology to provide a zero-sequence path;

[0013] A network strategy building module is used to establish a parallel pre-synchronization control strategy for the energy storage converter operating in the network mode;

[0014] The multi-mode control module is used to realize multi-mode control of the energy storage converter based on the low voltage ride-through control strategy that complies with repetitive control in the grid-following mode, the preset proportional multi-resonance control strategy in the off-grid mode, and the parallel pre-synchronization control strategy in the grid-forming mode.

[0015] In an exemplary embodiment of the present disclosure, a multi-mode control method for an energy storage converter is provided. The method includes: for the energy storage converter operating in a grid-following mode, establishing a low voltage ride-through control strategy based on repetitive control by creating a second-order infinite impulse response (IIR) filter; for the energy storage converter operating in an off-grid mode, using a three-phase four-bridge-arm topology to provide a zero-sequence path and establishing a preset proportional multi-resonance control strategy; for the energy storage converter operating in a grid-forming mode, establishing a parallel pre-synchronization control strategy; and achieving multi-mode control of the energy storage converter based on the low voltage ride-through control strategy that complies with repetitive control, the preset proportional multi-resonance control strategy, and the parallel pre-synchronization control strategy. The present disclosure can set any resonant frequency and filter radius to suppress the resonant peak, ensuring system stability. At the same time, compared with traditional pre-synchronization methods, it only utilizes its own frequency modulation characteristics and can complete pre-synchronization without additional control links, greatly saving control resources.

[0016] It is to be understood that the foregoing general description and the following detailed description are exemplary and explanatory only and are not restrictive of the disclosure. BRIEF DESCRIPTION OF THE DRAWINGS

[0017] The above and other features and advantages of the present disclosure will become more apparent by describing in detail example embodiments thereof with reference to the accompanying drawings.

[0018] Figure 1 A flow chart of a multi-mode control method for an energy storage converter according to an exemplary embodiment of the present disclosure is shown;

[0019] Figure 2 An equivalent circuit diagram of an energy storage converter in a control method according to an exemplary embodiment of the present disclosure is shown;

[0020] Figure 3 A flag bit judgment logic diagram in a control method according to an exemplary embodiment of the present disclosure is shown;

[0021] Figure 4 A logic diagram for determining the tolerance time of a drop greater than 0.8 according to an exemplary embodiment of the present disclosure is shown;

[0022] Figure 5 A logic diagram for determining the tolerance time for a drop of 0.15-0.8° is shown according to an exemplary embodiment of the present disclosure;

[0023] Figure 6 A logic diagram for determining the tolerance time of a drop of less than 0.15 is shown according to an exemplary embodiment of the present disclosure;

[0024] Figure 7 A mode switching logic diagram in a control method according to an exemplary embodiment of the present disclosure is shown;

[0025] Figure 8 A reactive power logic diagram for different depths of judgment in a control method according to an exemplary embodiment of the present disclosure is shown;

[0026] Figure 9 shows a limiting logic diagram in a control method according to an exemplary embodiment of the present disclosure;

[0027] Figure 10 A schematic diagram of the decomposition of the positive and negative sequence of the grid-connected voltage and current dq axes in a control method according to an exemplary embodiment of the present disclosure is shown;

[0028] Figure 11 A block diagram of positive and negative sequence current control in a control method according to an exemplary embodiment of the present disclosure is shown;

[0029] Figure 12 shows a composite repetitive control structure diagram in a control method according to an exemplary embodiment of the present disclosure;

[0030] Figure 13 shows the Bode diagram of the controlled object P(z) before and after compensation according to an exemplary embodiment of the present disclosure;

[0031] Figure 14 shows a phase-frequency characteristic curve diagram of a system with different k values ​​according to an exemplary embodiment of the present disclosure;

[0032] Figure 15 shows a root locus variation diagram for different k values ​​according to an exemplary embodiment of the present disclosure;

[0033] Figure 16 shows an open-loop amplitude-frequency curve diagram of repetitive control and compound control according to an exemplary embodiment of the present disclosure;

[0034] Figure 17FIG2 shows a block diagram of a VSG control algorithm according to an exemplary embodiment of the present disclosure;

[0035] Figure 18 FIG. 2 shows a general block diagram of a VSG according to an exemplary embodiment of the present disclosure;

[0036] Figure 19 A diagram showing a connection system of parallel pre-synchronous energy storage converters according to an exemplary embodiment of the present disclosure is shown;

[0037] Figure 20 FIG2 shows a schematic diagram of simplified pre-synchronization according to an exemplary embodiment of the present disclosure;

[0038] Figure 21 shows a main circuit diagram of a parallel PCS according to an exemplary embodiment of the present disclosure;

[0039] Figure 22 shows a block diagram without adding virtual impedance control according to an exemplary embodiment of the present disclosure;

[0040] Figure 23 FIG2 shows a VSG overall control block diagram of virtual stator impedance according to an exemplary embodiment of the present disclosure;

[0041] Figure 24 A schematic block diagram of a multi-mode control device for an energy storage converter according to an exemplary embodiment of the present disclosure is shown. DETAILED DESCRIPTION

[0042] Example embodiments will now be described more fully with reference to the accompanying drawings. However, example embodiments can be embodied in many forms and should not be construed as limited to the embodiments set forth herein; rather, these embodiments are provided so that this disclosure will be thorough and complete and will fully convey the concepts of the example embodiments to those skilled in the art. Like reference numerals in the drawings represent like or similar parts, and thus repetitive description thereof will be omitted.

[0043] In addition, the described features, structures or characteristics may be combined in any suitable manner in one or more embodiments. In the following description, many specific details are provided to provide a full understanding of the embodiments of the present disclosure. However, those skilled in the art will appreciate that the technical solutions of the present disclosure can be practiced without one or more of the specific details, or other methods, components, materials, devices, steps, etc. can be adopted. In other cases, well-known structures, methods and devices, implementations, materials or operations are not shown or described in detail to avoid blurring various aspects of the present disclosure.

[0044] The blocks shown in the accompanying drawings are merely functional entities and do not necessarily correspond to physically separate entities. Specifically, these functional entities may be implemented in software, or in one or more software-hardened modules, or in different networks and / or processor devices and / or microcontroller devices.

[0045] In this exemplary embodiment, a multi-mode control method for an energy storage converter is first provided; Figure 1 As shown in , the multi-mode control method of the energy storage converter may include the following steps:

[0046] Step S110, for the energy storage converter operating in the grid-following mode, a low voltage ride-through control strategy based on repetitive control is established by creating a second-order infinite impulse response (IIR) filter;

[0047] Step S120, for the energy storage converter operating in off-grid mode, a three-phase four-leg topology is used to provide a zero-sequence path, and a preset proportional multi-resonance control strategy is established;

[0048] Step S130, establishing a parallel pre-synchronization control strategy for the energy storage converter operating in the grid-forming mode;

[0049] Step S140 , implementing multi-mode control of the energy storage converter based on the low voltage ride-through control strategy that complies with repetitive control in the grid-following mode, the preset proportional multi-resonance control strategy in the off-grid mode, and the parallel pre-synchronization control strategy in the grid-forming mode.

[0050] In an exemplary embodiment of the present disclosure, a multi-mode control method for an energy storage converter is provided. The method includes: for the energy storage converter operating in a grid-following mode, establishing a low voltage ride-through control strategy based on repetitive control by creating a second-order infinite impulse response (IIR) filter; for the energy storage converter operating in an off-grid mode, using a three-phase four-bridge-arm topology to provide a zero-sequence path and establishing a preset proportional multi-resonance control strategy; for the energy storage converter operating in a grid-forming mode, establishing a parallel pre-synchronization control strategy; and achieving multi-mode control of the energy storage converter based on the low voltage ride-through control strategy that complies with repetitive control, the preset proportional multi-resonance control strategy, and the parallel pre-synchronization control strategy. The present disclosure can set any resonant frequency and filter radius to suppress the resonant peak, ensuring system stability. At the same time, compared with traditional pre-synchronization methods, it only utilizes its own frequency modulation characteristics and can complete pre-synchronization without additional control links, greatly saving control resources.

[0051] Next, a multi-mode control method for an energy storage converter in this exemplary embodiment will be further described.

[0052] In step S110 , a low voltage ride-through control strategy based on repetitive control may be established by creating a second-order infinite impulse response (IIR) filter when the energy storage converter operates in the grid-following mode.

[0053] In this exemplary embodiment, the method further comprises:

[0054] In the dq synchronous rotating coordinate system, the mathematical modeling of the energy storage converter and the control model in the grid-following mode are established;

[0055] Create a second-order infinite impulse response (IIR) filter and design a repetitive controller based on the second-order infinite impulse response (IIR) filter.

[0056] In this exemplary embodiment, the method further comprises:

[0057] Design and calculate the auxiliary link Q(z) for adjusting system performance, the compensation link S(z) for realizing amplitude and frequency characteristic correction, the feedforward coefficient k for compensating system phase lag, and the gain coefficient k for adjusting system response amplitude. r , to achieve repetitive controller design.

[0058] In this exemplary embodiment, the method further comprises:

[0059] The phase lag compensation coefficient is designed and calculated. Based on the phase lag compensation coefficient and the repetitive controller, a low voltage ride-through control strategy based on repetitive control is established.

[0060] In step S120 , a three-phase four-leg topology may be used to provide a zero-sequence path for the energy storage converter to operate in an off-grid mode, and a preset proportional multi-resonance control strategy may be established.

[0061] In this exemplary embodiment, the method further comprises:

[0062] Modeling the three-phase four-bridge-leg topology energy storage converter in a rotating coordinate system to generate a mathematical model of the three-phase four-bridge-leg topology energy storage converter;

[0063] Based on the mathematical model of energy storage converter, a mathematical model of energy storage converter under unbalanced load and nonlinear load is established;

[0064] Based on the mathematical model of the energy storage converter under unbalanced load and nonlinear load, a control strategy for the energy storage converter in off-grid mode is established.

[0065] In this exemplary embodiment, the method further comprises:

[0066] In a control system of an energy storage converter in an off-grid mode, a preset number of quasi-PR controllers are connected in parallel to establish a preset proportional multi-resonance control strategy for the energy storage converter in the off-grid mode.

[0067] In step S130 , a parallel pre-synchronization control strategy may be established for the energy storage converter operating in the grid-forming mode.

[0068] In the embodiment of this example, the method further includes: establishing a mathematical model of the energy storage converter in the grid-forming mode;

[0069] Based on the mathematical model of the energy storage converter in grid-connected mode, the current loop control parameters, voltage loop control parameters, active power-frequency loop parameters, and reactive power-voltage loop parameters are designed and calculated respectively;

[0070] Establish a small signal model of the active loop of the energy storage converter.

[0071] In this exemplary embodiment, the method further comprises:

[0072] Based on the virtual stator impedance, the parallel parameters of the energy storage converter in the grid-forming mode are corrected, and a parallel pre-synchronization control strategy is generated to achieve parallel pre-synchronization of the energy storage converter in the grid-forming mode.

[0073] In step S140, multi-mode control of the energy storage converter can be achieved based on the low voltage ride-through control strategy that complies with repeated control in the grid-following mode, the preset proportional multi-resonance control strategy in the off-grid mode, and the parallel pre-synchronization control strategy in the grid-forming mode.

[0074] In this exemplary embodiment, in response to the numerous requirements for low voltage ride-through in grid-following mode and the complexity of the control links, this disclosure proposes a simplified low voltage ride-through logic based on a table lookup method. The low voltage ride-through control strategy of the energy storage converter in grid-following mode is studied, and a low voltage ride-through control method based on composite repetitive control is proposed. This method eliminates the need for positive and negative sequence separation, reduces control complexity, and has better harmonic suppression capabilities in normal grid-following mode. In the design of the repetitive controller, a new second-order infinite impulse response (IIR) filter is proposed. It can set any resonant frequency and filter radius to suppress the resonant peak and ensure system stability.

[0075] To address the issue of three-phase voltage imbalance in off-grid mode, the disclosed energy storage converter utilizes a three-phase, four-leg topology to provide a zero-sequence path, achieving 100% unbalanced load capability. Proportional multi-resonance is employed to dramatically reduce output waveform distortion under nonlinear loads, ensuring power supply quality.

[0076] To address the complexities of virtual synchronous control parameters and pre-synchronization control design in grid-forming mode, effective parameter design guidance is provided based on national standards. This paper proposes a novel pre-synchronization method for virtual synchronous machine control in grid-forming mode for energy storage converters. Compared to traditional pre-synchronization methods, this method only utilizes the inherent frequency modulation characteristics of the converter itself, eliminating the need for additional control steps to achieve pre-synchronization, significantly saving control resources.

[0077] In the context of the low-carbon transformation of the global energy structure, the power grid system faces technical challenges brought about by the large-scale access of renewable energy. On the one hand, the high penetration rate of intermittent distributed power sources such as wind and solar power leads to the deterioration of power supply quality (voltage and current fluctuation rate is as high as ±12%); on the other hand, the problem of insufficient low voltage ride-through capability of the system is becoming increasingly prominent (when the voltage drops to 0.2pu, the risk of grid disconnection increases by 47%). This paper conducts an optimization study on the grid-following mode of the three-phase two-level energy storage converter. The specific implementation path is as follows: first, a mathematical model based on the dq coordinate system is established, and the interaction mechanism between the converter and the grid is revealed through coordinate transformation; based on the mathematical model, a grid-following control strategy is designed, and the key parameters of the LC filter are determined to provide a theoretical basis for subsequent simulations; then, the control method under low voltage ride-through conditions is focused on, and the use of composite repetitive control for low voltage ride-through is proposed to improve the harmonic suppression capability while reducing the control complexity.

[0078] In this exemplary embodiment, Figure 2 The topology shown is a single-stage two-level PCS system, in which the DC side uses energy storage medium and forms a three-phase inverter bridge through power switch tubes S1 to S6 to convert the DC bus voltage U dc Converted into a controllable three-phase AC voltage. After filtering inductor L a 、L b 、L c and filter capacitor C a 、C b 、C c After the LC filter network is formed, the output voltage is connected to the grid. In addition, the circuit topology also sets the damping resistor R d Used to suppress the resonance of the filter link. Based on the single-phase AC equivalent circuit corresponding to the three-phase topology, the PCS AC side output voltage is expressed as u o Indicates that the voltage on the grid side is e g , the current flowing through the output branch is represented by i o The entire PCS system realizes power conversion from DC to AC and adjusts u o With e g The amplitude and phase difference between them controls the output current i o , we can know that:

[0079] In the embodiment of this example, the dq coordinate coefficient-based mathematical modeling and network-following control include:

[0080] When the energy storage battery is charging and discharging, it exchanges energy with the grid. In the dq synchronous rotating coordinate system, the d-axis is strictly aligned with the grid voltage vector direction and rotates synchronously. In this way, the grid voltage component on the d-axis, e dEqual to the grid voltage amplitude E m , and the component e on the q axis q =0. In this manner, the dq coordinate system always rotates synchronously with the grid voltage vector, simplifying system power calculations. The instantaneous power calculation formula in the dq coordinate system can be used to derive the specific power flow and control relationship, enabling precise control of the charging or discharging power of the energy storage battery.

[0081]

[0082] At this time, the active power P is only related to the active current i d The reactive power Q is only related to the reactive current i q It is related to the active power and reactive power, that is, the decoupling control is realized.

[0083] Since the sampling voltage is the capacitor voltage, the filter capacitor and transformer impedance do not affect the control, and only the single inductor filtering case is considered.

[0084] according to Figure 2 And KVL theorem shows that:

[0085]

[0086] The transformation matrix from the abc coordinate system to the dq coordinate system (3s / 2r) is:

[0087]

[0088] After dq transformation of formula (3) by formula (4), we can get:

[0089]

[0090] Simplifying formula (5), we can get:

[0091]

[0092] The mathematical model of PCS grid connection contains i d 、i q Cross-coupling term ωLi d and ωLi q , while i d 、i q Also affected by the grid voltage e d 、e q The presence of nonlinear loads in real power grids generates harmonic currents that flow through line impedance, causing background harmonics to appear in the voltage at the point of common coupling (PCC). When the grid voltage at the PCC fluctuates and contains background harmonics, the PCS output current will be significantly distorted.

[0093] In order to achieve i d 、i q Independent closed-loop control and suppression of grid voltage fluctuations and background harmonics on grid current, adding i d 、i q Feedforward term ωLi d 、-ωLi q To eliminate the cross-coupling term to achieve i d 、i q Decoupling control; adding grid voltage feedforward term e d 、e q Suppress the impact of grid disturbances such as grid voltage fluctuations and background harmonics on grid current.

[0094] After adding the grid voltage feedforward compensation term and the current cross feedforward decoupling term, i d 、i q The complete decoupling between them can be controlled independently d 、i q After the dq axis is decoupled, the current inner loop can use a PI regulator to achieve zero-static-error tracking of the command current.

[0095] In the embodiment of this example, the filter circuit parameter design includes:

[0096] The parameters of the filter circuit have a significant impact on the implementation of the control strategy. In order to ensure the accuracy of the simulation model and control strategy verification, it is necessary to calculate the parameters of the grid-side filter.

[0097] The filter structure uses LC filter. The values ​​of filter inductor L1 and filter capacitor directly affect the output power quality of the grid-connected inverter.

[0098] Filter inductor design: Considering the inverter side inductor current ripple limit, the minimum inductance of the output filter inductor L1 is calculated according to formula (7):

[0099]

[0100] L 1min is the minimum inductance value, V bus is the bus voltage, F svm f is the normalized current ripple coefficient of SVM modulation, that is, the current ripple on the filter inductor when the modulation ratio is known under unit bus voltage, switching frequency and inverter side inductance; sw is the switching frequency, ΔI L is the ripple of the output inductor current (between 15% and 30%)*IN).

[0101] Take F svm =0.03Calculation: Set ΔI LThe value is 20% IN; press I N = 20% of 15, we get ΔI L =3A.

[0102]

[0103] Considering the total voltage drop limit of the filter, the total voltage drop U generated by the inductive impedance of the filter is L Less than 10% of the rated value of the grid voltage. Calculate the maximum inductance of the output filter inductor L1 according to formula (9).

[0104]

[0105] L 1max is the maximum filter inductance value, V N is the rated value of the grid voltage, f g is the grid frequency, I outmax is the maximum output current.

[0106] Calculate L 1max :

[0107]

[0108] According to formula (10), the value is between 770uH and 3.3mH, and 3mH is used in simulation.

[0109] Filter capacitor design: Designed through a resonant filter. If the resonant frequency is too high, it will lead to larger high-order harmonics. If the resonant frequency is too low, it will lead to a lower bandwidth of the current control loop and unsatisfactory system dynamics. To ensure the parameter f k To ensure the stability of the filter capacitor, it is usually selected to be less than half of the switching frequency and significantly higher than the base frequency, that is, its value should be at least greater than the grid base frequency and at the same time meet the requirement of being far below the switching frequency. The filter capacitor value is determined by the resonant frequency. Satisfying the relationship (11)

[0110]

[0111] General k Range is 10f n ≤f k ≤f c / 5f n Power frequency

[0112] Set the resonant frequency f k =1kHzSubstitute into formula (11) to get.

[0113]

[0114] According to formula (12), the value is 10uF.

[0115] Damping resistor design: To avoid oscillation, add passive damping. Usually the more suitable choice is C under resonant conditions. f The damping resistor has the same relative magnitude as the impedance. The damping resistor R is calculated according to formula (13): d value.

[0116]

[0117] calculate:

[0118]

[0119] Due to the parasitic resistance in the transmission circuit and the filter inductor and capacitor, R d Select a value of 1Ω.

[0120] In the embodiment of this example, the low voltage ride-through control method of the three-phase energy storage converter includes:

[0121] As new energy distributed power sources account for an increasing proportion in the power grid, when the voltage in the grid drops, the distributed power sources are required to have certain low voltage ride-through and voltage support capabilities to prevent large-scale disconnection of distributed power sources and large-scale active power loss, which will further worsen the grid fault.

[0122] (1) Low voltage ride-through requirements

[0123] According to the relevant requirements of the national standard GB / T 34120-2023, when a power system fault causes the voltage at the PCS grid connection point to drop, the low voltage ride-through voltage limit curve should be used for judgment. If the grid connection point voltage remains above or equal to the specified voltage contour line, the PCS must maintain continuous grid connection and cannot disconnect from the grid. If the voltage drops below the curve requirements, the PCS is allowed to disconnect from the grid.

[0124] Specifically, there are three requirements for the low voltage ride-through capability of PCS:

[0125] (1) When the voltage at the PCS grid connection point drops completely to zero, the PCS should maintain stable operation without disconnecting from the grid for at least 0.15 seconds;

[0126] (2) If the voltage at the PCS grid connection point is lower than the voltage contour line shown by curve 1 in the figure, the PCS can be allowed to disconnect from the grid;

[0127] (3) For PCS that are not disconnected from the grid during the fault, power recovery should be started immediately after the fault is cleared, and its active power should be quickly restored to the pre-fault level at a rate of change of not less than 30% of the rated power per second.

[0128] When a grid voltage sag occurs, if the PCS power remains unchanged while the AC side voltage decreases, without suppressing the AC output current, an AC overcurrent fault will occur in the PCS, causing shutdown. Therefore, one control goal is to operate at a reduced rating when a voltage sag occurs. In the four grid-following modes (constant voltage, constant current, constant DC power, and constant AC power), the power is generally lower under constant voltage control. In the constant current mode and the two constant power modes, the DC current setpoint value i dcref And power set value P ref Reduced to enable continuous operation during low voltage ride-through.

[0129] When the grid voltage drops asymmetrically, the three-phase grid current will be unbalanced. Therefore, another goal is to control the unbalanced current. When an asymmetrical voltage drop occurs, the negative sequence component of the grid current must be suppressed to prevent a larger fault.

[0130] (2) The specific requirements for dynamic reactive voltage support capability are as follows:

[0131] When the power system voltage drops due to a short circuit fault, the PCS should quickly activate the dynamic reactive voltage support function.

[0132] ① After the grid voltage drops, the energy storage converter needs to respond quickly and output the corresponding reactive current to the grid;

[0133] ② From the time the reactive current starts to respond until the voltage level returns to 0.85pu, the dynamic reactive current I provided by the PCS to the grid is T It must be able to accurately track the changing trend of the grid connection point voltage in real time and meet the following support requirements at the same time.

[0134]

[0135] Where U T is the per-unit voltage of the PCS grid connection point; I N is the PCS rated current.

[0136] In order to achieve dynamic reactive power support, it is necessary to collect the grid voltage in real time and set the reactive current set value as required to adjust the size of the reactive current. In order to prevent the reactive current from being too large and triggering the overcurrent protection, the rated current I N As per unit value, reactive current instruction i qref It is given according to its boundary, as shown in formula (16).

[0137]

[0138] Low voltage ride-through control requirements for three-phase energy storage converters: During low voltage ride-through, the voltage withstand capability requirements can be divided into two parts:

[0139] (1) Voltage tolerance can be divided into three areas:

[0140] (a): Drop depth is less than 0.15

[0141] (b): The drop depth is between 0.15 and 0.8

[0142] (c): Drop depth greater than 0.8

[0143] (2) The holding time of different intervals is different, and the corresponding flag bit is judged to facilitate the subsequent tolerance time calculation. Figure 3 shown.

[0144] (3) Figure 4 This is the logic diagram for judging the tolerance time when the drop depth is greater than 0.8. The tolerance time is calculated as shown in the figure. It can be seen from the figure that when the drop depth is greater than 0.8, it is 0.15s. The counter is used to count at the beginning of the crossing. If it exceeds 0.15s, the flag position is 0 and exits. If it is less than 0.15s, the counting continues.

[0145] Figure 5 This is a logic diagram for judging the tolerance time of a drop of 0.15-0.8. As shown in the figure, when the drop depth is in the range of 0.8 to 0.15, the tolerance time is linearly required according to different drop depths. For the convenience of calculation, the tolerance is rounded and the tolerance calculation is performed based on the closest depth. If it exceeds the limit, the calculation will be exited, and if it does not exceed the limit, the calculation will continue.

[0146] Figure 6 This is the logic diagram for judging the tolerance time when the drop depth is less than 0.15. As shown in the figure, when the drop depth is in the range of 0.8 to 0.15, when the drop is less than 0.15, all flag bits are cleared to 0, and the low penetration is exited at the same time, and normal mode is restored.

[0147] (4) Reactive power support capability is required in low voltage ride-through. The relationship between voltage and current is shown in Equation (17).

[0148] I T ≥1.6×(0.85-U T )I N (0.2≤U T ≤0.85)

[0149] I T ≥1.04×I N (U T <0.2)

[0150] I T =0(U T >0.85) (17)

[0151] Among them, U T is the per-unit value of the grid-connected point voltage, I N is the rated current.

[0152] It is divided into three parts as required:

[0153] (1) Switch from active / reactive mode to constant current mode according to the low wear flag bit, such as Figure 7 shown.

[0154] (2) Provide different reactive supports according to different intervals and establish the relationship between drop depth and reactive current. Figure 8 shown.

[0155] (3) According to the current amplitude limit, the part exceeding the maximum current is processed as the maximum value, such as Figure 9 shown.

[0156] Separation of positive and negative sequence voltage and current: When an asymmetric fault occurs in the power system, the AC side voltage and current of the energy storage grid-connected converter will show asymmetric characteristics. Based on the symmetrical component analysis method, the actual asymmetric three-phase voltage and current phasors can be analyzed into three independent components, namely positive sequence, negative sequence, and zero sequence components. The relationship between these components satisfies Equation (18). Considering that the energy storage converter is usually connected to the grid using a three-phase three-wire structure, the zero sequence component can be ignored. Therefore, in the system analysis, only the influence of the positive sequence and negative sequence components needs to be considered, thereby simplifying the complexity of the problem.

[0157]

[0158] In formulas (18) and (19), z a 、z b 、z c represents the original asymmetric three-phase phasor, and Represent the three-phase positive sequence components, represents the negative sequence component, is the three-phase zero-sequence component. Parameter a is the operator used for symmetrical component transformation, defined as a=e j120° According to the above definition, the three-phase asymmetric quantities can be mapped to the positive, negative, and zero-sequence coordinate systems respectively after the symmetrical component transformation. Subsequently, using the positive and negative sequence Park transformations shown in Equations (20) and (21), these components can be further mapped to the dq rotating coordinate system, thereby clearly distinguishing the positive and negative sequence phasor components, which is convenient for subsequent analysis and control.

[0159]

[0160] Where, Represents the DC mapping of the positive sequence component in the dq rotating coordinate system; Then it represents the DC mapping of the negative sequence component in the dq coordinate system. In addition, ω0 represents the angular velocity of the asymmetric quantity used for synchronous rotation coordinate transformation, Figure 10 The mapping relationship between the positive and negative sequence components of the grid voltage and current in their respective dq coordinate systems under asymmetric conditions is clearly demonstrated.

[0161] According to the principle of the symmetrical component method, when the grid voltage or current is asymmetric, it can be expanded into a combination of a DC component and a double-frequency fluctuation component in the synchronously rotating dq coordinate system. Specifically, if the dq synchronous coordinate system is fixed in the positive sequence rotation direction, the negative sequence component in the original asymmetric signal is manifested as a double-frequency AC fluctuation component superimposed on the positive sequence DC component; if the synchronous coordinate system rotates in the negative sequence direction, the positive sequence component at this time is manifested as a double-frequency AC fluctuation component superimposed on the negative sequence DC component. For the above two situations, Equation (6) gives the decomposition of the asymmetric signal in the positive sequence rotating reference coordinate system, and Equation (22) expresses the decomposition of the asymmetric signal in the negative sequence rotating coordinate system.

[0162]

[0163] In formulas (22) and (23), z d 、z q are the dq components of the asymmetric vector, f and g represent double frequency functions, where f(-2ω0t)=cos(-2ω0t), g(-2ω0t)=sin(-2ω0t), f(2ω0t)=cos(2ω0t), g(2ω0t)=sin(2ω0t).

[0164] A notch filter is a type of filter designed to suppress specific frequency harmonics and reduce the impact of interference on the system. When a fundamental signal is superimposed with a harmonic interference component of a specific frequency, a notch filter can precisely filter out that specific harmonic frequency component, retaining only the desired fundamental signal.

[0165] The transfer function of the notch filter can be expressed as:

[0166]

[0167] In formula (34), ω α is the filtering frequency, in order to filter out the double frequency interference signal, so ω α =2ω g =628rad / s

[0168] Based on the above analysis, it can be seen that in order to accurately achieve the positive and negative sequence separation and effective control of the AC side voltage and current, the following method can be adopted: first, the three-phase asymmetric voltage and current signals are subjected to Park coordinate transformation of the positive and negative sequences respectively, and the original abc stationary coordinate system phasors are projected into the dq rotating coordinate system; then, the doubled frequency AC component in the transformed signal is removed by a notch filter, and the corresponding positive and negative sequence DC components are extracted, thereby achieving accurate separation of the positive and negative sequence voltage and current on the grid side.

[0169] Negative-sequence current suppression: During grid-connected PCS operation, factors such as load imbalance, short-circuit faults, and non-full-phase operation can lead to three-phase grid voltage imbalance, which in turn leads to three-phase grid current imbalance. Grid voltage feedforward compensation within the inner current loop can suppress grid voltage disturbances on grid current. However, when the three-phase grid voltage is unbalanced, the PCS, which uses a PQ control strategy and targets constant output power, uses instantaneous power theory to determine that the inner current command generated by the outer power loop contains an unbalanced component, resulting in unbalanced three-phase current output by the PCS.

[0170] However, the power grid usually requires that the grid-connected inverter equipment's grid-connected current be well-balanced, so the unbalanced current should be compensated. According to the symmetrical component method, when the grid-connected current is unbalanced, only the fundamental grid-connected current is considered. The grid-connected current of a three-phase three-wire PCS can be decomposed into positive-sequence components and negative-sequence components as follows:

[0171]

[0172] The positive sequence component of current is expressed as:

[0173]

[0174] In formula (36), I g + is the amplitude of the positive sequence component of the grid current.

[0175] The negative sequence component of the grid current is expressed as

[0176]

[0177] Where, I g - is the amplitude of the negative sequence component of the grid current.

[0178] The positive and negative sequence components of the grid-connected current can be obtained by the positive sequence 3s / 2r coordinate transformation.

[0179]

[0180] According to the above equation, when the fundamental positive-sequence dq rotating coordinate system is used as a reference, the negative-sequence current component manifests as a fluctuation at double the frequency. Similarly, if the coordinate system rotates in the negative-sequence direction, the positive-sequence current component also manifests as an alternating variable at double the frequency. Because traditional PI controllers have insufficient gain near double the frequency, control strategies using a single rotating coordinate system are unable to effectively eliminate this double-frequency imbalance. Therefore, a dual synchronous rotating coordinate system is employed to achieve independent control of the positive and negative sequence currents.

[0181] Figure 11 In, T P abc-dq1 、T N abc-dq1 They are the transformation matrix from the abc three-phase stationary coordinate system to the fundamental positive sequence dq coordinate system and the transformation matrix from the abc three-phase stationary coordinate system to the fundamental negative sequence dq coordinate system. + d 、i + q are the dq axis components of the output current in the fundamental positive sequence dq coordinate system; i - d 、i - q are the dq axis components of the output current in the fundamental wave negative sequence dq coordinate system. A notch filter is used to filter out the double frequency signal to obtain the DC components of the positive and negative sequence currents. + dref 、u - qref and u - dref 、u - qref are the output instructions of the fundamental positive sequence and negative sequence current controllers respectively. The control strategy consists of two parts: positive sequence current regulation in the fundamental positive sequence dq coordinate system, and negative sequence current regulation in the fundamental negative sequence dq coordinate system. The fundamental negative sequence current instruction is set to i - dref =i - qref =0, thereby effectively suppressing the unbalanced influence of the system on the negative sequence component.

[0182] In the embodiment of this example, the low voltage ride-through method based on repetitive control includes:

[0183] As mentioned above, traditional PI control requires positive- and negative-sequence separation and negative-sequence current suppression, making control complex. A composite repetitive control approach is proposed for low-voltage ride-through (LVRT), eliminating the need for positive- and negative-sequence separation and negative-sequence current control, thus reducing control complexity. Because conventional repetitive controllers have a slow response, a composite repetitive controller is used to improve controller response speed and meet LVRT requirements. A second-order infinite impulse response (IIR) filter is also proposed, allowing for arbitrary resonant frequency and filter radius settings to suppress resonant peaks and enhance stability. The following is the controller analysis and design process.

[0184] Proportional resonance-repetitive control theory analysis: The transfer function of the control object P(s) of the inverter circuit system under no-load conditions can be simplified to:

[0185]

[0186] Based on formula (39), after discretization using the "zero-order holder (ZOH)" method, the P(z) transfer function expression of the control system is:

[0187]

[0188] The composite repetitive control structure adopted in this disclosure is as follows Figure 12 shown.

[0189] Figure 12 The repetitive control structure in the MATLAB software mainly consists of two key modules: the periodic error integration link and the compensation link. The periodic integration function is implemented by the internal model, which is realized by the transfer function z -N / (1-Q(z)z -N ) accumulates the error e(z) periodically, thereby eliminating the recurring periodic error components in the system and improving the accuracy and stability of the output waveform. When the condition Q(z) = 1 is met, the internal model has the ideal tracking ability for the periodic external disturbance signal, that is, it achieves zero steady-state error control. The model uses the advance compensation link z -N To correct the phase delay and amplitude attenuation caused by the control object and the filtering link, the system can accurately follow the reference signal. Compensation link C(z) = k r z mS(z) is composed of an advance correction link and a filtering network. The former uses the positive power of z to achieve phase lag compensation for the controlled object and the filter, thereby improving the frequency characteristics of the system; the latter is mainly used to adjust the amplitude response characteristics. Specifically, the advance link offsets the inherent phase delay and frequency characteristic deficiencies of the system by introducing the forward shift characteristic of the z domain, thereby optimizing the dynamic performance of the system and improving the response speed and accuracy. In addition, the filter part ensures that the system has appropriate gain characteristics and anti-interference capabilities in the key frequency band, making the overall control effect more robust and reliable. S(z) is usually designed by combining a low-pass filter and a notch filter to ensure that the control system maintains unity gain in the medium and low frequency bands, while effectively attenuating high-frequency harmonics and noise to ensure system stability and anti-interference capabilities.

[0190] When the system is stable, the tracking error e(z) of the composite control is:

[0191]

[0192] Let P o (z)=P(z) / [1+G pr (z)P(z)], H1(z)=1+G pr (z)P(z), H2(z)=1-z -N [Q(z)-C(z)P o (z)], so the eigenvalue polynomial H(z) of the system can be obtained as:

[0193] H(z)=H1(z) H2(z) (42)

[0194] To ensure stable system operation, the stability criterion of Equation (42) requires that all roots of the characteristic equations H1(z) = 0 and H2(z) = 0 lie within the unit circle. When the PR regulator alone controls the inverter, the stability of the closed-loop transfer function is determined by the characteristic polynomial H1(z); when the repetitive controller alone acts on the control object P(z), the closed-loop stability of the system is determined by the characteristic polynomial H2(z). Therefore, for the combined control method, it is first necessary to ensure that the system characteristic roots (i.e., roots satisfying H1(z) = 0) when the PR regulator alone acts are within the unit circle. Subsequently, the roots of the system characteristic polynomial H2(z), which only includes the repetitive control action, are also within the unit circle to ensure the stability of the entire composite control system. This also means that the control object P(z) is stable under the repetitive control alone.

[0195] When repeat control works alone:

[0196]

[0197] At the same time, the system error e(z) is:

[0198]

[0199] The characteristic equation of the system can be expressed as:

[0200] |z -N (Q(z)-z k s(z)k r P o (z))|<1 (46)

[0201] By z N =1, we can deduce that:

[0202] |Q(z)-z k s(z)k r P0(z)|<1 (47)

[0203] Proportional Resonant-Repetitive Control Controller Design:

[0204] (1) Proportional resonant controller design

[0205] In summary, in order to ensure the stability of the composite controller, it is necessary to design a proportional resonant controller first. The proportional resonant controller expression is:

[0206]

[0207] Where K p is the proportionality coefficient, K r is the resonant controller coefficient, and ω is the resonant frequency. This disclosure adopts the "ZOH" discrete method, and the discrete expression is:

[0208]

[0209] There are significant differences between the resonant controller and the proportional controller. The effect of the resonant controller is concentrated near a specific resonant frequency point, while the impact on the non-resonant frequency point is very limited. Therefore, when designing the controller, you can temporarily ignore the impact of the resonant control link and determine the appropriate proportional gain K based on the proportional controller alone. p Once the proportional parameters are set properly, the resonant controller parameters K can be further determined based on this. r .

[0210] (2) Proportional coefficient design

[0211] When the control object adopts only proportional control, based on equation (40), after discretization using the "zero-order holder (ZOH)" method, the open-loop transfer function expression of the control system is:

[0212]

[0213] According to formula (50), we can draw the root locus diagram and obtain K that keeps the system stable. p The value range is [0, 0.289]. To ensure a certain stability margin, the phase margin must be 30 to 60 degrees and the gain margin must be ≥ 6dB. K is selected based on the open-loop Bode diagram of the system under proportional action. p =0.14, the gain margin is 6.11dB, and there is sufficient phase margin.

[0214] (3) Resonance control parameter design

[0215] The frequency characteristics of the resonant controller are characterized by a significant ±90° phase mutation near the resonant frequency. This mutation increases the difficulty of analyzing the stability of the system in the frequency domain. Therefore, when designing a resonant controller, the root locus analysis method should be used to plot the change trajectory of the resonant parameter Kr based on the determined proportional controller coefficient to determine Kr. r The reasonable value range of ensures that the stability of the entire system is effectively guaranteed. The closed-loop continuous transfer function characteristic equation of the system can be derived from its open-loop transfer function, and the expression is:

[0216]

[0217] According to the basic principle of the root locus method, the system to be analyzed can be transformed into an equivalent structure with unit feedback. At this time, the equivalent open-loop transfer function of the system can be expressed as:

[0218]

[0219] Using the “ZOH” discretization method, the discrete transfer function of B(s) is obtained as shown in Equation (53), from which K is obtained r The root locus during the change.

[0220]

[0221] Through root locus analysis, the parameter K of the resonant controller is r The theoretical value range is 0 to 375, and the specific value needs to be determined through further in-depth analysis. r As K increases gradually, the dynamic response speed of the system is significantly improved, but the corresponding stability margin is continuously reduced. r As the value of K increases, the peak amplitude at the resonant frequency of the LC filter also increases. Considering that there may be parameter fluctuations and delay effects in the LC filter under actual working conditions, in order to ensure the robustness of the system and balance the relationship between dynamic performance and stability margin, K is finally determined. r is 300.

[0222] (4) Repeated controller design

[0223] The parameter design of the repetitive controller involves the following core elements: the auxiliary link Q(z) for adjusting the system performance, the compensation link S(z) for correcting the amplitude and frequency characteristics, the feedforward coefficient k for compensating the system phase lag, and the gain coefficient k for adjusting the system response amplitude. r .

[0224] 1) Design of Q(z)

[0225] The parameter Q(z) plays a vital role in the design of repetitive controllers. Reasonable selection of Q(z) can improve system stability and ensure the tracking target of zero steady-state error. Generally speaking, Q(z) can be set to a fixed constant close to but less than 1, or a low-pass filter can be used. When Q(z) is very close to 1, the steady-state performance of the system is better and the harmonic suppression effect is enhanced, but it is easy to cause the amplitude of the transfer function H(z) in the high-frequency range to exceed the stability limit, causing system instability. In contrast, although Q(z) in the form of a low-pass filter can better ensure stability, it will weaken the system's suppression performance for medium and high-frequency harmonics and increase the difficulty of controller design. Taking into account stability, design complexity and robustness, this disclosure chooses the fixed constant method and takes Q(z) = 0.95.

[0226] 2) Design of S(z)

[0227] The compensator S(z) link is generally implemented by a combination of a low-pass filter and a notch filter. When the low-pass filter cutoff frequency is selected to be slightly lower than the resonant frequency of the LC filter network itself, it can effectively reduce the system resonance peak and significantly suppress high-frequency signal components. Considering that the resonant point frequency of the LC filter is approximately 973Hz, this disclosure designs a second-order Butterworth low-pass filter with a cutoff frequency set at 1kHz. It is then converted into a discrete form using the zero-order hold (ZOH) discretization method. The discretized expression is as follows:

[0228]

[0229] like Figure 13 Figure 1 shows the Bode plots of the controlled object P(z) before and after compensation. Comparing curves a and b, we can see that while the low-pass filter has some effect on suppressing the resonant peak, the effect is still unsatisfactory. Therefore, a second-order infinite impulse response (IIR) filter is often required to eliminate the resonant peak without compromising system stability. The basic form of a second-order IIR filter is:

[0230]

[0231] It is known that the suppression frequency f0 = 900Hz and the sampling frequency f s=10kHz, filter radius r = 0.1, calculated ω o =0.5655. The transfer function of the second-order infinite impulse response (IIR) filter designed in this way is:

[0232]

[0233] Figure 13 Curve c shows the Bode plot of the controlled object P(z) after compensation through a combination of a low-pass filter and a notch filter. As can be seen from the curve, the notch filter effectively reduces the system's original resonant peaks. Furthermore, the system maintains the desired amplitude characteristics near the fundamental frequency and achieves near-0dB gain in other mid- and low-frequency ranges.

[0234] 3) Phase lag compensation coefficient design

[0235] Depend on Figure 14 It can be seen that except for the area near the fundamental frequency, due to the inherent phase lag characteristics of P(z) and the additional lag introduced by the fourth-order Butterworth filter, the system has a significant phase offset in the low frequency band. In order to improve the phase-frequency characteristics of the system, the leading link z is introduced k Perform phase compensation. Figure 14 As shown, when k=2 is selected, the system can approach the zero-phase characteristic in a wider frequency range, thereby having the harmonic compensation capability within the 1 kHz frequency band.

[0236] Proportional gain k of repetitive control r Directly affects the response speed and harmonic suppression capability of the control system. Selecting a suitable k r The value has an important influence on the stability of the system. After determining the phase compensation parameter k value, it is necessary to further determine k r The value range of different k r The parameter trajectory corresponding to the value is as follows Figure 15 shown.

[0237] Depend on Figure 15 It can be seen that the system is in a critical stable state near the fundamental frequency. This phenomenon is caused by the ±90° phase jump caused by the proportional resonant controller at the fundamental frequency. However, when the repetitive controller is designed independently, the gain amplitude of the controlled object P(s) at the fundamental frequency tends to infinitesimal. Under the parameter configuration of Q(z) = 0.95, theoretically, no matter k r The system can remain stable near the fundamental frequency regardless of the value of k. r As the gain increases, the error convergence speed and harmonic suppression effect of the system are improved, but at the same time, the system stability margin is gradually reduced. Considering the uncertainties such as control delay and modeling error in the actual system operation process, this paper selects a more conservative gain value, that is, kr =2 to balance stability and system performance.

[0238] According to the composite control parameters determined above, the present disclosure further draws the open-loop amplitude-frequency characteristic curves of repetitive control and composite control, such as Figure 16 As shown in the figure, the composite control strategy exhibits the same high-gain characteristics as repetitive control at the fundamental frequency, and maintains consistent harmonic suppression in the mid- and low-frequency bands. Furthermore, due to the introduction of the rapid dynamic response of the proportional resonant controller, the composite control strategy exhibits better transient performance and response speed.

[0239] In this example embodiment, the networking mode control strategy research includes:

[0240] In the process of building new power systems, the penetration rate of power electronic equipment has exceeded 63% (Global Energy Interconnection Development Report 2023), resulting in a 58% decrease in grid inertia compared to traditional systems (typical regional system inertia constant H = 2.1s → 0.9s), and a maximum rate of change of frequency (RoCoF) of 1.2 Hz / s (NERC standard limit is 0.5 Hz / s). To address this issue, this paper optimizes the grid control mode. This paper adopts virtual synchronous machine control. First, the typical characteristics and operating boundaries of the weak grid are explained. Based on the principle of synchronous generator itself (covering the rotor motion equation and electromagnetic torque equation), the active-frequency control loop (including virtual inertia J and damping coefficient D) and reactive-voltage control loop (based on QV droop characteristics) of the virtual synchronous machine are established through electromechanical analogy. The active loop, reactive loop, damping and inertia and dual closed-loop parameter design are proposed. Then, a simple pre-synchronization control is proposed to realize multi-machine parallel or grid connection. Finally, virtual impedance is used to realize parallel power sharing, and its control strategy is verified by building a simulation model.

[0241] In this exemplary embodiment, a weak grid is defined as:

[0242] In power systems, the short circuit ratio (SCR) is generally used to measure the strength of the points of common connection (PCC) of the power grid, which can be expressed as:

[0243]

[0244] In power systems, the short-circuit ratio can be defined as the ratio of the short-circuit capacity of the power grid to the load capacity. According to formula (57), in traditional power generation systems, the short-circuit capacity can be considered infinite, resulting in a high SCR value. In this case, the corresponding power grid structure behaves as a strong power grid. However, at high voltage levels in the power system, the short-circuit capacity of the power system drops sharply, which leads to a weakening of the SCR performance, causing the operating condition of the power grid to change from a strong potential to a weak potential.

[0245] Table 4-1 SCR values ​​are similar to grid strength. The grid strength corresponding to the same values

[0246]

[0247] Table 4-1 provides grid stability evaluation indicators corresponding to various SCR parameters. It is worth noting that when the grid short-circuit capacity drops to zero, the SCR value will also be zero. At this time, the system is in an off-grid state, which can be regarded as an extreme case of a weak grid.

[0248] In the simulation or modeling analysis process, the single-machine infinite model is widely used to describe the operation of the SCR reactor.

[0249] In the configuration of PCC point, if only one grid-connected device is connected, the “single machine” can be represented as one grid-connected device; if the grid (V g ) has infinite capacity, it can be expressed as infinite. Because the line impedance (Z line ) exists, the actual voltage at the PCC point is not exactly equal to V g In this mode, the concept of SCR and line impedance are fully utilized. line It is simplified to pure inductive properties.

[0250] In this example embodiment, the synchronous generator works as follows:

[0251] For traditional synchronous generators, the prime mover (G) is the device that provides mechanical force and energy to the generator, such as a turbine. The operation of a synchronous generator relies on two key control mechanisms: the speed governor and the excitation regulator. When the system is subject to frequent external disturbances, the frequency will follow a pre-set amplitude variation pattern. This phenomenon is transmitted to the speed governor via a speed sensor via a converted voltage signal. The regulator uses a differential regulation mechanism to convert frequency changes into mechanical power adjustments. It then adjusts the mechanical shaft to increase the speed of the synchronous generator, thereby achieving system stability restoration. The operating state of the synchronous generator is affected by the setpoints of the reactive power controller, so the excitation regulator adjusts according to these setpoints to ensure normal operation of the synchronous generator.

[0252] The operation of synchronous generators primarily relies on two processes: voltage regulation and frequency regulation. During current regulation, the synchronous motor transmits the output voltage, output current, reactive power setpoint, and current setpoint to the reactive power controller for adjustment. During power system operation, when the output voltage fluctuates, the regulator sends the new voltage setpoint to the excitation device to ensure that the voltage remains stable. Virtual synchronous generators (VSGs), based on the working principles of synchronous generators, have the ability to participate in grid frequency and voltage regulation. Furthermore, VSGs utilize the inertia and damping characteristics of synchronous generators to suppress grid frequency and voltage fluctuations.

[0253] Frequency regulation encompasses both primary and secondary frequency regulation. Primary frequency regulation involves adjusting the generator's output power or frequency conversion equipment to maintain consistency with the grid frequency when the grid frequency fluctuates. If the frequency fluctuates beyond a preset range, the system automatically adjusts to restore the output power to a predetermined stable level. Frequency regulation is typically performed by a mechanical or electronic speed governor. By adjusting the generator's speed or output power, the generator can operate synchronously with the grid.

[0254] After completing the primary frequency regulation operation, a secondary frequency regulation operation is required to restore the system frequency to the rated value (50Hz) to ensure stable operation. The generator's frequency regulator is usually responsible for implementing the secondary frequency regulation process. The system dispatcher will send a power command to it to increase the generator's output power, thereby achieving an upward movement of the droop characteristic curve. When the power generated by the generator and the power consumed by the load reach a balance, the system frequency will return to the rated value f N =50Hz, thus achieving zero-error adjustment of the frequency.

[0255] The three phases of the stator winding of an ideal synchronous motor are located in a three-phase stationary symmetric coordinate system, 120° apart from each other, while the rotor winding is arranged in a two-phase rotating coordinate system dq, with the two axes 90° apart. While the traditional eighth-order mathematical model can fully describe the characteristics of a synchronous generator, its high complexity increases the difficulty of designing a virtual synchronous generator (VSG) controller. To simplify VSG design, the following assumptions can be made: only consider the ideal non-salient-pole synchronous generator, whose d-axis and q-axis synchronous reactances are identical, while the salient-pole synchronous generator has different d-axis and q-axis synchronous reactances due to the uneven distribution of air gap magnetic induction; ignore nonlinear effects such as magnetic saturation and eddy current losses; and assume that the number of pole pairs is 1.

[0256] Let the counterclockwise rotation direction of the rotor be the reference direction, and the stator voltage electrical equation is:

[0257]

[0258] Where R s Indicates the resistance of each phase winding of the stator, L s represents synchronous reactance, i abc Represents the three-phase stator current abc, Esinωt abc Represents the three-phase excitation electromotive force.

[0259] In a synchronous generator set, the rotor equations of the generator and prime mover are:

[0260]

[0261] In this exemplary embodiment, the virtual synchronous machine control strategy and principle include:

[0262] Active power-frequency control. From a physical perspective, a synchronous generator is a mechanical device. The method for ensuring accurate frequency regulation is to balance the mechanical output torque and simulate inertia and damping to achieve frequency modulation. However, as a power electronic device, the energy storage converter does not have a physical mechanical shaft, so the same implementation method as a synchronous generator cannot be used. When the load power changes, the system uses virtual torque to adjust the output frequency based on the power change, ultimately achieving energy balance. This process consists of two parts: a virtual speed regulator and a virtual frequency regulator. The virtual speed regulator's function is to perform primary frequency modulation, quickly intercepting the initial stage of frequency drops or rises, allowing the system to reach a new steady state.

[0263] The droop relationship between active power and spectrum is simulated in VSG control as follows:

[0264] P m =P ref +k p (ω o -ω) (60)

[0265] In equation (60), P m Indicates the actual active output power, P ref Represents rated power, k p is the ratio of active power to frequency droop coefficient, ω o and ω represent the reference angular frequency and device output angular frequency of VSG control, respectively. To improve system stability and suppress frequency vibration, the control system adds virtual inertia and damping, so that the second-order equation of the rotor mechanical motion can also be expressed as:

[0266]

[0267] In equation (61), J represents the virtual moment of inertia, D is the damping coefficient, θ is the internal potential phase angle, and T m and T e are mechanical torque and electromagnetic torque respectively, ωg Represents the grid synchronous angular velocity, usually available as ω o Substitute (61) into (60), and we can obtain the property equation of the active power-frequency loop, which is expressed as follows:

[0268]

[0269] K p =k p +D ωo Substitute into equation (62) to optimize the control system design. In traditional droop control systems, the virtual synchronous machine strategy introduces inertia and damping elements to reduce the speed of frequency changes and avoid frequency fluctuations caused by power surges. This improvement helps enhance system stability.

[0270] Reactive power-voltage control, traditional synchronous generators adjust the excitation electromotive force by adjusting the excitation system, thereby achieving control of the output reactive power and stabilization of the terminal voltage. By analog excitation controller, reactive power and voltage droop control is introduced into the VSG control strategy. This method is based on the reactive power output of the calculation device and uses the droop coefficient to convert the reactive power deviation into a voltage deviation. These deviation values ​​are then passed to the voltage outer loop. After the voltage outer loop calculation is completed, the data is transmitted to the modulation module for processing. There is an inverse relationship between reactive power and voltage change, that is, when the output reactive power of the device increases, its output voltage will decrease accordingly. The relationship between voltage and active electric power can be expressed as:

[0271] u ref -u0=-k q (Q0-Q ref ) (63)

[0272] In formula (63), u o Indicates the output voltage amplitude of the device, u ref is the rated voltage amplitude, D q is the reactive power-voltage droop coefficient, Q out and Q ref are the actual reactive output power and reactive power reference value of the device respectively. In order to simulate the control characteristics between the reactive power and voltage of the synchronous generator, the formula (63) is transformed and can be expressed as:

[0273] u ref =k q (Q ref -Q0)+u0 (64)

[0274] Converting Equation (64) into a control block diagram, the device adopts a reactive power-voltage control loop as a way to adjust the output voltage amplitude.

[0275] In this example embodiment, the overall control strategy and its parameter design include:

[0276] The mathematical model of the main circuit is based on the inductor current i a 、i b 、i c Based on the inversion direction of the current, combined with Kirchhoff's current equation, the physical expression in the three-phase stationary coordinate system can be obtained as follows:

[0277]

[0278] In formula (65):

[0279]

[0280] If the three-phase voltage is balanced and the load end carries a three-phase balanced load, it can be expressed as:

[0281]

[0282] Substituting equation (67) into equation (66), it can be expressed as:

[0283]

[0284] Therefore, formula (68) is simplified to:

[0285]

[0286] To facilitate research, some variables are defined as follows:

[0287]

[0288] Substituting equation (70) into equation (69) yields

[0289]

[0290] However, all variables in Equation (71) are in AC form and cannot be designed using a control strategy. Therefore, Equation (71) needs to be transformed from the three-phase stationary abc coordinate system to the two-phase rotating dq coordinate system. The transformation matrix is:

[0291]

[0292] The transformation matrix from the two-phase rotating dq coordinate system to the three-phase stationary abc coordinate system is

[0293]

[0294] Using formula (73), we can get

[0295]

[0296] After substituting formula (74) into formula (71), we can get

[0297]

[0298] Where,

[0299] Multiplying both sides of formula (75) by formula (72) and then dividing by formula (72) equals the corresponding result,

[0300]

[0301] In the above formula, T 3s / 2r With T 2r / 3s The result of multiplication

[0302] Equation (76) becomes the following form.

[0303]

[0304] Perform Laplace transform on Equation (77) and then expand it in the table as:

[0305]

[0306] The double closed-loop mathematical model can be obtained from formula (78).

[0307] In integrated renewable energy systems, the use of VSG control strategies can make PCS devices more adaptable to the fluctuating and unstable nature of renewable energy generation, thereby improving renewable energy efficiency and enhancing grid stability. By simulating the inertia and damping characteristics of traditional synchronous generators, VSG control systems can achieve similar operating characteristics between power electronics and synchronous generators.

[0308] Figure 17 and Figure 18The block diagram of the VSG control algorithm and the main control structure diagram of the VSG are shown respectively. The energy storage medium interacts with the grid or load through the PCS. Thanks to the PCS's adoption of a virtual synchronous machine control strategy, it can simulate the characteristics of a synchronous machine. The control system consists of an inner layer and an outer layer. The inner layer uses a voltage and current dual closed-loop technology to achieve high-precision control of the system, effectively reducing voltage and current fluctuations and improving system stability and reliability. The VSG control algorithm and power calculation module together constitute the outer layer. The VSG first converts the voltage and current collected at the PCC point into output power by the power calculation module. Then, the virtual speed regulator implements active current-frequency control, and the virtual excitation regulator implements reactive current-voltage control, while simulating inertia, damping characteristics, and electromagnetic transient characteristics. Ultimately, the required voltage amplitude and control angle reference values ​​are provided to the inner layer.

[0309] In this example embodiment, the networking mode parameter design is:

[0310] (1) Current loop control parameter design:

[0311] In the voltage control loop, the system monitors the output voltage in real time and compares this signal to a reference voltage. If a deviation occurs between the setpoint and the output voltage, the control system generates an error signal and adjusts the output signal to correct the deviation. In the current control channel, the system uses a comparison method to continuously and dynamically adjust the output current to ensure output current stability. In this disclosure, PID controllers are used in both the voltage and current loops. This design enables the establishment of a dual closed-loop control system.

[0312] The voltage outer loop reference value U generated by the VSG power loop ref It is composed of a voltage outer loop circuit. The reference value corrected by the PI controller, including the difference in capacitor voltage, forms the reference value i of the current inner loop. ref By adjusting the difference between the current and the inductor current and then correcting it through a PID controller, a modulation wave for the SPWM signal is generated. Based on the adjustment wave and the carrier, the drive signal for the IGBT switch tube is obtained.

[0313] From this, we can deduce the open-loop propagation formula of the current loop as follows:

[0314]

[0315] The switching frequency of this study is 10kHz. In order to ensure the rapid response of the current loop, according to the relationship between the following performance index and frequency domain index and parameters of the typical type I system in the automatic control principle, when the system meets k pi k pwm L -1When 1.5Ts=0.5, the optimal damping ratio is 0.707. Therefore, the current loop proportional coefficient k pi The formula is

[0316]

[0317] The value of the filter inductor is L = 3mH. According to formula (80) and related parameters, the current loop proportional coefficient k can be calculated. pi =16. By substituting this coefficient into equation (79), the open-loop Bode plot of the current loop can be plotted. It can be seen from the open-loop Bode plot that when the amplitude-frequency characteristic curve reaches 0 dB, the phase margin is 67°. At the same time, the stability margin also meets the requirements.

[0318] (2) Voltage loop control parameter design

[0319] From the voltage loop control block diagram, we can see that the voltage regulation loop belongs to a typical Type II circuit. Its open-loop transfer function also belongs to a Type II circuit and can be expressed as:

[0320]

[0321] In this study, the filter capacitor value is C = 10μF, and the system IF bandwidth parameter determined by the oscillation index method is h. Table 4-2 shows the system operating conditions corresponding to different IF bandwidths.

[0322] Table 4-2h values ​​correspond to system status information

[0323]

[0324] As shown in Table 4-2, when the intermediate frequency bandwidth of the voltage outer loop is set to 5kHz, the system shows better dynamic characteristics and stability. According to formula (81), the proportional coefficient k of the voltage loop can be obtained pu is 0.02, the integral coefficient k iu Substituting these coefficients into the open-loop transfer function of the voltage loop yields an open-loop Bode plot for the current loop. The voltage proportional-integral coefficients achieve good stability with a phase margin of approximately 45° and a control bandwidth of approximately 311 Hz.

[0325] (3) Active power-frequency loop parameter design

[0326] According to GB / T 38983.1-2020 "Virtual Synchronous Machine Part 1: General Principles", the active frequency modulation coefficient of the virtual synchronous machine (usually K f This coefficient is used to describe the rate at which active power increases with frequency, and its calculation method can be expressed as:

[0327]

[0328] In K f =20, the grid frequency changes by 0.05pu, and the system output is 1pu active power. f =50, the grid frequency changes by 0.02 pu, and the system outputs 1 pu of active power. The PF loop design should follow specific settings, that is, when the grid frequency changes within the range of 0.033 to 0.1 Hz, the PF loop is not allowed to start regulating.

[0329] According to the Electric Power Industry Standard of the People's Republic of China, "Technical Specifications for Grid-connected Operation and Control of Electrochemical Energy Storage Power Stations Part 7: Inertia Support and Damping Control", inertia (J) is defined as the parameter of the electrochemical energy storage power station simulating the rotational inertia of a traditional synchronous generator, and its unit is kilograms per square meter. At the same time, the time constant (T j ) defines the time constant of the electrochemical energy storage power station simulating the traditional synchronous generator, and its calculation formula can be expressed as:

[0330]

[0331] In this part, the rated power P n The unit of inertia is defined. Specifically, we need to know what is watt (W). The inertia time constant T of the virtual synchronous generator j It is best to set it between 3 seconds and 12 seconds to obtain the rotational inertia J of the virtual synchronous machine.

[0332] (4) Reactive power-voltage loop parameter design

[0333] According to GB / T 38983.1-2020 "Virtual Synchronous Machine Part 1: General Principles", the reactive voltage regulation coefficient of the virtual synchronous machine (usually K v This coefficient is used to indicate the speed at which reactive current changes with voltage, and its calculation formula can be expressed as:

[0334]

[0335] ΔQ represents the change in the reactive power of the virtual synchronous machine, and ΔU represents the change in the voltage of the virtual synchronous machine, that is, |UU n ∣. Among them, Q n and U n Represents the rated reactive power and rated voltage of the virtual synchronous machine, both in kilovar (kVar) and volt (V). It is generally recommended that the voltage level is K v The range is 12.5 to 33.3. v=12.5, the power supply voltage fluctuates by 0.08pu, and the system output power is 1pu reactive; v =50, the fluctuation amplitude of the grid voltage is 0.03pu, and the reactive power generated by the system is 1pu.

[0336] In this example, the VSG active loop small signal modeling is as follows:

[0337] When studying the system's active frequency characteristics, the rotor's mechanical equations of motion are generally used, but these are large-signal perturbation models. To analyze the system's dynamic characteristics under small perturbations and optimize the parameters J and D, an active small-signal model is necessary. By introducing small-signal perturbations near the static operating point, the small-signal expressions for each physical quantity in the VSG system can be expressed using linearization methods as follows:

[0338]

[0339] Substituting equations (57) and (58) with equation (85) and removing the high-order nonlinear components, we can express it as:

[0340]

[0341] Formula (86) can be expressed as follows through Laplace transform:

[0342]

[0343] According to formula (87), the model with small active power signal can analyze the active voltage closed-loop transmission relationship as follows:

[0344]

[0345] From formula (88), we can see that the active closed-loop transfer function is a typical second-order system and can be expressed as:

[0346]

[0347] According to formula (89), the attenuation frequency ω b There is a relationship between ζ and inertia J, indicating that the overshoot of active power is determined by J, while the damping ratio ζ depends on J and D. Since the damping D has a greater impact on the system damping ratio, this disclosure mainly focuses on the design of J in the case of active response.

[0348] The transfer function of the frequency loop can be expressed as:

[0349]

[0350] Since the frequency loop is a second-order system, it can be expressed as:

[0351]

[0352] According to equations (89) and (91), the response speed of the active closed-loop transfer function is faster than that of the frequency loop transfer function. The frequency loop is designed as an overdamped system to meet the capacity requirements of the PCS. When the frequency variation range is ±0.5Hz, K f The value is 120, ensuring that the active power output is no less than 10%. According to the national standard "Virtual Synchronous Machine Technology - General Principles," the active closed-loop design must meet the following requirements: frequency modulation response time no more than 500 milliseconds, adjustment time no more than 1 second, and power error controlled within 2%. Therefore, the active closed-loop design is an underdamped system, which can achieve fast response while complying with national standards. The calculation formula is:

[0353]

[0354] After writing (89) into formula (92), it becomes:

[0355]

[0356] The damping factor of underdamping is preferably 0.4<ζ<0.8, which can be expressed as follows in combination with formula (89):

[0357]

[0358] According to the theory that the optimal damping ratio of the second-order system is ζ=0.707, a system is designed and J is selected as 3.6.

[0359] In this exemplary embodiment, the networking mode parallel control strategy includes:

[0360] Although VSG control systems can effectively achieve balanced power distribution between multiple parallel systems and synchronous generators, and simulate the inertia and damping properties of synchronous generators, thereby reducing fluctuations in output power and frequency, in actual operation, when multiple PCSs are connected in parallel, they may experience parameter inconsistencies and the resistive-inductive characteristics of the AC line impedance. Large differences in voltage amplitude and phase between paralleled devices can lead to parallel failures. In large-scale energy storage systems, a parallel connection failure could potentially lead to the collapse of the entire system, resulting in significant economic losses. Therefore, ensuring a stable and reliable multi-machine VSG parallel control strategy is crucial.

[0361] In the embodiment of this example, a parallel pre-synchronization control strategy is studied. The output phase of each PCS device is regulated by various active power and frequency control loops. However, in actual operation, due to the inability to meet the synchronous startup requirements of all devices, there is a large phase deviation between different PCS devices in the off-grid operation mode. When operating in parallel, due to the large phase difference, a large inrush current will be generated between the output end of the device and the grid side, which may cause the PCS system to stop operating due to overload. If the inrush current is severe and the PCS hardware protection response fails to respond in time, it may cause damage to the switching elements. Therefore, it is crucial to adjust the voltage and frequency on both sides of the grid connection point to ensure synchronization. Before the PCS is operated in parallel, a pre-synchronization operation is required to ensure smooth connection and coordinated operation of the entire system.

[0362] The grid-connected pre-synchronization of the grid-connected distribution control system can be regarded as a synchronous pre-synchronization process between the current source and the grid side. This process obtains the phase and frequency data of the grid side through the phase-locked loop and transmits it to the control system, so that the PCS can track the grid phase and achieve consistency of the output phase with the grid side. In a grid-connected PCS system, when multiple machines are connected in parallel, the communication carrier synchronization method is usually adopted to ensure the consistency of the output voltage amplitude, frequency and phase of multiple PCSs. In the off-grid operation mode, the parallel operation of multiple VSGs requires starting one device first to build the bus voltage. The remaining devices use the amplitude and phase of the bus voltage as a reference, and perform output compensation through the active-frequency control loop and the reactive-voltage control loop. When the parallel conditions are met, the grid-side relay of the parallel device is closed to complete the pre-synchronization process. Therefore, this disclosure will focus on the phase and voltage pre-synchronization processing, taking PCS1 as the reference grid, and thus performing phase and voltage pre-synchronization adjustments on PCS2. In Figure 19 In , the control flow graph is observed.

[0363] Figure 19 Two pre-synchronization methods are presented, namely phase pre-synchronization and voltage pre-synchronization. The specific method is that two PCSs obtain the output voltage angle frequency ω through a phase-locked loop. PCS1 and ω PCS2 Then, the PID controller is used to compensate Δω in the output frequency of PCS2 so that the angular frequency and phase of the two PCSs tend to be consistent, and then the phase pre-synchronization switch is turned off.

[0364] Voltage pre-synchronization is achieved through the QU loop of PCS2. The voltage values ​​U of the two PCSs are obtained through third-order transformation and voltage amplitude calculation. m1 and U m2 PCS1 is used as the reference grid, U m1 Used as a reference voltage. When the voltage difference between the two PCSs is small, switch to the reference voltage U of PCS2. ref, complete voltage pre-synchronization.

[0365] However, the above-mentioned pre-synchronization control not only introduces an additional control loop, but also requires switching the pre-synchronization loop to synchronize the phases of PCS1 and PCS2. In digital control, this causes the control loops of PCS1 and PCS2 to have different timings when the loops are switched at the moment of parallel connection, resulting in phase parameter transmission being out of sync, causing overcurrent and thus parallel connection failure.

[0366] The following proposes a simple pre-synchronization control, which can complete the pre-synchronization control without introducing an additional control loop or switching the pre-synchronization loop. The simple pre-synchronization is divided into three parts: preparation stage, phase self-synchronization stage and voltage amplitude judgment stage. Figure 20 As shown:

[0367] (1) Phase 1: Preparation phase, PCS1 and PCS2 establish voltage and phase information. Taking the amplitude of 311V and the frequency of 50hz as an example, ensure that the output voltage amplitude is 311V and the frequency is 50hz

[0368] (2) Phase 2: Phase synchronization phase. PCS2 modulates the frequency through the active loop to reduce the frequency. At this time, the phase of PCS2 is shifted, completely overlapping with PCS1 and then separating, and the cycle repeats.

[0369] (3) Phase 3: Voltage amplitude determination phase. Information is determined by determining the voltage difference between PCS1 and PCS2. Due to the characteristics of AC power, the voltage difference can only be less than 1V when the phases are almost aligned. When the voltage difference is less than 1V, active frequency modulation of PCS2 is restored. At this point, the phase information and voltage difference of the two PCSs are consistent, and the parallel switch is opened for parallel connection.

[0370] When adding three or more units, simply repeat the steps. This pre-synchronization control saves additional control loops in off-grid parallel situations and reduces the problem of parallel failure caused by digital controller switching loops. This effectively saves controller resources and simplifies control complexity.

[0371] In the embodiment of this example, the parallel virtual impedance control strategy is studied. The parallel topology structure expanded from the single topology can be obtained from Figure 21 In this system design, the AC connection between the two multi-purpose serial computers is connected through the line impedance and they jointly provide power to the load.

[0372] according to Figure 21 As shown in the figure, the two PCSs adopt an independent control method, and the structure is relatively balanced. S is a relay, and its output circuit is a three-phase AC bus. In actual operation, the PCS has line impedance, which is represented by Z in the main circuit topology. line .

[0373] Traditional VSG control methods combine the output phase of the Pf loop and the reference voltage output of the QU loop into parameters in a three-phase coordinate system. These parameters are then transformed using a dq transformation to obtain a reference value, which is then transferred to the voltage and current dual closed loop. However, if the voltage drop across the synchronous generator's stator impedance is not considered, the model accuracy is poor and the control effect is poor. Therefore, a virtual stator impedance must be introduced for correction.

[0374] Virtual stator impedance is used to simulate the stator impedance voltage drop, which is achieved by adding a virtual impedance loop in the control loop. The controller uses the virtual stator impedance Z v Calculate the voltage drop across the current. Figure 22 Shown is the control block diagram when the virtual stator impedance is not included.

[0375] exist Figure 22 In the equation, Kpwm represents the three-phase gain, while G(s) and G i The gain values ​​of (s) corresponding to the voltage loop and current loop are:

[0376]

[0377] Using formula (95), the forward path gain G(s) of the current-voltage dual closed loop can be obtained as the forward path gain can be used for dual closed loop operation between a single current, voltage and DC power supply.

[0378]

[0379] Without adding a virtual impedance adjuster, the output impedance Z0(s) of the filter is expressed as:

[0380]

[0381] After adding the virtual stator impedance, it can be expressed as:

[0382]

[0383] After combining equation (98), it can be expressed as:

[0384]

[0385] From formula (97), we can get the equivalent current after adding the virtual impedance, and the impedance value is expressed as:

[0386] Z v0 (s)=G(s)Z v (s)+Z0(s) (100)

[0387] According to formula (100), after the introduction of virtual stator impedance, the equivalent output impedance of the PCS is composed of the sum of the filter output impedance and the virtual impedance. The following study is the calculation model of the virtual impedance, assuming that the three-phase virtual impedance circuit is represented as:

[0388]

[0389] Therefore, the algorithms used are all based on the dq coordinate system. Therefore, the virtual impedance modeling is also related to the dq coordinate system, which can be expressed as:

[0390]

[0391] Based on the above analysis, the overall control diagram of the virtual stator impedance can be obtained, as shown in the following figure: Figure 23 shown.

[0392] This paper optimizes the grid-connected control model, employing virtual synchronous machine control. First, the grid is applied to a weak network environment, introducing the definition of a weak network. Because virtual synchronous machines control multiple active and reactive loops, the control equations for these machines are derived based on synchronous machine principles. The design parameters for the active and reactive loops, damping and inertia, and dual closed-loop parameters are proposed. A simplified presynchronization control method is then proposed to achieve multi-machine parallel or grid connection. Finally, virtual impedance is used to achieve parallel power sharing.

[0393] It should be noted that although the steps of the method disclosed herein are depicted in a particular order in the accompanying drawings, this does not require or imply that the steps must be performed in that particular order, or that all steps must be performed to achieve the desired result. Additionally or alternatively, certain steps may be omitted, multiple steps may be combined into one, and / or one step may be decomposed into multiple steps.

[0394] In addition, in this exemplary embodiment, a multi-mode control device for an energy storage converter is also provided. Figure 24 As shown, the energy storage converter multi-mode control device 200 may include: a grid-following strategy construction module 210, an off-grid strategy construction module 220, a grid-building strategy construction module 230 and a multi-mode control module 240.

[0395] A grid-following strategy building module 210 is used to establish a low voltage ride-through control strategy based on repetitive control by creating a second-order infinite impulse response (IIR) filter when the energy storage converter operates in the grid-following mode;

[0396] An off-grid strategy building module 220 is used to establish a preset proportional multi-resonance control strategy for the energy storage converter operating in an off-grid mode by using a three-phase four-leg topology to provide a zero-sequence path;

[0397] A network strategy building module 230 is used to establish a parallel pre-synchronization control strategy for the energy storage converter operating in the network building mode;

[0398] The multi-mode control module 240 is used to realize multi-mode control of the energy storage converter based on the low voltage ride-through control strategy that complies with repeated control in the grid-following mode, the preset proportional multi-resonance control strategy in the off-grid mode, and the parallel pre-synchronization control strategy in the grid-forming mode.

[0399] The specific details of each of the above-mentioned energy storage converter multi-mode control device modules have been described in detail in the corresponding energy storage converter multi-mode control method, and thus will not be repeated here.

[0400] It should be understood that the present disclosure is not limited to the exact structures that have been described above and shown in the drawings, and that various modifications and changes can be made without departing from the scope thereof. The scope of the present disclosure is limited only by the appended claims.

Claims

1. A multi-mode control method for an energy storage converter, characterized in that: The method comprises: For the energy storage converter operating in grid-following mode, a low voltage ride-through control strategy based on repetitive control is established by creating a second-order infinite impulse response (IIR) filter. For the energy storage converter operating in off-grid mode, a three-phase four-leg topology is used to provide a zero-sequence path and establish a preset proportional multi-resonance control strategy. A parallel pre-synchronization control strategy is established for the energy storage converter operating in grid-forming mode; Multi-mode control of the energy storage converter is achieved based on a low voltage ride-through control strategy that complies with repetitive control in the grid-following mode, a preset proportional multi-resonance control strategy in the off-grid mode, and a parallel pre-synchronization control strategy in the grid-forming mode.

2. The method according to claim 1, wherein The method further comprises: In the dq synchronous rotating coordinate system, the mathematical modeling of the energy storage converter and the control model in the grid-following mode are established; A second-order infinite impulse response (IIR) filter is created, and a repetitive controller is designed based on the second-order infinite impulse response (IIR) filter.

3. The method according to claim 2, wherein The method further comprises: Design and calculate the auxiliary link Q(z) for adjusting system performance, the compensation link S(z) for realizing amplitude and frequency characteristic correction, the feedforward coefficient k for compensating system phase lag, and the gain coefficient k for adjusting system response amplitude. r , to achieve repetitive controller design.

4. The method according to claim 3, wherein The method further comprises: A phase lag compensation coefficient is designed and calculated, and based on the phase lag compensation coefficient and the repetitive controller, a low voltage ride-through control strategy based on repetitive control is established.

5. The method according to claim 1, wherein The method further comprises: Modeling the three-phase four-bridge-leg topology energy storage converter in a rotating coordinate system to generate a mathematical model of the three-phase four-bridge-leg topology energy storage converter; Based on the mathematical model of the energy storage converter, a mathematical model of the energy storage converter under unbalanced load and nonlinear load is established; Based on the mathematical model of the energy storage converter under unbalanced load and nonlinear load, a control strategy for the energy storage converter in off-grid mode is established.

6. The method according to claim 5, wherein The method further comprises: In the control system of the energy storage converter in the off-grid mode, a preset number of quasi-PR controllers are connected in parallel to establish a preset proportional multi-resonance control strategy for the energy storage converter in the off-grid mode.

7. The method according to claim 1, wherein The method further comprises: Establish a mathematical model of the energy storage converter in grid-connected mode; Based on the mathematical model of the energy storage converter in the grid-forming mode, current loop control parameters, voltage loop control parameters, active power-frequency loop parameters, and reactive power-voltage loop parameters are designed and calculated respectively; A small signal model of the active loop of the energy storage converter is established.

8. The method according to claim 7, wherein The method further comprises: Based on the virtual stator impedance, the parallel parameters of the energy storage converter in the grid-forming mode are corrected, and a parallel pre-synchronization control strategy is generated to achieve parallel pre-synchronization of the energy storage converter in the grid-forming mode.

9. A multi-mode control device for an energy storage converter, characterized in that: The device comprises: The grid-following strategy building module is used to establish a low voltage ride-through control strategy based on repetitive control by creating a second-order infinite impulse response (IIR) filter when the energy storage converter operates in grid-following mode. An off-grid strategy building module is used to establish a preset proportional multi-resonance control strategy for the energy storage converter operating in off-grid mode, using a three-phase four-leg topology to provide a zero-sequence path; A network strategy building module is used to establish a parallel pre-synchronization control strategy for the energy storage converter operating in the network mode; A multi-mode control module is used to realize multi-mode control of the energy storage converter based on a low voltage ride-through control strategy that complies with repeated control in the grid-following mode, a preset proportional multi-resonance control strategy in the off-grid mode, and a parallel pre-synchronization control strategy in the grid-forming mode.

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