Phase-out angle determination method and system, variable frequency phase-out control method and chip

Abstract

The application relates to a phase shift angle determination method and system, a variable frequency phase shift control method and a chip. The method comprises the following steps: determining a first current value of a resonant inductor at a first time and a second current value of the resonant inductor at a second time based on a general solution of a time domain differential equation of a resonant inductor current and a resonant capacitor voltage of a DBSRC equivalent circuit and symmetry of the DBSRC equivalent circuit; obtaining a first relationship of an external phase shift angle with respect to an input voltage, an output voltage and a switching frequency of a switching tube of the DBSRC based on the first current value when a primary bridge circuit meets ZVS soft switching requirements; obtaining a second relationship of the external phase shift angle with respect to the input voltage, the output voltage and the switching frequency based on the second current value when a secondary bridge circuit meets ZVS soft switching requirements; and determining an expression of the external phase shift angle with respect to the switching frequency, a steady voltage value and the input voltage according to the first relationship, the second relationship and the steady voltage value of an output end of the DBSRC. The scheme is simple, accurate and highly transplantable.

Description

Technical Field

[0001] This application relates to the field of dual-bridge series resonant converter circuit technology, and in particular to an external phase angle determination method and system, a frequency conversion phase shift control method and chip. Background Technology

[0002] The Double Bridge Series Resonant Converter (DBSRC) circuit involves controlling the outward phase angle in order to optimize output gain and efficiency.

[0003] In related technologies, DBSRC efficiency optimization methods mainly include circulating current power optimization, peak current optimization, or effective current optimization. However, the calculation of these optimization methods is usually quite complex. Generally, it is necessary to use a computer with the help of multiple mathematical calculation software to solve the problem and obtain the external phase shift angle. In addition, it is necessary to solve the problem with multiple mathematical calculation software under the condition that the inductance and other parameters of the circuit are known. Therefore, when the circuit parameters are changed, it is necessary to recalculate, and the method has low portability.

[0004] Furthermore, in related technologies, the derivation of the external phase shift angle is performed by approximating the fundamental wave and conducting frequency domain analysis. When the circuit switching frequency fs differs significantly from the circuit resonant frequency fr, the influence of higher harmonics is ignored in the fundamental wave equivalence, which leads to a decrease in the calculation accuracy of voltage and current, thereby affecting the control effect. Summary of the Invention

[0005] Therefore, it is necessary to provide a simple, accurate, and highly portable method and system for determining the external phase angle, a frequency conversion phase shift control method, and a computer chip, which can be applied to DBSRC to achieve output gain and efficiency optimization.

[0006] In a first aspect, embodiments of this application provide a method for determining the external phase shift angle, applied to a dual-bridge series resonant converter circuit, the method comprising:

[0007] Based on the general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit of the dual-bridge series resonant converter circuit, the general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit, the first current value of the resonant inductor at the first moment and the second current value of the resonant inductor at the second moment are determined.

[0008] Based on the first current value when the primary-side bridge circuit meets the ZVS soft-switching requirements, a first relationship is obtained between the external phase angle and the input voltage, output voltage, and switching frequency of the switching transistor in the dual-bridge series resonant converter circuit.

[0009] Based on the second current value when the secondary bridge circuit meets the ZVS soft switching requirement, a second relationship between the external phase shift angle and the input voltage, the output voltage, and the switching frequency is obtained;

[0010] Based on the first relation, the second relation, and the steady-state voltage value at the output of the dual-bridge series resonant converter circuit, determine the expression for the external phase shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage;

[0011] Wherein, the first moment is the moment when the first upper switch of the first left bridge arm of the primary bridge circuit starts to turn on, and the second moment is the moment when the second upper switch of the second left bridge arm of the secondary bridge circuit starts to turn on during the conduction of the first upper switch.

[0012] Based on the same inventive concept, in a second aspect, embodiments of this application provide a frequency conversion phase shift control method applied to a dual-bridge series resonant converter circuit, the control method comprising:

[0013] The output of the dual-bridge series resonant converter circuit is sampled to obtain the actual voltage value of the output at the current moment.

[0014] Based on the actual voltage value and the PI control principle, the first switching frequency of the switching transistor when the voltage at the output terminal reaches the steady-state voltage value is obtained;

[0015] Based on the first switching frequency and the preset maximum switching frequency, the required switching frequency of the dual-bridge series resonant converter circuit in the frequency conversion phase shift control process is obtained.

[0016] Based on the required switching frequency, the input voltage at the input terminal of the dual-bridge series resonant converter circuit at the current moment, the steady-state voltage value at the output terminal, and the expression based on the method for determining the external phase shift angle described in the first aspect above, the required external phase shift angle in the frequency conversion phase shift control process at the current moment is obtained.

[0017] Based on the preset duty cycle of the switching transistor, the required switching frequency, and the required outward phase shift angle, the PWM drive signal of the switching transistor of the primary bridge circuit and the PWM drive signal of the switching transistor of the secondary bridge circuit are obtained.

[0018] Based on the same inventive concept, in a third aspect, embodiments of this application provide an external phase angle determination system, comprising:

[0019] A dual-bridge series resonant converter circuit includes a primary-side bridge circuit, a resonant network, and a secondary-side bridge circuit. The input terminal of the primary-side bridge circuit is connected to an input power supply that provides the input voltage. The input terminal of the resonant network is connected to the output terminal of the primary-side bridge circuit. The input terminal of the secondary-side bridge circuit is connected to the output terminal of the resonant network. The output terminal of the secondary-side bridge circuit provides an output voltage to the load. The resonant network includes a transformer, a resonant inductor, and a resonant capacitor.

[0020] The execution unit is configured to acquire the input voltage and the output voltage of the output terminal of the secondary bridge circuit, and to execute the steps of the method described in the first aspect above when the output voltage reaches a set steady-state voltage value.

[0021] Based on the same inventive concept, in a fourth aspect, embodiments of this application provide a computer chip, including a memory and a processor, wherein the memory stores a computer program, and the processor executes the computer program to implement the steps of the method described in the first and second aspects above.

[0022] The aforementioned method and system for determining the external phase shift angle, the frequency conversion phase shift control method, and the computer chip firstly determine the first current value of the resonant inductor at the first moment and the second current value of the resonant inductor at the second moment based on the general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit, the general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit. Then, based on the first current value of the primary-side bridge circuit when meeting the ZVS soft-switching requirement, a first relationship between the external phase shift angle and the input voltage, output voltage, and switching frequency of the DBSRC is obtained. Furthermore, based on the second current value of the secondary-side bridge circuit when meeting the ZVS soft-switching requirement, a second relationship between the external phase shift angle and the input voltage, output voltage, and switching frequency is obtained. Finally, based on the first and second relationships and the steady-state voltage value at the output of the DBSRC, an expression for the external phase shift angle with respect to the switching frequency, steady-state voltage value, and input voltage is determined.

[0023] Therefore, compared to the fundamental equivalent and frequency domain analysis in related technologies, the embodiments of this application use the time domain method for formula derivation, which can ensure high accuracy of optimization. Compared to the DBSRC efficiency optimization methods in related technologies, the embodiments of this application obtain a physical measurement expression for the outer phase shift angle, which can adapt to DBSRC with different parameters. For DBSRC circuits with different circuit parameters, the outer phase shift angle can be directly obtained from the known input voltage, required output voltage, and switching frequency of the current circuit. The calculation of the outer phase shift angle is relatively simple and does not require solving with mathematical software such as MATLAB or MathCAD as in the prior art. It has high portability in DBSRC with different parameters and can be applied to single-chip microcomputers such as DSP (Digital Signal Processor) and ARM (Advanced RISC Machines) to perform calculations in each control cycle, ensuring the real-time performance of circuit control. Furthermore, the embodiments of this application use critical ZVS for optimization, and ensure the soft switching conditions of the circuit through real-time calculation, thereby improving the efficiency of the circuit while reducing the amount of calculation. Attached Figure Description

[0024] To more clearly illustrate the technical solutions in the embodiments of this application or the conventional technology, the drawings used in the description of the embodiments or the conventional technology will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0025] Figure 1A This is a schematic diagram of one embodiment of a dual-bridge series resonant converter circuit;

[0026] Figure 1B This is a second schematic diagram of the topology of a dual-bridge series resonant converter circuit according to an embodiment;

[0027] Figure 1C for Figure 1A DBSRC equivalent circuit diagram;

[0028] Figure 2 for Figure 1A The waveforms of some driving signals, resonant inductor current, and resonant capacitor voltage in the dual-bridge series resonant converter circuit are shown.

[0029] Figure 3 This is a flowchart illustrating an embodiment of a method for determining the outer phase angle;

[0030] Figure 4 This is a flowchart illustrating step S301 in an embodiment of the method for determining the outward phase angle;

[0031] Figure 5 This is one of the flowcharts illustrating step S302 in the method for determining the outward phase angle according to an embodiment;

[0032] Figure 6 This is a second flowchart illustrating step S302 in an embodiment of the method for determining the outward phase angle.

[0033] Figure 7 This is a flowchart illustrating a frequency conversion phase shift control method according to one embodiment;

[0034] Figure 8 This is a control principle diagram involving a frequency conversion phase shift control method according to one embodiment;

[0035] Figure 9 This is a schematic diagram of the circuit operation state obtained through simulation based on the external phase angle determination method in one embodiment;

[0036] Figure 10 This is a schematic diagram of the output gain obtained by simulation based on the external phase angle determination method in one embodiment. Detailed Implementation

[0037] To facilitate understanding of this application, a more complete description will be provided below with reference to the accompanying drawings, which illustrate embodiments of the present application. However, the present application can be implemented in many different forms and is not limited to the embodiments described herein. Rather, these embodiments are provided so that the disclosure of this application will be thorough and complete.

[0038] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs. The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the application.

[0039] It is understood that the terms "first," "second," etc., used in this application may be used to describe various elements herein, but these elements are not limited by these terms. These terms are only used to distinguish one element from another. For example, without departing from the scope of this application, a first resistor may be referred to as a second resistor, and similarly, a second resistor may be referred to as a first resistor. Both the first resistor and the second resistor are resistors, but they are not the same resistor. For example, in embodiments of this application, a first current value may be referred to as a second current value, and similarly, a second current value may be referred to as a first current value. Both the first current value and the second current value are current values, but they are not the same current value.

[0040] It is understood that the term "connection" in the following embodiments should be understood as "electrical connection," "communication connection," etc., if the connected circuits, modules, units, etc., have electrical signal or data transmission with each other.

[0041] It is understandable that "at least one" refers to one or more, and "multiple" refers to two or more. "At least a part of an element" refers to part or all of an element.

[0042] When used herein, the singular forms of “a,” “an,” and “the” may also include the plural forms unless the context clearly indicates otherwise. It should also be understood that the terms “comprising / including” or “having,” etc., specify the presence of the stated features, wholes, steps, operations, components, parts, or combinations thereof, but do not preclude the possibility of the presence or addition of one or more other features, wholes, steps, operations, components, parts, or combinations thereof. Meanwhile, the term “and / or” as used in this specification includes any and all combinations of the associated listed items.

[0043] Reference Figure 1A This is an exemplary topology of a dual-bridge series resonant converter circuit provided in this application. Figure 1B It is a resonant push-pull voltage multiplier topology. Figure 1A Taking this as an example, the dual-bridge series resonant converter circuit includes a primary-side bridge circuit, a secondary-side bridge circuit, a resonant network (including a transformer, a resonant inductor Ls, and a resonant capacitor Cs), an output capacitor, and an output resistor Ro. The input terminal of the primary-side bridge circuit is connected to the input power supply providing the input voltage, and the input terminal of the resonant network is connected to the output terminal of the primary-side bridge circuit. In some exemplary embodiments, the primary winding of the transformer is connected to the output terminal of the primary-side bridge circuit, and the input terminal of the secondary-side bridge circuit is connected to the output terminal of the resonant network. The output terminal of the secondary-side bridge circuit provides the output voltage to the load. In some exemplary embodiments, the two ends of the secondary winding of the transformer are connected to the input terminal of the secondary-side bridge circuit through the resonant inductor Ls and the resonant capacitor Cs, respectively. Of course, the connection method of the resonant inductor Ls and the resonant capacitor Cs is not limited to this, and this application does not specifically limit the connection method of the resonant inductor Ls and the resonant capacitor Cs.

[0044] In some exemplary embodiments, the primary-side bridge circuit includes a first upper switch Q1, a first lower switch Q2, a third upper switch Q4, and a third lower switch Q3. The first upper switch Q1 and the first lower switch Q2 are connected in series to form the first left arm of the primary-side bridge circuit, and the third upper switch Q4 and the third lower switch Q3 are connected in series to form the first right arm of the primary-side bridge circuit. The first left arm and the first right arm are connected in parallel. The two ends of the primary winding of the transformer are respectively connected to the midpoints of the first left arm and the first right arm. The midpoint of the first left arm is the connection point of the first upper switch Q1 and the first lower switch Q2, and the midpoint of the first right arm is the connection point of the third upper switch Q4 and the third lower switch Q3.

[0045] In some exemplary embodiments, the secondary bridge circuit includes a second upper switch Q5, a second lower switch Q6, a fourth upper switch Q8, and a fourth lower switch Q7. The second upper switch Q5 and the second lower switch Q6 are connected in series to form the second left arm of the secondary bridge circuit, and the fourth upper switch Q8 and the fourth lower switch Q7 are connected in series to form the second right arm of the secondary bridge circuit. The second left arm and the second right arm are connected in parallel. The two ends of the secondary winding of the transformer are respectively connected to the midpoints of the second left arm and the second right arm. The midpoint of the second left arm is the connection point of the second upper switch Q5 and the second lower switch Q6, and the midpoint of the first right arm is the connection point of the fourth upper switch Q8 and the fourth lower switch Q7.

[0046] Figure 1C for Figure 1A The DBSRC equivalent circuit of the dual-bridge series resonant converter circuit is shown below. The DBSRC equivalent circuit is an RC circuit. (Refer to the following...) Figure 1A and Figure 2 The driving signals and circuit parameters of each switch are explained below. Vi represents the input voltage at the input terminal of the dual-bridge series resonant converter circuit, and Vo represents the output voltage at the output terminal of the dual-bridge series resonant converter circuit. The output capacitor and output resistor Ro are connected in parallel to the output terminal of the dual-bridge series resonant converter circuit. The capacitor connected in parallel with each switch is the parasitic capacitance Coss of the switch. In the primary-side bridge circuit, the driving signals of the control terminals of the first upper switch Q1 and the third lower switch Q3 are the same and are both denoted as the first PWM driving signal g1. The waveform of the first PWM driving signal g1 is shown below. Figure 2 The first waveform diagram is shown. The control terminal drive signals of the first lower switch Q2 and the third upper switch Q4 are the same and are both the second PWM drive signal g2. The first PWM drive signal g1 and the second PWM drive signal g2 drive each other complementaryly.

[0047] In the secondary bridge circuit, the drive signals at the control terminals of the second upper switch Q5 and the fourth lower switch Q7 are the same and are both denoted as the third PWM drive signal g3. The waveform of the third PWM drive signal g3 is shown below. Figure 2 The second waveform diagram is shown; and the outward phase shift angle between the first PWM drive signal g1 and the third PWM drive signal g3 is θ. The drive signals of the control terminals of the second lower switch Q6 and the fourth upper switch Q8 are the same and are both denoted as the fourth PWM drive signal g4. The third PWM drive signal g3 and the fourth PWM drive signal g4 are complementary. In the embodiments of this application, after ignoring the dead time, the duty cycle of each PWM drive signal of each switch in the primary bridge circuit and the secondary bridge circuit is 50%, and the inward phase shift angle of both the primary bridge circuit and the secondary bridge circuit is 0; the switching frequency of each switch in the primary bridge circuit and the secondary bridge circuit is fs, and the switching frequency fs is a variable parameter.

[0048] In one exemplary embodiment, refer to Figure 1C and Figure 3 This paper provides a method for determining the outward phase angle, which is applied, for example but not limited to, to... Figure 1A and Figure 1B An exemplary dual-bridge series resonant converter circuit, the method may include the following steps S301 to S304.

[0049] S301, based on the general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit of the dual-bridge series resonant converter circuit, the general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit, determine the first current value of the resonant inductor at the first moment and the second current value of the resonant inductor at the second moment; wherein, the first moment is the moment when the first upper switch of the first left bridge arm of the primary bridge circuit starts to turn on, and the second moment is the moment when the second upper switch of the second left bridge arm of the secondary bridge circuit starts to turn on during the conduction of the first upper switch.

[0050] like Figure 2 In the diagram, the first time point t0 is the moment when the first upper-side switch Q1 begins to conduct. The second time point t1 is the moment when the second upper-side switch Q5 begins to conduct during the conduction of the first upper-side switch Q1. The third time point t2 is the moment when the first upper-side switch Q1 begins to turn off during the conduction of the second upper-side switch Q5. The first time point t0 and the second time point t1 are the start and end times of the first time period (t0~t1), respectively, and the start and end times of the second time period (t1~t2) are the second time point t1 and the third time point t2, respectively.

[0051] And, iL0 is: the first current value of the resonant inductor Ls at the first time t0. iL1 is: the second current value of the resonant inductor Ls at the second time t1. iL2 is: the current value of the resonant inductor Ls at the third time t2. Uc0 is: the first voltage value of the resonant capacitor Cs at the first time t0. Uc1 is: the second voltage value of the resonant capacitor Cs at the second time t1. Uc2 is: the voltage value of the resonant capacitor Cs at the third time t2.

[0052] S302, based on the first current value of the primary-side bridge circuit when meeting the ZVS soft-switching requirements, obtains the first relationship between the external phase angle and the input voltage, output voltage, and switching frequency of the dual-bridge series resonant converter circuit.

[0053] S303, based on the second current value of the secondary bridge circuit when meeting the ZVS soft switching requirements, obtains the second relationship between the external phase angle and the input voltage, output voltage, and switching frequency.

[0054] S304. Based on the first relation, the second relation, and the steady-state voltage value at the output of the dual-bridge series resonant converter circuit, determine the expression for the external phase shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage.

[0055] In this embodiment, the ZVS soft switch is specifically a zero voltage switch (ZVS).

[0056] Compared to the fundamental equivalent and frequency domain analysis in related technologies, the above-described method for determining the outer phase shift angle uses a time-domain method for formula derivation, ensuring high accuracy in optimization. Compared to DBSRC efficiency optimization methods in related technologies, this application provides a physical expression for the outer phase shift angle, adaptable to DBSRCs with different parameters. For DBSRC circuits with different circuit parameters, the outer phase shift angle can be directly obtained using the known input voltage, required output voltage, and switching frequency of the current circuit. The calculation of the outer phase shift angle is relatively simple and does not require solving using mathematical software such as MATLAB or MathCAD as in existing technologies. It has high portability in DBSRCs with different parameters and can be applied to DSP, ARM, and other microcontroller chips for calculation in each control cycle, ensuring the real-time performance of circuit control. Furthermore, this application uses critical ZVS for optimization, ensuring the soft-switching conditions of the circuit through real-time calculation, reducing computational load while improving circuit efficiency.

[0057] In one exemplary embodiment, in conjunction with reference to Figure 2 and Figure 4 The above step S301 may include the following steps S401 to S404.

[0058] S401, Obtain the general solution: Obtain the first general solution iL1(t) of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit in the first time period (t0~t1) and the second general solution iL2(t) in the second time period (t1~t2). Obtain the third general solution Uc1(t) of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit in the first time period (t0~t1) and the fourth general solution Uc2(t) in the second time period (t1~t2).

[0059] In step S401, the first general solution iL1(t) and the third general solution Uc1(t) are both linear functions with the first current value iL0 and the first voltage value Uc0 of the resonant capacitor voltage at the first time t0 as first-type coefficients. The second general solution iL2(t) and the fourth general solution Uc2(t) are both linear functions with the second current value iL1 and the second voltage value Uc1 of the resonant capacitor voltage at the second time t1 as second-type coefficients.

[0060] It is also worth noting that the first general solution iL1(t) and the second general solution iL2(t) are both inductor current functions of the resonant inductor Ls at different times with respect to the input voltage Vi, the output voltage Vo, and the outward phase shift angle θ. The third general solution Uc1(t) and the fourth general solution Uc2(t) are both capacitor voltage functions of the resonant capacitor Cs at different times with respect to the input voltage Vi, the output voltage Vo, and the outward phase shift angle θ.

[0061] Reference Figure 2 , Figure 2 The third waveform in the diagram is the waveform of the resonant inductor current I(Ls) with respect to time t. Figure 2 The fourth waveform is the waveform of the resonant capacitor voltage V(Cs) with respect to time t. Specifically, the first general solution iL1(t) and the second general solution iL2(t) correspond to... Figure 2 The third waveform diagram shows the resonant inductor current with respect to time t in the first time period (t0~t1) and the second time period (t1~t2). The third general solution Uc1(t) and the fourth general solution Uc2(t) correspond to... Figure 2 The fourth waveform in the figure shows the resonant inductor voltage with respect to time t during the first time period (t0~t1) and the second time period (t1~t2).

[0062] In step S401, based on the DBSRC equivalent circuit, the first general solution iL1(t), the second general solution iL2(t), the third general solution Uc1(t), and the fourth general solution Uc2(t) can be directly obtained according to the principle of a second-order series RLC circuit. The specific first general solution iL1(t), second general solution iL2(t), third general solution Uc1(t), and fourth general solution Uc2(t) are as follows:

[0063] The first general solution iL1(t) is:

[0064]

[0065] The second general solution iL2(t) is:

[0066]

[0067] The third general solution Uc1(t) is:

[0068]

[0069] The fourth general solution Uc2(t) is:

[0070]

[0071] In the above general solutions, Vi is the input voltage, Ls represents the inductance of the resonant inductor, Cs represents the capacitance of the resonant capacitor, Ntr is the turns ratio of the transformer in the dual-bridge series resonant converter circuit, Vo is the output voltage, cos is the cosine function, and sin is the sine function.

[0072] S402, Obtain the current and voltage values ​​of the general solution: Obtain the first general solution current value of the first general solution iL1(t) at the first time t0, the second general solution current value of the first general solution iL1(t) at the second time t1, the third general solution current value of the second general solution iL2(t) at the third time t2, the first general solution voltage value of the third general solution Uc1(t) at the first time t0, the second general solution voltage value of the third general solution Uc1(t) at the second time t1, and the third general solution voltage value of the fourth general solution Uc2(t) at the third time t2.

[0073] In step S402, to analyze one cycle, let the first time t0 be the starting time, which is 0, i.e., t0 = 0. The second time t1 is determined by the outward phase angle θ and the switching frequency fs; specifically, when t0 is the starting time, the second time t1 is the time difference from the starting time, and the time difference corresponds exactly to the time interval value corresponding to the outward phase angle θ. That is, when t0 = 0, t1 = θ / (2πfs), and the third time t2 is T / 2 = 1 / (2fs), where T is one switching cycle.

[0074] Thus, substituting t=t0 into the first general solution iL1(t) above, we obtain the first general solution current value iL1(t=t0) at the first time t0. Since the initial time t0=0, as shown above, iL0 is the first current value of the resonant inductor Ls at the first time t0. Combined with the waveform diagram... Figure 2It can be seen that iL1(t=t0=0)=iL0. Similarly, substituting t=t1 into the first general solution iL1(t) above, we obtain the second general solution current value iL1(t=t1) at the second time t1. As shown before, iL1 is the second current value of the resonant inductor Ls at the second time t1. Combined with the waveform diagram... Figure 2 It can be seen that iL1(t=t1)=iL1. Similarly, substituting t=t2 into the second general solution iL2(t) above, we obtain the third general solution current value iL2(t=t2) at the third time t2. As mentioned before, iL2 is the current value of the resonant inductor Ls at the third time t2. Combined with the waveform diagram... Figure 2 It can be seen that iL2(t=t2)=iL2.

[0075] Thus, substituting t=t0 into the third general solution Uc1(t) above, we obtain the first general solution voltage value Uc1(t=t0) at the first time t0. Since the initial time t0=0, as shown before, Uc0 is the first voltage value of the resonant capacitor Cs at the first time t0. Combined with the waveform diagram... Figure 2 It can be seen that Uc1(t=t0=0)=Uc0. Similarly, substituting t=t1 into the third general solution Uc1(t) above, we obtain the second general solution voltage value Uc1(t=t1) at the second time t1. As shown before, Uc1 is the second voltage value of the resonant capacitor Cs at the second time t1. Combined with the waveform diagram... Figure 2 It can be seen that Uc1(t=t1)=Uc1. Similarly, substituting t=t2 into the fourth general solution Uc2(t) above, we obtain the third general solution voltage value Uc2(t=t2) of the fourth general solution Uc2(t) at the third time t2. As mentioned before, Uc2 is the voltage value of the resonant capacitor Cs at the third time t2. Combined with the waveform diagram... Figure 2 It can be seen that Uc2(t=t2)=Uc2.

[0076] S403, determine the symmetry conditions of the resonant inductor current and the resonant capacitor voltage of the DBSRC equivalent circuit: the third solution current value iL2 (t=t2) and the first solution current value iL1 (t=t0=0) are opposites of each other, and the third solution voltage value Uc2 (t=t2) and the first solution voltage value Uc1 (t=t0=0) are opposites of each other.

[0077] That is, iL2(t=t2) = -iL1(t=t0), which is iL2 = -iL0.

[0078] Similarly, Uc2(t=t2) = -Uc1(t=t0), which means Uc2 = -Uc0.

[0079] S404, based on the general solution current and voltage values ​​and the symmetry condition, a series of multivariate linear equations are obtained simultaneously. By solving these equations, the first current value iL0 and the first voltage value Uc0 of the first type of coefficients, as well as the second current value iL1 of the first type of coefficients and the second voltage value Uc1 of the second type of coefficients, are obtained, as follows:

[0080] The first current value iL0 of the first type of coefficient is:

[0081]

[0082] The second current value iL1 of the second type of coefficient is:

[0083]

[0084] The first voltage value Uc0 of the first type of coefficient is:

[0085]

[0086] The second voltage value Uc1 of the second type of coefficient is:

[0087]

[0088] In the above coefficient values ​​of current and voltage, Vi is the input voltage, Ls is the inductance of the resonant inductor, Cs is the capacitance of the resonant capacitor, Ntr is the turns ratio of the transformer in the dual-bridge series resonant converter circuit, Vo is the output voltage, fs is the switching frequency, θ is the outward phase shift angle, tan is the tangent function, sin is the sine function, and cos is the cosine function.

[0089] In this embodiment, the time-domain method is used for formula derivation, ensuring high accuracy in optimization. Compared to related technologies' DBSRC efficiency optimization methods, this embodiment obtains multiple multivariate linear equations by acquiring the general solution, obtaining the general solution current and voltage values, determining symmetry conditions, and solving them in parallel. This yields a physical expression for the external phase shift angle, adaptable to DBSRC with different parameters, exhibiting high portability. Furthermore, the calculation of the external phase shift angle is relatively simple and applicable to DSP, ARM, and other microcontroller chips, allowing calculations within each control cycle and ensuring real-time circuit control. Additionally, this embodiment employs critical ZVS for optimization, ensuring soft-switching conditions through real-time calculation, thus improving circuit efficiency while reducing computational load.

[0090] In one exemplary embodiment, refer to Figure 5 Step S302 may include the following steps S501~S502.

[0091] S501. Obtain the first equation satisfied by the first current value \(i_{L0}\) when the primary bridge circuit meets the critical condition for ZVS soft-switching requirement: \(i_{L0} = 0\).

[0092] Among them, when the parasitic capacitance \(C_{oss}\) of the switching tube is small enough, the first current value \(i_{L0}\) when the primary bridge circuit meets the ZVS soft-switching requirement is \(i_{L0}<0\), that is:

[0093]

[0094] Normalized phase-shift angle \((0 < D < 1)\), resonance frequency In the above formula (1), all parameters are greater than 0. Therefore, according to the above formula (1), it can be deduced that:

[0095] (2)

[0096] S502. Obtain the first relational expression according to the first equation.

[0097] That is, when the primary bridge circuit meets the critical condition for ZVS soft-switching requirement, that is, \(i_{L0} = 0\), that is, when the inequality (2) is equal to 0, the first relational expression obtained is:

[0098] ,

[0099] In the above first relational expression, \(D_{min}\) is the minimum normalized external phase-shift angle, \(f_r\) is the resonance frequency, \(f_s\) is the switching frequency, \(V_i\) is the input voltage, \(V_o\) is the output voltage, \(N_{tr}\) is the turns ratio of the transformer of the double-bridge series resonance conversion circuit, \(\sin\) is the sine function, and \(asin\) is the arcsine function.

[0100] In the embodiment of the present application, calculating the minimum normalized value \(D_{min}\) of the external phase-shift angle based on the above first relational expression can enable all power tubes in the circuit to achieve ZVS turn-on, thereby improving the efficiency of the circuit.

[0101] In an exemplary embodiment, referring to Figure 6 , step S302 may further include the following steps S601 to S602.

[0102] S601. Obtain the second equation satisfied by the second current value \(i_{L1}\) when the secondary bridge circuit meets the critical condition for ZVS soft-switching requirement: \(i_{L1} = 0\).

[0103] Among them, when the parasitic capacitance \(C_{oss}\) of the switching tube is small enough, the first current value \(i_{L1}\) when the secondary bridge circuit meets the ZVS soft-switching requirement is \(i_{L1}>0\), that is:

[0104] (3)

[0105] Normalized phase shift angle (0 < D < 1), resonant frequency All parameters in the above formula (3) are greater than 0. Therefore, according to the above formula (3), it can be deduced that:

[0106] (4)

[0107] S602, obtain the second relational expression according to the second equation.

[0108] That is, when the secondary bridge circuit meets the critical condition for ZVS soft-switching requirement, that is, iL1 = 0, that is, when the inequality (4) is equal to 0, the second relational expression obtained therefrom is:

[0109] , ;

[0110] In the above second relational expression, Dmin is the minimum normalized external phase shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo is the output voltage, Ntr is the turn ratio of the transformer of the dual-bridge series resonant conversion circuit, sin is the sine function, and asin is the arcsine function.

[0111] In the embodiment of the present application, calculating the minimum normalized value Dmin of the external phase shift angle based on the above second relational expression can enable all power transistors in the circuit to achieve ZVS turn-on, thereby improving the efficiency of the circuit.

[0112] In an exemplary embodiment, the steady-state voltage value Vo_set refers to the stable output of the output voltage provided by the resonant conversion circuit to the load. For simplicity of calculation, ZVS is only optimized after the circuit enters the steady state (when the circuit enters the steady state, it means that after PI control, the output voltage Vo has reached the set steady-state voltage value Vo_set, and this is when the circuit enters the steady state). At this time, Vo = Vo_set, and Vo_set is the output voltage of the DAB circuit required by the setting. Based on the first relational expression and the second relational expression, determine the expression of the external phase shift angle with respect to the switching frequency, steady-state voltage value, and input voltage, including:

[0113] , _set

[0114] , _set

[0115]

[0116] (5)

[0117] Where M can be understood as an intermediate quantity, used to replace the relationship between input voltage, steady-state voltage setpoint and turns ratio; Dmin is the minimum normalized outward phase shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo_set is the steady-state voltage value, Ntr is the turns ratio of the transformer in the dual-bridge series resonant converter circuit, sin is the sine function, and asin is the arcsine function. These are parameters that are related to both the resonant frequency fr and the switching frequency fs.

[0118] In summary, once the circuit reaches steady state, that is, when the external phase shift angle D > Dmin, all switching transistors can meet the ZVS requirement.

[0119] The scheme of this application embodiment calculates the minimum normalized value Dmin of the external phase shift angle based on formula (5), which enables all power transistors in the circuit to achieve ZVS turn-on, thereby improving the efficiency of the circuit (because the derivation of formula (5) requires meeting the ZVS soft switching requirement, thus improving the transmission efficiency of the circuit). The external phase shift angle in formula (5) is only related to the output gain and is not related to the output load (i.e., the circuit power), thereby reducing the complexity of the calculation and reducing the dependence of the control on the current sampling accuracy. The calculation of the external phase shift angle is relatively simple and can be performed by single-chip microcomputers such as DSP and ARM in each control cycle, ensuring the real-time performance of the circuit control.

[0120] Based on the above, in this embodiment of the application, the first expression for the output power Po of the output terminal of the dual-bridge series resonant converter circuit can be determined based on the piecewise calculus of the first general solution iL1(t) and the second general solution iL2(t). The gain of the output voltage Vo of the output terminal can be determined based on the first expression for the output power Po and the second expression for the output power Po of the output terminal determined based on Ohm's law.

[0121] The first expression for the output power Po is:

[0122]

[0123] The first expression for the output power Po above simplifies to:

[0124]

[0125] The second expression for the output power Po is:

[0126]

[0127] Based on the fact that the output power in the first expression is equal to the output power in the second expression, the gain of the output voltage Vo can be derived as follows:

[0128] (6)

[0129] Once the circuit reaches steady state, i.e., when the external phase shift angle D > Dmin, all switches can meet the ZVS requirement, and the gain of the output voltage Vo can be simplified as follows:

[0130] (7)

[0131] Since M is a constant less than 1, When fs > fr, 0 < θr < π / 2. From the simplified formula (7) of the output voltage Vo above, it can be seen that Vo decreases monotonically with fs. Therefore, the steady-state voltage value Vo_set can be obtained by adjusting fs through closed loop, which shows the feasibility of realizing the circuit output gain through closed loop control.

[0132] Based on the same inventive concept, in one exemplary embodiment, combined with Figure 7 and Figure 8 A frequency conversion phase shift control method is provided, which is applied, for example but not limited to, to... Figure 1A and Figure 1B An exemplary dual-bridge series resonant converter circuit, the method may include the following steps S701 to S705.

[0133] S701 samples the output of the dual-bridge series resonant converter circuit to obtain the actual voltage value of the output at the current moment.

[0134] S702, based on the actual voltage value and PI control principle, obtains the first switching frequency of the switching transistor when the output voltage reaches the steady-state voltage value.

[0135] S703, based on the first switching frequency fs1 and the preset maximum switching frequency fs_max, obtains the required switching frequency fs of the dual-bridge series resonant converter circuit in the frequency conversion phase shift control process.

[0136] S704 obtains the required external phase shift angle in the frequency conversion phase shift control process at the current moment based on the required switching frequency, the input voltage at the input terminal of the dual-bridge series resonant converter circuit at the current moment, the steady-state voltage value at the output terminal, and the expression of the method for determining the external phase shift angle based on any embodiment.

[0137] S705 obtains the PWM drive signals of the primary bridge circuit's switching transistors and the secondary bridge circuit's switching transistors based on the preset duty cycle, required switching frequency, and required outward phase shift angle.

[0138] in, Figure 8In this equation, Vout is the actual output voltage value of the DBSRC output terminal, and Vo-set is the set steady-state voltage value of the DBSRC output terminal. The DBSRC output voltage Vout is controlled by PI so that the DBSRC output voltage Vout reaches Vo-set, and the first switching frequency fs1 is obtained. The required switching frequency fs in the frequency-shifting phase control of the DBSRC is obtained by subtracting the first switching frequency fs1 from the maximum allowed switching frequency fs_max of the DBSRC. The input voltage Vi of the DBSRC and the set steady-state voltage value Vo-set of the DBSRC are substituted into the above formula (5) to obtain the minimum normalized value Dmin of the external phase angle θ. Then, an appropriate angle margin d0 is added to Dmin to compensate for the effects of dead time and parasitic capacitance Coss, etc., to obtain the required external phase angle D of the external phase angle θ in the frequency-shifting phase control. Optionally, the duty cycle of all switching transistors can be set to 50%. Based on the switching frequency fs of the switching transistors and the required external phase shift angle D, the PWM drive signals of the primary-side switching transistors Q1, Q2, Q3, and Q4 of the DBSRC transformer and the secondary-side switching transistors Q5, Q6, Q7, and Q8 of the transformer can be obtained.

[0139] In addition, when the output load is changed to a power supply, the DBSRC circuit supports bidirectional power transfer: when D < 1, power is transferred from output to output, and when D > 1, power is transferred from output to input.

[0140] Optional, refer to Figure 8 Step S704 may include: inputting the required switching frequency, the current input voltage and the steady-state voltage value into the expression to obtain the initial outward phase angle Dmin; and obtaining the required outward phase angle D based on the sum of the initial outward phase angle Dmin and the preset angle value d0.

[0141] In this embodiment of the application, the required switching frequency fs obtained from the output is substituted into the above formula (5) using PI closed-loop control to calculate the minimum phase shift angle Dmin (i.e., the initial external phase shift angle Dmin). An appropriate angle margin d0 is added to Dmin to compensate for the effects of dead time and parasitic capacitance Coss, etc., where d0 is the experimental test value, such as 0.02. The size of d0 is not limited in this embodiment of the application. The required PWM drive signal can be generated according to the required switching frequency fs and the required external phase shift angle D to make the circuit meet the requirements of output gain and ZVS.

[0142] For example, the technical solutions of the embodiments of this application are simulated below:

[0143] The circuit parameters of a DBSRC circuit C are as follows: the inductance value Ls of the resonant inductor is 64 μH, the capacitance value Cs of the resonant capacitor is 80 nF, the turns ratio Ntr of the transformer is 1 / 8, the output resistance Ro at the output end of the DBSRC circuit is 53.3 Ω, and the steady-state voltage value Vo_set is 400 V. A closed-loop control model is built according to the solution of the embodiment of the present application, and the simulation waveform is as Figure 9 shown.

[0144] From Figure 9 it can be seen that when Vi < Ntr × Vo_set, iL0 ≈ 0 and iL1 > 0, that is, all switching tubes can be turned on with ZVS, and the primary switching tube is in critical ZVS; when Vi ≥ Ntr × Vo_set, iL0 < 0 and iL1 ≈ 0, that is, all switching tubes can be turned on with ZVS, and the secondary switching tube is in critical ZVS.

[0145] Based on the above parameters, the function curve of the output gain Vo × Ntr / Vi versus the switching frequency fs is plotted as Figure 10 shown, Figure 10 where the horizontal axis is the frequency and the vertical axis is the output voltage gain. The steady-state voltage Vo_set at the output end of the resonant conversion circuit in the above formula is a fixed value, so M is a constant. Among them, Figure 10 the first function curve GM(fs) of Figure 10 is plotted according to the above formula (7) of the output voltage gain and the parameters of the above DBSRC circuit C, Figure 10 the second function curve G1(fs) in is plotted according to the above formula (6) of the output voltage gain, the expression of the external phase-shift angle of the embodiment of the present application, and the parameters of the above DBSRC circuit C, Figure 10 the third function curve G2(fs) in is plotted according to the above formula (6) of the output voltage gain, the first relational expression about the external phase-shift angle, and the parameters of the above DBSRC circuit C.

[0146] It can be seen from Figure 10 that the second function curve G1(fs) shows that the output voltage gain is a monotonically decreasing function with respect to the switching frequency fs, and the third function curve G2(fs) shows that the output voltage gain is a monotonically increasing function with respect to the switching frequency, that is, G is not monotonic in the full gain range and needs to be segmented for PI regulation. The first function curve GM(fs) shows that the output voltage gain is monotonically decreasing with respect to the switching frequency fs, so a single PI can be used for regulation, and the PI control regulation is more convenient. Figure 9 When the switching frequency in changes from 70 kHz to 150 kHz, the output gain GM is adjusted between 0.25 and 1.5, which can meet the design requirements of wide-range DC-DC such as lithium batteries.

[0147] Based on the same inventive concept, this application also provides an external phase angle determination system, which includes:

[0148] The dual-bridge series resonant converter circuit includes a primary-side bridge circuit, a resonant network, and a secondary-side bridge circuit. The input terminal of the primary-side bridge circuit is connected to the input power supply that provides the input voltage. The input terminal of the resonant network is connected to the output terminal of the primary-side bridge circuit. The input terminal of the secondary-side bridge circuit is connected to the output terminal of the resonant network. The output terminal of the secondary-side bridge circuit provides the output voltage to the load. The resonant network includes a transformer, a resonant inductor, and a resonant capacitor.

[0149] The execution unit is used to acquire the input voltage and the output voltage of the secondary bridge circuit, and is also used to execute the steps of the method for determining the external phase angle as in any embodiment when the output voltage reaches the set steady-state voltage value.

[0150] The external phase angle determination system provided in this application embodiment and the external phase angle determination method in any of the above embodiments belong to the same inventive concept, can solve the same technical problem, and thus achieve the same technical effect. Repeated content will not be repeated here.

[0151] Based on the same inventive concept, this application also provides a computer chip, including a memory and a processor. The memory stores a computer program, and the processor executes the computer program to implement the steps of the method for determining the outward phase angle of any embodiment.

[0152] Based on the same inventive concept, this application also provides a computer chip, including a memory and a processor. The memory stores a computer program, and the processor executes the computer program to implement the steps of the frequency conversion phase shift control method of any embodiment.

[0153] Based on the same inventive concept, this application also provides a computer device, including a memory and a processor, wherein the memory stores a computer program, and the processor executes the computer program to implement the steps of the method for determining the outward phase angle of any embodiment.

[0154] Based on the same inventive concept, this application also provides a computer device, including a memory and a processor, wherein the memory stores a computer program, and the processor executes the computer program to implement the steps of the frequency conversion phase shift control method of any embodiment.

[0155] Based on the same inventive concept, embodiments of this application also provide a computer-readable storage medium storing a computer program thereon, wherein when the computer program is executed by a processor, it implements the steps of the method for determining the outward phase angle of any embodiment.

[0156] Based on the same inventive concept, embodiments of this application also provide a computer-readable storage medium storing a computer program thereon, wherein the computer program, when executed by a processor, implements the steps of the frequency conversion phase shift control method of any embodiment.

[0157] Based on the same inventive concept, embodiments of this application also provide a computer program product, including a computer program that, when executed by a processor, implements the steps of the method for determining the outward phase angle of any embodiment.

[0158] Based on the same inventive concept, embodiments of this application also provide a computer program product, including a computer program that, when executed by a processor, implements the steps of the frequency conversion phase shift control method of any embodiment.

[0159] In the description of this specification, references to terms such as "some embodiments," "other embodiments," etc., indicate that a specific feature, structure, material, or characteristic described in connection with that embodiment or example is included in at least one embodiment or example of this application. In this specification, the illustrative descriptions of the above terms do not necessarily refer to the same embodiments or examples.

[0160] The technical features of the above embodiments can be combined in any way. For the sake of brevity, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0161] The embodiments described above are merely illustrative of several implementation methods of this application, and while the descriptions are specific and detailed, they should not be construed as limiting the scope of this application. It should be noted that those skilled in the art can make various modifications and improvements without departing from the concept of this application, and these modifications and improvements all fall within the protection scope of this application. Therefore, the protection scope of this application should be determined by the appended claims.

Claims

1. A method for determining the external phase shift angle, applied to a dual-bridge series resonant converter circuit, characterized in that, The method includes: Based on the general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit of the dual-bridge series resonant converter circuit, the general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit, the first current value of the resonant inductor at the first moment and the second current value of the resonant inductor at the second moment are determined. When the primary-side bridge circuit meets the critical condition for ZVS soft switching, the first current value iL0 satisfies the first equation: iL0=0; based on the first equation, the first relational expression is obtained. When the secondary bridge circuit meets the critical condition for satisfying the ZVS soft switching requirement, the second current value iL1 satisfies the second equation: iL1=0; based on the second equation, the second relationship is obtained. Based on the first relation, the second relation, and the steady-state voltage value at the output of the dual-bridge series resonant converter circuit, determine the expression for the external phase shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage; The first relation is: ， ； The second relation is: ， ； Wherein, Dmin is the minimum normalized outward phase shift angle, fr is the resonant frequency, fs is the switching frequency, Vi is the input voltage, Vo is the output voltage, Ntr is the turns ratio of the transformer in the dual-bridge series resonant converter circuit, sin is the sine function, and asin is the arcsine function; the first moment is the moment when the first upper switch of the first left bridge arm of the primary-side bridge circuit starts to turn on, and the second moment is the moment when the second upper switch of the second left bridge arm of the secondary-side bridge circuit starts to turn on during the conduction of the first upper switch.

2. The method according to claim 1, characterized in that, The determination of the first current value of the resonant inductor at a first time moment and the second current value of the resonant inductor at a second time moment, based on the general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit based on the dual-bridge series resonant converter circuit, the general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit, and the symmetry of the DBSRC equivalent circuit, includes: Obtaining the general solutions: Obtain the first general solution of the time-domain differential equation of the resonant inductor current of the DBSRC equivalent circuit in the first time period and the second general solution in the second time period. Obtain the third general solution of the time-domain differential equation of the resonant capacitor voltage of the DBSRC equivalent circuit in the first time period and the fourth general solution in the second time period. The first general solution and the third general solution are both linear functions of the first current value and the first voltage value of the resonant capacitor voltage at the first moment with first type coefficients. The second general solution and the fourth general solution are both linear functions of the second current value and the second voltage value of the resonant capacitor voltage at the second moment with second type coefficients. The first moment and the second moment are the start moment and the end moment of the first time period, respectively. The start moment of the second time period is the second moment. The end moment of the second time period is recorded as the third moment. The third moment is the moment when the first upper switch starts to turn off during the conduction process of the second upper switch. Obtain the current and voltage values ​​of the solution: Obtain the first current value of the first solution at the first time, the second current value of the first solution at the second time, the third current value of the second solution at the third time, the first voltage value of the third solution at the first time, the second voltage value of the third solution at the second time, and the third voltage value of the fourth solution at the third time, respectively. Wherein, the first time t0 is the starting time t0=0, the second time t1 is determined by the external phase shift angle and the switching frequency, and the third time t2 is T / 2, where T is one switching cycle. The symmetry conditions for the resonant inductor current and the resonant capacitor voltage of the DBSRC equivalent circuit are determined as follows: the third through current value is the opposite of the first through current value, and the third through voltage value is the opposite of the first through voltage value. Based on the current and voltage values ​​of the solution and the symmetry condition, multiple multivariate linear equations are obtained simultaneously, and the first type of coefficient values ​​and the second type of coefficient values ​​are obtained by solving the equations.

3. The method according to claim 2, characterized in that, Both the first general solution and the second general solution are functions of the inductor current at different times with respect to the input voltage, the output voltage, and the outward phase shift angle; Both the third and fourth general solutions are capacitance-voltage functions of the resonant capacitor at different times with respect to the input voltage, the output voltage, and the outward phase shift angle.

4. The method according to claim 2, characterized in that, The first current value iL0 of the first type of coefficient is: ； The second current value iL1 of the second type of coefficient is: ； Wherein, Vi is the input voltage, Ls represents the inductance value of the resonant inductor, Cs represents the capacitance value of the resonant capacitor, Ntr is the turns ratio of the transformer in the dual-bridge series resonant converter circuit, Vo is the output voltage, fs is the switching frequency, θ is the outward phase shift angle, tan is the tangent function, sin is the sine function, and cos is the cosine function.

5. The method according to claim 1, characterized in that, When the output voltage is the set steady-state voltage value, based on the first and second relationships, the expression for the external phase shift angle with respect to the switching frequency, the steady-state voltage value, and the input voltage is determined, including: ； ； ， ； ， ； Where Vo_set is the steady-state voltage value.

6. A frequency conversion phase shift control method applied to a dual-bridge series resonant converter circuit, characterized in that, The control method includes: The output of the dual-bridge series resonant converter circuit is sampled to obtain the actual voltage value of the output at the current moment. Based on the actual voltage value and the PI control principle, the first switching frequency of the switching transistor when the voltage at the output terminal reaches the steady-state voltage value is obtained; Based on the first switching frequency and the preset maximum switching frequency, the required switching frequency of the dual-bridge series resonant converter circuit in the frequency conversion phase shift control process is obtained. Based on the required switching frequency, the input voltage at the input terminal of the dual-bridge series resonant converter circuit at the current moment, the steady-state voltage value at the output terminal, and the expression of the method for determining the external phase shift angle according to any one of claims 1 to 5, the required external phase shift angle in the frequency conversion phase shift control process at the current moment is obtained. Based on the preset duty cycle of the switching transistor, the required switching frequency, and the required outward phase shift angle, the PWM drive signal of the switching transistor of the primary bridge circuit and the PWM drive signal of the switching transistor of the secondary bridge circuit are obtained.

7. The method according to claim 6, characterized in that, Based on the required switching frequency, the current input voltage, the steady-state voltage value at the output terminal, and the expression, the required external phase shift angle in the frequency conversion phase shift control process at the current moment is obtained, including: The required switching frequency, the current input voltage, and the steady-state voltage value are input into the expression to obtain the initial outward phase shift angle; The required outward phase angle is obtained by summing the initial outward phase angle and the preset angle value.

8. A system for determining an outward phase angle, characterized in that, include: A dual-bridge series resonant converter circuit includes a primary-side bridge circuit, a resonant network, and a secondary-side bridge circuit. The input terminal of the primary-side bridge circuit is connected to the input power supply that provides the input voltage, the input terminal of the resonant network is connected to the output terminal of the primary-side bridge circuit, the input terminal of the secondary-side bridge circuit is connected to the output terminal of the resonant network, and the output terminal of the secondary-side bridge circuit provides the output voltage to the load. The resonant network includes a transformer, a resonant inductor, and a resonant capacitor; An execution unit is configured to acquire the input voltage and the output voltage of the output terminal of the secondary bridge circuit, and to execute the steps of the method as described in any one of claims 1 to 5 when the output voltage reaches a set steady-state voltage value.

9. A computer chip comprising a memory and a processor, wherein the memory stores a computer program, characterized in that, When the processor executes the computer program, it implements the steps of the method according to any one of claims 1 to 7.

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