A control method of a bridgeless PFC single-stage LLC resonant converter

Abstract

The application relates to the technical field of power electronics, in particular to a control method of a bridgeless PFC single-stage LLC resonant converter, which comprises simultaneously collecting an output voltage and a bus voltage, respectively calculating the difference from a respective preset reference voltage, inputting the difference into independent PI controllers respectively, obtaining corresponding output frequencies respectively, then performing weighted processing on the output frequencies to obtain an actual control frequency, and generating a switching tube control signal of the converter according to the actual control frequency. The application solves the problem that a traditional single-stage converter is difficult to simultaneously ensure the stability of an output voltage and the safety of a bus voltage due to the adoption of a single control loop, effectively improves the overall performance and operation reliability of the converter under complex working conditions such as grid fluctuation and load mutation. Moreover, by integrating multiple independent magnetic elements onto a composite magnetic core, the volume, weight and component quantity of the converter are significantly reduced, so that the power density is improved and the manufacturing cost is reduced.

Description

Technical Field

[0001] This application relates to the field of power electronics technology, and in particular to a control method for a bridgeless PFC single-stage LLC resonant converter. Background Technology

[0002] Currently, with the booming development of the new energy vehicle industry, on-board chargers (OBC) and DC charging piles, as core energy replenishment equipment, are increasingly becoming research hotspots. To improve charging efficiency, reduce equipment size, and lower costs, high-power-density, high-efficiency AC / DC converter topologies have become a key technological direction pursued by the industry. Among numerous topologies, LLC resonant converters, due to their ability to achieve zero-voltage turn-on (ZVS) of the switching transistors, significantly reduce switching losses and exhibit great advantages in high-frequency, high-efficiency applications.

[0003] To achieve efficient DC / DC conversion while meeting power factor correction (PFC) standards, a two-stage architecture is typically employed, consisting of a separate PFC correction circuit in the front stage and a separate LLC resonant circuit in the rear stage. While this two-stage architecture is mature and stable, it suffers from drawbacks such as reduced overall efficiency due to the need for two energy conversions, a larger number of components, more complex control, and higher cost and size. To address these issues, integrating the PFC and LLC stages to form a single-stage converter has become an important technological trend.

[0004] However, existing single-stage converters still face challenges in their control strategies. To ensure precise battery charging management, the control system must implement closed-loop control of the output voltage. Therefore, most solutions employ a single closed-loop control strategy with the output voltage as the sole feedback variable. The drawback of this strategy is that the intermediate bus capacitor voltage lacks effective direct control, and its voltage stability relies entirely on the energy transfer balance of the LLC stage. When the input grid voltage fluctuates, or when the load characteristics, such as those of the power battery at different charging stages, undergo abrupt changes, this passive balance is easily disrupted, leading to significant ripple or even instability in the bus voltage. This not only affects the performance of the PFC stage but also, in turn, interferes with the stability of the output voltage and even threatens the safety of power devices.

[0005] Furthermore, at the physical implementation level, even with a single-stage topology, the input inductor required for power factor correction, the resonant inductor required for LLC resonance, and the main transformer are typically still discrete magnetic components. This layout not only occupies valuable board space, increases assembly complexity and cost, but also makes it difficult to control stray parameters between discrete magnetic components, potentially affecting the overall electromagnetic compatibility (EMC) performance of the system and limiting further improvements in converter power density. Therefore, there is room for improvement. Summary of the Invention

[0006] This application provides a control method for a bridgeless PFC single-stage LLC resonant converter, which improves converter efficiency while reducing cost and space occupation.

[0007] This application provides a bridgeless PFC single-stage LLC resonant converter, applied to a bridgeless PFC single-stage LLC resonant converter, wherein the control method of the bridgeless PFC single-stage LLC resonant converter includes:

[0008] Collect output voltage V o Calculate the output voltage V o With preset output reference voltage V oref While measuring the difference V1, the bus voltage V is also collected. Cbus Calculate the bus voltage V Cbus With the preset bus reference voltage V Cbusref The difference V2;

[0009] The difference V1 is input to the first PI controller for proportional-integral control to obtain the first output frequency f1. At the same time, the difference V2 is input to the second PI controller for proportional-integral control to obtain the second output frequency f2.

[0010] The first output frequency f1 and the second output frequency f2 are weighted to obtain the actual control frequency f;

[0011] The actual control frequency f is input to the drive signal generator to obtain the control signal for switch S3 and the control signal for switch S4.

[0012] By adopting the above technical solution and setting up two independent PI control loops for the output voltage and the bus voltage, the stability of the output load and the stability of the input bus voltage can be adjusted separately, thus forming a dual closed-loop feedback system. By weighting the frequencies of the two loop outputs, the focus of the control strategy can be dynamically adjusted according to the actual operating state of the converter. This allows the converter to simultaneously achieve voltage stabilization of the output voltage and effective control of the bus voltage under a single control core, improving the overall performance and stability of the system.

[0013] In a preferred embodiment, this application can be further configured such that: the weighted processing of the first output frequency f1 and the second output frequency f2 to obtain the actual control frequency f specifically includes:

[0014] Determine the bus voltage V Cbus Is it greater than the preset bus reference voltage V? Cbusref ;

[0015] If the bus voltage V CbusGreater than the preset bus reference voltage V Cbusref Then, using the formula f = f1 + K p The actual control frequency f is calculated using f2, where K p The preset emergency coefficient is used to adjust the influence weight of the bus voltage, f is the actual control frequency, f1 is the first output frequency, and f2 is the second output frequency;

[0016] If the bus voltage V Cbus Not greater than the preset bus reference voltage V Cbusref The actual control frequency f can be calculated using the formula f = f1.

[0017] By adopting the above technical solution, the weighted processing is further concretized into a dynamic priority control strategy based on a safety threshold. When the bus voltage is normal, the system control is completely handed over to the output voltage loop to ensure the best power supply quality and dynamic response to the load. Once the bus voltage is detected to exceed the safety threshold, the control of the bus voltage loop is immediately intervened with a higher emergency weight to achieve rapid suppression of bus overvoltage. Without sacrificing conventional performance, the survivability and safety of the converter under extreme conditions such as grid fluctuations or load changes are greatly improved.

[0018] In a preferred embodiment, this application can be further configured such that the bridgeless PFC single-stage LLC resonant converter includes:

[0019] The circuit consists of a pre-amplifier input circuit, a transformer, and a secondary rectifier circuit.

[0020] The pre-amplifier input circuit includes switching transistors S1, S2, S3, and S4. The source of switching transistor S1 and the drain of switching transistor S2 are connected to the AC input source V. g The positive terminal of the switch S3 and the drain of the switch S4 are connected to the AC input source v. g The negative electrode;

[0021] The first output terminal of the preamplifier input circuit is connected to the first input terminal of the transformer, and the second output terminal of the preamplifier input circuit is connected to the second input terminal of the transformer.

[0022] The secondary rectifier circuit includes synchronous rectifier diodes SR1 and SR2. The drain of synchronous rectifier diode SR1 is connected to the first output terminal of the transformer, and the drain of synchronous rectifier diode SR2 is connected to the third output terminal of the transformer.

[0023] In the bridgeless PFC single-stage LLC resonant converter, the second output terminal of the transformer is the first output terminal, and the source of the synchronous rectifier SR2 is the second output terminal.

[0024] By adopting the above technical solution, the traditional diode rectifier bridge is replaced by a bridgeless PFC preamplifier circuit composed of switching transistors, which fundamentally eliminates the fixed conduction loss of the two diodes and lays the foundation for high-efficiency power factor correction. At the same time, by using synchronous rectifiers SR1 and SR2 on the secondary side, the rectification loss on the output side is also greatly reduced.

[0025] In a preferred embodiment, this application can be further configured such that the pre-amplifier input circuit also includes:

[0026] Output capacitor C bus and resonant capacitor C r ;

[0027] The drain of the switching transistor S1 is connected to the drain of the switching transistor S3, and the output capacitor C bus One end is connected to the connection point between the drain of the switching transistor S1 and the drain of the switching transistor S3, and the source of the switching transistor S2 and the source of the switching transistor S4 are connected. The output capacitor C bus The other end is connected to the junction of the source of the switching transistor S2 and the source of the switching transistor S4;

[0028] The resonant capacitor C r One end is connected to the source of the switching transistor S2, the source of the switching transistor S4, and the output capacitor C. bus The resonant capacitor C is connected at the connection point. r The other end is connected to the second input terminal of the transformer.

[0029] By adopting the above technical solution and specifying the circuit positions of the output capacitor as the bus energy storage element and the resonant capacitor as the key element of the resonant network, the circuit topology is further clarified. This layout allows the output capacitor to effectively smooth the DC voltage after rectification in the PFC stage, providing a stable input for the subsequent LLC stage. At the same time, the resonant capacitor, resonant inductor, and magnetizing inductor together form a resonant cavity, which is the core for achieving efficient energy transfer and soft-switching operation, ensuring the prerequisite for high-performance operation of the converter.

[0030] In a preferred embodiment, this application can be further configured such that the pre-amplifier input circuit also includes:

[0031] Input inductance L f Resonant inductor L r and transformer magnetizing inductance L m ;

[0032] The source of the switching transistor S1 and the drain of the switching transistor S2 are connected through the input inductor L. f Connect AC input source vg The positive electrode;

[0033] The first output terminal of the pre-amplifier input circuit is connected to the resonant inductor L. r Connected to the first input terminal of the transformer;

[0034] The transformer magnetizing inductance L m One end is connected to the resonant inductor L r The transformer is connected to the first input terminal of the transformer, and the transformer's magnetizing inductance L m The other end is connected to the resonant capacitor C r Connect to the second input terminal of the transformer.

[0035] By adopting the above technical solution, all the magnetic components required to achieve power factor correction and LLC resonance are completely defined. The input inductor is used to smooth the input current, so that the input current can be effectively shaped, thereby achieving high power factor correction. The resonant inductor and the magnetizing inductor together determine the operating characteristics of the LLC resonant network, which together constitute the unique resonant network of the LLC resonant converter. This network can achieve efficient power conversion and output voltage regulation over a wide range of input voltage and load, providing a guarantee for the excellent electrical performance of the entire single-stage converter.

[0036] In a preferred embodiment, this application can be further configured such that the control method also includes:

[0037] The input inductor L is integrated using magnetic integration technology. f The resonant inductor L r The transformer is integrated onto the same EEI-type composite magnetic core, and a corresponding magnetic circuit model is established, wherein the input inductance L... f The primary and secondary windings of the transformer are both wound on the central column of a second E-type magnetic core, and the central column of the first E-type magnetic core is provided with a first air gap.

[0038] By adopting the above technical solution, and integrating multiple independent magnetic elements onto a composite magnetic core, the size, weight, and number of components of the converter can be significantly reduced, thereby increasing power density and reducing manufacturing costs. By setting independent air gaps on different magnetic pillars, the independence of the magnetic characteristics of the input inductor and transformer can be ensured while achieving structural integration, and the unity of structural compactness and high performance can be achieved.

[0039] In a preferred embodiment, this application can be further configured such that: the resonant inductor L r It is formed by the leakage inductance between the primary and secondary windings of the transformer.

[0040] By adopting the above technical solution, and by cleverly utilizing the inherent leakage inductance of the transformer itself as the resonant inductor, a separate physical inductor component is eliminated. This design not only further simplifies the circuit structure and reduces cost and size, but also reduces parasitic losses and electromagnetic interference caused by additional components and their connecting lines, making the entire resonant network more compact and efficient.

[0041] In a preferred embodiment, the present application may be further configured such that: the secondary winding has a center tap, the two ends of the secondary winding respectively constitute the first output terminal and the third output terminal of the transformer, and the center tap constitutes the second output terminal of the transformer.

[0042] By adopting the above technical solution, a full-wave rectifier circuit is formed by using a center-tapped secondary winding in conjunction with two synchronous rectifier tubes. Compared with the bridge rectifier circuit that requires four rectifier devices, the voltage drop of one semiconductor device is reduced in the current path during each working cycle, which greatly reduces the conduction loss on the secondary side.

[0043] In a preferred embodiment, this application may be further configured such that the source of the synchronous rectifier SR1 is connected to the source of the synchronous rectifier SR2.

[0044] By adopting the above technical solution, the common source connection of the two synchronous rectifier diodes is clarified. This connection allows the two devices to be easily controlled by the same driver chip, simplifying the design complexity of the driver circuit. At the same time, connecting the common source to the negative terminal of the output also makes the circuit layout simpler, which helps to reduce the parasitic inductance of the output circuit and thus improve the ripple characteristics of the output voltage.

[0045] In a preferred embodiment, this application may be further configured to include an RC filter circuit disposed between the first output terminal and the second output terminal.

[0046] By adopting the above technical solution and adding an RC filter circuit at the output end, the high-frequency voltage ripple after synchronous rectification can be further absorbed and filtered out, effectively improving the smoothness and purity of the output DC voltage, and ensuring that more stable and high-quality power is provided to the downstream load.

[0047] In summary, this application includes at least one of the following beneficial technical effects:

[0048] 1. By setting two independent PI control loops for the output voltage and the bus voltage, the stability of the output load and the stability of the input bus voltage can be adjusted separately, thus forming a dual closed-loop feedback system. By weighting the frequency of the output of the two loops, the focus of the control strategy can be dynamically adjusted according to the actual operating state of the converter. This allows the converter to simultaneously achieve voltage stabilization of the output voltage and effective control of the bus voltage under a single control core, improving the overall performance and stability of the system.

[0049] 2. By integrating multiple independent magnetic elements onto a composite magnetic core, the size, weight, and number of components of the converter can be significantly reduced, thereby increasing power density and reducing manufacturing costs. By setting independent air gaps on different magnetic pillars, the independence of the magnetic characteristics of the input inductor and transformer can be ensured while achieving structural integration, thus achieving a balance between structural compactness and high performance. Attached Figure Description

[0050] Figure 1 This is a flowchart illustrating a control method for a bridgeless PFC single-stage LLC resonant converter in an embodiment of this application.

[0051] Figure 2 This is a control block diagram of the voltage dual-loop control mode in the embodiments of this application;

[0052] Figure 3 This is a circuit diagram of the bridgeless PFC single-stage LLC resonant converter in the embodiments of this application;

[0053] Figure 4 This is a waveform diagram of the bridgeless PFC single-stage LLC resonant converter in the embodiments of this application;

[0054] Figure 5 This is the equivalent circuit diagram of the working mode from t0 to t1 of the bridgeless PFC single-stage LLC resonant converter in the positive half-power frequency period in the embodiments of this application;

[0055] Figure 6 This is the equivalent circuit diagram of the working mode from t1 to t2 of the bridgeless PFC single-stage LLC resonant converter in the embodiment of this application during the positive half-power frequency period;

[0056] Figure 7 This is the equivalent circuit diagram of the working mode from t2 to t3 of the bridgeless PFC single-stage LLC resonant converter in the embodiment of this application during the positive half-power frequency period;

[0057] Figure 8 This is the equivalent circuit diagram of the working mode from t3 to t4 of the bridgeless PFC single-stage LLC resonant converter in the embodiment of this application during the positive half-power frequency period;

[0058] Figure 9This is the equivalent circuit diagram of the working mode from t4 to t5 of the bridgeless PFC single-stage LLC resonant converter in the embodiment of this application during the positive half-power frequency period;

[0059] Figure 10 This is the equivalent circuit diagram of the working mode from t5 to t6 of the bridgeless PFC single-stage LLC resonant converter in the positive half-power frequency period in the embodiments of this application;

[0060] Figure 11 This is the equivalent circuit diagram of the working mode from t6 to t7 of the bridgeless PFC single-stage LLC resonant converter in the positive half-power frequency period of the embodiment of this application;

[0061] Figure 12 This is a magnetic integration structure diagram of the EEI type composite magnetic core in the embodiments of this application;

[0062] Figure 13 This is a schematic diagram of the magnetic integrated circuit of the bridgeless PFC single-stage LLC resonant converter in the embodiments of this application. Detailed Implementation

[0063] The following embodiments will help those skilled in the art to further understand the function of this application, but do not limit this application in any way. It should be noted that those skilled in the art can make several modifications and improvements without departing from the concept of this application. These all fall within the protection scope of this application.

[0064] In the following description, specific details such as particular system architectures and techniques are set forth for illustrative purposes and not for limitation, in order to provide a thorough understanding of the embodiments of this application. However, those skilled in the art will understand that this application may also be implemented in other embodiments without these specific details. In other instances, detailed descriptions of well-known systems, apparatuses, circuits, and methods have been omitted so as not to obscure the description of this application with unnecessary detail.

[0065] It should be understood that, when used in this application specification and the appended claims, the term "comprising" indicates the presence of the described features, integrals, steps, operations, elements and / or components, but does not exclude the presence or addition of one or more other features, integrals, steps, operations, elements, components and / or a collection thereof.

[0066] The present application will be further described in detail below with reference to the accompanying drawings.

[0067] Please refer to Figures 1 to 2 , Figure 1 This is a flowchart illustrating the control method of a bridgeless PFC single-stage LLC resonant converter in an embodiment of this application. Figure 2 This is a control block diagram of the voltage dual-loop control mode in the embodiments of this application. The control method specifically includes the following steps:

[0068] S1, Acquire output voltage V o Calculate the output voltage V o With preset output reference voltage V oref While measuring the difference V1, the bus voltage V is also collected. Cbus Calculate the bus voltage V Cbus With the preset bus reference voltage V Cbusref The difference V2.

[0069] Specifically, high-precision voltage sensors or voltage sampling circuits are installed at the output of the converter and across the DC bus capacitor, respectively. The output signals of the sensors or the analog voltage signals are connected to the analog-to-digital converter (ADC) channel of the digital controller or digital signal processor (DSP). To ensure the real-time performance and accuracy of the two voltage signals, the sampling frequency of the ADC is set much higher than the operating switching frequency of the converter, such as more than ten times. For example, if the converter operates at 100kHz, the sampling frequency can be set to 1MHz to capture instantaneous voltage changes. Furthermore, synchronous triggering sampling is used for the two ADC channels to avoid phase errors caused by sampling time differences, thereby providing accurate and delay-free state input for subsequent dual-loop control. The preset output reference voltage V... oref It is set according to the charging voltage required by the new energy battery at the current stage. For example, this value is precisely fixed during the constant voltage charging stage, while the preset bus reference voltage V Cbusref This is a fixed value determined based on the grid voltage range and converter design margin, designed to maintain stable operation of the PFC stage. For example, when charging a new energy vehicle battery with a rated charging voltage of 400V, V... oref It may be set to 400V, while V Cbusref A safety value, such as 450V, is set based on the peak value of the input AC voltage. The output voltage error and the difference V1 and V2 are obtained by performing error subtraction operation inside the digital signal processor. These two difference signals directly reflect the degree of deviation between the current state and the desired state of the system and are the direct input for subsequent closed-loop control.

[0070] S2. Input the difference V1 to the first PI controller for proportional-integral control to obtain the first output frequency f1. At the same time, input the difference V2 to the second PI controller for proportional-integral control to obtain the second output frequency f2.

[0071] Specifically, two parallel proportional-integral (PI) controllers are implemented using software algorithms in a digital signal processor (DSP). The first PI controller focuses on precise control of the output voltage, and its proportional parameter K... p1 and integration parameter K i1The tuning objective of the first controller is to minimize steady-state error and optimize load dynamic response to meet the stringent voltage accuracy requirements of the battery charging protocol. The second PI controller focuses on stabilizing the bus voltage, and its proportional parameter K... p2 and integration parameter K i2 The tuning objective is to quickly suppress bus voltage fluctuations caused by grid voltage dips or surges. By inputting the difference V1 to the first PI controller and the difference V2 to the second PI controller, the two controllers operate independently and output the corresponding frequency adjustment values ​​to obtain the first output frequency f1 and the second output frequency f2. Here, f1 mainly carries the power regulation information for the load, while f2 mainly carries the suppression information for input disturbances.

[0072] S3. Weight the first output frequency f1 and the second output frequency f2 to obtain the actual control frequency f.

[0073] Specifically, the weighted processing here is a strategy that merges two independent control objectives. It aims to dynamically or statically allocate the control weights of the two frequency regulation quantities according to the overall operating state of the converter. For example, in one implementation, fixed weighting coefficients can be set for f1 and f2, and they can be linearly superimposed to prioritize the basic stability of the converter system between output accuracy and bus stability. In another more complex implementation, this weighting process can be dynamically changed, adjusting their respective weights according to other parameters in the system, such as the rate of change of input current or load current, thereby achieving an adaptive cooperative control that enables the converter to exhibit optimal comprehensive performance under different operating conditions.

[0074] S4. Input the actual control frequency f to the drive signal generator to obtain the control signal of switch S3 and the control signal of switch S4.

[0075] Specifically, after receiving the calculated actual control frequency f, the dedicated PWM (Pulse Width Modulation) module inside the digital controller, i.e., the drive signal generator, immediately calculates the switching period T=1 / f in real time based on this frequency. Within this period, it generates two complementary drive signals, i.e., control signals, with a constant duty cycle of 50%. These two signals are used to drive the upper arm switch S3 and the lower arm switch S4 in the converter's half-bridge topology, respectively. To absolutely avoid a bridge arm short circuit fault caused by the simultaneous conduction of switches S3 and S4 at the moment of switching, a microsecond-level dead time must be forcibly inserted into this pair of complementary drive signals. That is, after the turn-off signal of one switch is issued, a short delay is made before the turn-on signal of the other switch is issued, ensuring that the other switch is turned on only after one switch is completely turned off. This strictly prevents shoot-through short circuits between the upper and lower arms, greatly improving the operational reliability of the power circuit.

[0076] Based on the above embodiments, as an optional embodiment, in step S3, the first output frequency f1 and the second output frequency f2 are weighted to obtain the actual control frequency f, specifically including:

[0077] S31. Determine the bus voltage V Cbus Is it greater than the preset bus reference voltage V? Cbusref .

[0078] Specifically, this step involves a routine logic check for switching control modes. During each digital control cycle, the real-time acquired bus voltage V is... Cbus With the set bus reference voltage V Cbusref The comparison result directly determines whether to adopt a voltage single-loop control mode with output voltage as the core or a voltage dual-loop control mode with bus voltage as the coordinator in the current control cycle. It constitutes a core state branch judgment point in the control algorithm.

[0079] S32. If the bus voltage V Cbus Greater than the preset bus reference voltage V Cbusref Then, using the formula f = f1 + K p The actual control frequency f is calculated using f2, where K p The preset emergency coefficient is used to adjust the influence weight of the bus voltage. f is the actual control frequency, f1 is the first output frequency, and f2 is the second output frequency.

[0080] Specifically, when the bus voltage is detected to be slightly higher than its target operating point, it usually indicates that the energy input from the grid side temporarily exceeds the energy transferred to the output side, causing energy to accumulate on the bus capacitor. At this point, the voltage dual-loop control mode is activated. In this mode, the second output frequency f2, representing the bus voltage regulation requirement, is multiplied by a preset gain coefficient, which is the preset emergency coefficient K used to adjust the influence weight of the bus voltage. p Then, it is superimposed with the first output frequency f1, which is mainly used to control the output voltage, to jointly determine the final actual control frequency f. p This is a flexibly configurable adjustable gain used to finely adjust the response strength of the bus voltage control loop and its weighting on the total frequency. It is typically set between 0.5 and 2.0, such as 1.5 or 2.0. This determines the intervention strength of the bus voltage control loop, allowing for the reduction or increase of the converter's actual control frequency f. By appropriately increasing the actual control frequency f, the voltage gain of the LLC converter is actively reduced, thereby actively reducing energy transfer from input to output. This releases the charge accumulated on the bus capacitors, allowing the bus voltage V to decrease. Cbus It can smoothly fall back to a safe range.

[0081] S33, If the bus voltage V Cbus Not greater than the preset bus reference voltage V Cbusref Then, the actual control frequency f can be calculated using the formula f = f1.

[0082] Specifically, when the bus voltage V Cbus At its set reference value, i.e., the preset bus reference voltage V Cbusref At or below this level, it indicates that the energy flow is in equilibrium or the output demand is high. In this case, the bus voltage is considered stable and requires no additional intervention. Therefore, the system switches to a single-loop voltage control mode with the output voltage as the sole core objective. In this mode, the second output frequency f2, representing bus voltage regulation, is temporarily disabled or its weight is reset to zero, thus achieving dynamic decoupling between the bus voltage control loop and the output voltage control loop. This design avoids unnecessary mutual influence or control contention between the two PI controllers under normal operating conditions due to minor disturbances, ensuring stable control of the output voltage V. o Absolute priority and maximum stability of control is an efficient strategy for dynamically adjusting the priority of control objectives.

[0083] Based on the above embodiments, as an optional embodiment, refer to... Figure 3 , Figure 3 This is a circuit diagram of a bridgeless PFC single-stage LLC resonant converter in an embodiment of this application. The bridgeless PFC single-stage LLC resonant converter includes:

[0084] The circuit consists of a pre-amplifier input circuit, a transformer, and a secondary rectifier circuit.

[0085] The preamplifier input circuit includes switches S1, S2, S3, and S4. Switches S1 and S2 form the PFC power frequency bridge arm, while switches S3 and S4 form the PFC and LLC multiplexed high frequency bridge arm. The source of switch S1 and the drain of switch S2 are connected to the AC input source V. g The positive terminal of transistor S3, the source of transistor S3, and the drain of transistor S4 are connected to the AC input source V. g The negative electrode;

[0086] The first output terminal of the preamplifier input circuit is connected to the first input terminal of the transformer, and the second output terminal of the preamplifier input circuit is connected to the second input terminal of the transformer.

[0087] The secondary rectifier circuit includes synchronous rectifier diodes SR1 and SR2. The drain of synchronous rectifier diode SR1 is connected to the first output terminal of the transformer, and the drain of synchronous rectifier diode SR2 is connected to the third output terminal of the transformer.

[0088] In a bridgeless PFC single-stage LLC resonant converter, the second output terminal of the transformer is the first output terminal, and the source of the synchronous rectifier diode SR2 is the second output terminal.

[0089] Specifically, switching transistors S1 and S2 form the PFC power frequency bridge arm, which is controlled by the controller based on the AC input source v. g The polarity of the switching transistors is used to periodically turn on the switching transistors S1 or S2 at the power frequency, such as 50Hz or 60Hz, thereby replacing the traditional rectifier bridge and avoiding the conduction losses of the rectifier bridge. Specifically, the switching transistors S1 and S2 alternately turn on during the positive and negative half-cycles of the AC input voltage to achieve the basic rectification function of the input AC power. The switching transistors S3 and S4 form a high-frequency bridge arm for PFC and LLC multiplexing. On the one hand, the switching transistors S3 and S4 cooperate with the switching transistors S1 and S2 to perform the PFC function as part of the Boost converter circuit. Through high-frequency switching, the input current is shaped into a sine wave in phase with the input voltage. On the other hand, they also form the half-bridge inverter circuit of the LLC resonant converter. They receive the actual control frequency f calculated by the aforementioned control method and chop the DC bus voltage into a high-frequency square wave voltage to drive the subsequent resonant network and high-frequency transformer. In the secondary rectifier circuit, synchronous rectifier diodes SR1 and SR2 are used instead of traditional diodes for rectification because synchronous rectifier diodes such as MOSFETs have extremely low on-resistance, which can significantly reduce the conduction loss of the rectification stage. This is crucial for improving the overall efficiency of the converter under heavy load, especially high-current charging conditions. By using precise drive signals from the controller or dedicated driver chip, the converter switches synchronously with the secondary voltage of the transformer, ensuring that it turns on at the correct time, thereby efficiently converting the high-frequency AC power induced on the secondary side of the transformer into a stable DC power that is ultimately provided to the load.

[0090] Based on the above embodiments, as an optional embodiment, the pre-amplifier input circuit further includes:

[0091] Output capacitor C bus and resonant capacitor C r ;

[0092] The drain of switching transistor S1 is connected to the drain of switching transistor S3, and the output capacitor C bus One end is connected to the drain of switching transistor S1 and the drain of switching transistor S3, and the source of switching transistor S2 and the source of switching transistor S4 are connected. The output capacitor C bus The other end is connected to the junction of the source of switch S2 and the source of switch S4;

[0093] Resonant capacitor C r One end is connected to the source of switching transistor S2, the source of switching transistor S4, and the output capacitor C. bus The connection point is connected to the resonant capacitor C.r The other end is connected to the second input terminal of the transformer.

[0094] Specifically, the output capacitor C bus That is, the bus capacitor is connected in parallel across the entire DC bus, specifically across the upper and lower ends of the PFC and LLC multiplexed high-frequency bridge arm, and more specifically across the upper and lower ends of switching transistors S3 and S4, and the output capacitor C. bus As an energy buffer unit, it absorbs the pulsating DC energy obtained from the PFC stage rectification and provides a stable and smooth DC voltage source for the subsequent LLC inverter. Its capacitance directly affects the bus voltage ripple and the system's dynamic response capability. Resonant capacitor C r Then, with the subsequent resonant inductance L r and transformer magnetizing inductance L m Together they form the core of the LLC resonant network. It is connected in series between the lower end of the PFC and LLC multiplexed high-frequency bridge arm and the primary winding of the transformer. Through this connection, it converts the square wave voltage output by the PFC and LLC multiplexed high-frequency bridge arm into an approximately sinusoidal resonant current, creating conditions for achieving zero-voltage switching (ZVS) of the switching transistor, thereby greatly reducing switching losses and improving the efficiency of the converter at high frequencies.

[0095] Based on the above embodiments, as an optional embodiment, the pre-amplifier input circuit further includes:

[0096] Input inductance L f Resonant inductor L r and transformer magnetizing inductance L m ;

[0097] The source of switch S1 and the drain of switch S2 are connected through the input inductor L. f Connect AC input source v g The positive electrode;

[0098] The first output terminal of the preamplifier input circuit is connected to the resonant inductor L. r Connect to the first input terminal of the transformer;

[0099] Transformer magnetizing inductance L m One end is connected to the resonant inductor L r Connected to the first input terminal of the transformer, the transformer magnetizing inductance L m The other end is connected to the resonant capacitor C r Connect to the second input terminal of the transformer.

[0100] Specifically, the input inductance L fAlso known as a PFC inductor or boost inductor, it is connected in series between the AC input source and the PFC power frequency bridge arm, i.e., switching transistors S1 and S2. It is an essential component for achieving boost power factor correction. Driven by high-frequency switching, it stores and releases energy, forcing the input current waveform to follow the input voltage waveform, thereby achieving a high power factor. Resonant inductor L r With resonant capacitor C r Transformer magnetizing inductance L m Together, they form the LLC resonant cavity that determines the converter's gain characteristics, where the resonant inductor L... r It is usually connected in series in the resonant network, and its value is related to the resonant capacitance C. r Together, they determine the resonant frequency of the circuit and are the main channel for energy transfer; the transformer's magnetizing inductance L m It is a parasitic parameter of the transformer itself or an independent external inductor. It is connected in parallel across the primary winding of the transformer. It not only participates in the resonance process, but more importantly, it provides the necessary excitation current to achieve zero-voltage switching under light load or no-load conditions, thereby extending the operating range of ZVS to full load and ensuring the efficient operation of the converter under various operating conditions.

[0101] The circuit structure of the bridgeless PFC single-stage LLC resonant converter is symmetrical, and the AC input source v g The circuit operates similarly during the positive and negative half-cycles. The difference lies in the fact that during the positive half-cycle, the AC input source v... g The input voltage polarity is positive on the left and negative on the right. Switch S1 is constantly on, and switches S3 and S4 can be considered as the main switch and freewheeling transistor of the PFC circuit, respectively. During the negative half-cycle of the power frequency, the AC input source v... g The input voltage polarity is negative on the left and positive on the right. Switch S2 is always on, and switches S4 and S3 can be considered as the main switch and freewheeling transistor of the PFC circuit, respectively. However, due to the input inductance L... f The inductor current is in discontinuous current mode (DCM), and switches S3 and S4 are always turned on and off with a duty cycle of approximately 50%. Therefore, this has no impact on the operation of the LLC. (Refer to...) Figure 4 , Figure 4 This is a waveform diagram of a bridgeless PFC single-stage LLC resonant converter in an embodiment of this application, where i Lf For the input inductor current, i Lr For the resonant inductor current, i Lm For the excitation inductor current, taking the positive half-cycle of the power frequency as an example, the bridgeless PFC single-stage LLC resonant converter includes the following operating modes:

[0102] Mode 1: In this state, both switches S3 and S4 are off, and the AC input source v g By input inductor L f The body diode or parasitic capacitance path of the constantly conducting switching transistors S1 and S3 affects the DC bus capacitance C. bus An initial pre-charge is performed; this is a passive charging process, lasting extremely shortly. Its main function is to prepare the battery for charging before the high-frequency switching action begins, at point C. bus A basic DC voltage is established across the two ends to provide the necessary energy reserve for the normal startup of the LLC resonant converter and to avoid excessive inrush current during zero-voltage startup. The equivalent circuit diagram of the converter is shown below. Figure 5 As shown.

[0103] Mode 2: In this state, switch S3 is on and switch S4 is off. The circuit simultaneously performs two processes: PFC boost energy storage and LLC energy transfer. In the PFC path, the AC input source v... g A positive voltage is applied to the input inductor L f Above, its current i Lf The energy from the grid rises linearly with an approximately constant slope, storing it in the form of a magnetic field in the input inductor L. f In the LLC path, the DC bus voltage C bus Through the conducting switch S3 and resonant inductor L r The primary winding of the transformer charges the resonant capacitor Cr, at which point the resonant inductor current i Lr Greater than the magnetizing inductor current i Lm The transformer couples energy from the primary side to the secondary side. The synchronous rectifier diode SR2 on the secondary side senses a positive voltage and turns on, supplying power to the load and the output filter capacitor C. o The equivalent circuit diagram of the converter for charging is as follows: Figure 6 As shown.

[0104] Mode 3: In this state, switch S3 remains on and switch S4 remains off. The energy storage process of the PFC section is exactly the same as in Mode 2, i.e., the input inductor current i... Lf It continues to increase linearly. However, in the LLC section, as resonance progresses, the resonant inductor current i... Lr The magnitude has decayed to less than the excitation inductor current i Lm This causes the current directions on the primary and secondary sides of the transformer to reverse or decrease to zero, thus stopping the transformer from transferring energy to the secondary side. At this time, the current in the synchronous rectifier diode SR2 on the secondary side naturally drops to zero, and the controller can precisely turn it off at this moment, achieving zero-current turn-off and greatly reducing its turn-off losses. At this moment, the energy of the LLC resonant cavity is mainly in the resonant inductor L. r Resonant capacitor Cr And excitation inductance L m Internal exchanges occur between them, while the output depends entirely on the output filter capacitor C. o The equivalent circuit diagram of the converter that supplies power to the load is as follows: Figure 7 As shown.

[0105] Mode 4: In this state, both switches S3 and S4 are off, and the circuit enters the dead time. After switch S3 is turned off, the PFC path stores data in the input inductor L. f The energy in the circuit is freewheeled through the body diodes of switches S1 and S4, and then flows to the DC bus capacitor, i.e., the output capacitor C. bus Charging causes current i Lf It decreases at a constant slope. Simultaneously, the energy stored in the LLC resonant cavity begins to charge and discharge the junction capacitance of the PFC and LLC multiplexed high-frequency bridge arm, and the resonant inductor current i... Lr This process charges the parasitic capacitance of switch S3, causing its terminal voltage to rise rapidly, and simultaneously discharges the parasitic capacitance of switch S4, causing its terminal voltage to drop rapidly. This creates the necessary conditions for switch S4 to achieve zero-voltage turn-on in the next stage. The equivalent circuit diagram of the converter is shown below. Figure 8 As shown.

[0106] Mode 5: In this state, switch S3 is off and switch S4 is on. Thanks to the complete discharge of the parasitic capacitance of switch S4 in Mode 4, the voltage across its drain and source is zero when the drive signal arrives, thus achieving zero-voltage turn-on and almost zero switching losses. Input inductor L f The energy that has not yet been fully released continues to flow to the output capacitor C. bus Charge until current i Lf The current drops to zero. Simultaneously, the LLC resonant cavity begins its second half-cycle of energy transfer, and the resonant inductor current i... Lf The current flows in reverse, transferring energy to the secondary side through the transformer. At this time, the synchronous rectifier diode SR1 senses a forward voltage and conducts, supplying power to the load R. o Power supply and output filter capacitor C o The equivalent circuit diagram of the converter for charging is as follows: Figure 9 As shown.

[0107] Mode 6: In this state, switch S3 remains off and switch S4 remains on. Since the PFC of this converter operates in discontinuous mode, at the beginning or during this stage, the input inductor current i Lf The current has dropped completely to zero, and the PFC section has finished operating in this cycle. The LLC resonant section has entered a resonant state similar to mode 3, with the resonant inductor current i... Lr The magnitude decreases again to less than the magnetizing inductor current iLm This causes the transformer to stop transmitting energy, and the current in the synchronous rectifier diode SR1 drops to zero. The controller then turns it off at this moment to achieve zero-current shutdown. The output of the entire system is again entirely controlled by the output filter capacitor C. o The equivalent circuit diagram of its converter is as follows: Figure 10 As shown.

[0108] Mode 7: In this state, switch S3 remains off, switch S4 is off, and after switch S4 is turned off, the resonant inductor current i flowing in the LLC resonant cavity... Lr The reverse charging and discharging operation is initiated on the junction capacitance of the PFC and LLC multiplexed high-frequency bridge arm. Specifically, the resonant inductor current i Lr The parasitic capacitance of switch S4 is charged, causing its terminal voltage to rise from zero. Simultaneously, the parasitic capacitance of switch S3 is discharged, causing its terminal voltage to rise from the DC bus voltage value, i.e., the output capacitor C. bus The voltage value begins to decrease, thus reducing the voltage across switch S3 to zero before the start of the next switching cycle. This creates the perfect prerequisite for switch S3 to achieve zero-voltage turn-on at time t7, thereby completing a full high-frequency switching cycle. During this period, no current flows through the PFC section, and the converter output continues to be supplied by the output filter capacitor C. o The converter is independently powered to the load, and its equivalent circuit diagram is as follows: Figure 11 As shown.

[0109] Based on the above embodiments, as an optional embodiment, the control method further includes:

[0110] Using magnetic integration technology to integrate the input inductor L f Resonant inductor L r The transformer is integrated onto the same EEI-type composite magnetic core, and a corresponding magnetic circuit model is established, wherein the input inductance L... f The primary and secondary windings of the transformer are wound on the central column of the first E-type magnetic core, and the central column of the first E-type magnetic core is provided with a first air gap. The secondary windings of the transformer are both wound on the central column of the second E-type magnetic core, and the central column of the second E-type magnetic core is provided with a second air gap.

[0111] Specifically, to address the power density bottleneck caused by incomplete magnetic integration, the original three independent magnetic components, namely the input inductor L... f Resonant inductor L r The transformer is integrated onto an EEI-type composite magnetic core, eliminating redundant skeleton, shell, and some core material. The magnetic integration structure diagram of this EEI-type composite magnetic core is shown below. Figure 12 As shown, this composite magnetic core structure combines two E-type magnetic cores with one I-type magnetic core to form a shared magnetic circuit system. The input inductance L of the PFC section...f Right now Figure 5 N in PFC The winding is wound on the center post of the first E-type magnetic core, and a single-layer flat winding method can be used. The primary winding N of the LLC section transformer... P and secondary winding N S1 N S2 It is then wound on the central post of the second E-type magnetic core, using a multi-layer wrapping winding method, i.e., a sandwich structure, to enhance coupling. To achieve effective decoupling and independent control of the PFC inductor function and the LLC transformer function, a first air gap and a second air gap are respectively opened on the central posts of the two E-type magnetic cores. The core function of the first air gap is to adjust the input inductance L. f The inductance value is adjusted to prevent magnetic saturation under high input current, ensuring stable operation of the PFC stage, while the second air gap is used to adjust the transformer's magnetizing inductance L. m The inductance value is adjusted to prevent transformer core saturation, thereby ensuring that the LLC resonant cavity operates in the preset optimal resonance state.

[0112] Based on the above embodiments, as an optional embodiment, the resonant inductor L r It is formed by the leakage inductance between the primary and secondary windings of the transformer.

[0113] Specifically, leakage inductance is an inherent physical characteristic formed because the magnetic flux generated by the primary winding of a transformer cannot be completely coupled to the secondary winding. This leakage inductance can be actively controlled and utilized by designing the transformer winding structure, such as using a sandwich structure with the primary and secondary windings wound in layers, or adjusting the thickness of the insulation layer between them, so that its value is equal to the resonant inductance L required for the LLC resonant cavity design. r Value. In this way, a discrete, physical resonant inductor component is no longer needed, which greatly reduces the number of magnetic components, lowers the footprint of the circuit board and the manufacturing cost, and is a key technical means to achieve high integration and miniaturization of the converter.

[0114] Based on the above embodiments, as an optional embodiment, the secondary winding has a center tap, the two ends of the secondary winding constitute the first output terminal and the third output terminal of the transformer, respectively, and the center tap constitutes the second output terminal of the transformer.

[0115] Specifically, to optimize the efficiency of the secondary rectifier circuit, the transformer's secondary winding is designed with a center tap. This means that a tap is drawn at the physical and electrical center point of the secondary winding, and the two ends of the secondary winding, together with this center tap, serve as the transformer's output. In practical connection, the two ends of the secondary winding are connected to synchronous rectifier diodes SR1 and SR2, respectively, while the center tap serves as the positive output terminal of the rectifier circuit. This configuration constitutes a highly efficient full-wave rectifier circuit. Its advantage lies in the voltage drop loss of only one synchronous rectifier diode per conduction cycle, reducing conduction losses by half compared to a full-bridge rectifier circuit. Furthermore, full-wave rectification requires only two switching transistors, simplifying the design and drive control of the secondary circuit. It is suitable for applications with low output voltage and high output current, significantly improving overall efficiency. (Refer to...) Figure 13 , Figure 13 This is a schematic diagram of the magnetic integrated circuit of the bridgeless PFC single-stage LLC resonant converter in the embodiments of this application.

[0116] Please refer to Figure 3 and Figure 13 , Figure 3 This is a circuit diagram of the bridgeless PFC single-stage LLC resonant converter in an embodiment of this application. Figure 13 This is a schematic diagram of the magnetic integrated circuit of the bridgeless PFC single-stage LLC resonant converter in the embodiments of this application.

[0117] Based on the above embodiments, as an optional embodiment, the source of synchronous rectifier SR1 is connected to the source of synchronous rectifier SR2.

[0118] Specifically, this common-source configuration allows the sources of the two synchronous rectifier diodes SR1 and SR2 to jointly form the common ground or negative output terminal of the secondary rectifier circuit. This allows the drive circuits driving these two synchronous rectifier diodes (N-channel MOSFETs) to share the same reference ground, greatly simplifying the design complexity and reducing implementation costs. Compared to the common-drain structure requiring floating ground drive, the common-source configuration allows for the use of simpler, cheaper, and faster ground drive chips or circuits, while also improving the anti-interference capability of the drive signal and the reliability of the entire secondary rectifier system.

[0119] Based on the above embodiments, as an optional embodiment, an RC filter circuit disposed between the first output terminal and the second output terminal is also included.

[0120] Specifically, to further suppress output voltage ripple and high-frequency noise, an RC filter circuit is set between the first and second output terminals. The RC filter circuit typically consists of a filter resistor and a filter capacitor connected in series or parallel. By utilizing the low impedance of the capacitor to high-frequency signals and the energy dissipation of the resistor, it provides additional absorption and attenuation of high-frequency switching noise caused by the switching action of the switching transistor and synchronous rectifier diode, as well as parasitic capacitance. This filter structure can effectively filter out residual high-frequency components in the circuit, thereby providing a cleaner and more stable DC output voltage, which is crucial for powering precision electronic equipment or loads with stringent power quality requirements.

[0121] It should be understood that the sequence number of each step in the above embodiments does not imply the order of execution. The execution order of each process should be determined by its function and internal logic, and should not constitute any limitation on the implementation process of the embodiments of this application.

[0122] Those skilled in the art will clearly understand that, for the sake of convenience and brevity, the above-described division of functional units and modules is used as an example. In practical applications, the above functions can be assigned to different functional units and modules as needed, that is, the internal structure of the device can be divided into different functional units or modules to complete all or part of the functions described above.

[0123] The above-described embodiments are only used to illustrate the technical solutions of this application, and are not intended to limit them. Although this application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of this application, and should all be included within the protection scope of this application.

Claims

1. A control method for a bridgeless PFC single-stage LLC resonant converter, characterized in that, A control method for a bridgeless PFC single-stage LLC resonant converter includes: Collect output voltage V o Calculate the output voltage V o With preset output reference voltage V oref While measuring the difference V1, the bus voltage V is also collected. Cbus Calculate the bus voltage V Cbus With the preset bus reference voltage V Cbusref The difference V2; The difference V1 is input to the first PI controller for proportional-integral control to obtain the first output frequency f1. At the same time, the difference V2 is input to the second PI controller for proportional-integral control to obtain the second output frequency f2. The first output frequency f1 and the second output frequency f2 are weighted to obtain the actual control frequency f; The actual control frequency f is input to the drive signal generator to obtain the control signal for switch S3 and the control signal for switch S4.

2. The control method according to claim 1, characterized in that, The weighted processing of the first output frequency f1 and the second output frequency f2 to obtain the actual control frequency f specifically includes: Determine the bus voltage V Cbus Is it greater than the preset bus reference voltage V? Cbusref ; If the bus voltage V Cbus Greater than the preset bus reference voltage V Cbusref Then, using the formula f = f1 + K p The actual control frequency f is calculated using f2, where K p The preset emergency coefficient is used to adjust the influence weight of the bus voltage, f is the actual control frequency, f1 is the first output frequency, and f2 is the second output frequency; If the bus voltage V Cbus Not greater than the preset bus reference voltage V Cbusref The actual control frequency f can be calculated using the formula f = f1.

3. The control method according to claim 1, characterized in that, The bridgeless PFC single-stage LLC resonant converter includes: The circuit consists of a pre-amplifier input circuit, a transformer, and a secondary rectifier circuit. The pre-amplifier input circuit includes switching transistors S1, S2, S3, and S4. The source of switching transistor S1 and the drain of switching transistor S2 are connected to the AC input source V. g The positive terminal of the switch S3 and the drain of the switch S4 are connected to the AC input source v. g The negative electrode; The first output terminal of the preamplifier input circuit is connected to the first input terminal of the transformer, and the second output terminal of the preamplifier input circuit is connected to the second input terminal of the transformer. The secondary rectifier circuit includes synchronous rectifier diodes SR1 and SR2. The drain of synchronous rectifier diode SR1 is connected to the first output terminal of the transformer, and the drain of synchronous rectifier diode SR2 is connected to the third output terminal of the transformer. In the bridgeless PFC single-stage LLC resonant converter, the second output terminal of the transformer is the first output terminal, and the source of the synchronous rectifier SR2 is the second output terminal.

4. The control method according to claim 3, characterized in that, The pre-amplifier input circuit also includes: Output capacitor C bus and resonant capacitor C r ; The drain of the switching transistor S1 is connected to the drain of the switching transistor S3, and the output capacitor C bus One end is connected to the connection point between the drain of the switching transistor S1 and the drain of the switching transistor S3, and the source of the switching transistor S2 and the source of the switching transistor S4 are connected. The output capacitor C bus The other end is connected to the junction of the source of the switching transistor S2 and the source of the switching transistor S4; The resonant capacitor C r One end is connected to the source of the switching transistor S2, the source of the switching transistor S4, and the output capacitor C. bus The resonant capacitor C is connected at the connection point. r The other end is connected to the second input terminal of the transformer.

5. The control method according to claim 4, characterized in that, The pre-amplifier input circuit also includes: Input inductance L f Resonant inductor L r and transformer magnetizing inductance L m ; The source of the switching transistor S1 and the drain of the switching transistor S2 are connected through the input inductor L. f Connect AC input source v g The positive electrode; The first output terminal of the pre-amplifier input circuit is connected to the resonant inductor L. r Connected to the first input terminal of the transformer; The transformer magnetizing inductance L m One end is connected to the resonant inductor L r The transformer is connected to the first input terminal of the transformer, and the transformer's magnetizing inductance L m The other end is connected to the resonant capacitor C r Connect to the second input terminal of the transformer.

6. The control method according to claim 5, characterized in that, The control method further includes: The input inductor L is integrated using magnetic integration technology. f The resonant inductor L r The transformer is integrated onto the same EEI-type composite magnetic core, and a corresponding magnetic circuit model is established, wherein the input inductance L... f The primary and secondary windings of the transformer are both wound on the central column of a second E-type magnetic core, and the central column of the first E-type magnetic core is provided with a first air gap.

7. The control method according to claim 6, characterized in that, The resonant inductor L r It is formed by the leakage inductance between the primary and secondary windings of the transformer.

8. The control method according to claim 6, characterized in that, The secondary winding has a center tap, and the two ends of the secondary winding respectively constitute the first output terminal and the third output terminal of the transformer, and the center tap constitutes the second output terminal of the transformer.

9. The control method according to claim 6, characterized in that, The source of the synchronous rectifier SR1 is connected to the source of the synchronous rectifier SR2.

10. The control method according to claim 6, characterized in that, It also includes an RC filter circuit disposed between the first output terminal and the second output terminal.

Images