Wireless ranging method and device for power line carrier dual-mode communication system

Abstract

The application discloses a wireless ranging method and device of a power carrier dual-mode communication system, and belongs to the technical field of power carrier communication. The method comprises the following steps: receiving wireless signals sent by a sending end successively, and performing frequency domain transformation on the wireless signals to obtain frequency domain response signals; obtaining an impulse response signal of the frequency domain response signals in a time domain; calculating a multi-path gravity center of the impulse response signal under a multi-path channel condition, and determining a synchronization timing deviation corresponding to the wireless signals based on a deviation between the multi-path gravity center and a main path; calibrating sampling clock frequencies of the sending end and the receiving end based on the synchronization timing deviations corresponding to the wireless signals received successively, and performing a ranging operation after the calibration to obtain a ranging result. The application can realize accurate and stable wireless ranging for each node in the power carrier dual-mode communication system.

Description

Technical Field

[0001] This invention belongs to the field of power line carrier communication technology, and particularly relates to a wireless ranging method and device for a power line carrier dual-mode communication system. Background Technology

[0002] In the Internet of Things (IoT) and smart grids, power line carrier dual-mode communication systems are commonly used in scenarios such as automatic meter reading, distribution network monitoring, and smart city terminal management. Wireless ranging between concentrators, agent nodes, and user terminals in a power line carrier dual-mode communication system can be used to calculate the coverage area of ​​power signals, optimize the network architecture, and improve communication efficiency.

[0003] However, in complex propagation environments such as cities, industrial parks, and indoor spaces, wireless signals are easily affected by various factors during transmission, and ranging accuracy still faces challenges. Summary of the Invention

[0004] This invention aims to at least solve one of the technical problems existing in related technologies. To this end, this invention proposes a wireless ranging method and apparatus for a power line carrier dual-mode communication system, so as to achieve accurate and stable ranging for each node in the power line carrier dual-mode communication system.

[0005] In a first aspect, the present invention provides a wireless ranging method for a power line carrier dual-mode communication system, the method comprising:

[0006] It receives wireless signals transmitted sequentially by the transmitting end and performs frequency domain transformation on the wireless signals to obtain a frequency domain response signal;

[0007] Obtain the impulse response signal in the time domain of the frequency domain response signal;

[0008] The multipath centroid of the impulse response signal is calculated under multipath channel conditions, and the synchronization timing deviation corresponding to the wireless signal is determined based on the deviation between the multipath centroid and the main path.

[0009] Based on the synchronization timing deviations corresponding to the successively received wireless signals, the sampling clock frequencies of the transmitting and receiving ends are calibrated, and a ranging operation is performed after calibration to obtain the ranging result.

[0010] In a second aspect, the present invention provides a wireless ranging device for a power line carrier dual-mode communication system, the device comprising:

[0011] The signal module is used to receive wireless signals transmitted sequentially by the transmitter and to perform frequency domain transformation on the wireless signals to obtain frequency domain response signals.

[0012] The processing module is used to acquire the impulse response signal of the frequency domain response signal in the time domain;

[0013] The calculation module is used to calculate the multipath centroid of the impulse response signal under multipath channel conditions, and determine the synchronization timing deviation corresponding to the wireless signal based on the deviation between the multipath centroid and the main path.

[0014] The ranging module is used to calibrate the sampling clock frequencies of the transmitting and receiving ends based on the synchronization timing deviations corresponding to the successively received wireless signals, and then perform ranging operations to obtain the ranging results.

[0015] Thirdly, the present invention provides an electronic device including a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein the processor executes the computer program to implement the wireless ranging method of the power line carrier dual-mode communication system as described in the first aspect above.

[0016] Thirdly, the present invention provides a non-transitory computer-readable storage medium having a computer program stored thereon, wherein the computer program, when executed by a processor, implements the wireless ranging method of the power line carrier dual-mode communication system as described in the first aspect above.

[0017] Fourthly, the present invention provides a chip comprising a processor and a communication interface, the communication interface being coupled to the processor, the processor being used to run computer programs or instructions to implement the wireless ranging method of the power line carrier dual-mode communication system as described in the first aspect above.

[0018] Fifthly, the present invention provides a computer program product, including a computer program that, when executed by a processor, implements the wireless ranging method of the power line carrier dual-mode communication system as described in the first aspect above.

[0019] The wireless ranging method, apparatus, electronic device, non-transitory computer-readable storage medium, chip, and computer program product of the power line carrier dual-mode communication system provided by the present invention receive wireless signals transmitted sequentially by the transmitting end and perform frequency domain transformation, which can fully utilize the frequency domain characteristics of the signal and improve the extraction accuracy of signal amplitude and phase information; then, the frequency domain response signal is converted into a time domain impulse response signal in the time domain to fully observe the time domain characteristics of the signal under multipath channel conditions; under the multipath channel conditions of the signal, for the wireless signal in the time domain, the multipath centroid of the time domain impulse response signal on multiple propagation paths is calculated, and the synchronization timing deviation of the wireless signal is obtained from the deviation of the calculated multipath centroid from the main path. Under the condition of weakened multipath interference, the actual arrival time of the wireless signal is accurately calculated, providing a basis for clock calibration and wireless ranging calculation at the transmitting and receiving ends; finally, the sampling clock frequency of the transmitting and receiving ends is dynamically calibrated using the synchronization timing deviation of the two sets of wireless signals received sequentially, effectively eliminating the cumulative error caused by multipath effect and clock drift, significantly improving the overall accuracy and stability of ranging, and is particularly suitable for high-noise and strong multipath scenarios of power line carrier dual-mode communication systems. After clock calibration, ranging operations are performed on nodes in power line carrier dual-mode communication, and multiple ranging results are processed by a filter to finally obtain accurate and reliable ranging results.

[0020] Additional aspects and advantages of the invention will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. Attached Figure Description

[0021] The above and / or additional aspects and advantages of the present invention will become apparent and readily understood from the description of the embodiments taken in conjunction with the following drawings, in which:

[0022] Figure 1 This is a flowchart illustrating the wireless ranging method of a power line carrier dual-mode communication system provided in some embodiments of the present invention;

[0023] Figure 2 This is a schematic diagram of cyclic shifting in the wireless ranging method of the power line carrier dual-mode communication system provided in some embodiments of the present invention;

[0024] Figure 3 This is a phase frequency response curve of the infinite impulse response filter of the wireless ranging method for a power line carrier dual-mode communication system provided in some embodiments of the present invention.

[0025] Figure 4 This is a signal subcarrier phase distribution diagram of the wireless ranging method for a power line carrier dual-mode communication system provided in some embodiments of the present invention;

[0026] Figure 5This is a schematic diagram of the signal compensation phase of the wireless ranging method for a power line carrier dual-mode communication system provided in some embodiments of the present invention;

[0027] Figure 6 This is a phase distribution diagram of the signal subcarrier after compensation in the wireless ranging method of the power line carrier dual-mode communication system provided in some embodiments of the present invention;

[0028] Figure 7 This is a schematic diagram of the structure of the wireless ranging device of the power line carrier dual-mode communication system provided in some embodiments of the present invention;

[0029] Figure 8 This is a schematic diagram of the structure of an electronic device provided in some embodiments of the present invention. Detailed Implementation

[0030] The technical solutions of the embodiments of the present invention will be clearly described below with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of the present invention. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention are within the scope of protection of the present invention.

[0031] Unless otherwise defined, all technical and scientific terms used in this invention have the same meaning as commonly understood by one of ordinary skill in the art to which this invention pertains; the terminology used in the specification of this application is for the purpose of describing particular embodiments only and is not intended to limit the invention; the terms "comprising" and "having," and any variations thereof, in the specification, claims, and foregoing drawings are intended to cover non-exclusive inclusion. The terms "first," "second," etc., in the specification, claims, or foregoing drawings are used to distinguish different objects, not to describe a particular order or hierarchy.

[0032] In this invention, the reference to "embodiment" means that a specific feature, structure, or characteristic described in connection with an embodiment may be included in at least one embodiment of the invention. The appearance of this phrase in various places in the specification does not necessarily refer to the same embodiment, nor is it a separate or alternative embodiment mutually exclusive with other embodiments. It will be explicitly and implicitly understood by those skilled in the art that the embodiments described in this invention can be combined with other embodiments.

[0033] In this invention, the term "and / or" is merely a description of the relationship between related objects, indicating that three relationships can exist. For example, A and / or B can represent: A existing alone, A and B existing simultaneously, or B existing alone. Additionally, in this invention, the character " / " generally indicates that the preceding and following related objects have an "or" relationship.

[0034] In this invention, "multiple" refers to two or more (including two), similarly, "multiple groups" refers to two or more (including two groups), and "multiple pieces" refers to two or more (including two pieces).

[0035] In smart grids, smart meters often have multiple complex and diverse nodes, including, but not limited to, concentrator terminals, agent nodes, and user terminals. The distance between these nodes plays a crucial role in power line carrier dual-mode communication. On one hand, the distance between nodes helps determine the coverage area of ​​the power line carrier signal, optimize the network topology, and improve communication efficiency. On the other hand, accurate measurement of the distance between nodes facilitates timely and rapid location and fault analysis of faulty nodes.

[0036] In power line carrier dual-mode communication systems, communication links often suffer from complex multipath channels and sampling clock deviations, leading to significant ranging errors. Existing methods mostly employ a single phase compensation strategy, neglecting the impact of multipath interference on synchronization timing, and lack an effective sampling clock frequency calibration mechanism. Therefore, ranging accuracy is low in complex power environment conditions.

[0037] In view of this, embodiments of the present invention provide a wireless ranging method for a power line carrier dual-mode communication system. Based on multipath noise suppression, nonlinear phase compensation and sampling clock frequency calibration, it can achieve high-precision and low-error wireless ranging in scenarios with strong multipath interference, high noise and complex power environment.

[0038] The wireless ranging method of the power line carrier dual-mode communication system provided by the present invention will be described in detail below with reference to the accompanying drawings, through specific embodiments and application scenarios.

[0039] The wireless signals used in this embodiment of the invention are employed for inter-node communication and signal transmission in a power line carrier dual-mode communication system. Specifically, these wireless signals include Orthogonal Frequency Division Multiplexing (OFDM) signals, as well as OFDM variant signals that have the same or similar signal architecture as OFDM signals and are adaptable to the method of this invention, achieving the technical effects achievable by this invention. These wireless signals enable communication connections and signal transmission between nodes in the power line carrier dual-mode communication system, providing a basic signal carrier for the implementation of the wireless ranging method in the power line carrier dual-mode communication system.

[0040] The wireless ranging method for a power line carrier dual-mode communication system provided in this embodiment of the invention can be executed by an electronic device or a functional module or entity in an electronic device that can implement the wireless ranging method of the power line carrier dual-mode communication system.

[0041] The electronic device can be a power line communication terminal, a concentrator device, a smart meter, a communication module embedded device, or a communication test terminal, etc. Alternatively, the electronic device can also be a device with computing capabilities or an intelligent robot, used to execute the processing steps of the wireless ranging method of the power line carrier dual-mode communication system in the embodiments of the present invention.

[0042] The following describes the wireless ranging method of the power line carrier dual-mode communication system provided in this embodiment of the invention, taking the receiving electronic device as the execution subject as an example.

[0043] Figure 1 This is a flowchart illustrating the wireless ranging method of a power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 1 As shown, the wireless ranging method of the power line carrier dual-mode communication system includes steps 110 to 140.

[0044] Step 110: Receive the wireless signals sent sequentially by the transmitting end, and perform frequency domain transformation on the wireless signals to obtain the frequency domain response signal.

[0045] In a dual-mode communication system, any node can be selected as the transmitter, and the corresponding node that receives the wireless signal can be selected as the receiver.

[0046] The transmitting end sends two sets of known modulated wireless signals sequentially. The communication mode in the dual-mode communication system can be, for example, high-speed power line communication (HPLC) or high-speed radio frequency (HRF).

[0047] In a dual-mode communication system, the signal transmitted by the transmitter typically includes a Short Training Field (STF) and a Long Training Field (LTF). The STF is used for synchronization between the receiver and transmitter, while the LTF is used for channel state estimation. The STF and LTF are transmitted sequentially. The receiver receives the complete wireless signal, comprising data including the STF and LTF, and stores them separately in its receive buffer.

[0048] Signal synchronization is a prerequisite for subsequent signal processing, and its accuracy directly determines the reliability of ranging, demodulation, and other stages. Therefore, the first step is to use the STF (Signal Frequency Response) from the complete wireless signal to synchronize the wireless signal and determine the starting position of valid data transmission. After completing the wireless signal synchronization, the LTF (Large Frequency Response) sequence is extracted from the complete wireless signal based on the determined starting position. The channel frequency domain response is then obtained through the LTF sequence, and stable channel characteristics are extracted from the signal.

[0049] It should be noted that in a complete wireless signal, both STF and LTF are known sequences. In some embodiments provided by this invention, the "wireless signal" used for frequency domain transformation, time domain transformation, and ranging refers to the LTF sequence in the complete wireless signal.

[0050] The receiving end sequentially receives these two sets of wireless signals within a preset time window. After receiving the signals, synchronization is required to determine the accurate signal reception time. Therefore, the synchronization result using the STF in the wireless signal directly affects the accuracy of ranging. During actual signal transmission, the presence of noise can lead to inaccurate synchronization results, affecting subsequent ranging results for the nodes.

[0051] Therefore, in order to extract more stable feature information from the received raw time-domain signal, the receiver further performs a frequency-domain transformation on the LTF sequences in these two sets of signals, converting them from the time domain to a frequency-domain response signal. If the LTF sequence consists of multiple OFDM symbols, after completing synchronization and extracting the LTF sequence, the receiver performs a frequency-domain transformation on each OFDM symbol in the sequence to obtain the frequency-domain response corresponding to each OFDM symbol; subsequently, a weighted average or arithmetic average is performed on the multiple independent frequency-domain responses to finally obtain the averaged frequency-domain response signal. Frequency-domain analysis can better utilize the structural characteristics of the signal in the frequency domain, which is helpful for subsequent synchronization timing calculations.

[0052] For example, frequency domain transformation can be implemented using Fast Fourier Transform (FFT) or other equivalent algorithms, and the specific method can be flexibly selected according to hardware resources and performance requirements.

[0053] Step 120: Obtain the impulse response signal of the frequency domain response signal in the time domain.

[0054] After obtaining the frequency domain response signal, the receiver converts the frequency domain response signal back to the time domain. This is because wireless signals are often affected by attenuation and interference as they pass through the channel, causing changes. Directly analyzing the received time domain signal makes it difficult to distinguish between the signal's inherent characteristics and the characteristics introduced by the wireless channel. This conversion effectively separates the two types of characteristics.

[0055] For example, time-domain transformation can be implemented by inverse fast Fourier transform (IFFT) or other equivalent algorithms, and the specific method can be flexibly selected according to hardware resources and performance requirements.

[0056] Step 130: Calculate the multipath centroid of the impulse response signal under multipath channel conditions, and determine the synchronization timing deviation corresponding to the wireless signal based on the deviation between the multipath centroid and the main path.

[0057] Multipath channel conditions refer to the channel environment state in which a wireless signal, during its transmission from the transmitter to the receiver, cannot propagate through a single direct path due to obstacles in the transmission environment (such as walls, the ground, etc.), but instead reaches the receiver through multiple different paths (such as direct, reflected, scattered, diffracted, etc.). Compared to power line channels, although power line channels are also subject to some interference, their transmission environment is relatively closed and stable, and the degree of multipath propagation is much lower than that of wireless channels.

[0058] In this regard, the superposition of multipath signals in a wireless channel complicates the time-domain characteristics of the received signal, making it difficult to extract the dominant path signal and the synchronization timing point, thus resulting in synchronization timing deviation. Therefore, when using wireless signals for ranging at nodes in a power line carrier dual-mode communication system, the impact of multipath effects in the wireless channel must be fully considered.

[0059] When the receiver divides the multiple propagation paths of the time-domain impulse response signal, it selects the path with the shortest propagation distance and the least attenuation as the primary path. Paths that experience reflection and scattering during transmission, resulting in longer propagation distances but still maintaining a certain amplitude, are designated as secondary paths.

[0060] In a power line carrier dual-mode communication system, synchronization timing deviation refers to the deviation between the start time of the sampling clock at the receiving end and the actual start time of the signal at the transmitting end. In other words, when synchronization timing deviation exists, the sampling window used by the receiving end to acquire the signal will deviate from the true time position of the signal.

[0061] The multipath centroid is the weighted average location of the energy of the multipath components in a multipath channel, comprehensively reflecting the center of the energy distribution of the multipath signal in the time domain. The multipath centroid can be obtained, for example, through an effective path that includes both the primary and secondary paths.

[0062] When the starting point of the sampling window deviates from the actual position of the main path, the time range of the multipath signal acquired by the receiver changes due to the offset of the sampling window. The centroid of the multipath, as the weighted average position of the multipath energy, will move accordingly with the offset of the sampling window. The magnitude of this movement is related to the time delay and energy of each multipath component.

[0063] Therefore, the deviation between the centroid of the multipath and the main diameter directly reflects the magnitude of the synchronization timing error. In other words, by observing the positional change of the centroid of the multipath relative to the main diameter, the synchronization timing error can be inferred.

[0064] Specifically, for the frequency domain response signals, the receiver performs time-domain transformation on multiple frequency domain response signals, converting the wireless signals in the frequency domain back to time domain signals to obtain the corresponding time-domain impulse response signals. Subsequently, the sampling point positions corresponding to the multipath centroid and the main path are calculated, and the actual propagation time of the signal from the transmitter to the receiver in the main path and multipath centroid is extracted, thereby calculating the synchronization timing deviation corresponding to each wireless signal.

[0065] Step 140: Based on the synchronization timing deviations corresponding to the successively received wireless signals, calibrate the sampling clock frequencies of the transmitting and receiving ends, and perform ranging operations after calibration to obtain the ranging results.

[0066] After calculating the synchronization timing deviation, the receiving end uses the synchronization timing deviation corresponding to the two wireless signals to dynamically calibrate the sampling clock frequencies of both the receiving and transmitting ends to ensure the accuracy and stability of subsequent ranging results. The sampling clock frequency refers to the clock signal frequency used by the transmitting and receiving ends for signal sampling, which is used to achieve signal transmission and reception synchronization and ranging at both ends.

[0067] Specifically, the receiver compares the synchronization timing deviations of the two sets of wireless signals received sequentially. Since the transmission time interval between the two signals is known, the receiver can calculate the frequency offset of its local sampling clock relative to the transmitter based on the difference in timing deviations.

[0068] Based on the calculation results, the receiver can choose different calibration strategies:

[0069] For devices that support hardware-level adjustment, the output frequency of the local oscillator can be directly corrected to keep the sampling clock at the receiving end synchronized with the clock at the transmitting end.

[0070] For devices with limited hardware resources or those that do not support frequency adjustment, software compensation can be used to effectively correct errors caused by clock skew by adjusting the position of sampling points, interpolation, or resampling. This adaptive calibration method based on synchronization timing deviation does not rely on additional hardware calibration signals, thus maintaining long-term ranging accuracy in dynamically changing wireless environments.

[0071] Therefore, by keeping the time base consistent with the receiver, the impact of sampling clock deviation on ranging accuracy can be effectively eliminated, significantly improving the accuracy and robustness of ranging.

[0072] After the sampling clock is calibrated, the receiver recalculates the propagation time of the wireless signal using the corrected time base, and obtains the actual distance between the transmitter and receiver accordingly.

[0073] The wireless ranging method for a power line carrier dual-mode communication system provided in this invention, by receiving and performing frequency domain transformation on wireless signals transmitted sequentially by the transmitting end, can fully utilize the frequency domain characteristics of the signals and improve the accuracy of extracting signal amplitude and phase information. The frequency domain response signal is then converted into a time-domain impulse response signal to fully observe the time-domain characteristics of the signal under multipath channel conditions. Under multipath channel conditions, for the wireless signal in the time domain, the multipath centroid of the time-domain impulse response signal on multiple propagation paths is calculated, and the deviation between the calculated multipath centroid and the main path is used to obtain the synchronization timing deviation of the wireless signal. Under weakened multipath interference, the actual arrival time of the wireless signal is accurately calculated, providing a basis for clock calibration and wireless ranging calculation at the transmitting and receiving ends. Finally, the synchronization timing deviation of the two sets of wireless signals received sequentially is used to dynamically calibrate the sampling clock frequencies of the transmitting and receiving ends, effectively eliminating the cumulative error caused by multipath effects and clock drift, significantly improving the overall accuracy and stability of ranging, and is particularly suitable for high-noise, strong multipath scenarios in power line carrier dual-mode communication systems. After clock calibration, ranging operations are performed on nodes in power line carrier dual-mode communication, and multiple ranging results are processed by a filter to finally obtain accurate and reliable ranging results.

[0074] In response to the characteristics of strong noise interference and abundant multipath components in power communication scenarios, the wireless ranging method of the power line carrier dual-mode communication system provided in this embodiment of the invention can maintain stable synchronization performance and high-precision ranging capability even under limited hardware conditions.

[0075] When power line carrier dual-mode communication systems are applied in urban and industrial park communication scenarios, the wireless signal propagates through numerous reflections, scattering, and diffraction paths due to the high-rise buildings and dense glass curtain walls in urban environments. These propagation paths have varying propagation lengths and times (i.e., delays), making it easy for the time-domain impulse response signal to separate in the time domain. If the multipath centroid of the original impulse response signal is directly calculated, it will be affected by the more distant secondary paths, resulting in a shift and failing to accurately reflect the signal's concentration area.

[0076] Based on this, in some embodiments, the multipath centroid of the impulse response signal is calculated under multipath channel conditions. The main contents include: cyclically shifting the impulse response signal to make the primary and secondary paths converge in the time domain; and calculating the multipath centroid based on the converged primary and secondary paths. The multipath centroid is used to characterize the concentration position of signal energy in the time domain to evaluate the impact of multipath effects on the synchronization timing position.

[0077] To address interference in the frequency domain response signal, the receiver performs a time-domain transformation on the compensated frequency domain response signal, converting it to the time domain to obtain the time-domain impulse response signal. At this point, the primary and secondary paths of the signal are distributed at different times in the time domain due to propagation delay differences. Therefore, a cyclic shift operation is performed on the compensated time-domain impulse response signal. Based on the time-domain position of the primary path, taking the propagation path corresponding to the energy peak as an example, by adjusting the time-domain starting point of the signal, the effective paths, including the primary and secondary paths, are clustered, reducing dispersion caused by time delay differences.

[0078] Signal energy is the cumulative signal amplitude over time, which can be represented by the amplitude of the pulse or the square of the pulse amplitude in the time-domain impulse response. Cyclic shift refers to using a fixed sampling point as the starting position and sequentially moving all sampling points of the entire signal sequence forward or backward a certain number of positions. Sampling points moved outside the sequence boundary re-enter from the other side of the sequence, thus clustering all sampling points within a range. This adjusts the relative positions of different paths in the signal in the time domain, causing the major and minor paths to converge in the time domain, facilitating subsequent processing. For example, Figure 2 This is a schematic diagram of cyclic shifting in the wireless ranging method of a power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 2 As shown, the horizontal axis represents the sampling points, and the vertical axis represents the signal energy value (the square of the pulse amplitude in the time-domain impulse response). Each vertical line in the figure corresponds to a propagation path, and the dot represents the energy peak point of the path. After cyclic shifting, the tallest vertical line in the figure is the main path, corresponding to the propagation path with the strongest energy and shortest delay. Several taller vertical lines to the right of the main path correspond to propagation paths with weaker energy and longer delays, while the scattered, low vertical lines at the bottom correspond to interference paths. After cyclic shifting, the main path is now located at the center, while the secondary paths converge towards the main path according to their energy values.

[0079] After this, the receiver calculates the multipath centroid based on the aggregated primary and secondary paths.

[0080] In the above embodiments, the signal energy of the effective path is enhanced by time-domain aggregation, improving the stability of channel characteristics and providing a more stable channel reference for synchronization timing deviation estimation and ranging algorithms. Specifically, cyclic shifting can fully capture the energy distribution characteristics of the primary and secondary paths, resulting in more stable synchronization timing estimation and improved ranging accuracy even under significant multipath interference.

[0081] In calculating the multipath centroid, the signal energy contained in the effective paths differs. To ensure the multipath centroid accurately reflects the concentrated position of wireless signal energy in the time domain and assesses the impact of multipath effects on synchronization timing, a suitable method is needed. The essence of the multipath centroid is the representation of the concentrated position of signal energy in the time domain, reflecting the dominant time domain position of the effective signal energy in the multipath channel, thereby assessing the impact of multipath on synchronization timing offset. Directly averaging all impulse response sampling points will confuse signal energy with noise energy on the effective paths, causing the centroid to deviate from the true dominant signal position.

[0082] To address the aforementioned issues, in some embodiments, the multipath centroid is calculated based on the aggregated primary and secondary paths. Effective paths are then selected from the aggregated primary and secondary paths, and the signal energy of the effective paths is obtained. Effective sampling points on the effective paths that are within the target range are selected, and the signal energy corresponding to sampling points outside the target range is set to zero. The signal energy of the effective paths within the target range is used as the weighting factor to perform a weighted average on the signal energy corresponding to each effective sampling point, thereby obtaining the multipath centroid.

[0083] The effective path refers to the set of propagation paths that include the primary path and secondary paths, carrying effective information about the wireless signal. Based on this, a target range is set for the effective path, and further wireless signal propagation paths that can carry the true information of the wireless signal are extracted.

[0084] The receiver performs signal energy detection on the compensated time-domain impulse response signal and identifies the sampling point with the highest signal energy as the principal path position. The signal energy corresponding to sampling points outside the target range on the effective path is set to zero. For the selected effective paths, the receiving end extracts the sampling point positions of each path. ( =1, 2, ... M ), M The number of effective paths within the window and the corresponding signal energy. The centroid of the multipath is calculated using the weighted average formula:

[0085] (1)

[0086] Among them, the molecule of formula (1) The sum of the products of the effective path positions and the signal energy, with the denominator being... The result represents the weighted center position of the signal energy of the effective paths within the window, where the total signal energy is the effective path signal energy. It's easy to understand that in calculating the multipath centroid, only sampling points within the target range are calculated; sampling points outside the range have zero intensity. The weight is the energy intensity; the greater the energy intensity of a sampling point, the greater its contribution to the multipath centroid. By setting the denominator to the total intensity within the range, a weighted average is achieved in the final result calculation.

[0087] It should be noted that the above values ​​are for illustrative purposes and are not intended to limit the invention.

[0088] In the above embodiments, by using the aggregated primary and secondary paths to filter effective paths and calculate the multipath centroid, the signal propagation paths containing effective information are effectively obtained. By using the signal energy of the effective paths, the effective sampling points corresponding to the effective paths are retained, limiting the range of effective paths participating in the calculation. This reduces the amount of computation while ensuring calculation accuracy, and removes information contained in paths irrelevant to the calculation. By performing weighted averaging on the signal energy corresponding to each effective sampling point, the calculation accuracy is further improved, avoiding deviation of the multipath centroid and ensuring more accurate multipath centroid calculation, thereby enhancing the robustness of subsequent time-domain correction and synchronous timing estimation.

[0089] In multipath centroid calculation, determining the effective path is the core basis for calculation. Using all effective paths may introduce interference and increase the computational load on the receiver. To avoid this, effective paths can be filtered. This allows for the extraction of signal propagation paths that have a real impact on wireless ranging in complex multipath environments, reducing the impact of interference and redundant information on multipath centroid calculation.

[0090] Therefore, in some embodiments, the effective path is selected from the aggregated primary and secondary paths by selecting a path within a preset window range centered on the primary path; or the path with signal energy greater than a preset threshold is determined as the effective path.

[0091] The receiving end first processes the signals from multiple propagation paths of the wireless signal under multipath propagation conditions, obtaining data such as the signal energy intensity information of the wireless signal on each propagation path and the time when the path signal arrives at the receiving end. Based on preset filtering rules, it selects the effective path. The time when the path signal arrives at the receiving end is also called the signal delay value on the corresponding propagation path.

[0092] The selection and rules for effective paths can be based on the main path with the strongest signal energy, extracting the time delay value of the wireless signal within the main path. Set a time window threshold , The setting can be based on the signal propagation period or on the maximum time delay value of the signal in each path.

[0093] Set the time window threshold to The process iterates through the primary and secondary paths, classifying paths with delay values ​​within a specified range as valid paths and those outside this range as interference paths—paths carrying insufficient effective information. Additionally, a threshold for signal energy value can be set, for example, based on channel noise levels or a preset threshold for minimum effective signal energy. Then, it iterates through the signal energy values ​​of each propagation path and selects those with energy values ​​greater than or equal to a preset threshold. The path is determined to be a valid path if it is less than or equal to a preset threshold. The path was determined to be an interference path.

[0094] It should be noted that the above effective path screening methods can be used individually or in combination.

[0095] After this, the receiving end integrates all the selected effective paths and records the time delay and energy values ​​of each effective path to obtain the effective path set, providing data for subsequent calculation of the multipath centroid.

[0096] In the above embodiments, by setting different effective path setting rules, based on time windows or energy thresholds, some effective paths are selectively screened out, paths with low contribution to the calculation are excluded, and signal propagation paths with high contribution to the multipath centroid of the calculated signal are retained. This reduces redundant information, lowers the computational load, and ensures the accuracy of multipath centroid calculation.

[0097] In power line carrier dual-mode communication systems, synchronization timing deviations can negatively impact signal transmission and reception to varying degrees, especially in multipath channels where the impact is more complex. To improve the accuracy of synchronization timing estimation, this invention determines the synchronization timing deviation based on the deviation between the multipath centroid and the main path, quantifying the degree of interference of multipath effects on the synchronization timing of dual-mode communication systems.

[0098] Therefore, in some embodiments, the synchronization timing deviation corresponding to the wireless signal is determined based on the deviation between the multipath centroid and the main path. This mainly includes: mapping the main path position and the multipath centroid position back to the original time axis, and calculating the sampling point offset between the mapped multipath centroid and the main path position; converting the sampling point offset into a time deviation according to the sampling time interval to obtain the synchronization timing deviation corresponding to the wireless signal.

[0099] The sampling point offset is determined based on the sampling time interval, i.e., the reciprocal of the sampling clock. It represents the deviation between the actual sampling time and the ideal sampling time at the receiving end, and is usually measured as a multiple of the sampling period or a specific time unit. When the multipath centroid is located after the main path, the sampling point offset is positive, indicating a synchronization lag; when the multipath centroid is located before the main path, the sampling point offset is negative, indicating an early synchronization; and when the sampling point offset is zero, it indicates an accurate synchronization.

[0100] In a power line carrier dual-mode communication system, the offset of the sampling point manifests as a synchronization timing deviation. The relationship between the deviation between the multipath centroid and the main path and the synchronization timing deviation in some embodiments of this invention is explained below.

[0101] Let the synchronization timing deviation be... b When the sampling window of the receiving end is precisely aligned with the main path, it is considered that there is no synchronization timing deviation. b =0. Let the position of the principal diameter be... , representing the actual time delay of the primary path. The location of the multipath centroid is... , representing the weighted average delay of multipath propagation, is determined by the distribution of the multipath paths. Here, delay refers to the actual time it takes for a signal emitted from the transmitter to reach the receiver after traveling through different energy levels and propagation paths.

[0102] Calculate the difference between the actual delay of the primary path and the weighted average delay of the multipath. At this point, the difference is only related to the multipath channel distribution characteristics, and not to the synchronization timing deviation. b Irrelevant.

[0103] When there is a synchronization timing deviation, if b >0 indicates that sampling is too early, and the overall sampling window is advanced; if b A value less than 0 indicates that sampling is too late, the overall sampling window is lagging, and the sampling window is offset relative to the principal diameter. b Units. At this point, the position of the main diameter within the window becomes... The multipath centroid is a weighted average of the multipath energies. When the sampling window shifts, the magnitude of the multipath centroid's movement depends on the distribution of the multipaths, expressed by a coefficient. k Measurement, among which, Therefore, the location of the centroid of the multipath is... The difference between the position of the main diameter and the position of the centroid of the multi-diameter is calculated using the following formula:

[0104] (2)

[0105] It should be noted that the coefficient The physical meaning of this is the degree to which the multipath centroid follows the synchronization deviation. Specifically, the more the energy distribution of the multipath is biased towards one side, the weaker the centroid's following ability to the sampling window. Conversely, if the energy distribution of the multipath is balanced, the centroid's following ability to the sampling window is relatively strong, ultimately leading to... At this point, the difference directly reflects the synchronization timing deviation.

[0106] If the wireless signal is in an outdoor environment, the multipath propagation is dominated by the reflection path behind the main path. For example, in the outdoor environment, the main path is direct, while the multipath propagation is caused by reflections from buildings, resulting in greater time delay. In this case, the center of gravity of the multipath is closer to the rear of the main path. When the offset of the sampling window is... At this time, the center of gravity hardly moves. This is because the energy of the multipath signal is concentrated at the rear, and the window offset has a relatively small impact on the center of gravity. Difference If the energy of the front and rear parts of the multipath signal is balanced, the centroid of the multipath is more sensitive to the movement of the sampling window. , .

[0107] When using a power line carrier dual-mode communication system for wireless ranging, the use of filters causes a nonlinear phase deviation in the wireless signal. To compensate for this phase deviation, nonlinear phase compensation can be used, depending on the system's specific requirements and the accuracy specifications.

[0108] Specifically, the receiver shifts the entire time-domain impulse response signal so that the energy peaks of the primary and secondary path signals are as close as possible to the sampling point corresponding to the multipath centroid. Based on the multipath centroid and the characteristics of each effective path, the required phase adjustment value is calculated.

[0109] When the synchronization timing deviation corresponds to 1 / n The phase deviation between adjacent subcarriers is calculated using the following formula for each sampling point. :

[0110] (3)

[0111] in, The interval between adjacent subcarriers, The baseband sampling frequency, n Let be the sampling point scaling factor corresponding to the synchronization timing deviation. Let the total number of samples obtained be... k A phase deviation, using the following formula for k The phase deviations are averaged for calculation:

[0112] (4)

[0113] in, The average phase deviation is used to calculate the required correction for the sample point offset using the following formula:

[0114] (5)

[0115] in, This is the sampling point offset that needs to be corrected. The sampling point offset can be an integer or a non-integer, for example, it could be 0.3 sampling points. For instance, if... =0.3, then the sampling time of the receiving end's sampling clock is adjusted forward or backward by 0.3 sampling periods. Where, when When the value is positive, the corresponding sampling is too late, and it needs to be adjusted forward by 0.3 sampling periods. When the value is negative, the sampling is too early and needs to be adjusted backward by 0.3 sampling periods. Through the above calculation, nonlinear phase compensation can be achieved.

[0116] After calculating the sampling point offset, the sampling point offset is calculated. The product of the product and the sampling time interval yields the corresponding synchronization timing deviation, where the sampling time interval is... ,Right now:

[0117] (6)

[0118] It should be noted that the above values ​​are examples and are not intended to limit the present invention.

[0119] In the above embodiments, by correlating the synchronization positioning time difference with the deviation between the multipath centroid and the main path, and through conversion such as sampling point offset, the deviation between the multipath centroid and the main path is converted into a time quantization value that can be directly used for synchronization correction. The receiving end can directly adjust the sampling time based on this. This method is intuitive and easy to operate, requires no complex model fitting, and the synchronization timing deviation obtained by this method is more accurate and stable, which can further improve the effect of subsequent processing.

[0120] The nonlinear phase compensation of signals in some embodiments of the present invention will be described below with reference to the accompanying drawings. Figure 3 This is a phase frequency response curve of the infinite impulse response filter in the wireless ranging method of the power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 3 As shown, the horizontal axis represents the frequency of the input signal, measured in Hertz (Hz), and the signal range is 0 to 3. 10 5The vertical axis represents the phase shift of different frequency components of the signal after passing through the infinite impulse response filter, measured in radians (rad). As can be observed from the graph, the phase changes at different frequencies are not linear. Specifically, the closer to zero frequency, the more linear the phase characteristics; however, as the frequency increases and moves further away from zero frequency, the degree of phase distortion becomes increasingly severe. This is because when a filter is deployed in the system, directly using the average phase difference of all adjacent subcarriers of the OFDM signal to estimate the timing synchronization deviation introduces a large error because the phase of the edge subcarriers has already been significantly distorted due to the filter characteristics. Therefore, when using a system with a deployed filter for wireless ranging, nonlinear phase compensation can be used to reduce the error.

[0121] Furthermore, Figure 4 This is a signal subcarrier phase distribution diagram of the wireless ranging method for a power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 4 As shown, the horizontal axis represents the subcarrier number, ranging from 0 to 70. The vertical axis represents the phase of each subcarrier, in radians (rad). The two sides of the figure represent empty subcarriers, which can be ignored. Effective subcarriers are concentrated in positions 7 to 59. The phase distribution shows that the phase changes of adjacent subcarriers caused by timing synchronization deviations are not ideally linear, with particularly significant phase distortion in edge subcarriers. This distortion characteristic closely matches the phase response curve of the filter, essentially representing interference from the filter's nonlinear phase on the signal.

[0122] Figure 5 This is a schematic diagram of the signal compensation phase of the wireless ranging method for a power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 5 As shown in the figure, the horizontal axis represents the frequency of the input signal, measured in Hertz (Hz), and the signal range is from 0 to 3 Hz. 10 5 The vertical axis represents phase, measured in radians (rad). Lines ending in rectangles represent the nonlinear phase response of the original filter, exhibiting the filter's inherent phase distortion characteristics. Lines ending in triangles represent the compensated target phase, while lines ending in circles represent the compensated phase value curve, which is the correction amount required to counteract the filter's nonlinear effects and restore the phase to the target state.

[0123] Figure 6 This is a phase distribution diagram of the signal subcarrier after compensation in the wireless ranging method of the power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 6As shown in the figure, the horizontal axis represents the subcarrier index, ranging from 0 to 70. The vertical axis represents the phase, in radians (rad). After phase compensation, by correcting the nonlinear phase deviation introduced by the filter, the phases of all effective subcarriers 7-59 are linearly correlated and can fully participate in the calculation of the average phase difference between adjacent subcarriers, expanding the amount of available data and making the estimation of synchronization timing deviation more accurate and stable.

[0124] In dual-mode wireless ranging systems, the receiver and transmitter typically use independent local sampling clocks. Due to factors such as hardware accuracy, temperature variations, or long-term operation, their sampling clock frequencies are difficult to synchronize perfectly. Even if the frequency deviation is small, over time, the receiver will develop systematic errors when calculating the arrival time of the wireless signal, leading to inaccurate synchronization timing deviation calculations and consequently affecting the accuracy of the ranging results.

[0125] To address this issue, this invention utilizes the synchronization timing deviation between two sets of successively received wireless signals to accurately estimate the sampling clock frequency deviation between the receiver and transmitter, and dynamically calibrates the receiver's sampling clock frequency based on this, thereby effectively suppressing the cumulative error caused by clock drift.

[0126] Therefore, in some embodiments, the sampling clock frequency of the receiving end is calibrated based on the synchronization timing deviations corresponding to the successively received wireless signals. The receiving time interval between the successively received wireless signals is determined based on the synchronization timing deviations corresponding to the successively received wireless signals; the transmitting time interval between the successively transmitted wireless signals is obtained, and the sampling clock frequency deviation between the receiving end and the transmitting end is determined based on the deviation between the transmitting time interval and the receiving time interval; the sampling clock frequency of the receiving end is calibrated based on the sampling clock frequency deviation.

[0127] The sampling clock frequency refers to the number of times the wireless signal is sampled per unit time. The sampling clock frequency deviation is caused by the different vibration frequencies of the crystals due to inconsistencies between the crystals at the transmitting and receiving ends, temperature changes, etc.

[0128] Specifically, the receiving end first determines the reception time interval between the two sets of wireless signals received sequentially based on the synchronization timing deviation. Since the synchronization timing deviation reflects the actual arrival time of the signal at the receiving end, the difference between the two deviations can be used to accurately calculate the reception time interval.

[0129] Meanwhile, the time interval between the two wireless signal transmissions at the transmitting end is a known value, which the receiving end can directly obtain from the received wireless signal, system configuration, or upper-layer control commands.

[0130] The receiver compares the transmission time interval of two signals with the reception time interval to determine the sampling clock frequency deviation between the receiver and the transmitter. If the receiver's sampling clock frequency is too high, the reception time interval will be shorter than the transmission time interval; conversely, if the sampling frequency is too low, the reception time interval will be longer. Based on this deviation relationship, the receiver can dynamically calculate the actual offset of its own sampling clock.

[0131] For example, the receiving end can calculate the difference between the sending time interval and the receiving time interval to obtain the sampling clock frequency deviation.

[0132] After obtaining the sampling clock frequency deviation, the receiver calibrates its local sampling clock based on this deviation. For systems with hardware supporting dynamic frequency adjustment, the sampling frequency output by the local oscillator can be directly corrected to ensure clock synchronization between the receiver and transmitter. For terminal devices with limited hardware resources, calibration can be achieved through software compensation, such as adjusting the sampling point position, interpolating resampling, or phase rotation, to achieve an equivalent clock alignment effect. Through this calibration mechanism, the receiver can maintain time synchronization with the transmitter for a long period without incurring additional hardware overhead.

[0133] For example, when calibrating the reference clocks of the transmitter and receiver, it is assumed that the transmitter knows the receiver's reception time interval. However, in practical engineering applications, due to the clock frequency deviation between the transmitting and receiving ends, the counting at the receiving end... From the sender's perspective, it is actually ,in It is the clock frequency deviation between the transmitting and receiving ends, usually measured in parts per million (PPM).

[0134] To calibrate the clock frequency deviation between the transmitting and receiving ends, the transmitting end sends two signals sequentially, and the difference between the transmission timestamps of the two signals is recorded as... The difference between the timestamps of the two received signals at the receiving end is denoted as... The clock frequency deviation between the transmitting and receiving ends is calculated using the following formula. :

[0135] (7)

[0136] In the above embodiments, by deriving the receiving time interval based on the synchronization timing deviation and combining it with the transmitting time interval, the sampling clock frequency deviation between the receiving and transmitting ends is accurately calculated, which can dynamically compensate for the accumulated error caused by clock drift. This method does not require the introduction of an additional hardware reference signal, reducing system complexity, and supports both hardware-level adjustment and software-level compensation, providing greater flexibility. Subsequently, by calibrating the sampling clock frequency, the error caused by the clock difference between the receiving and transmitting ends is eliminated, allowing the receiving end to maintain a consistent time reference with the transmitting end. This significantly improves the accuracy of the synchronization timing deviation calculation, enabling high-precision ranging to be maintained under different devices and channel conditions, thereby improving the accuracy and stability of the entire wireless ranging system.

[0137] In wireless ranging systems, to estimate the sampling clock frequency deviation between the receiver and transmitter using sequentially received wireless signals, it is necessary to accurately obtain the time interval between the two received signals. However, due to factors such as multipath interference, noise, and sampling clock offset, relying solely on the original sample point counts often introduces accumulated errors, leading to inaccurate time interval calculations.

[0138] To improve the accuracy of time interval estimation, this invention introduces a sampling point compensation mechanism based on synchronization timing deviation when calculating the receiving time interval. By performing calibration compensation based on the original sampling point count values, synchronization errors caused by multipath and noise can be eliminated, thereby obtaining a more accurate receiving time interval.

[0139] Therefore, in some embodiments, in the process of determining the reception time interval of the wireless signals received by the receiver based on the synchronization timing deviation corresponding to the successively received wireless signals, the sampling point count values ​​corresponding to the synchronization positions of the successively received wireless signals are obtained respectively. Based on the synchronization timing deviation corresponding to the successively received wireless signals, the sampling point count values ​​corresponding to each wireless signal are compensated to obtain a first calibration count value and a second calibration count value. Based on the first calibration count value and the second calibration count value, the reception time interval of the wireless signals received by the receiver is determined.

[0140] After receiving two sets of wireless signals, the receiver first acquires the original sampling point count values ​​corresponding to the respective synchronization positions of the two sets of signals. The sampling point count value refers to the value obtained by accumulating the number of sampling actions triggered by the receiver with a fixed starting point in time when sampling the wireless signals. This sampling point count value records the cumulative number of sampling points since the start of sampling and serves as an initial reference characterizing the signal arrival time.

[0141] Since the synchronization timing deviation reflects the time offset caused by the influence of the channel and equipment during the propagation of the wireless signal, the receiver needs to compensate for the respective sampling point count values ​​based on the synchronization timing deviation of the two sets of wireless signals. By correcting the original sampling point count values, interference from factors such as multipath, noise, and clock drift on the signal synchronization position can be eliminated, resulting in the first calibration count value and the second calibration count value.

[0142] The first calibration count value is the result obtained after compensating for the synchronization timing deviation of the wireless signal received by the receiver for the sampling point count value at the synchronization position. Similarly, the second calibration count value is the result obtained after compensating for the synchronization timing deviation of the wireless signal received by the receiver for the sampling point count value at the synchronization position.

[0143] Finally, the receiver determines the precise reception time interval between the two received wireless signals based on the first and second calibration count values. Compared to directly using the original sample point count values, this method significantly improves the accuracy of the reception time interval calculation, providing a reliable data basis for subsequent estimation and calibration of sampling clock frequency deviation.

[0144] It should be noted that the compensation method can be flexibly selected in different application scenarios. For example, when the system has low requirements for hardware resources, linear interpolation can be used to achieve fast compensation; while in high-precision scenarios, multipath suppression and nonlinear phase compensation methods can be combined to improve the accuracy of synchronization timing deviation estimation, thereby improving the reliability of the received time interval calculation.

[0145] For example, the receiving end records the sampling point count values ​​corresponding to the synchronization positions of the signals received successively, denoted as . and The current sampling point count is calculated based on the sampling clock at the receiving end, without considering timing deviation. Subsequently, the number of sampling points corresponding to the synchronization timing deviation is calculated. Let the synchronization timing deviations of the successively received signals be... and The number of sampling points corresponding to the deviations in the received signals are respectively and , This is the baseband sampling frequency, also known as the sampling clock frequency.

[0146] Following this, the original count value is compensated to obtain a first calibration count value and a second calibration count value. The first calibration count value... Second calibration count value The time interval between receiving signals sequentially. The following formula is used to calculate:

[0147] (8)

[0148] In the above embodiments, by introducing a synchronization timing deviation compensation mechanism when calculating the receiving time interval at the receiving end, the deviation of the original sampling point count values ​​can be effectively corrected, reducing the impact of multipath effects, noise, and sampling clock drift on the time interval estimation. By using the calibrated sampling point count values ​​to calculate the receiving time interval, the accuracy of the time reference can be significantly improved, providing more reliable basic data for sampling clock frequency deviation estimation, thereby improving the overall accuracy and stability of the wireless ranging system.

[0149] In wireless ranging systems, ranging accuracy depends not only on the accurate estimation of synchronization timing deviation and effective calibration of the sampling clock frequency, but also on the round-trip time measurement of the ranging signal. Traditional one-way ranging methods are susceptible to the effects of terminal clock asynchrony, leading to error accumulation.

[0150] To this end, the present invention introduces a timestamp marker between the transmitting and receiving ends through a two-way ranging mechanism. Combined with a preset signal propagation speed, it can obtain high-precision ranging results without the need for precise synchronization of the local clocks at both ends.

[0151] Accordingly, in some embodiments, a ranging operation is performed after calibration to obtain a ranging result. Specifically, one of the transmitting and receiving ends sends a ranging signal to the other end at a first timestamp and receives a feedback signal returned by the other end at a second timestamp. Based on the time interval between the first and second timestamps, the ranging result is calculated according to a preset signal propagation speed.

[0152] After completing the sampling clock frequency calibration at the receiving end, the dual-mode communication system enters the ranging phase.

[0153] Either the sending or receiving end can initiate a ranging request. For example, when the sending end initiates ranging, it records a first timestamp on its local clock and sends a ranging signal to the receiving end at that time. Upon receiving the ranging signal, the receiving end immediately generates a feedback signal and returns it to the sending end. When the sending end receives the feedback signal, it records a second timestamp.

[0154] The first timestamp and the second timestamp are time recording parameters used to mark the time of wireless signal interaction for ranging. The first timestamp is the time record corresponding to the wireless signal sent by the transmitter for ranging, and the second timestamp is the time record corresponding to the feedback signal sent by the receiver.

[0155] Since both timestamps are based on the same local clock, precise synchronization of the absolute times at both ends is unnecessary. The one-way propagation time between the receiver and transmitter can be obtained by dividing the time interval between the first and second timestamps by two. Combined with a preset signal propagation speed (e.g., the speed of electromagnetic waves in free space), the actual distance between the two ends can be calculated.

[0156] For example, suppose the clock reference of the transmitting end is from Counting begins, the receiver's clock reference starts from... The counting begins when the transmitting end first sends a signal to the receiving end, which can be calculated as the interval between the transmission time and its reference clock. With counting time The sum of + The time when the receiver receives the first signal is... .

[0157] in, This is the time after the receiving end receives the first signal sent by the transmitting end, after synchronization timing deviation compensation; let this time be the accurate synchronization time point. In actual ranging, and Since the two signals cannot be guaranteed to be completely equal, the flight time of a wireless signal is often estimated using two segments of the wireless signal. The flight time refers to the total propagation time of the wireless ranging signal from one end to the other end, after which it is fed back to the transmitting end. In other words, after receiving the first signal, the receiving end sends a feedback signal to the transmitting end after a certain period of time; this time is [missing information]. In this process, the interval between the receiver receiving the signal sent by the transmitter and the receiver sending a feedback signal is... It can be adjusted according to the actual time requirements of distance measurement. For example, when the real-time requirements of distance measurement are high, a shorter time frame can be used. .in, .

[0158] After this, the sending end receives the feedback signal sent by the receiving end, at which time the time is... The time of flight of the wireless signal during ranging can then be calculated using the following formula:

[0159] (9)

[0160] Similarly, the above formula can also be described as:

[0161] (10)

[0162] in, It refers to the receiving time interval at the receiving end. The receiving end receives the feedback signal, compensates for the count value at the corresponding sampling point, and obtains the corrected result. .

[0163] After obtaining the flight time of the aforementioned wireless signal, and combining this with the propagation speed of the wireless signal, the actual distance between the current ranging nodes can be determined. The propagation speed of the wireless signal typically refers to the speed of electromagnetic wave propagation in a medium; in air, the propagation speed of a wireless signal is approximately 3... 10 8 Meters per second (m / s) allows for adjustment of the wireless signal propagation parameters based on actual measurement accuracy requirements.

[0164] In the above embodiments, high-precision ranging is achieved through a two-way ranging mechanism. Since the time interval calculation is based on the calibrated sampling clock frequency, the cumulative error caused by terminal clock asynchrony in traditional one-way ranging methods is avoided. Combined with a preset signal propagation speed, the distance between the two ends can be calculated quickly and accurately, significantly improving the adaptability and reliability of the wireless ranging system.

[0165] Furthermore, the above ranging method is based on the propagation time and speed of the wireless signal, and the ranging process relies on the timestamp information of the same end, which greatly reduces the computational complexity. During the calculation process, there is no need to perform complex signal correlation operations or large-scale matrix processing, which significantly reduces the demand for computing power resources and storage resources of power equipment, and can run efficiently in terminal devices with limited hardware resources.

[0166] In particular, in typical application scenarios such as automatic meter reading of power equipment, smart grid terminals and Internet of Things sensing nodes, since these devices are usually low in power consumption, limited in computing power and limited in storage, the embodiments of the present invention can achieve high-precision ranging without increasing hardware overhead, taking into account both low power consumption and high performance. It is especially suitable for power communication and Internet of Things ranging applications in environments with high multipath occurrence, such as dense urban buildings and complex indoor layouts.

[0167] In practical ranging processes, the results of a single measurement are often subject to chance. This can be due to multiple factors, such as clock jitter or offset between the receiving and transmitting nodes, and changes in the channel environment. Therefore, in practical applications, it is necessary to process the wireless ranging results to obtain more accurate results.

[0168] Therefore, in some embodiments, the calculation of the ranging result further includes: acquiring the ranging results of multiple round trip ranging measurements, extracting the ranging results of a preset number of times to calculate the average value, using the average value as the initial value for filtering, and for each ranging result after the preset number of ranging results, performing a weighted average of the initial value for filtering and the current ranging result based on the filtering coefficient to obtain the updated ranging result. The larger the filtering coefficient, the greater the impact of the new ranging result on the final ranging result.

[0169] Specifically, the average value of a preset number of ranging results is calculated and used as the initial value of the filter to avoid errors caused by the lack of historical data during the initial input. Afterward, for each new ranging result acquired, a weighted average is calculated between this result and the previous initial value based on the filter coefficients.

[0170] The filter coefficient reflects the degree of influence of the newly obtained ranging result on the final ranging result, and can be adjusted according to actual engineering calculations. When the filter coefficient is larger, the new ranging result has a greater impact on the final result, and the algorithm is more sensitive to environmental changes, making it suitable for fast-moving dynamic scenarios. When the filter coefficient is smaller, the final ranging result is more robust to short-term fluctuations, making it suitable for static or quasi-static ranging scenarios.

[0171] After weighted averaging, the updated prediction results are output, and this process is continued until the obtained ranging results meet the needs of the actual project.

[0172] For example, the average of the first 10 distance measurement results is taken and used as the initial value. The ranging result is updated using the following formula:

[0173] (11)

[0174] in, These are historical estimates. This is the current new measurement value. This is the updated filtered estimate. It is the filter coefficient and ,when When it is close to 1, the new measured value The weights are close to 1, and the historical estimates are... The weight is close to 0, which can quickly track changes in new measurements, but noise will be retained to some extent; when When it is close to 0, the new measured value The weight is close to 0, historical estimate The weight is close to 1, which has a strong suppression effect on noise in new measurements and makes the filtering result more stable, but it is slower to respond to real dynamic changes.

[0175] In the filtering process described above, the first iteration uses... As the previous estimate, every new test result obtained thereafter... The value is compared with the previous estimate using formula (11). Integration and continuous output Continue until the result is smooth.

[0176] In the above embodiments, by introducing a weighted average filtering strategy based on multiple round-trip ranging results, the influence of random noise and outliers can be effectively suppressed, significantly improving the smoothness and stability of the final ranging result. Since the calculation process involves only a small number of addition, subtraction, multiplication, and division operations, the computational complexity and workload are greatly reduced. There is no need to store large amounts of historical data, significantly reducing the computational and storage resource requirements for low-power terminals such as power equipment. Updates can be completed simply by saving the current filter value and the latest ranging result. This is particularly suitable for resource-constrained scenarios such as automatic power meter reading, IoT communication, and smart grid terminals, as well as low-power terminal devices with limited computing power, such as automatic power meter reading devices, IoT communication nodes, and smart grid sensing terminals.

[0177] The wireless ranging method for a power line carrier dual-mode communication system provided in this embodiment of the invention can be executed by a wireless ranging device for the power line carrier dual-mode communication system. This embodiment of the invention uses the example of a wireless ranging device for the power line carrier dual-mode communication system executing the wireless ranging method for the power line carrier dual-mode communication system to illustrate the wireless ranging device for the power line carrier dual-mode communication system provided in this embodiment of the invention.

[0178] This invention also provides a wireless ranging device for a power line carrier dual-mode communication system, which is applied to electronic devices.

[0179] Figure 7 This is a schematic diagram of the structure of a wireless ranging device in a power line carrier dual-mode communication system provided in some embodiments of the present invention. For example... Figure 7 As shown, the wireless ranging device of this power line carrier dual-mode communication system includes a signal module 700, a processing module 701, a calculation module 702, and a ranging module 703. Wherein:

[0180] The signal module 700 is used to receive wireless signals sent sequentially by the transmitter and to perform frequency domain transformation on the wireless signals to obtain a frequency domain response signal.

[0181] Processing module 701 is used to acquire the impulse response signal of the frequency domain response signal in the time domain;

[0182] The calculation module 702 is used to calculate the multipath centroid of the impulse response signal under multipath channel conditions, and determine the synchronization timing deviation corresponding to the wireless signal based on the deviation between the multipath centroid and the main path.

[0183] The ranging module 703 is used to calibrate the sampling clock frequencies of the transmitting and receiving ends based on the synchronization timing deviations corresponding to the successively received wireless signals, and to perform ranging operations after calibration to obtain the ranging results.

[0184] The wireless ranging device for a power line carrier dual-mode communication system provided by the present invention receives wireless signals transmitted sequentially by the transmitting end and performs frequency domain transformation, which can fully utilize the frequency domain characteristics of the signal and improve the extraction accuracy of signal amplitude and phase information. Then, the frequency domain response signal is converted into a time domain impulse response signal in the time domain to fully observe the time domain characteristics of the signal under multipath channel conditions. Under multipath channel conditions, for the wireless signal in the time domain, the multipath centroid of the time domain impulse response signal on multiple propagation paths is calculated, and the deviation between the calculated multipath centroid and the main path is used to obtain the synchronization timing deviation of the wireless signal. Under weakened multipath interference, the actual arrival time of the wireless signal is accurately calculated, providing a basis for clock calibration and wireless ranging calculation at the transmitting and receiving ends. Finally, the synchronization timing deviation of the two sets of wireless signals received sequentially is used to dynamically calibrate the sampling clock frequencies of the transmitting and receiving ends, effectively eliminating the cumulative error caused by multipath effects and clock drift, significantly improving the overall accuracy and stability of ranging, and is particularly suitable for high-noise, strong multipath scenarios in power line carrier dual-mode communication systems. After clock calibration, ranging operations are performed on nodes in power line carrier dual-mode communication, and multiple ranging results are processed by a filter to finally obtain accurate and reliable ranging results.

[0185] In some embodiments, the processing module is further configured to cyclically shift the impulse response signal to converge the primary and secondary paths in the time domain; calculate the multipath centroid based on the converged primary and secondary paths; wherein the multipath centroid is used to characterize the concentration location of signal energy in the time domain to evaluate the impact of multipath effects on the synchronization timing position.

[0186] In some embodiments, the processing module is further configured to: filter effective paths from the aggregated primary and secondary paths and obtain the signal energy of the effective paths; select effective sampling points on the effective paths that are within the target range and set the signal energy corresponding to the sampling points outside the target range to zero; and perform weighted averaging on the signal energy corresponding to each effective sampling point using the signal energy of the effective paths within the target range as the weighting weight to obtain the multipath centroid.

[0187] In some embodiments, the processing module is further configured to select a path within a preset window range as an effective path, centered on the main path; or to determine a path with signal energy greater than a preset threshold as an effective path.

[0188] In some embodiments, the processing module is further configured to map the main path position and the multipath centroid position back to the original time axis, and calculate the sampling point offset between the mapped multipath centroid and the main path position; convert the sampling point offset into a time deviation according to the sampling time interval to obtain the synchronization timing deviation corresponding to the wireless signal.

[0189] In some embodiments, the ranging module is further configured to: determine the reception time interval of the wireless signals received sequentially by the receiving end based on the synchronization timing deviation corresponding to the sequentially received wireless signals; obtain the transmission time interval of the wireless signals transmitted sequentially by the transmitting end; determine the sampling clock frequency deviation between the receiving end and the transmitting end based on the deviation between the transmission time interval and the reception time interval; and calibrate the sampling clock frequency of the receiving end based on the sampling clock frequency deviation.

[0190] In some embodiments, the ranging module is further configured to acquire the sampling point count values ​​corresponding to the synchronization positions of the successively received wireless signals; compensate the sampling point count values ​​corresponding to each wireless signal based on the synchronization timing deviations corresponding to the successively received wireless signals to obtain a first calibration count value and a second calibration count value; and determine the reception time interval of the successively received wireless signals by the receiving end based on the first calibration count value and the second calibration count value.

[0191] In some embodiments, the ranging module is further configured to send a ranging signal from one end of the transmitting end to the other end at a first timestamp, and receive a feedback signal returned by the other end at a second timestamp; and calculate the ranging result based on the time interval between the first timestamp and the second timestamp, according to a preset signal propagation speed.

[0192] In some embodiments, the ranging module is further configured to acquire ranging results from multiple round trips; extract ranging results from a preset number of times to calculate an average value, and use the average value as an initial value for filtering; for each ranging result after the preset number of ranging results, perform a weighted average of the initial value for filtering and the current ranging result based on the filtering coefficient to obtain an updated ranging result; wherein, the larger the filtering coefficient, the greater the impact of the new ranging result on the final ranging result.

[0193] The wireless ranging device in the power line carrier dual-mode communication system of this invention can be an electronic device or a component of an electronic device, such as an integrated circuit or a chip. The electronic device can be a terminal device or a server.

[0194] The wireless ranging device of the power line carrier dual-mode communication system provided in this embodiment of the invention can realize the various processes implemented in each method embodiment. To avoid repetition, it will not be described again here.

[0195] Figure 8This is a schematic diagram of the structure of an electronic device provided in some embodiments of the present invention. In some embodiments, such as... Figure 8 As shown, this embodiment of the invention also provides an electronic device 800, including a processor 801, a memory 802, and a computer program stored in the memory 802 and executable on the processor 801. When the program is executed by the processor 801, it implements the various processes of the above-described method embodiments and achieves the same technical effects. To avoid repetition, it will not be described again here.

[0196] It should be noted that the electronic devices in the embodiments of the present invention include the mobile electronic devices and non-mobile electronic devices described above.

[0197] This invention also provides a non-transitory computer-readable storage medium storing a computer program. When executed by a processor, the computer program implements the various processes of the wireless ranging method embodiment of the above-described power line carrier dual-mode communication system and achieves the same technical effect. To avoid repetition, it will not be described again here.

[0198] The processor is the processor in the electronic device described in the above embodiments. The readable storage medium includes computer-readable storage media, such as computer read-only memory (ROM), random access memory (RAM), magnetic disk, or optical disk.

[0199] This invention also provides a computer program product, including a computer program that, when executed by a processor, implements the wireless ranging method of the above-described power line carrier dual-mode communication system.

[0200] The processor is the processor in the electronic device described in the above embodiments. The readable storage medium includes computer-readable storage media, such as computer read-only memory (ROM), random access memory (RAM), magnetic disk, or optical disk.

[0201] This invention also provides a chip, which includes a processor and a communication interface. The communication interface is coupled to the processor. The processor is used to run programs or instructions to implement the various processes of the wireless ranging method embodiment of the above-described power line carrier dual-mode communication system, and can achieve the same technical effect. To avoid repetition, it will not be described again here.

[0202] It should be understood that the chip mentioned in the embodiments of the present invention may also be referred to as a system-on-a-chip, system chip, chip system, or system-on-a-chip, etc.

[0203] It should be noted that, in this document, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such a process, method, article, or apparatus. Without further limitations, an element defined by the phrase "comprising one..." does not exclude the presence of other identical elements in the process, method, article, or apparatus that includes that element. Furthermore, it should be noted that the scope of the methods and apparatuses in the embodiments of the present invention is not limited to performing functions in the order shown or discussed, but may also include performing functions substantially simultaneously or in the reverse order, depending on the functions involved. For example, the described methods may be performed in a different order than described, and various steps may be added, omitted, or combined. Additionally, features described with reference to certain examples may be combined in other examples.

[0204] The embodiments of the present invention have been described above with reference to the accompanying drawings. However, the present invention is not limited to the specific embodiments described above. The specific embodiments described above are merely illustrative and not restrictive. Those skilled in the art can make many other forms under the guidance of the present invention without departing from the spirit and scope of the claims, and all of these forms are within the protection scope of the present invention.

[0205] The above are merely preferred embodiments of the present invention and are not intended to limit the present invention. Various modifications and variations can be made to the present invention by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A wireless ranging method for a power line carrier dual-mode communication system, characterized in that, The method comprises: receiving wireless signals sent by the sending end in succession, and performing frequency domain transformation on the wireless signals to obtain frequency domain response signals; obtaining an impulse response signal of the frequency domain response signal in the time domain; performing cyclic shift on the impulse response signal to make the main path and the secondary path gather in the time domain; calculating a multipath center of gravity based on the gathered main path and secondary path, and determining a synchronization timing deviation corresponding to the wireless signal based on a deviation between the multipath center of gravity and the main path; the multipath center of gravity represents a concentrated position of signal energy in the time domain; based on the synchronization timing deviations corresponding to the wireless signals received in succession, respectively, calibrating the sampling clock frequency of the sending end and the receiving end, and performing a ranging operation after calibration to obtain a ranging result.

2. The method of claim 1, wherein, The method comprises: selecting effective paths from the gathered main path and secondary path, and obtaining signal energy of the effective paths; selecting effective sampling points in a target range of the effective paths, and setting signal energy of sampling points outside the target range to zero; performing weighted average processing on signal energy of each effective sampling point by taking signal energy of the effective paths in the target range as a weighted weight to obtain the multipath center of gravity.

3. The method of claim 2, wherein, The method comprises: selecting paths in a preset window range as effective paths with the main path as the center; or determining paths with signal energy greater than a preset threshold as effective paths.

4. The method according to any one of claims 1 to 3, characterized in that, The method comprises: mapping the main path position and the multipath center of gravity position back to an original time axis, and calculating a sampling point offset between the mapped multipath center of gravity and the main path position; converting the sampling point offset into a time deviation according to a sampling time interval to obtain the synchronization timing deviation corresponding to the wireless signal.

5. The method of claim 1, wherein, The method comprises: determining a receiving time interval of the receiving end for receiving the wireless signals in succession based on the synchronization timing deviations corresponding to the wireless signals received in succession; obtaining a sending time interval of the sending end for sending the wireless signals in succession; determining a sampling clock frequency deviation between the receiving end and the sending end based on a deviation between the sending time interval and the receiving time interval; calibrating the sampling clock frequency of the receiving end based on the sampling clock frequency deviation.

6. The method of claim 5, wherein, The method comprises: obtaining sampling point count values corresponding to the synchronization positions of the wireless signals received in succession, respectively; compensating the sampling point count values corresponding to each wireless signal based on the synchronization timing deviations corresponding to the wireless signals received in succession to obtain first calibration count values and second calibration count values; determining the receiving time interval of the receiving end for receiving the wireless signals in succession based on the first calibration count values and the second calibration count values.

7. The method of claim 1, wherein, The method comprises: sending a ranging signal from one of the sending end and the receiving end to the other end at a first time stamp, and receiving a feedback signal returned by the other end at a second time stamp; calculating a ranging result according to a preset signal propagation speed based on a time interval between the first time stamp and the second time stamp.

8. The method according to claim 1 or 7, characterized in that, The method further comprises: obtaining ranging results of multiple round-trip rangings; extracting ranging results of a preset number of times to calculate an average value, and taking the average value as a filtering initial value; for each ranging result after the ranging results of the preset number of times, performing weighted averaging on the filtering initial value and the current ranging result based on a filtering coefficient to obtain an updated ranging result; wherein the greater the filtering coefficient, the greater the influence of the new ranging result on the final ranging result.

9. A wireless ranging device for a power line carrier dual mode communication system, comprising: The apparatus comprises: a signal module configured to receive wireless signals sent by the sending end in sequence, and perform frequency domain transformation on the wireless signals to obtain frequency domain response signals; a processing module configured to obtain an impulse response signal of the frequency domain response signal in time domain; a calculation module configured to perform cyclic shift on the impulse response signal to make a main path and a secondary path gather in time domain; a ranging module configured to calculate a multi-path barycenter based on the gathered main path and secondary path, and determine a synchronization timing deviation corresponding to the wireless signal based on a deviation between the multi-path barycenter and the main path; the multi-path barycenter represents a concentrated position of signal energy in time domain; based on synchronization timing deviations corresponding to the wireless signals received in sequence, calibrate sampling clock frequencies of the sending end and the receiving end, and perform a ranging operation after the calibration to obtain a ranging result.

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