Bidirectional DC-DC high-gain multi-cavity parallel converter and control method thereof

Abstract

The invention discloses a bidirectional DC-DC high-gain multi-cavity parallel converter and a control method thereof.The primary side of the converter adopts at least two bridge circuits to be connected in parallel, natural current sharing of input current is achieved through a collaborative multi-cavity resonance module, the input current is output to a plurality of resonance modules sharing an integrated magnetic core, and symmetrical cavities are formed in the magnetic core to contain module transformers; magnetic interference is avoided; the secondary side is connected in series and laminated through a transformer winding, and then is sequentially subjected to full-bridge rectification, two-stage voltage regulation and series voltage doubling, so that the traditional single-cavity gain bottleneck is broken through, and ripples are filtered by matching with an output capacitor. In order to ensure all-working-condition soft switching, the invention also provides a control module for the converter, and the control module is used for cooperatively regulating and controlling internal and external phase shifting angles based on a fundamental wave phasor model and calculating and compensating a fundamental wave error in combination with a time domain numerical value. The invention further provides an integrated magnetic coupling design which supports linear expansion of power levels, is suitable for high-voltage and high-power scenes, and meets the THD requirement of grid-connected current and the bidirectional flow requirement of efficient energy.

Description

Technical Field

[0001] This invention relates to a converter, and more particularly to a bidirectional DC-DC high-gain multi-cavity parallel converter. Background Technology

[0002] With the rapid development of new energy power generation, electric vehicles, and energy storage technologies, the demands on power density, efficiency, and voltage gain of power electronic converters are becoming increasingly stringent. In DC / DC conversion scenarios, traditional single-cavity converters, limited by the voltage and current ratings of the devices, struggle to meet the requirements of megawatt-level power transmission and high-voltage output. Multi-cavity parallel connection reduces stress on single-cavity devices through input-side current sharing, effectively increasing system power capacity; multi-stage series voltage stacking on the secondary side eliminates the need for high-ratio transformers to achieve high voltage gain, significantly optimizing topology compactness.

[0003] However, existing multi-cavity parallel converters still face many technical bottlenecks that urgently need to be addressed in practical applications:

[0004] First, the collaborative control mechanism is imperfect. Although the existing multi-cavity parallel scheme can achieve current sharing among modules, it does not fully consider the coupling relationship between secondary series voltage superposition and two-stage voltage regulation, which makes it difficult for the system to maintain efficient operation over a wide input and output range. In particular, voltage fluctuations and power distribution imbalances are prone to occur when switching operating conditions.

[0005] Secondly, the soft-switching range is limited. Traditional solutions often use the fundamental frequency approximation method, ignoring the influence of higher harmonics on the resonant current, which leads to significant errors in the current calculation at the switching moment, making it difficult to achieve stable soft switching across the entire load range. At the same time, the wide range of fluctuations in the switching frequency further restricts the improvement of efficiency and power density.

[0006] Third, the ripple suppression capability is insufficient in high-gain scenarios. Some solutions improve voltage gain by coupling inductors, but the output ripple suppression effect is limited and it is difficult to adapt to bidirectional power flow requirements, thus failing to meet the stringent requirements of new energy storage systems for ripple coefficient and bidirectional operating mode.

[0007] In addition, existing multi-cavity magnetic core components are mostly distributed, which can easily lead to asymmetry in the magnetic circuits of each module and exacerbate the problem of uneven current distribution; while the topology design with independent drive of multiple bridge arms not only increases the control complexity, but also significantly increases the system size and cost. Summary of the Invention

[0008] Objective of this invention: This invention provides a bidirectional DC-DC high-gain multi-cavity parallel converter. Through topological innovation of multi-module parallel input and multi-stage series voltage multiplier output, combined with phase-shift angle cooperative modulation, time-domain numerical calculation compensation, and integrated magnetic coupling design, it can simultaneously achieve front-end multi-cavity parallel current sharing, secondary-side multi-stage series voltage superposition, and two-stage voltage regulation in high-power scenarios. Furthermore, it features a wide soft-switching range and low output ripple, achieving highly efficient energy conversion. Another objective of this invention is to provide a control method based on the aforementioned converter.

[0009] Technical solution: The bidirectional DC-DC high-gain multi-cavity parallel converter includes: primary side circuit, multiple resonant modules, secondary side circuit and output capacitor;

[0010] The primary circuit includes an input capacitor and at least two parallel bridge circuits; the input capacitor is connected in parallel with the input terminal of each bridge circuit to the input terminal of the primary circuit, and the output terminal of the bridge circuit is connected in parallel with the primary input terminal of the transformer of the plurality of resonant modules, through which the input current is evenly distributed to each of the resonant modules.

[0011] Each of the resonant modules includes a transformer and a composite resonant network; the multiple transformers share an integrated magnetic core, and the integrated magnetic core has multiple symmetrically distributed chambers inside, each chamber accommodating the primary winding and secondary winding of a transformer; the composite resonant network is electrically connected to the primary winding of the corresponding transformer, together forming a resonant circuit for energy transfer;

[0012] The secondary circuit includes a full-bridge rectifier unit, a two-stage voltage regulation module, and a series voltage multiplier output circuit connected in sequence; the secondary windings of the multiple transformers are connected in series to achieve voltage superposition, and the superimposed voltage is multiplied by the full-bridge rectifier unit, the two-stage voltage regulation module, and the series voltage multiplier output circuit in sequence.

[0013] The output capacitor is connected in parallel to the output terminal of the series voltage multiplier output circuit to filter out high-frequency ripple in the output voltage.

[0014] Optionally, the bridge circuit adopts a half-bridge circuit; the half-bridge circuit includes two switching transistors and two capacitors, the two switching transistors are connected in series to form a first bridge arm, the two capacitors are connected in series to form a second bridge arm, and the output terminal of the two bridge arms connected in parallel is connected in parallel to the primary input terminal of the transformer of the plurality of resonant modules.

[0015] Optionally, the bridge circuit is a full-bridge circuit; the full-bridge circuit includes four switching transistors S1, S2, S3, and S4 and an output bus capacitor; wherein, switching transistors S1 and S3 are the upper arm switching transistors of the full bridge, and switching transistors S2 and S4 are the lower arm switching transistors of the full bridge; switching transistors S1 and S2 are connected in series to form the left arm of the full bridge, and switching transistors S3 and S4 are connected in series to form the right arm of the full bridge; after the left arm and the right arm are connected in parallel, their two ends are respectively connected to the positive and negative terminals of the input bus capacitor, and the common midpoint of the left arm and the right arm is respectively connected to the primary input terminal of the transformer of the plurality of resonant modules.

[0016] Optionally, the bidirectional DC-DC high-gain multi-cavity parallel converter further includes a control module; the control module is used to coordinately regulate the inner phase shift angle and the outer phase shift angle based on the fundamental phasor model; the inner phase shift angle is the phase shift angle of the driving signals of the leading and lagging bridge arms in the full-bridge rectifier unit, and the outer phase shift angle is the phase shift angle between the alternating square wave voltage output by the bridge circuit and the alternating square wave voltage output by the full-bridge rectifier unit.

[0017] Optionally, the fundamental phasor model is:

[0018]

[0019] Among them, V dcL V is the primary power supply voltage. dcH ω is the secondary power supply voltage; s =2πfs is the switching angular frequency, fs is the switching frequency; θ is the inner phase shift angle of the leading and lagging bridge arm drive waveforms on the secondary side.

[0020] Optionally, the control module is also used to establish the differential equation of the multimode resonant circuit using a time-domain numerical calculation method, solve for the accurate value of the resonant current at the switching moment, and compensate for the fundamental wave approximation error by adjusting the reactive component or phase shift angle of the resonant current through closed-loop adjustment, so as to ensure zero-voltage turn-on of the primary and secondary switching transistors.

[0021] Optionally, the composite resonant network includes a resonant inductor and a resonant capacitor; the resonant inductor is connected in series with the primary winding of the corresponding transformer, and the resonant capacitor is connected in parallel with the series-connected resonant inductor and the primary winding of the transformer; the resonant inductor and the resonant capacitor work together with the primary winding of the transformer to realize energy transfer and assist the soft switching of the primary switching transistor.

[0022] Optionally, when the converter is in the forward operating mode, the regulation of the control module satisfies the following: the average value of the active component of the resonant current during the switching cycle is equal to the input current of the primary circuit; during the dead time before the primary switch is turned on, the direction of the resonant current is negative and its magnitude meets the charging and discharging requirements of its output capacitor; the phase shift angle α of the secondary switch is ≥ β, where β is the phase difference between the secondary resonant current and the secondary switch drive signal.

[0023] Optionally, when the converter is in reverse operating mode, the control module's regulation satisfies the following: the output current of the secondary circuit is controllable by adjusting the active component of the resonant current; during the dead time before the secondary switch is turned on, the direction of the resonant current is positive and its magnitude meets the charging and discharging requirements of its output capacitor; the phase shift angle β of the primary switch is 0, where β is the phase difference between the primary resonant current and the drive signal, and the dead time of the primary switch is increased.

[0024] Optionally, the integrated magnetic core is made of high-frequency, low-loss materials, such as nanocrystalline alloys or amorphous alloys, which can meet the requirements of high power density.

[0025] Optionally, the power can be linearly expanded by increasing or decreasing the number of the multiple resonant modules, and the gain of the series voltage multiplier output circuit increases with the increase of the number of transformer secondary windings connected in series.

[0026] The present invention also provides a control method based on the above-mentioned bidirectional DC-DC high-gain multi-cavity parallel converter, comprising:

[0027] Based on the fundamental phasor model, the inner and outer phase shift angles are coordinated and controlled; the inner phase shift angle is the phase shift angle of the driving signals of the leading and lagging bridge arms in the full-bridge rectifier unit, and the outer phase shift angle is the phase shift angle between the alternating square wave voltage output by the bridge circuit and the alternating square wave voltage output by the full-bridge rectifier unit.

[0028] Optionally, the method further includes:

[0029] By employing time-domain numerical calculation methods, the differential equations of the multimode resonant circuit are established, and the precise value of the resonant current at the switching moment is obtained by solving them. By adjusting the reactive component or phase shift angle of the resonant current in a closed loop, the approximate error of the fundamental wave is compensated, ensuring that the primary and secondary switching transistors turn on with zero voltage.

[0030] Optionally, the method further includes: when the converter is operating in forward mode, setting the average value of the active component of the resonant current during the switching cycle to be equal to the input current of the primary circuit; during the dead time before the primary switch is turned on, setting the direction of the resonant current to be negative and its magnitude to meet the charging and discharging requirements of its output capacitor; setting the phase shift angle of the secondary switch to α≥β, where β is the phase difference between the secondary resonant current and the secondary switch drive signal.

[0031] Optionally, the method further includes: when the converter is operating in reverse mode, adjusting the active component of the resonant current to control the output current of the secondary circuit; during the dead time before the secondary switch is turned on, setting the direction of the resonant current to be positive and its magnitude to meet the charging and discharging requirements of its output capacitor; setting the phase shift angle β of the primary switch to 0 and increasing the dead time of the primary switch, wherein the phase shift angle β is the phase difference between the primary resonant current and the driving signal.

[0032] Beneficial effects: Compared with the prior art, the present invention has the following significant advantages:

[0033] 1. High gain characteristics: By combining the secondary winding series superposition with the voltage multiplier circuit, the voltage gain is significantly higher than that of the traditional single-module topology, eliminating the need for high-frequency transformer multi-turn ratio design and reducing core size and loss;

[0034] 2. Current sharing optimization: The symmetrical cavity layout of the integrated magnetic core ensures the symmetry of the magnetic circuit of each resonant module. Combined with the natural current sharing characteristics of the bridge circuit, no complex current sharing control algorithm is required.

[0035] 3. Wide-range soft switching: The combination of composite resonant network and phase-shift control strategy enables zero-voltage turn-on of primary and secondary side switches over a wide load range, reducing switching losses;

[0036] 4. Two-way flexibility: The control strategy is optimized for both forward and reverse working modes, adapting to two-way energy transfer scenarios such as charging and discharging of energy storage systems;

[0037] 5. High scalability: The modular design supports flexible expansion of power and voltage levels, adapting to diverse needs from small and medium power home appliances to high power industrial equipment.

[0038] 6. High power density: Through optimization of core materials and structural innovation, the amount of core used is reduced by 40% and the volume is reduced by 35%, significantly improving power density;

[0039] 7. Long device lifespan: Magnetic shielding is achieved through the integrated magnetic core structure design, which reduces EMI radiation, and combined with current sharing control, extends the device lifespan. Attached Figure Description

[0040] Figure 1 This is a circuit diagram of a bidirectional DC-DC high-gain multi-cavity parallel converter;

[0041] Figure 2 The equivalent circuit diagram of the front-end converter in forward operating mode;

[0042] Figure 3(a) shows the phase shift angle relationship of the alternating square wave voltage in the forward operating mode, and Figure 3(b) shows the phase shift angle relationship of the alternating square wave voltage in the DC / DC reverse operating mode.

[0043] Figure 4 The working waveform of the converter under extended phase-shift modulation rectification mode;

[0044] Figure 5 The steady-state waveform and corresponding phasor diagram in the time domain of the converter in forward operating mode;

[0045] Figure 6 The steady-state waveform and corresponding phasor diagram in the time domain of the converter in reverse operating mode;

[0046] Figure 7 This is a schematic diagram of the closed-loop regulation of Irq in the forward operating mode of the converter;

[0047] Figure 8 The modulation principle of the phase shift angle α in the forward operating mode of the converter;

[0048] Figure 9 For closed-loop regulation f in the forward operating mode of the converter s Control diagram;

[0049] Figure 10 This is a topology diagram of an integrated magnetically coupled bidirectional DC-DC high-gain multi-cavity parallel converter. Detailed Implementation

[0050] The technical solution of the present invention will be further described below with reference to the accompanying drawings.

[0051] like Figure 1 As shown, a bidirectional DC-DC high-gain multi-cavity parallel converter includes: a primary-side circuit, multiple resonant modules, a secondary-side circuit, and an output capacitor.

[0052] The primary circuit includes an input capacitor C. dcL A bridge circuit with at least two parallel connections is used to evenly distribute the input current to multiple resonant modules compatible with various topologies. An input capacitor is connected in parallel with the input terminal of each bridge circuit to the primary input terminal for filtering the input voltage; the output terminals of the bridge circuits are connected in parallel to drive all resonant modules.

[0053] The multiple resonant modules adopt an integrated magnetic core symmetrical cavity layout. In one embodiment, the magnetic core body has an E-type structure with multiple symmetrically distributed cavities inside, each cavity accommodating a winding. Each cavity contains a transformer and a composite resonant network, the composite resonant network including a resonant inductor and a capacitor.

[0054] The secondary circuit achieves high gain through a series voltage multiplier output circuit, specifically comprising a full-bridge rectifier unit, a two-stage voltage regulation module, and a series voltage multiplier output circuit connected in sequence. The secondary windings of the multiple transformers are connected in series to achieve voltage superposition. The superimposed voltage is then multiplied sequentially by the full-bridge rectifier unit, the two-stage voltage regulation module, and the series voltage multiplier output circuit, and finally output through the output capacitor C. dcH The filtered output.

[0055] Taking the first cavity as an example, the input voltage first passes through the input capacitor C. dcL Filtering. The primary side of the circuit adopts a half-bridge structure, consisting of a set of switches S1 and S2 and capacitors C1 and C2 forming two arms of the half-bridge. The output of the two arms is connected in parallel to the multiple resonant modules. Capacitors C1 and C2 form the second arm, sharing the input voltage and reducing the withstand voltage of the switching transistors to half of the input voltage. The secondary side uses S3-S6 to form a full-bridge rectifier circuit connected to a two-stage voltage regulator circuit, and then outputs the voltage through capacitor C. dcH The filtered output.

[0056] The bidirectional DC-DC high-gain multi-cavity parallel converter also includes a control module for coordinating the inner phase shift angle θ and the outer phase shift angle based on the fundamental phasor model. The inner phase shift angle θ is the phase shift angle of the driving waveforms of the leading and lagging arms of the secondary-side full-bridge rectifier unit. The outer phase shift angle... This represents the phase difference between the square wave output from the primary-side bridge circuit and the square wave output from the secondary-side full-bridge rectifier unit. Power control is achieved through phase shifting, as shown in Figures 3(a) and 3(b).

[0057] In DC / DC forward operating mode, the two primary-side switches conduct alternately, generating a square wave voltage V at the midpoint AB of the bridge arm. ab A square wave voltage V is generated at the midpoint CD of the secondary bridge arm. cd .

[0058] Assuming the transformer turns ratio is 1:n, where n is the number of turns on the secondary side divided by the number of turns on the primary side, the equivalent circuit diagram of the upstream converter is as follows: Figure 2 As shown.

[0059] For v respectively ab (ω s t) and nv cd (ω s t) Performing Fourier decomposition yields the fundamental component expressions for both, i.e., the fundamental phasor model:

[0060]

[0061] In the formula: V dcL V is the primary power supply voltage. dcH ω is the secondary power supply voltage;s =2πfs is the switching angular frequency, fs is the switching frequency; θ is the inner phase shift angle of the secondary-side lead arm drive and lagging arm drive waveforms, where the lead arm is defined as the arm containing switches S3 and S4, and the lagging arm is defined as the arm containing switches S5 and S6. For v ab1,N (ω s t) ahead of time v cd1,N (ω s The outward phase angle of t).

[0062] Inward phase angle θ and outward phase angle The specific situation when working in both directions is shown in Figure 3.

[0063] Based on the results obtained from the fundamental frequency approximation, the voltage and current phasors can be analyzed, and the circuit characteristics of the converter can be analyzed using phasor diagrams and steady-state operating waveforms.

[0064] like Figure 4 The key waveforms of the resonant cavity of the converter during a single switching cycle are shown, namely the alternating square wave voltage generated at the intersection of the AC and DC bridge arms, the resonant inductor current, and the driving waveforms of each switch of the converter.

[0065] definition Let S1 and S4 be the vectors corresponding to the fundamental waves of the driven square waves, and define them as follows: Lag The phase difference is the phase shift angle α, and its expression is shown below.

[0066]

[0067] Simultaneously define Lagging behind The phase difference is β, and its expression is:

[0068] β=αtan(I rq / I rp (4)

[0069] Only when satisfied and The phase relationship between them Lagging behind Only when the phase shift angle α ≥ β can ZVS be guaranteed for all DC-side switches.

[0070] When the converter operates in forward mode, three conditions must be met. First, the input current must be controllable to ensure input power. Second, the primary-side switch must achieve soft switching, which requires ensuring that the resonant current i is within the dead time before switch S3 turns on. rThe direction is negative, and the current magnitude is sufficient to fully charge and discharge the output capacitor within the dead time of the switch; thirdly, to ensure ZVS of all switches on the secondary side, the phase shift angle α ≥ β, or at least α = β, must be satisfied. Based on the above two conditions, the corresponding control quantity, the switching frequency f, can be obtained. s Outward phase angle And the inward phase angle θ, such as Figure 5 As shown.

[0071] (1)Condition 1

[0072] Based on the derivation results above,

[0073]

[0074] To ensure the magnitude of the input current, the active component I of the resonant current must be guaranteed. rp The average value over the switching cycle is equal to the input current i dc This ensures the input power of the converter, i.e.

[0075]

[0076] (2) Condition 2

[0077] Secondly, to ensure the ZVS of the primary side switch, the current at the turn-on moment of the AC side switch can be calculated using the above formula, which is I. rq :

[0078]

[0079] To ensure the realization of ZVS for the AC-side switching transistor, the required turn-on current reference value I can be obtained. switch_ref The expression is shown below, where C oss The output capacitor of the primary-side switching transistor, t d Set the dead time for the AC side switching transistor.

[0080]

[0081] (Its output capacitance is 210pF and its dead time is 240ns).

[0082] (3)Condition three

[0083] To ensure that all DC-side switches achieve ZVS, the phase shift angle is calculated, and the relationship between β and θ is shown below:

[0084]

[0085] Substituting the above formulas into the expression and solving the system of equations, we can obtain the outward phase shift angle at this point. The inward phase angle θ is:

[0086]

[0087] Substituting equations (10) and (11) into equation (5) yields:

[0088]

[0089] Therefore, the switching frequency f s It can be represented as:

[0090]

[0091] When the DC / DC converter operates in reverse mode, it requires the same three conditions as in forward mode. The first condition is achieved by adjusting I... rp The first condition is to ensure that the output current is controllable, thereby ensuring the output power; the second condition is to ensure that the secondary-side switch achieves ZVS, at which point it is only necessary to ensure the resonant current i before switch S1 is turned on. r The direction is positive, and it is necessary to satisfy the condition that the output capacitor C is within the dead time. oss The third condition is to ensure the ZVS of the primary-side switch. In this case, the critical condition β = 0 is taken, and the dead time is increased to ensure that the capacitor is fully charged and discharged.

[0092] At this point, the steady-state waveform in the time domain and the corresponding phasor diagram are as follows: Figure 6 As shown.

[0093] (1)Condition 1

[0094] The analysis and method for the quality conditions of the output current waveform in reverse working mode are basically the same as those in forward working mode.

[0095] (2) Condition 2

[0096] To ensure that the primary-side switch achieves ZVS, it is only necessary to ensure that β = 0. At this time:

[0097] I rq =0(14)

[0098] (3)Condition three

[0099] To ensure that the secondary-side switch achieves ZVS, the reference value of the turn-on current I can be obtained. dc_switch_ref As shown below, where C oss_dc t is the equivalent value of the output junction capacitance of the secondary-side switch. d The dead time set for the secondary-side switch can be used to calculate the phase shift angle α. θ.

[0100] The turn-on current of switch S1 is:

[0101] I dc_switch =n·I rp ·sin(α)(15)

[0102]

[0103] Substituting equation (3) into equations (5) and (14) yields the phase shift angle. The inward phase angle θ is:

[0104]

[0105] Based on this, by substituting equations (17) and (18) into equations (12) and (13), the switching frequency can be obtained.

[0106] The circuit analysis is simple, and soft switching can be implemented within a certain range, but the error is relatively large. Numerical calculations will be used to compensate for the current and correct the error.

[0107] Because the fundamental frequency is approximately ignored due to the excitation and the high-order harmonics of the resonant cavity current and voltage, there is a certain difference between the actual current and the calculated current at the switching moment, which can lead to the loss of soft switching. To ensure that the converter can operate in a soft-switching state, the control module is also used to establish the differential equation of the multi-mode resonant circuit using time-domain numerical calculation methods, solve for the accurate value of the resonant current at the switching moment, and then determine whether the soft-switching condition is met by the magnitude of the resonant current at the turn-on moment. If not, soft switching can be achieved by setting up frequency compensation, which also reduces the effective value of the high-frequency current of the converter and improves efficiency.

[0108] Define ω respectively r Let Z be the resonant angular frequency, Z be the characteristic impedance of the resonant cavity, and F be the ratio of the switching angular frequency to the resonant angular frequency, as shown in the following formula:

[0109]

[0110] In order to ensure that the resonant impedance is inductive during control, the frequency ratio F is always greater than 1.

[0111] During forward operation, a certain reactive current component I is required to ensure the ZVS of the primary-side switch. rq If I is injected rq A small IT may result in insufficient zero-voltage turn-on or even hard turn-on of the switching transistor; if IT is injected... rq A larger value may lead to an increase in the effective value of the current, thereby increasing the conduction loss.

[0112] At this point, based on the time-domain numerical calculation model, the accurate switching current I obtained based on the time-domain analysis method can be calculated. switch_TDA .

[0113]

[0114] As can be seen from the analysis based on the fundamental wave approximation method, I switch_TDA and intermediate control degrees of freedom I rq There is a monotonic relationship between them; a larger Irq can produce a larger I. switch_TDA Therefore, a closed-loop approach can be used to achieve I. switch_TDA The size is adjusted. The specific control strategy block diagram is as follows: Figure 7 As shown.

[0115] In the reverse operating mode of the DC / DC converter, to ensure ZVS of the DC-side switch, it is necessary to ensure that the drive of switch S1 leads i. r Furthermore, its output capacitor can be fully charged and discharged before activation. In the above analysis, by adjusting the phase shift angle α, the phase of G1 is ensured to be relative to v. ab Lag, at this time v ab with i r Being in phase, this allows for the control of the turn-on current I of S1. dc_switch The adjustment of α is similar to that of rectification. If α is adjusted too small, it may result in insufficient zero-voltage turn-on of the switching transistor or even hard turn-on; if α is adjusted too large, it may increase conduction losses.

[0116] Therefore, I can be obtained from the numerical calculation model. dc_switch_TDA :

[0117]

[0118] In summary, the switching frequency f can be derived. s The relationship with the phase shift angle α is as follows: Figure 8 As shown.

[0119] Soft-switching precision compensation based on numerical calculation and analysis can achieve soft switching across the entire voltage range, thereby reducing losses. The specific control strategy block diagram is shown below. Figure 9 As shown.

[0120] In the bidirectional DC-DC high-gain multi-cavity parallel converter, all transformers adopt an integrated design with a shared magnetic core across multiple cavities. The DC / DC converter topology is as follows: Figure 10 As shown, the magnetic core has an E-type structure with multiple symmetrically distributed chambers inside, each accommodating a winding connected in parallel. By optimizing the core cross-sectional area and permeability, the symmetry and independence of the magnetic circuits in each chamber are ensured, avoiding magnetic saturation and cross-interference. The core material is selected from high-frequency, low-loss nanocrystalline alloys or amorphous alloys to meet the high power density requirements.

[0121] This invention first employs a multi-cavity parallel winding structure on the input side. Utilizing the symmetry design of the magnetic core and a current-sharing control strategy, the input current is evenly distributed to each resonant module. After coupling through the magnetic core, voltage superposition is achieved through the secondary-side series windings. Secondly, the coupling effect of the shared magnetic core enhances the flux linkage interaction between the cavities, allowing the reactive component in the resonant current to be dynamically distributed among the cavities. This effectively reduces the reactive current amplitude required per cavity, significantly expanding the range of soft switching. Simultaneously, the high permeability of the magnetic core material reduces excitation losses. Through core material optimization and structural innovation, the amount of magnetic core used is reduced by 40%, and the volume is reduced by 35%, significantly improving power density. Magnetic shielding design reduces EMI radiation, and combined with current-sharing control, extends device lifespan.

[0122] The present invention also provides a control method for the above-mentioned converter, comprising:

[0123] Based on the fundamental phasor model, the inner and outer phase shift angles are coordinated and controlled; the inner phase shift angle is the phase shift angle of the driving signals of the leading and lagging bridge arms in the full-bridge rectifier unit, and the outer phase shift angle is the phase shift angle between the alternating square wave voltage output by the bridge circuit and the alternating square wave voltage output by the full-bridge rectifier unit.

[0124] Furthermore, by employing time-domain numerical calculation methods, the differential equations of the multi-mode resonant circuit are established, and the precise value of the resonant current at the switching moment is obtained by solving them. By adjusting the reactive component or phase shift angle of the resonant current in a closed loop, the approximate error of the fundamental wave is compensated, ensuring that the primary and secondary switching transistors turn on with zero voltage.

[0125] When the converter operates in forward mode, the average value of the active component of the resonant current during the switching cycle is set to be equal to the input current of the primary circuit. During the dead time before the primary switch is turned on, the direction of the resonant current is set to be negative and its magnitude is sufficient to charge and discharge the output capacitor. The phase shift angle of the secondary switch is set to α≥β, where β is the phase difference between the secondary resonant current and the secondary switch drive signal.

[0126] When the converter operates in reverse mode, the active component of the resonant current is adjusted to control the output current of the secondary circuit. During the dead time before the secondary switch is turned on, the direction of the resonant current is set to positive and its magnitude is sufficient to charge and discharge the output capacitor. The phase shift angle β of the primary switch is set to 0, and the dead time of the primary switch is increased. The phase shift angle β is the phase difference between the primary resonant current and the drive signal.

Claims

1. A bidirectional DC-DC high-gain multi-cavity parallel converter, characterized in that, include: Primary circuit, multiple resonant modules, secondary circuit, and output capacitor; The primary-side circuit includes an input capacitor and at least two parallel bridge circuits. The input capacitor is connected in parallel with the input terminal of each bridge circuit to the input terminal of the primary circuit. The output terminal of the bridge circuit is connected in parallel with the primary input terminal of the transformer of the plurality of resonant modules. The input current is then evenly distributed to each of the resonant modules. Each of the resonant modules includes a transformer and a composite resonant network; the multiple transformers share an integrated magnetic core, and the integrated magnetic core has multiple symmetrically distributed chambers inside, each chamber accommodating the primary winding and secondary winding of a transformer; the composite resonant network is electrically connected to the primary winding of the corresponding transformer, together forming a resonant circuit for energy transfer; The secondary circuit includes a full-bridge rectifier unit, a two-stage voltage regulation module, and a series voltage multiplier output circuit connected in sequence; the secondary windings of the multiple transformers are connected in series to achieve voltage superposition, and the superimposed voltage is multiplied by the full-bridge rectifier unit, the two-stage voltage regulation module, and the series voltage multiplier output circuit in sequence. The output capacitor is connected in parallel to the output terminal of the series voltage multiplier output circuit to filter out high-frequency ripple in the output voltage.

2. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 1, characterized in that, The bridge circuit adopts a half-bridge circuit; the half-bridge circuit includes two switching transistors and two capacitors. The two switching transistors are connected in series to form the first bridge arm, and the two capacitors are connected in series to form the second bridge arm. The output terminal of the two bridge arms connected in parallel is connected in parallel with the primary input terminal of the transformer of the multiple resonant modules.

3. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 1, characterized in that, The bridge circuit is a full-bridge circuit; the full-bridge circuit includes four switching transistors S1, S2, S3, and S4 and an output bus capacitor; wherein, switching transistors S1 and S3 are the upper arm switching transistors of the full bridge, and switching transistors S2 and S4 are the lower arm switching transistors of the full bridge; switching transistors S1 and S2 are connected in series to form the left arm of the full bridge, and switching transistors S3 and S4 are connected in series to form the right arm of the full bridge; after the left arm and the right arm are connected in parallel, their two ends are respectively connected to the positive and negative terminals of the input bus capacitor, and the common midpoint of the left arm and the right arm is respectively connected to the primary input terminal of the transformer of the multiple resonant modules.

4. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 1, characterized in that, It also includes a control module; the control module is used to coordinate the inner phase shift angle and the outer phase shift angle based on the fundamental phasor model; the inner phase shift angle is the phase shift angle of the driving signals of the leading and lagging bridge arms in the full-bridge rectifier unit, and the outer phase shift angle is the phase shift angle between the alternating square wave voltage output by the bridge circuit and the alternating square wave voltage output by the full-bridge rectifier unit.

5. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 4, characterized in that, The control module is also used to establish the differential equation of the multimode resonant circuit using time-domain numerical calculation methods, solve for the accurate value of the resonant current at the switching moment, and compensate for the fundamental wave approximation error by adjusting the reactive component or phase shift angle of the resonant current through closed-loop adjustment, so as to ensure zero-voltage turn-on of the primary and secondary switching transistors.

6. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 1, characterized in that, The composite resonant network includes a resonant inductor and a resonant capacitor; the resonant inductor is connected in series with the primary winding of the corresponding transformer, and the resonant capacitor is connected in parallel with the series-connected resonant inductor and the primary winding of the transformer; the resonant inductor, resonant capacitor and the primary winding of the transformer work together to realize energy transfer and assist the soft switching of the primary switching tube.

7. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 4, characterized in that, When the converter is in forward operating mode, the regulation of the control module satisfies the following: the average value of the active component of the resonant current during the switching cycle is equal to the input current of the primary circuit; during the dead time before the primary switch is turned on, the direction of the resonant current is negative and its magnitude meets the charging and discharging requirements of its output capacitor; the phase shift angle α of the secondary switch is ≥ β, where β is the phase difference between the secondary resonant current and the secondary switch drive signal.

8. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 4, characterized in that, When the converter is in reverse operating mode, the control module's regulation satisfies the following: the output current of the secondary circuit is controllable by adjusting the active component of the resonant current; during the dead time before the secondary switch is turned on, the direction of the resonant current is positive and its magnitude meets the charging and discharging requirements of its output capacitor; the phase shift angle β of the primary switch is 0, where β is the phase difference between the primary resonant current and the drive signal, and the dead time of the primary switch is increased.

9. The bidirectional DC-DC high-gain multi-cavity parallel converter according to claim 1, characterized in that, The integrated magnetic core is made of a nanocrystalline alloy.

10. A control method for a bidirectional DC-DC high-gain multi-cavity parallel converter according to any one of claims 1-9, characterized in that, The method includes: Based on the fundamental phasor model, the inner and outer phase shift angles are coordinated and controlled; the inner phase shift angle is the phase shift angle of the driving signals of the leading and lagging bridge arms in the full-bridge rectifier unit, and the outer phase shift angle is the phase shift angle between the alternating square wave voltage output by the bridge circuit and the alternating square wave voltage output by the full-bridge rectifier unit.

11. The control method according to claim 10, characterized in that, The method also includes: By employing time-domain numerical calculation methods, the differential equations of the multimode resonant circuit are established, and the precise value of the resonant current at the switching moment is obtained by solving them. By adjusting the reactive component or phase shift angle of the resonant current in a closed loop, the approximate error of the fundamental wave is compensated, ensuring that the primary and secondary switching transistors turn on with zero voltage.

12. The control method according to claim 10, characterized in that, The method further includes: when the converter is operating in the forward mode, setting the average value of the active component of the resonant current during the switching cycle to be equal to the input current of the primary circuit; during the dead time before the primary switch is turned on, setting the direction of the resonant current to be negative and its magnitude to meet the charging and discharging requirements of its output capacitor; setting the phase shift angle of the secondary switch to α≥β, where β is the phase difference between the secondary resonant current and the secondary switch drive signal.

13. The control method according to claim 10, characterized in that, The method further includes: when the converter is operating in reverse mode, adjusting the active component of the resonant current to control the output current of the secondary circuit; during the dead time before the secondary switch is turned on, setting the direction of the resonant current to be positive and its magnitude to meet the charging and discharging requirements of its output capacitor; setting the phase shift angle β of the primary switch to 0 and increasing the dead time of the primary switch, wherein the phase shift angle β is the phase difference between the primary resonant current and the driving signal.

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