High-precision temperature detection system based on AC impedance bridge

Abstract

The invention relates to a high-precision temperature detection system based on an alternating-current impedance bridge. The high-precision temperature detection system comprises a double-resistance bridge circuit, a lock-in amplifier and a differential circuit, the double-resistor bridge circuit comprises a resistive temperature sensor and a standard resistor which are connected in series, and the phase-locked amplifier is a digital phase-locked amplifier realized based on an FPGA (Field Programmable Gate Array). Wherein the initial potential of a first voltage measurement end arranged at the midpoint of the double-resistor bridge circuit is zero by adjusting the two excitation signals which are loaded to the double-resistor bridge circuit and have the same frequency and opposite phases before temperature measurement; in the temperature measuring process, the lock-in amplifier detects the potential of the first voltage measuring end and the voltage at the two ends of the standard resistor through the differential circuit, then the resistance value of the resistance-type temperature sensor is calculated by combining the resistance value of the standard resistor, and then the temperature of the resistance-type temperature sensor is reduced. According to the invention, the anti-interference capability can be improved, and the measurement precision of 0.1 mK level can be reached near 273K.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of high-precision temperature detection and weak signal detection, and particularly relates to a high-precision temperature detection system based on an alternating current impedance bridge. BACKGROUND

[0002] In the modern industry, scientific research and medical field, a high-precision temperature measurement system is of great significance to product quality control and experimental data accuracy. Traditional temperature detection technology usually adopts a single-resistance measurement method, but has obvious deficiencies in weak temperature change detection and anti-interference performance. Specifically, a traditional temperature detection circuit is usually built based on a platinum resistance. The platinum metal has stable chemical properties, small long-term drift and high stability. In addition, the resistance-temperature relationship of the platinum resistance is close to linear, and has the advantage of high precision. The temperature measurement accuracy of a platinum resistance temperature sensor PT1000 can reach ±0.3K (Class B, 273.15K), and the temperature measurement error of some models of platinum resistance temperature sensors can be further controlled within ±0.1℃ (or ±0.1K). However, the actual temperature measurement accuracy of a temperature measurement system is not only determined by the temperature measurement accuracy of the platinum resistance temperature sensor itself. For example, the resistance of the wire itself for measuring voltage and current, the contact resistance, electromagnetic interference, and the change of the distribution capacitance of the coaxial cable caused by the slight disturbance of the mechanical environment, and various factors can all have a negative impact on the actual temperature measurement accuracy of the temperature measurement system. Therefore, although the temperature measurement accuracy of the PT1000 platinum resistance temperature sensor itself can reach ±0.1K, the traditional temperature detection technology using the single-resistance measurement method may be difficult to actually achieve this performance indicator.

[0003] To overcome the above technical problems, a measurement scheme combining a platinum resistance temperature sensor with a Wheatstone bridge is proposed. For example, reference can be made to the following document: Niu G, Tang E, Han W, et al. High-precision AC impedance bridge temperature measurement system based on high-frequency excitation[J]. Navigation, Positioning and Timing, 2018, 5(05): 101-106. DOI:10.19306 / j.cnki.2095-8110.2018.05.017. In this scheme, a Wheatstone bridge is constructed by three low-temperature drift precision standard resistors and a platinum resistance temperature sensor. Based on the bridge balance principle, the small changes of the platinum resistance temperature sensor can be detected with a higher resolution, thereby improving the measurement accuracy. At the same time, the current and voltage at both ends of the platinum resistance temperature sensor are measured by combining the four-wire method, which can further eliminate the interference of contact resistance and wire resistance. Further, in some newer temperature measurement technologies, a lock-in amplifier is used to provide an AC excitation signal for the bridge, and digital signal processing is performed through phase-sensitive detection algorithms, and finally the bridge output voltage representing the temperature is obtained to realize temperature measurement. This scheme can further improve the temperature measurement accuracy and anti-interference ability. For example, under the environmental conditions where the small disturbance of the mechanical environment causes the distributed capacitance of the coaxial cable to change, the temperature measurement stability can be improved by an order of magnitude, reaching 0.01℃. However, when higher temperature measurement accuracy is required, such as an order of magnitude improvement on the basis of 0.01℃ or even higher accuracy, the anti-interference ability and measurement accuracy of the above scheme will face great challenges. For example, the Wheatstone bridge is composed of three low-temperature drift precision standard resistors and a platinum resistance temperature sensor. Before actual measurement, the resistance box of the bridge needs to be adjusted to achieve bridge balance, and then the target object is measured for temperature. However, in the process of ultra-high precision measurement, the temperature drift problem of the standard resistor will still affect the measurement result, and the three standard resistors will introduce a lot of complexity. It is actually difficult to maintain the accurate matching of multiple resistors, which limits the bridge balance ability and may induce nonlinear response problems. These problems will make it difficult to further improve the temperature measurement accuracy and anti-interference ability.

[0004] On the other hand, a digital lock-in amplifier is a high-precision instrument for detecting weak signals, and its core advantage lies in realizing noise suppression and signal extraction through digital processing technology. With the development of digital lock-in amplifier technology, its detection sensitivity and detection resolution for weak electrical signals have reached a very high level. However, the detection resolution of weak electrical signals cannot be directly translated into temperature measurement accuracy. Because simply improving the detection resolution of the detection end cannot directly solve the system error of the temperature measurement circuit itself, nor can it solve the error introduced by the parameter drift of each electronic component caused by environmental interference. For example, in the measurement scheme mentioned in the foregoing, which combines a platinum resistance temperature sensor with a Wheatstone bridge, a lock-in amplifier is introduced for phase-sensitive detection to extract weak electrical signals in the temperature measurement circuit, but its temperature measurement accuracy is only 0.01℃. If it is necessary to further improve the accuracy by one order of magnitude or even higher accuracy on the basis of 0.01℃, the anti-interference ability and measurement accuracy of the above-mentioned scheme will face great challenges.

[0005] Finally, in the application scenario of ice point detection, which is a typical application scenario requiring detection of small temperature changes, a conventional temperature measurement system is difficult to achieve a measurement accuracy of 0.1mK near 273K, and is easily disturbed by environmental noise, resulting in unstable measurement results. Moreover, the traditional measurement circuit for ice point detection is insufficient in compensating for the resistance of the sensor connection line, which will introduce significant errors in long-distance temperature monitoring applications. In addition, the existing systems generally have low signal-to-noise ratio and poor common-mode rejection capability when processing weak signals, which makes it difficult to meet the requirements of high-precision temperature measurement, especially when there is a large amount of electromagnetic interference in the environment, the performance of the measurement system will be significantly reduced.

[0006] In summary, there is an urgent need for a high-precision temperature detection solution based on platinum resistance temperature sensors with higher temperature measurement accuracy and better anti-interference ability. SUMMARY

[0007] The purpose of the present application is to overcome the shortcomings of the prior art, and to provide a high-precision temperature detection solution based on platinum resistance temperature sensors with higher temperature measurement accuracy and better anti-interference ability.

[0008] The application provides a high-precision temperature detection system based on an alternating current impedance bridge, which comprises an alternating current impedance bridge, a phase-locked amplifier and a differential circuit, characterized in that the alternating current impedance bridge is a double-resistance bridge circuit comprising a resistance temperature sensor and a standard resistance in series; the phase-locked amplifier is a digital phase-locked amplifier based on FPGA, which comprises a signal generator module for generating two digital excitation signals in the form of sinusoidal waves with the same frequency and opposite phases by direct digital synthesis technology; a digital-to-analog conversion module for converting the digital excitation signals into analog signals and loading the converted two excitation signals into both ends of the double-resistance bridge circuit; an analog-to-digital conversion module for receiving the differential detection signal output by the differential circuit and converting it into a digital signal; and a digital phase-locked calculation core module for receiving the digitized differential detection signal and calculating the amplitude and phase of the differential detection signal based on correlation degree.

[0009] The digital phase-locked amplifier is also used for adjusting the virtual zero point of the double-resistance bridge circuit before measuring the temperature, and the virtual zero point adjustment comprises fine-tuning the amplitude ratio of the two excitation signals to make the signal amplitude at the output end of the first differential circuit always lower than a preset threshold.

[0010] The digital phase-locked calculation core module comprises a quadrature demodulation module, a low-pass filter module and an amplitude and phase calculation module.

[0011] The phase-locked amplifier loads the two excitation signals into both ends of the double-resistance bridge through a coaxial line, and the double-resistance bridge uses a coaxial line to realize the series connection of the resistance temperature sensor and the standard resistance.

[0012] The coaxial line loaded with the two excitation signals is a cable with a double shielding structure of aluminum foil and tinned copper mesh; and the shielding cable connecting the differential circuit and the double resistance bridge is a four-core twisted shielding line.

[0013] The resistance temperature sensor is a PT1000 platinum resistance, and the standard resistance is a metal foil resistance.

[0014] The detection system further comprises a host computer, which is configured to issue working parameters to the phase-locked amplifier, the working parameters including the frequency, peak-to-peak voltage, phase, sampling rate and filter parameters of the two excitation signals; and the host computer is further configured to obtain a temperature measurement result according to an interpolation method index of a sensor resistance-temperature relationship.

[0015] The differential circuit adopts a two-stage amplifier architecture, which comprises an input buffer stage and a differential amplification stage, and an isolation stage is arranged between the input buffer stage and the differential amplification stage; the differential circuit is implemented based on an instrument amplifier chip, the instrument amplifier chip has a common-mode rejection ratio (CMRR) > 120 dB and an open-loop gain (G) > 120 dB; the isolation stage comprises a signal conversion module and a signal reconstruction module, wherein the signal conversion module has an electrical isolation function, and the signal reconstruction module is configured to reconstruct the signal passing through the signal conversion module, so as to ensure that the weak signal from the input buffer stage is transmitted to the differential amplification stage faithfully after passing through the signal conversion module.

[0016] The digital phase-locked amplifier is further configured to adjust a virtual zero point of the double resistance bridge circuit before measuring the temperature, the virtual zero point adjustment comprising: adjusting the combination of the amplitude ratio and the phase difference of the two excitation signals in two dimensions, so that the amplitude of the measured signal at the output end of the first differential circuit is always lower than a preset threshold; or adjusting the combination of the amplitude ratio and the phase difference of the two excitation signals in two dimensions to find an extreme point at which the measured signal at the output end of the first differential circuit is closest to zero, and then determining the preferred combination of the amplitude ratio and the phase difference.

[0017] The resistance temperature sensor is configured to detect the freezing point of an ice-water mixture.

[0018] Compared with the prior art, the present application has at least one of the following technical effects:

[0019] 1. In some embodiments of the present application, a double resistance bridge is used instead of a Wheatstone bridge, a virtual zero point is set instead of the physical ground of the traditional bridge, and a dual excitation of a high-precision digital lock-in amplifier is used, which simplifies the complexity of the resistance network of the alternating current impedance bridge, thereby suppressing the system tolerance caused by the resistance value drift of different resistances due to various environmental factors, and effectively improving the temperature measurement accuracy of ultra-high-precision temperature monitoring.

[0020] 2. In some embodiments of the present application, the measurement accuracy of ice point detection can be improved, and the measurement result instability caused by environmental noise interference can be suppressed. The actual measurement shows that the temperature measurement accuracy of ±0.1mK is achieved near 273K (i.e. near 0℃).

[0021] 3. In some embodiments of the present application, the virtual zero point is adjusted by adjusting the amplitude ratio of the dual excitation, which can compensate for the sensor connection line resistance and various interference factors, thereby improving the anti-interference ability and temperature measurement accuracy, while taking into account the zero adjustment efficiency, so that the zero adjustment process before temperature measurement can be completed in a short time. The zero adjustment process of the traditional Wheatstone bridge often needs to adjust the resistance value of the adjustable resistor in the resistance box, and the adjustment accuracy and convenience of the zero adjustment process before temperature measurement of the present application are significantly better than the scheme based on the Wheatstone bridge.

[0022] 4. In some other embodiments of the present application, the amplitude ratio and phase difference are combined for two-dimensional adjustment to comprehensively compensate for various random interference factors such as temperature measurement circuit and its connection line resistance, parasitic capacitance, and the like, thereby better improving the anti-interference ability and temperature measurement accuracy. BRIEF DESCRIPTION OF DRAWINGS

[0023] Figure 1 A high-precision temperature detection system logic structure diagram based on a double resistance bridge and a lock-in amplifier in an embodiment of the present application is shown;

[0024] Figure 2 A sub-module decomposition schematic diagram of a high-precision temperature detection system based on a double resistance bridge and a lock-in amplifier in an embodiment of the present application is shown;

[0025] Figure 3 A double bridge circuit structure schematic diagram of the present application is shown;

[0026] Figure 4 A circuit structure schematic diagram of a high-precision temperature detection system based on a double resistance bridge and a lock-in amplifier in an embodiment of the present application based on a dual excitation mode is shown;

[0027] Figure 5 A differential circuit design schematic diagram in an embodiment of the present application is shown;

[0028] Figure 6A flowchart of a high-precision temperature detection system based on a dual-resistance bridge and a lock-in amplifier according to an embodiment of the present invention is shown.

[0029] Figure 7 The circuit structure diagram of a high-precision temperature detection system based on a single-excitation method using a dual-resistance bridge and a lock-in amplifier is shown in the comparative experiment. Detailed Implementation

[0030] To better understand this application, various aspects of this application will be described in more detail with reference to the accompanying drawings. It should be understood that these detailed descriptions are merely illustrative of exemplary embodiments of this application and are not intended to limit the scope of this application in any way. Throughout the specification, the same reference numerals refer to the same elements. The expression "and / or" includes any and all combinations of one or more of the associated listed items.

[0031] It should be noted that in this specification, the terms "first," "second," etc., are used only to distinguish one feature from another and do not imply any limitation on the features. Therefore, without departing from the teachings of this application, the first subject discussed below may also be referred to as the second subject.

[0032] In the accompanying drawings, the thickness, size, and shape of the objects have been slightly exaggerated for ease of illustration. The drawings are for illustrative purposes only and are not drawn to scale.

[0033] It should also be understood that the terms "comprising," "including," "having," "containing," and / or "comprising," when used in this specification, indicate the presence of the stated features, integrals, steps, operations, elements, and / or components, but do not exclude the presence or addition of one or more other features, integrals, steps, operations, elements, components, and / or combinations thereof. Furthermore, when expressions such as "at least one of..." appear after a list of listed features, they modify the entire listed feature, not individual elements in the list. Additionally, when describing embodiments of this application, the word "may" is used to mean "one or more embodiments of this application." And the term "exemplary" is intended to refer to an example or illustration.

[0034] As used herein, the terms “basically,” “approximately,” and similar terms are used as terms of approximation rather than terms of degree, and are intended to describe inherent biases in measured or calculated values ​​that will be recognized by those skilled in the art.

[0035] Unless otherwise defined, all terms (including technical and scientific terms) used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs. It will be further understood that terms, such as those defined in commonly used dictionaries, should be interpreted as having a meaning that is consistent with their meaning in the context of the relevant art and will not be interpreted in an idealized or overly formal sense unless expressly so defined herein.

[0036] It should be noted that the embodiments in the present application and the features in the embodiments can be combined with each other without conflict.

[0037] The present application will be further described below in conjunction with the drawings and specific embodiments.

[0038] As shown in Figure 1 , Figure 2 , according to one embodiment of the present application, a high-precision temperature detection system based on a double-resistance bridge and a phase-locked amplifier is provided, and the logic structure thereof includes an upper computer, a digital phase-locked amplifier (which is realized in the form of an FPGA circuit), a double-resistance bridge circuit and a differential circuit. The upper computer and the FPGA circuit are connected through an Ethernet, and the digital phase-locked amplifier, the double-resistance bridge circuit and the differential circuit are connected with each other through a shielded cable.

[0039] As shown in Figure 3 , in the present embodiment, the double-resistance bridge circuit is composed of a resistance temperature sensor and a standard resistance connected in series. Referring to Figure 1 and Figure 2 , in the present embodiment, the digital phase-locked amplifier is realized based on an FPGA, and includes an FPGA master module, a signal generator module, a digital-to-analog conversion module, an analog-to-digital conversion module and a digital phase-locked calculation core module. The digital phase-locked calculation core module can include a quadrature demodulation module, a low-pass filter module and an amplitude and phase calculation module. Among them, the signal generator module is used to generate two digital excitation signals in the form of sinusoidal waves with the same frequency and opposite phases through a direct digital synthesis technology (the two digital excitation signals correspond to Figure 2two excitation signals are loaded to the two ends of the double resistance bridge circuit. The analog-digital conversion module is used to receive the differential detection signal output by the differential circuit and convert it into a digital signal. The digital phase-locked calculation core module is used to receive the digitized differential detection signal and calculate the amplitude and phase of the differential detection signal based on the correlation degree. The concept of correlation degree can be referred to CN201110380805.X, which will not be described here. The principle of detecting the phase, frequency and amplitude is: based on the reference signal (usually a signal with the same waveform and frequency as the excitation signal) and the input measured signal (i.e. the differential detection signal), search for the phase-frequency combination with the maximum correlation degree in the phase space and frequency space, so as to obtain the accurate phase, frequency and amplitude of the measured signal. In this embodiment, the FPGA module is used to realize the above-mentioned correlation degree-based detection. Specifically, the digital phase-locked calculation core module realizes the correlation degree calculation through two multipliers and a low-pass filter (integrator), and directly outputs two direct current components: in-phase component (X) and quadrature component Y. The amplitude R and the phase θ are calculated through the two components: R=sqrt(X 2 +Y 2 ), θ=atan2(Y,X). The sqrt() represents the square root function. The atan2() is a function for calculating the inverse tangent value of a given coordinate point.

[0040] Further, referring to Figure 4 , the phase-locked amplifier has two input and two output ports, which are: the first input port AD1, the second input port AD2, the first output port DA1, and the second output port DA2. After receiving the instructions from the host computer, the phase-locked amplifier can generate excitation with set frequency, peak-to-peak value and phase, and output to the double resistance bridge through the first output port DA1 (or DA1 channel) and the second output port DA2 (or DA2 channel). The double resistance bridge outputs a differential voltage signal proportional to the temperature. The first input port AD1 and the second input port AD2 (or AD1 channel and AD2 channel) of the phase-locked amplifier receive the analog voltage signal from the differential circuit, and then sample and digitize the amplified analog signal.

[0041] Still referring to Figure 4In the embodiment, the differential circuit can include a first differential circuit and a second differential circuit. In the double resistance bridge circuit, a first voltage measurement end is arranged at the midpoint of the connecting line of the resistance temperature sensor and the standard resistance, the first voltage measurement end is connected to one input end of the first differential circuit through a shielded cable, the other input end of the first differential circuit is physically grounded, and the two input ends of the second differential circuit are respectively connected to the two ends of the standard resistance and the connection is also realized through the shielded cable. The differential circuit converts the voltage signals at the two ends of the standard resistance and the temperature sensor into single-ended differential signals, is the interface between the double resistance bridge circuit and the FPGA circuit (i.e. the digital lock-in amplifier), and can significantly improve the performance and reliability of the entire temperature detection system.

[0042] In particular, in the present embodiment, no physical ground terminal is provided in the double-resistance bridge circuit, and instead, the two excitation signals (for the sake of description, the two excitation signals are referred to as dual-excitation signals) of the same frequency and opposite phase loaded to the double-resistance bridge circuit are used to make the initial potential of the first voltage measurement terminal zero before measuring the temperature; during the temperature measurement, the phase-locked amplifier detects the signal amplitude and phase of the first differential circuit and the second differential circuit respectively, and then combines the resistance value of the standard resistor to calculate the resistance value of the resistance temperature sensor, and further restores the temperature of the resistance temperature sensor. The dual-excitation signals are the two excitation signals output by the first output port DA1 (or DA1 channel) and the second output port DA2 (or DA2 channel) of the digital phase-locked amplifier. In the present embodiment, the double-resistance bridge is used instead of the Wheatstone bridge, the virtual zero point is used instead of the physical ground of the traditional bridge, and the dual-excitation of the high-precision digital phase-locked amplifier is used, which effectively improves the temperature measurement accuracy of the ultra-high-precision temperature monitoring. The traditional Wheatstone bridge needs to rely on the accurate proportional relationship between the four resistors to achieve bridge balance. However, during actual temperature measurement, especially when ultra-high-precision temperature measurement is required, the circuit topology of such a bridge is complex, the impedance characteristics are inconsistent, and in a certain temperature measurement environment, the four resistors used to build the Wheatstone bridge may deviate in impedance, and such deviation may be different. Under the influence of such a relatively complex circuit topology, the entire temperature measurement circuit has a nonlinear system tolerance. In some prior art, although the resistance drift (such as temperature drift of resistance impedance) can be compensated based on some empirical formula on the host computer, the effect of such compensation is limited. The temperature measurement accuracy of the temperature measurement circuit based on the Wheatstone bridge is usually only 0.01°C. When higher temperature measurement accuracy is required, for example, when the accuracy is improved by one order of magnitude or more based on 0.01°C, the temperature measurement circuit based on the Wheatstone bridge is usually difficult to achieve. In the present embodiment, the double-resistance bridge replaces the four-resistor Wheatstone bridge, and the dual-excitation of the high-precision digital phase-locked amplifier forms a virtual zero point in the bridge to replace the physical ground of the traditional bridge. This design simplifies the circuit topology, improves the symmetry of the circuit, suppresses the nonlinear system error caused by the resistance deviation in the bridge caused by various environmental disturbances. On the other hand, the ultra-high control accuracy of the digital phase-locked amplifier is fully utilized to participate in the zeroing process of the virtual zero point combined with the actual measured waveform, thereby compensating for various disturbances and effectively improving the temperature measurement accuracy. Experiments show that the temperature measurement system based on dual-excitation and double-resistance bridge reaches a measurement accuracy of ±0.1mK near 273K. In the following, the mechanism of the temperature measurement system will be further described in combination with specific experiments.

[0043] Further, in one embodiment, the host computer is configured to send on-off instructions, process collected data, and display temperature information in real time. The host computer sends working parameters to the lock-in amplifier through a communication interface, including excitation signal frequency, peak-to-peak voltage, phase, sampling rate, and filter parameters. The effective working frequency of the embodiment is 10 Hz to 10 kHz, and the effective working range of the peak-to-peak voltage is 0.2 V to 0.8 V. After receiving the instructions from the host computer, the lock-in amplifier can generate an excitation signal with the set frequency, peak-to-peak voltage, and phase, and output it to the double-resistance bridge through the DA1 channel and the DA2 channel. The bridge outputs a differential voltage signal proportional to the temperature. The differential circuit amplifies the weak differential signal output by the bridge, suppresses common-mode noise, and improves signal quality to prepare for subsequent lock-in amplifier processing.

[0044] The first differential circuit is used to measure the voltage difference across the PT1000 sensor, convert the analog differential voltage signal to a single-ended signal, and provide it to the AD1 channel of the lock-in amplifier. This eliminates common-mode interference and only focuses on the actual voltage difference across the PT1000 sensor, enhancing the system's noise suppression capability. The second differential circuit is used to measure the voltage difference across the standard resistor, and the output is provided to the AD2 channel of the lock-in amplifier.

[0045] In one embodiment, the first differential circuit and the second differential circuit can have the same structure. Both can be collectively referred to as differential circuits. By matching the appropriate input impedance, the differential circuit can reduce signal reflection and distortion, and through differential processing, the system can distinguish smaller temperature changes, thereby improving the resolution of the entire temperature measurement system and ensuring the quality of the signal output from the double-resistance bridge to the lock-in amplifier. The AD1 channel and the AD2 channel of the lock-in amplifier receive analog voltage signals from the differential circuit, and then the ADC module samples and digitizes the amplified analog signals.

[0046] Further, as Figure 5As shown, in some embodiments of the present application, the differential circuit can include an input buffer stage, an isolation stage and a differential amplification stage in sequence. The input buffer stage is used to provide a higher input impedance and a lower output impedance for the subsequent circuit to buffer the signal from the sensor or the AC impedance bridge. The input buffer stage can prevent the load variation (such as resistance network adjustment) from affecting the output characteristics of the signal from the sensor or the AC impedance bridge, and reduce the influence of resistance matching error on the common mode rejection ratio. In this embodiment, the input buffer stage is implemented based on two negative feedback circuits, the positive input terminal of the first operational amplifier U1 is used as the first buffer input terminal in_1 for connecting the first signal output terminal of the double resistance AC impedance bridge, and the positive input terminal of the second operational amplifier U2 is used as the second buffer input terminal in_2 for connecting the second signal output terminal of the double resistance AC impedance bridge. The negative input terminal of the first operational amplifier U1 is physically grounded through the first grounding resistor R1. The negative input terminal of the second operational amplifier U2 is physically grounded through the second grounding resistor R2. The output terminal of the first operational amplifier U1 is connected to the negative input terminal of the first operational amplifier U1 through the first feedback resistor R3. The output terminal of the second operational amplifier U2 is connected to the negative input terminal of the second operational amplifier U2 through the second feedback resistor R4. The output terminals of the first operational amplifier U1 and the second operational amplifier U2 are respectively used as the two output terminals of the input buffer stage, and are connected to the next stage of the differential circuit. In this embodiment, the next stage is the isolation stage. Figure 5 In the following, the power connection terminals of the operational amplifier are not described again.

[0047] The isolation stage is a high-precision signal transmission module with electrical isolation function. In an embodiment, the isolation stage can be implemented based on the AD210 isolation amplifier chip. The isolation stage can suppress common mode interference and ground loop noise through electrical isolation, and at the same time, ensure the transmission of the required signal to the next stage.

[0048] In other embodiments, the isolation stage described above can also be designed by itself. For example, in an embodiment, the isolation stage can include a mutual inductor for realizing the electrical isolation function and a signal pickup amplification module. The mutual inductor receives the output of the input buffer stage, and in this embodiment, the mutual inductor is used to suppress common mode interference and ground loop noise based on the electrical isolation function. The signal pickup amplification module is used to pick up and amplify the weak signal output by the mutual inductor, so as to ensure that the weak signal from the input buffer stage can be transmitted to the next stage, i.e. the differential amplification stage, with high fidelity after passing through the mutual inductor. In specific implementation, an operational amplifier coupled with the mutual inductor can be used to realize high-fidelity transmission. It should be noted that, Figure 5 The isolation stage in the above embodiment is only a schematic diagram, which roughly shows the logical structure of the isolation stage. When the signal from the input buffer stage has two paths, the isolation stage is also designed to have two paths, each of which has a mutual inductor and an operational amplifier coupled with the mutual inductor.Figure 5 The fourth operational amplifier U4) is shown. The operational amplifier can be implemented by an AD8221 chip (i.e., an AD8221 instrument amplifier chip), two output terminals of the transformer can be connected to positive and negative input terminals of the AD8221 chip respectively to realize coupling, and an output terminal of the AD8221 chip can be connected to the negative input terminal of the AD8221 chip through the fourth feedback resistor R7.

[0049] In the isolation stage of the present application, the transformer can also be replaced by an optical coupling component. The optical coupling component realizes electrical isolation of the temperature measurement circuit through photoelectric conversion and electro-optical conversion, and then picks up and amplifies the weak signal output by the optical coupling component, so as to ensure that the weak signal from the input buffer stage can be transmitted to the next stage with high fidelity after passing through the transformer. In the implementation scheme of the above isolation stage, the transformer or the optical coupling component can be regarded as a signal conversion module with electrical isolation function, and the signal pickup and amplification module can be understood as a signal reconstruction module, which is used to reconstruct the signal passing through the signal conversion module, so as to ensure that the weak signal from the input buffer stage can be transmitted to the next stage with high fidelity after passing through the transformer.

[0050] In an embodiment of the present application, the differential amplification stage is implemented based on a negative feedback circuit. The positive input terminal and the negative input terminal of the third operational amplifier U3 are connected to two output signals of the isolation stage respectively. The positive input terminal of the third operational amplifier U3 is connected to an input resistor R5, the first signal is input to the positive input terminal of the third operational amplifier U3 through the input resistor R5, and the second signal is directly input to the negative input terminal of the third operational amplifier U3. The output terminal of the third operational amplifier U3 is connected to the negative input terminal of the third operational amplifier U3 through the third feedback resistor R6. Moreover, the output terminal of the third operational amplifier U3 is the output terminal of the differential amplification stage, and is also the output terminal of the entire differential circuit. In the present embodiment, the first operational amplifier U1, the second operational amplifier U2 and the third operational amplifier U3 can use the same type of operational amplifier, for example, an AD8221 instrument amplifier chip. In the present embodiment, the common-mode rejection ratio CMRR of the AD8221 instrument amplifier chip is >120 dB, and the open-loop gain G is >120 dB.

[0051] According to an embodiment of the present application, the digital phase-locked amplifier is implemented by using a JFM7K325T series FPGA chip, which internally integrates complete digital phase-locked amplifier functions, including an ADC module and a DAC module for signal acquisition and processing, a DDS signal generator module for reference signal generation, a quadrature modulation module, a low-pass filter module and an amplitude and phase calculation module for phase-sensitive detection, and a core module for data communication with the upper computer.

[0052] The ADC module uses the AD9268 dual-channel 16-bit high-speed ADC chip of Analog Devices Company, and the maximum sampling rate can reach 125 MSPS, effectively supporting high-frequency signal detection. The JFM7K325T chip has a high-speed serial transceiver and a configurable I / O interface, and its LVDS interface can directly realize high-speed data transmission with the differential digital output interface of the AD9268, with an effective bandwidth of up to 62.5 MHz. The ADC interface module realizes high-speed parallel data acquisition and digital down-conversion processing, and contains a digital filter for suppressing ADC quantization noise. At the same time, in order to ensure measurement accuracy, the on-chip reference voltage source of AD9268 adopts low-temperature drift and low-noise design, with a temperature coefficient better than 2.5ppm / K, maintaining high-precision data conversion within the full operating temperature range (233.15K to 358.15K).

[0053] The DDS signal generator module makes full use of the built-in DSP48 unit of the JFM7K325T chip, adopts a 48-bit phase accumulator structure, and the frequency resolution reaches 0.001Hz, with a phase noise lower than -120dBc / Hz@1kHz. The digital sine wave output by the DDS module is converted into an analog signal by a 16-bit high-precision DAC, which is used to drive the bridge circuit. In order to improve the flexibility of the system, the DDS module supports multiple waveform outputs (sine wave, square wave, triangle wave), and can dynamically adjust the amplitude, frequency and phase parameters.

[0054] The quadrature demodulation module is the core part of the FPGA phase-locked amplifier, which uses the built-in 300+ high-performance DSP operation units of the JFM7K325T chip to realize high-precision digital multiplication operation. This structure realizes the multiplication operation of the input signal and the reference signal, with an operation speed of up to 500MHz and a precision of up to 18 bits, meeting the requirements of high-precision phase-locked detection. The I and Q signals after quadrature demodulation are filtered by a digital low-pass filter to filter out high-frequency components.

[0055] The low-pass filter module makes full use of the rich on-chip RAM resources of the JFM7K325T chip, and adopts a reconfigurable IIR+FIR hybrid structure. The filter order and cutoff frequency can be dynamically configured by the host computer. The IIR filter provides fast primary filtering, while the FIR filter ensures linear phase characteristics and stable group delay. The filter bandwidth can be adjusted in the range of 0.001Hz to 100Hz to adapt to different response speed requirements of measurement scenarios.

[0056] The amplitude and phase calculation module is realized based on the optimized CORDIC algorithm, fully utilizes the parallel computing capability of the JFM7K325T chip, reduces the FPGA resource occupation by 30% compared with the traditional implementation method, and improves the calculation precision. The module calculates the amplitude and phase of the I and Q signals, and corrects the inherent error of the CORDIC algorithm through a lookup table, so that the calculation result precision reaches 18 effective digits.

[0057] Reference Figure 3 According to one embodiment of the present application, the double resistance bridge circuit is composed of a temperature sensing element PT1000 and a standard resistance R. The temperature sensing element PT1000 (also referred to as PT1000 sensor) adopts a Midas+ series M310 model platinum resistance temperature sensor produced by Heraeus Company in Germany, with a package size of 0805 and a nominal resistance of 1000Ω (0℃). Here, PT1000 is the model of the sensor, i.e. PT1000 platinum resistance. The PT1000 sensor has high precision, high stability and good linearity, with a temperature coefficient of 3850ppm / K. The double resistance bridge circuit can adopt a traditional four-wire connection mode. In this embodiment, the double resistance bridge circuit is provided with four BNC connectors, namely a first connector BNC_1, a second connector BNC_2, a third connector BNC_3 and a fourth connector BNC_4. Under the four-wire connection mode, the first connector BNC_1 and the fourth connector BNC_4 form a first pair of connection terminals for providing excitation current. The second connector BNC_2 and the third connector BNC_3 form a second pair of connection terminals for voltage measurement. The BNC connector can also be replaced by an SMA connector. The four-wire connection mode can effectively eliminate the influence of lead resistance on measurement accuracy, and is particularly suitable for high-precision temperature measurement applications. The standard resistance R adopts a Y17451K00000T9R model metal foil resistance produced by Vishay Precision Group Inc Company in the United States, with a nominal resistance of 1000Ω and a typical temperature coefficient of ±2ppm / ℃, having extremely high precision and stability. The metal foil resistance is manufactured by special foil technology, with extremely low noise, excellent long-term stability and extremely small temperature coefficient. Figure 4In the dual excitation mode, the first output port DA1 is connected to the first joint BNC_1 through a cable, and the second output port DA2 is connected to the second joint BNC_2 through a cable. The first input end of the first differential circuit is connected to the double resistance bridge through the fourth joint BNC_4, and the second input end of the first differential circuit is physically grounded, for detecting the potential change of the midpoint of the double resistance bridge, and then obtaining the resistance change of the temperature sensor. The first input end and the second input end of the second differential circuit are connected to the double resistance bridge through the fourth joint BNC_4 and the second joint BNC_2 respectively, for detecting the signal between the standard resistance R. In specific implementation, a three-way joint can be connected to the fourth joint BNC_4, so that the fourth joint BNC_4 can be connected to the first input end of the second differential circuit and the first input end of the first differential circuit through two cables respectively.

[0058] In the following, a single excitation mode will also be mentioned. Referring to Figure 7 , in the single excitation mode, the first output port DA1 is connected to the first joint BNC_1 through a cable, and the second joint BNC_2 is physically grounded. The first input end and the second input end of the first differential circuit are connected to the third joint BNC_3 and the fourth joint BNC_4 respectively, for detecting the voltage signal between the temperature sensor pt1000 (i.e. the temperature sensing element PT1000 in Figure 3 ). The first input end of the second differential circuit is connected to the fourth joint BNC_4, and the second input end of the second differential circuit is physically grounded, for detecting the voltage signal between the standard resistance R. In specific implementation, a three-way joint can be connected to the fourth joint BNC_4, so that the fourth joint BNC_4 can be connected to the first input end of the second differential circuit and the first input end of the first differential circuit through two cables respectively.

[0059] The PT1000 sensor and the standard resistance R form a double resistance bridge structure. When the ambient temperature changes, the resistance of PT1000 changes accordingly, while the standard resistance R remains stable. By measuring the voltage difference between the two ends of the bridge, the resistance change of the PT1000 sensor can be accurately calculated, and then the accurate temperature value can be obtained.

[0060] Referring to Figure 6In one embodiment of the present application, the temperature measurement process of the high-precision temperature detection system based on the AC impedance bridge comprises the following steps. System initialization: the host computer starts and configures system parameters, including excitation signal frequency, amplitude, phase, sampling rate and filter parameters; signal generation: the host computer sends instructions to the FPGA circuit to set the parameters of the phase-locked amplifier, and the FPGA generates a high-precision sine wave as the excitation signal through the direct digital synthesis (DDS) technology to ensure the stability of the signal frequency and phase; bridge excitation: the generated excitation signal is applied to the bridge circuit composed of the PT1000 sensor and the standard resistor; signal conditioning: the differential circuit amplifies and conditions the weak signal output by the bridge to improve the signal-to-noise ratio; data acquisition: the FPGA collects the conditioned signal through a high-speed AD interface and performs digital processing; phase-locked demodulation: the phase-locked amplifier performs quadrature demodulation on the collected signal and the reference signal to separate the in-phase and quadrature components; signal filtering: a digital low-pass filter is applied to the demodulated in-phase and quadrature components to filter out high-frequency noise; data processing: the amplitude and phase information are transmitted to the host computer through Ethernet for impedance calculation to obtain the resistance change, and the temperature value is converted according to the characteristic curve of the PT1000 sensor; display and storage: the host computer displays, records and analyzes the received temperature data in real time.

[0061] It should be noted that in the above embodiment, the double-resistor bridge circuit does not have a physical ground terminal, but a virtual zero point is set instead of a physical ground. Specifically, before measuring the temperature, the two excitation signals (i.e., the dual excitation signals) of the same frequency and opposite phase loaded into the double-resistor bridge circuit can be adjusted so that the initial potential of the first voltage measurement terminal is zero, thereby obtaining the virtual zero point. During the temperature measurement process, the phase-locked amplifier detects the signal amplitude and phase of the output terminals of the first differential circuit and the second differential circuit, respectively, and then calculates the resistance value of the resistive temperature sensor in combination with the resistance value of the standard resistor, and further restores the temperature of the resistive temperature sensor. Further, the adjustment of the virtual zero point includes: by adjusting the amplitude ratio of the two excitation signals, the signal amplitude of the output terminal of the first differential circuit is adjusted to approach zero stably. In specific implementation, the signal amplitude of the output terminal of the first differential circuit is always lower than a preset threshold.

[0062] The application provides a high-precision temperature detection based on a dual-bridge and a phase-locked amplifier with dual excitation, which improves the self-balancing and anti-interference ability of the temperature detection circuit and is suitable for a more severe measurement environment. The dual excitation mode is adopted, and the excitation signals with a phase difference of 180 degrees are provided through two digital output ports (i.e. a first output port DA1 and a second output port DA2) of the FPGA. Compared with the single excitation mode, the dual excitation mode simultaneously applies the signals with opposite phases to the two arms of the bridge, thereby forming a true near-zero potential balance driving. This design significantly reduces the sensitivity of the system to external interference and improves the amplitude of the effective signal. The balance degree of the bridge circuit has an important influence on the temperature detection precision. When the bridge is in a balanced state, the output signal is close to zero, which provides an accurate reference point. The higher the balance degree, the more accurate the reference point, and the smaller the error of the temperature detection starting point. In the ideal state of complete balance, even a small temperature change can cause a detected output change, thereby improving the temperature detection resolution. When the change of the resistance of the measured temperature sensor relative to the standard resistance is large, the amplitude ratio of the dual excitation can be adjusted to easily restore the zero balance of the bridge, so that the resistance change that determines the deviation from the zero imbalance can be effectively amplified and measured with higher precision. When the bridge is completely balanced, the common-mode interference is cancelled out, thereby improving the signal-to-noise ratio. A well-balanced bridge can obtain sufficient output signals with a smaller excitation current, thereby reducing the measurement error caused by the self-heating effect of the sensor. In summary, the balance degree of the temperature detection bridge directly determines the basic precision, sensitivity and stability of the temperature detection system, and is a key factor affecting the temperature detection precision.

[0063] It should be noted that some temperature detection systems based on the Wheatstone bridge in the prior art also use a phase-locked amplifier as a signal detection device. However, the role of the phase-locked amplifier is to extract a weak signal (such as a platinum resistance voltage signal) by phase-sensitive detection, and the interference ability of the system tolerance of the temperature detection circuit itself is insufficient. If the temperature detection circuit itself has a system tolerance, the output signal itself will carry this system tolerance, so even if the phase-locked amplifier itself has high sensitivity, the entire temperature detection system cannot achieve corresponding temperature detection precision. In practice, this scheme often relies on a specific compensation algorithm to modify and compensate the directly measured data to obtain high-precision temperature measurement results, and the temperature detection precision is often limited to 0.01℃.

[0064] However, the Wheatstone bridge has four resistors, and the deviations generated by the four resistors when encountering various interference factors are different, and the change in the proportional relationship often presents a nonlinear characteristic, and the compensation algorithm may fail when higher precision measurement is required. On the other hand, the premise of using such a compensation algorithm to compensate for the tolerance of the temperature measurement circuit system is to accurately understand the main interference factors causing the tolerance of the temperature measurement circuit system and quantitatively study the mechanism of action. However, in many cases, various interference factors naturally existing in the temperature measurement system and the surrounding environment are often very complex, and sometimes it is even impossible to determine what the interference factors causing the tolerance of the temperature measurement circuit system are, at which time it is difficult to use the compensation algorithm to improve the temperature measurement precision.

[0065] In this application, the digital phase locked amplifier is not only used to detect weak signals, but also deeply intervenes in the temperature measurement circuit itself through the design of dual excitation and virtual zero point. The zero setting process based on the actual measured waveform is essentially compensating for the system tolerance caused by various interference factors of the temperature measurement circuit (mainly the double resistance bridge) by means of the extremely high control accuracy of the digital phase locked amplifier. Moreover, this compensation can be performed without specific analysis of the various interference factors causing the system tolerance, and even if there are unknown interference factors, as long as the zero setting is based on the actual measured waveform, the interference factors can be suppressed. Therefore, the scheme of the present application can have higher temperature measurement precision and wider application scenarios compared with the traditional compensation algorithm based scheme.

[0066] The inventor sets up a zero point detection test environment. A thermostat is used to ensure constant external temperature. An ice water mixture is used to act on the measured object in the sample pool. A small volume of liquid is used for temperature measurement, and a large number of measurements are repeated to obtain statistically meaningful measured data.

[0067] Experiments show that compared with the traditional single excitation method, the dual excitation method proposed in the present application has enhanced anti-interference ability and improved measurement precision stability. Specifically, the dual excitation structure greatly enhances the anti-interference ability of the system, and when external common mode interference simultaneously affects the two signal paths of the circuit, these noises will be effectively suppressed in the dual amplifier. In the test, the inventor sets up an experimental environment with 50Hz power frequency interference, and the actual temperature measurement results show that the suppression ability of the dual excitation method to the power frequency interference is improved by about 20dB compared with the single excitation method, which enables the system to maintain high precision measurement in harsh industrial environments. Through the dual excitation method, the detection sensitivity of the system to small temperature changes is significantly improved.

[0068] On the other hand, in the absence of actively imposed disturbance variables to the temperature measurement system, long time span temperature measurements are performed using the single excitation mode and the dual excitation mode respectively, to compare the effects of the two excitation modes on the tolerance of the temperature measurement circuit system in the natural state (i.e. without actively imposed disturbance variables). Specifically, the sample cell is placed in a thermostat, and a small volume sample cell of 50 ml is used to ensure that the sample temperature remains constant. An ice water mixture is added to the sample cell. An intermittent data acquisition method is used to balance data density and total time. In the test, the test sampling rate is set to 2 kHz, i.e. one data is collected every 0.5 ms, and each continuous temperature measurement is 10 s, obtaining a total of 20,000 data. After each temperature measurement, a period of time is allowed to elapse before the next temperature measurement is performed. The total number of temperature measurements N is generally not less than 100. The intermittent time between adjacent two temperature measurements is set to 1 minute, 10 minutes, 30 minutes, etc. To ensure that the actual temperature of the measured object, i.e. the ice water mixture in the sample cell, remains unchanged (it can be understood that the actual temperature of the ice water mixture fluctuates far less than 0.1 mK, i.e. it can be considered that the temperature remains unchanged), the total time span of continuous use of the sample cell is not more than 4 hours. When more than 4 hours, the ice water mixture in the sample cell needs to be replaced, and the virtual zero point of the temperature measurement system needs to be adjusted. Then a new measurement is started. In order to better compare the differences between the dual excitation mode and the single excitation mode, the test data and test times of the dual excitation mode are set to 4 times of the single excitation mode, while the intermittent time between adjacent two temperature measurements is approximately the same as the single excitation mode. In other words, more times and longer time span tests are performed on the dual excitation mode in the test, in order to better observe whether the dual excitation mode can indeed better suppress the system tolerance (such as the system tolerance caused by various disturbance factors naturally existing in the temperature measurement system and the surrounding environment).

[0069] The experimental results show that, in the same measurement environment, the dual excitation mode can keep the measurement accuracy at ±0.1mK for a longer time at 273K, reducing the measurement error and meeting the demand for long-time high-precision temperature detection. Specifically, when using single excitation mode and not actively applying disturbance variables to the temperature measurement system, in the long-time temperature monitoring process, it is found that the measurement value floats between 1.24E-04 and 2.01E-04 (unit: °C), and some data readings show that the temperature measurement result deviates from the actual temperature of the measured object (i.e. ice-water mixture) by more than ±0.1mK. It can be reasonably inferred that the temperature measurement system was disturbed by one or more factors in the corresponding time period (such as electromagnetic interference in the experimental environment, small disturbance of the mechanical environment causing disturbance of the electrical characteristics of the coaxial cable, etc. Note that these disturbance factors are not active disturbance variables, but natural disturbance factors existing in the temperature measurement system and the surrounding environment). At the same time, in the same measurement environment, using the dual excitation mode, the temperature measurement results of a large number of collected data always keep within ±0.1mK. That is, as long as no active disturbance variables are applied, even if some disturbance factors naturally exist in the temperature measurement system and the surrounding environment, the dual excitation mode can still guarantee a temperature measurement accuracy of ±0.1mK. From the above comparative experiment, it can be inferred that the dual excitation mode can effectively improve the anti-interference ability of the temperature measurement system when performing ultra-high precision temperature measurement (temperature measurement accuracy reaches 0.001°C or higher).

[0070] Moreover, under the dual excitation mode, a plurality of groups of monitoring results in some time periods keep within 1.01E-04 (unit: °C), which indicates that in the corresponding time period (it is presumed that the disturbance factors disappear in this time period), the tolerance is actually close to the level of ±0.05mK. This indicates that if the natural disturbance factors existing in the temperature measurement system and the surrounding environment can be effectively eliminated, the measurement accuracy of the temperature measurement system of the present application is expected to reach a level close to ±0.05mK.

[0071] To further verify the stability and robustness of the dual-excitation method in a long time span, a grouping analysis was performed on all the dual-excitation test data. Through grouping analysis, the details can be avoided to be hidden due to the large amount of data. The grouping basis is directly derived from the above experimental control conditions. Specifically, to ensure the absolute constancy of the temperature of the ice-water mixture in the sample cell, the total time span of each measurement is strictly controlled within 4 hours. If it exceeds 4 hours, the sample needs to be replaced and the system needs to be recalibrated, which naturally forms an independent measurement batch. A large amount of data obtained by the dual-excitation method is divided into 4 groups according to the corresponding measurement time batch, and each group represents an independent measurement period completed within a 4-hour time window. This grouping method helps to evaluate whether the dual-excitation method can maintain high precision in the face of possible slow-changing background interference (such as slight drift of the environmental electromagnetic field, slow relaxation of cable stress, etc.) in different measurement periods. The analysis results show that among the 4 independent data groups, 3 groups of data show high consistency, and the 99.7% data tolerance of the 3 groups of data is better than 1.4E-04℃, which is significantly improved compared with the single-excitation method (the 99.7% data tolerance of the single-excitation method is about 1.7E-04℃). This shows that the improvement effect of the dual-excitation is not accidental, but has high repeatability and stability. Although the other group of data has a slightly larger deviation, all data points are still strictly controlled within the tolerance band of ±0.1mK (2.0E-04℃), which is still better than the single-excitation mode. The inventors preliminarily speculate that the appearance of this group of data may be due to the presence of relatively large random interference factors in the experimental environment (for example, changes in cable stress or other factors may cause parasitic capacitance in the temperature measurement circuit, thereby affecting the temperature measurement data). Experiments show that even in batches that encounter relatively strong interference, the dual-excitation mechanism can still ensure that the system precision does not break the design upper limit of ±0.1mK, proving its reliable fault tolerance capability. Through this grouping analysis based on the experimental time batch, it is proved that the dual-excitation method can effectively improve the anti-interference ability and precision stability of the system in multiple independent experimental periods.

[0072] Further, in one embodiment of the present application, in order to suppress the influence of relatively large random interference factors (for example, cable stress changes or other factors can cause parasitic capacitance in the temperature measurement circuit), in the process of adjusting the virtual zero point, the adjustment of the phase difference is introduced, that is, the two-dimensional adjustment of the amplitude ratio and the phase difference is used instead of the amplitude ratio adjustment in the foregoing embodiment. The zero adjustment process based on two-dimensional adjustment not only compensates for the resistance value changes of the resistance network in the alternating current impedance bridge (for example, resistance value drift caused by various environmental factors), but also compensates for the capacitance changes of the alternating current impedance bridge and its connecting cable (for example, parasitic capacitance caused by cable stress changes or other factors). Specifically, in an ideal state, the dual excitation signals should be two signals of the same frequency and opposite direction, and the phase difference between the two signals is 180°. However, some random interference factors can cause the capacitance of the alternating current impedance bridge and its connecting cable to change, which can cause the phase difference between the two excitation signals output by the DDS signal generator module of the digital phase-locked amplifier to be inconsistent with the phase difference of the dual excitation actually acting on the dual-resistance alternating current impedance bridge. In order to make the phase difference of the dual excitation actually acting on the dual-resistance alternating current impedance bridge reach or very close to the ideal state of 180°, in the present embodiment, the phase difference between the two excitation signals output by the DDS signal generator module is allowed to be fine-tuned within a small range (for example, within a range of ±0.01°). In this way, the zero adjustment process is changed from the single amplitude ratio adjustment in the foregoing embodiment to two-dimensional adjustment of the amplitude ratio and the phase difference, that is, to find the preferred combination of the amplitude ratio and the phase difference, so that the measured signal at the output end of the first differential circuit (hereinafter referred to as the measured signal) is stably close to zero. The criterion for determining stable close to zero can be based on the threshold of one or more parameters (for example, peak-to-peak voltage, etc.). When the corresponding parameters of the measured signal waveform are all less than the preset threshold, it can be considered that the criterion of stable close to zero is met. When it is difficult to find the amplitude ratio and phase difference combination that makes the corresponding parameters of the measured signal waveform all less than the threshold, the preferred amplitude ratio and phase difference combination can also be determined by finding the extreme point of the measured signal closest to zero, so as to complete the zero adjustment of the virtual zero point. This method takes into account the zero adjustment efficiency and the suppression effect on the tolerance, and avoids too long zero adjustment time. In the present embodiment, the two-dimensional adjustment of the amplitude ratio and the phase difference is used instead of the amplitude ratio adjustment in the foregoing embodiment. When there are relatively large random interference factors in the temperature measurement circuit and its surrounding environment, using the two-dimensional adjustment scheme of the present embodiment to adjust the virtual zero point can better suppress the influence of random interference factors.

[0073] In another embodiment of the present application, in order to suppress the influence of relatively large random interference factors, when selecting the connecting cable, a cable with a relatively short length (e.g. less than 0.1 m) can be used to reduce the parasitic capacitance generated in the temperature measurement circuit caused by cable stress changes, thereby reducing the system tolerance. This scheme can be combined with the aforementioned amplitude ratio-based adjustment to zero the virtual zero point, which has relatively high efficiency, and at the same time, since the influence of some relatively large random interference factors is avoided, the system tolerance of the temperature measurement circuit is also better suppressed.

[0074] The above experiments show that the present application achieves a temperature measurement accuracy of ±0.1 mK near 273 K. However, it should be noted that this does not mean that the interference suppression and system tolerance reduction capability of the present application scheme only works near 273 K. Since the accuracy of a general temperature control box or other actively temperature-controllable device is limited, for example, a temperature control box can only achieve a temperature control accuracy of 0.01℃. Therefore, when performing higher accuracy tests, the inventors use an ice-water mixture as the sample to be measured, which is a temperature-constant substance. If a solid-liquid mixture of other substances is used as the sample to be measured, and the temperature is kept constant at the phase transition temperature, the performance of the temperature measurement system of the present application at other temperatures can also be measured. Due to limited experimental conditions, only experimental results based on ice-water mixtures are provided. However, in principle, the present application scheme also has the ability to reduce the system tolerance of the temperature measurement circuit near other temperatures.

[0075] It should be noted that although platinum resistance temperature sensors are used in the AC impedance bridge in each of the embodiments described above, the present application is not limited thereto, and in other embodiments, other types of resistance temperature sensors can also be used, for example, MF501 semiconductor temperature sensors can be used instead of platinum resistance temperature sensors.

[0076] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present application and are not limiting. Although the present application has been described in detail with reference to the embodiments, those skilled in the art should understand that modifications or equivalent replacements to the technical solutions of the present application do not deviate from the spirit and scope of the present application, and they should be included in the scope of the claims of the present application.

Claims

1. A high-precision temperature detection system based on an AC impedance bridge, comprising: AC impedance bridge, lock-in amplifier and differential circuit, characterized in that, The AC impedance bridge is a dual resistance bridge circuit, which includes a resistive temperature sensor and a standard resistor connected in series. The lock-in amplifier is a digital lock-in amplifier implemented based on FPGA. It includes a signal generator module for generating two sinusoidal digital excitation signals of the same frequency and opposite phase using direct digital synthesis technology; a digital-to-analog converter module for converting the digital excitation signals into analog signals and loading the two converted excitation signals onto the two ends of the dual-resistance bridge circuit; an analog-to-digital converter module for receiving the differential detection signal output by the differential circuit and converting it into a digital signal; and a digital lock-in calculation core module for receiving the digitized differential detection signal and calculating the amplitude and phase of the differential detection signal based on the correlation. The differential circuit includes a first differential circuit and a second differential circuit. In the dual-resistance bridge circuit, a first voltage measurement terminal is set at the midpoint of the connection line between the resistive temperature sensor and the standard resistor. The first voltage measurement terminal is connected to one input terminal of the first differential circuit through a shielded cable. The other input terminal of the first differential circuit is physically grounded. The two input terminals of the second differential circuit are respectively connected to the two ends of the standard resistor, and this connection is also achieved through the shielded cable. The dual-resistance bridge circuit does not have a physical ground terminal. Instead, before measuring the temperature, the two excitation signals with the same frequency and opposite phase applied to the dual-resistance bridge circuit are adjusted so that the initial potential of the first voltage measurement terminal is zero. During the temperature measurement process, the lock-in amplifier detects the amplitude and phase of the output signals of the first differential circuit and the second differential circuit respectively, and then calculates the resistance value of the resistive temperature sensor by combining it with the resistance value of the standard resistor, thereby restoring the temperature of the resistive temperature sensor.

2. The high-precision temperature detection system based on an AC impedance bridge according to claim 1, characterized in that, The digital lock-in amplifier is also used to adjust the virtual zero point of the dual-resistance bridge circuit before measuring the temperature. The adjustment of the virtual zero point includes: fine-tuning the amplitude ratio of the two excitation signals so that the signal amplitude at the output of the first differential circuit is always lower than a preset threshold.

3. The high-precision temperature detection system based on an AC impedance bridge according to claim 2, characterized in that, The core module of the digital phase-locked loop calculation includes an orthogonal demodulation module, a low-pass filter module, and an amplitude and phase calculation module.

4. The high-precision temperature detection system based on an AC impedance bridge according to claim 2, characterized in that, The lock-in amplifier loads the two excitation signals to both ends of the dual-resistance bridge via a coaxial line, and in the dual-resistance bridge, the resistive temperature sensor and the standard resistor are connected in series using a coaxial line.

5. The high-precision temperature detection system based on an AC impedance bridge according to claim 4, characterized in that, The coaxial cable for loading the two excitation signals is a cable with a double-shielded structure of aluminum foil and tin-plated copper mesh; The shielded cable connecting the differential circuit and the dual-resistance bridge is a four-core twisted shielded cable.

6. The high-precision temperature detection system based on an AC impedance bridge according to claim 2, characterized in that, The resistive temperature sensor is a PT1000 platinum resistance thermometer; the standard resistor is a metal foil resistor.

7. The high-precision temperature detection system based on an AC impedance bridge according to claim 2, characterized in that, It also includes a host computer, which is used to send operating parameters to the lock-in amplifier. The operating parameters include the frequency, peak-to-peak voltage, phase, sampling rate and filter parameters of the two excitation signals. The host computer is also used to obtain the temperature measurement result by interpolation based on the sensor resistance-temperature relationship.

8. The high-precision temperature detection system based on an AC impedance bridge according to claim 2, characterized in that, The differential circuit employs a two-stage amplifier architecture, comprising an input buffer stage and a differential amplifier stage, with an isolation stage between them. The differential circuit is implemented using an instrumentation amplifier chip with a common-mode rejection ratio (CMRR) > 120 dB and an open-loop gain (G) > 120 dB. The isolation stage includes a signal conversion module and a signal reconstruction module. The signal conversion module provides electrical isolation, and the signal reconstruction module reconstructs the signal after passing through it, ensuring that the weak signal from the input buffer stage is transmitted faithfully to the differential amplifier stage after passing through the signal conversion module.

9. The high-precision temperature detection system based on an AC impedance bridge according to claim 1, characterized in that, The digital lock-in amplifier is also used to adjust the virtual zero point of the dual-resistance bridge circuit before measuring the temperature. The adjustment of the virtual zero point includes: adjusting the combination of the amplitude ratio and phase difference of the two excitation signals in two dimensions so that the amplitude of the measured signal at the output of the first differential circuit is always lower than a preset threshold; or adjusting the combination of the amplitude ratio and phase difference of the two excitation signals in two dimensions to find the extreme point that makes the measured signal at the output of the first differential circuit closest to zero, thereby determining the preferred combination of amplitude ratio and phase difference.

10. The high-precision temperature detection system based on an AC impedance bridge according to claim 2, characterized in that, The resistive temperature sensor is used to detect the freezing point of the ice-water mixture.

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