Quadrature imbalance compensation method, signal receiving link and signal transceiving system

Abstract

A quadrature imbalance compensation method, signal receiving link, and signal transceiver system are disclosed. The method includes: receiving multiple calibration signals, wherein the calibration signals are non-orthogonally mismatched signals; demodulating the received calibration signals using an orthogonal mixer to obtain a baseband calibration signal; acquiring the amplitude and phase information of the baseband calibration signal at a frequency position, wherein is the baseband signal frequency corresponding to the calibration signal; calculating a compensation coefficient at the frequency position based on the amplitude and phase information of the baseband calibration signal at the frequency position; receiving a signal to be processed, and compensating the signal to be processed using the calculated compensation coefficient.

Description

[0001] This application claims priority to Chinese patent application No. 2024117545165, filed on November 30, 2024, entitled "Synchronization Method, Integrated Circuit, Electromagnetic Wave Device and Terminal Equipment", the contents of which shall be construed as incorporated herein by reference. Technical Field

[0002] This disclosure relates to, but is not limited to, the field of electromagnetic wave device technology, and particularly to an orthogonal unbalance compensation method, a signal receiving link, and a signal transceiver system. Background Technology

[0003] Theoretically, quadrature modulation and demodulation (collectively referred to as quadrature mixing) can provide virtually unlimited image rejection without requiring additional hardware and filters. However, this only applies to theoretical, ideal quadrature modulation and demodulation. In practice, quadrature modulation and demodulation are typically implemented using analog circuits. The quadrature carrier signals required for quadrature modulation and demodulation are difficult to match precisely in analog circuits as they are in digital signal processing (i.e., equal amplitude and 90-degree phase difference). When the gains of the I and Q channels are not perfectly matched, or when the phase difference between them is not precisely 90°, quadrature imbalance (IQ Imbalance) occurs. Due to the presence of quadrature imbalance, the image rejection capability of quadrature modulation and demodulation deteriorates.

[0004] Figure 1 This diagram illustrates the impact of quadrature imbalance on the downmixing process at the receiver. Figure 1 As shown, the spectrum of the modulated signal received by the receiver is in the frequency range. Positionally conjugate symmetric, due to the presence of orthogonal imbalance (amplitude mismatch and / or phase mismatch), the local oscillator signal at frequency... The position generates a small component, which, after down-mixing by an orthogonal demodulator and filtering by a low-pass filter, yields the desired signal and its mirror signal at the zero-frequency position, as shown in the figure. How to suppress or compensate for the mirror signal is a technical problem that urgently needs to be solved by those skilled in the art. Summary of the Invention

[0005] The following is an overview of the subject matter described in detail herein. This overview is not intended to limit the scope of the claims.

[0006] This disclosure provides an orthogonal imbalance compensation method, which may include: receiving multiple calibration signals, wherein the calibration signals may be non-orthogonal mismatched signals (or non-ideal orthogonal signals), i.e., non-ideal orthogonal signals or IQ mismatched signals; demodulating the received calibration signals using an orthogonal mixer to obtain a baseband calibration signal; and acquiring the baseband calibration signal in... Amplitude and phase information at frequency location, where, The baseband signal frequency corresponding to the calibration signal; based on the baseband calibration signal in Amplitude and phase information at frequency location, calculated The compensation coefficient for frequency position; receiving the signal to be processed, and compensating the signal to be processed using the calculated compensation coefficient (optionally, the compensation operation in the embodiments of this application can be implemented by means of digital signal processing).

[0007] This disclosure provides a signal receiving link, including a signal receiving channel, a computing unit, and a compensation unit, wherein: the signal receiving channel is configured to receive multiple calibration signals, the calibration signals being non-orthogonal mismatched signals; the received calibration signals are demodulated using an orthogonal mixer to obtain a baseband calibration signal; and is also configured to receive a signal to be processed; the computing unit is configured to acquire the baseband calibration signal in... Amplitude and phase information at frequency location, where, The baseband signal frequency corresponding to the calibration signal; based on the baseband calibration signal in Amplitude and phase information at frequency location, calculated The compensation coefficient for frequency position; the compensation unit is configured to compensate the signal to be processed using the calculated compensation coefficient.

[0008] This disclosure also provides a signal transceiver system, including a signal transmission link and a signal reception link, wherein: the signal transmission link includes a transmitting-end baseband processing module and a real-number mixer, wherein the transmitting-end baseband processing module is configured to output a calibration signal or a signal to be processed; the real-number mixer is configured to perform up-mixing processing on the calibration signal or the signal to be processed; the signal reception link includes a local oscillator signal generator, a switch, a quadrature mixer, and a receiving-end baseband processing module, wherein the local oscillator signal generator is configured to output two quadrature local oscillator signals to the quadrature mixer; the switch is configured to select one of the two quadrature local oscillator signals according to a control signal from a control terminal and output it to the real-number mixer; the quadrature mixer is configured to demodulate the received calibration signal or the signal to be processed to obtain a baseband calibration signal or a baseband signal to be processed; the receiving-end baseband processing module is configured to calculate a compensation coefficient based on the amplitude and phase information (such as amplitude information and / or phase information) of the baseband calibration signal, and use the compensation coefficient to compensate the baseband signal to be processed.

[0009] This disclosure also provides an integrated circuit, including a signal receiving link as described in any embodiment of this disclosure, or a signal transceiver system as described in any embodiment of this disclosure.

[0010] This disclosure also provides an electromagnetic wave device, comprising: a carrier; an integrated circuit as described in any of the present disclosure, disposed on the carrier; an antenna disposed on the carrier, or the antenna and the integrated circuit are integrated into a single device disposed on the carrier; the antenna includes a transmitting antenna and a receiving antenna; wherein the integrated circuit is connected to the antenna and is used to transmit electromagnetic wave signals and / or receive electromagnetic wave signals.

[0011] This disclosure also provides a user terminal device, including: a device body; and an electromagnetic wave device disposed on the device body as described in any embodiment of this disclosure; wherein the electromagnetic wave device is used for target detection and / or wireless communication to provide reference information to the operation of the device body.

[0012] The quadrature imbalance compensation method, signal receiving link, and signal transceiver system of this disclosure receive multiple calibration signals (which are non-orthogonally mismatched), demodulate the received calibration signals using an orthogonal mixer, and then adjust the signal based on the baseband calibration signal. Amplitude and phase information at frequency location, calculated The compensation coefficient for frequency position is simple and easy to implement, enabling real-time (on-the-fly) quadrature imbalance compensation. Furthermore, by making the local oscillator signal at the transmitting end one of the local oscillator signals input to the IQ mixer at the receiving end, image suppression can be achieved without the need for an additional local oscillator signal for calibration, and without the need for frequency misalignment between the local oscillator signals at the transmitting and receiving ends during calibration, thus reducing resource overhead.

[0013] After reading and understanding the accompanying diagrams and detailed descriptions, other aspects can be understood. Attached Figure Description

[0014] The accompanying drawings are provided to further illustrate the technical solutions of this disclosure and form part of the specification. They are used together with the embodiments of this disclosure to explain the technical solutions of this disclosure and do not constitute a limitation on the technical solutions of this disclosure. The shapes and sizes of the components in the drawings do not reflect actual proportions and are only intended to illustrate the content of this disclosure.

[0015] Figure 1 This is a schematic diagram illustrating the impact of quadrature imbalance on the downmixing process at the receiver. Figure 2 A schematic diagram of the signal model for quadrature demodulation at the receiver when the signal is quadrature unbalanced. Figure 3 This is a schematic diagram illustrating the principle of a calibration method in related technologies; Figure 4 This is a flowchart illustrating an orthogonal imbalance compensation method as an exemplary embodiment of this disclosure; Figure 5 This is a schematic diagram of the structure of a signal transceiver system as an exemplary embodiment of the present disclosure; Figure 6A and Figure 6B This is a schematic diagram illustrating the spectrum transformation relationship of two signals, as an exemplary embodiment of this disclosure. Figure 7 This is a schematic diagram of the structure of a signal receiving link, which is an exemplary embodiment of the present disclosure. Figure 8 This is a schematic diagram of another signal receiving link structure as an exemplary embodiment of the present disclosure; Figure 9 This is a schematic diagram of the structure of a signal transceiver system as an exemplary embodiment of the present disclosure; Figure 10 This is a schematic diagram of a signal transceiver system including a compensation unit according to an embodiment of the present disclosure; Figure 11 This is a schematic diagram of a signal transceiver system including a BIST unit according to an embodiment of this disclosure; Figure 12 This is a schematic diagram of a signal transceiver system including an auxiliary circuit unit and a BIST unit according to an embodiment of the present disclosure; Figure 13 This is a schematic diagram of the structure of a mixer according to an embodiment of the present disclosure; Figure 14 This is a schematic diagram of the structure of a compensation unit in an embodiment of this disclosure. Detailed Implementation

[0016] To make the objectives, technical solutions, and advantages of this disclosure clearer, the embodiments of this disclosure will be described in detail below with reference to the accompanying drawings. It should be noted that, unless otherwise specified, the embodiments and features described in this disclosure can be arbitrarily combined with each other.

[0017] Unless otherwise defined, the technical or scientific terms used in the embodiments of this disclosure shall have the ordinary meaning understood by one of ordinary skill in the art to which this disclosure pertains. The terms "first," "second," and similar terms used in the embodiments of this disclosure do not indicate any order, quantity, or importance, but are merely used to distinguish different components. Terms such as "comprising" or "including" indicate that the element or object preceding the word encompasses the elements or objects listed following the word and their equivalents, but do not exclude other elements or objects.

[0018] Figure 2 This is a schematic diagram of the signal model for quadrature demodulation at the receiver when the signal is quadrature unbalanced. (Example:) Figure 2 As shown, This is the frequency domain expression of the received modulated signal. This represents the passband frequency response of channel I. The amplitude gain of the I-channel mixer. This represents the phase response of the I-channel mixer. This represents the baseband frequency response of channel I; This is the passband frequency response of the Q channel. For the amplitude gain of the Q-channel mixer, This refers to the phase response of the Q-channel mixer. This is the Q-channel baseband frequency response; (f) is the frequency domain expression of the passband output signal of channel I. This is the frequency domain expression for the output signal of the I-channel mixer. This is the frequency domain expression of the output signal of the I-channel low-pass filter (LPF). (f) is the frequency domain expression of the I-channel baseband output signal; (f) is the frequency domain expression of the Q-channel passband output signal. This is the frequency domain expression for the output signal of the Q-channel mixer. This is the frequency domain expression for the output signal of the Q-channel low-pass filter (LPF). (f) represents the frequency domain expression of the Q-channel baseband output signal, where ADC is an analog-to-digital converter and DSP is a digital signal processor. In this embodiment, the passband includes all circuits from the high-frequency signal to the mixer, and the baseband includes all circuits from the mixer to the ADC.

[0019] according to Figure 2 The signal model shown has the following expected signal. With mirror signal The frequency response function can be expressed as: (1); (2).

[0020] Where ~ represents the baseband equivalent form, and * represents complex conjugate. It should be noted that... Figure 2 In this process, the signal modeling for the LPF, ADC, and all intermediate frequency processing of the I and Q channels has been abstracted to the I channel baseband frequency response. Q-channel baseband frequency response middle.

[0021] The RIR of a quadrature demodulator is defined as the ratio of the power of the image signal component to the power of the desired signal component. RIR measures the quadrature demodulator's ability to suppress unwanted image signal components. The smaller the RIR, the stronger the quadrature demodulator's ability to suppress image signal components. Its minimum value is 0. Therefore, we can obtain: (3).

[0022] in, (4); (5); (6); (7); (8); (9).

[0023] in, RIR for quadrature demodulator, This represents the ratio of the cascaded gain of the I-channel passband, mixer, and baseband to the cascaded gain of the Q-channel passband, mixer, and baseband. This represents the cascaded gain of the I-channel passband, mixer, and baseband. This represents the cascaded gain of the Q-channel passband, mixer, and baseband. This represents the difference between the phase responses of the I-channel passband, mixer, and baseband, and the phase responses of the Q-channel passband, mixer, and baseband. This represents the phase response of the I-channel passband, mixer, and baseband. This indicates the phase response of the Q-channel passband, mixer, and baseband. This indicates the phase response.

[0024] Explanation of equations (3) to (9): (1) The cascaded gain and phase response of the I and Q channels need to be equal; otherwise, the channels will be unbalanced and the mirror signal components will not be completely suppressed.

[0025] (2) The cascaded gain and phase response of each channel should be taken into account. This also means that in each channel, the baseband frequency response can compensate for the distortion caused by passband imbalance or mixer imbalance, and vice versa.

[0026] (3) Digital compensation techniques can be used to compensate for baseband, mixer and passband imbalances in quadrature demodulators, even if these imbalances are frequency-dependent.

[0027] The following section will present a specific model of the baseband signal demodulated by the quadrature demodulator. This represents the baseband equivalent form of the signal required by the quadrature demodulator. This is the frequency offset. This represents the phase offset. Mathematically, this desired signal can be expressed as: (10).

[0028] Therefore, the baseband signal demodulated by the quadrature demodulator It can be represented as: (11).

[0029] In summary, while quadrature mixers theoretically offer virtually unlimited image rejection, providing numerous practical advantages, especially in designing low-cost, low-power, and compact systems, in practice, quadrature mixer architectures are highly sensitive to mismatches (I / Q imbalance) between their two signal paths and the LO signal, severely impacting their image rejection performance. Imbalance caused by any part of the channel (baseband imbalance, mixer LO imbalance, or passband imbalance) can theoretically be compensated for. This is the basis of imbalance compensation techniques. In practice, digital baseband technology is often used to compensate for I / Q imbalance caused by front-end cascading.

[0030] Based on the baseband signal model demodulated by the quadrature demodulator shown in equation (11), to consider how to calibrate, we must first know how to perform compensation. Ideally, a clean signal should be obtained after compensation. To eliminate unwanted mirror signal components observed at the demodulator output, some form of post-compensation (or correction) should be applied.

[0031] A simple post-compensation operation is shown in the following equation: (12).

[0032] in, and These represent the demodulated baseband signals during post-compensation. and its conjugate signal The compensation coefficient, This is the compensated signal.

[0033] To eliminate unwanted image signal frequency components, it should be made ,if The optimal solution is: (13).

[0034] From equation (13), we can see that the function and Compensation can be achieved simply by estimating the ratio between them, without needing to estimate them separately.

[0035] Figure 3 This is a schematic diagram illustrating the principle of a calibration method in related technologies. For example... Figure 3 As shown, the transmitter sends a single-tone (or single-frequency) signal, which is located at the LO frequency of the demodulator. of At the frequency offset, and with an initial phase of θ relative to the LO signal, the time-domain expression of this single-tone signal is: (14). Among them, 2 This represents the signal amplitude.

[0036] right Performing a Fourier transform, we get: (15).

[0037] Its baseband equivalent form can be expressed as: (16). Among them, This is the Dirac function.

[0038] Based on the demodulator distortion model given in equation (11), the Fast Fourier Transform (FFT) can be used to obtain the distortion at f= The peak value of the desired signal component at f= The ratio of the peak values ​​of the mirror signal components at that location, that is, as shown in the figure. Figure 3 As shown, calculate and The ratio of their complex conjugates is obtained as follows: (17); It can be seen that by using frequency f= Using a single-tone signal as the input signal to the demodulator, the frequency f can be estimated. The frequency response value of the post-compensation filter at that point. If the imbalance of the demodulator can be considered frequency-independent (in the narrowband case), then... It can be directly used for frequency-independent inphase quadrature mismatch (FID-IQM).

[0039] However, this simple calibration method has the following implementation difficulties: (1) If the above calibration method is used to send a device with For a single-tone signal, if the transmitting end also has an orthogonal upmixer, then the LO at the transmitting end and the LO at the receiving end need to have a certain frequency difference.

[0040] (2) If the auxiliary circuit is used for transmission, although the transmitter does not need an orthogonal upmixer, an additional LO is still required.

[0041] The related technology also provides another calibration method, but this calibration method has the following disadvantages: (1) it requires a large amount of observation data, and (2) it obtains the optimal solution of a frequency point by solving a non-convex optimization problem, and the calibration algorithm is too complicated.

[0042] like Figure 4 As shown, this disclosure provides an orthogonal unbalance compensation method, including: Step 401: Receive multiple calibration signals, which are non-orthogonal mismatched signals; use an orthogonal mixer to demodulate the received calibration signals to obtain the baseband calibration signal; Step 402: Obtain the baseband calibration signal. Amplitude and phase information at frequency location, where, The baseband signal frequency corresponding to the calibration signal; based on the baseband calibration signal in Amplitude and phase information at frequency location, calculated Compensation coefficient for frequency position; Step 403: Receive the signal to be processed and compensate the signal to be processed using the calculated compensation coefficient.

[0043] The quadrature imbalance compensation method provided in this disclosure involves receiving multiple calibration signals at a receiving end, wherein the calibration signals are non-quadrature mismatched signals; demodulating the received calibration signals using a quadrature mixer to obtain a baseband calibration signal; and then applying the baseband calibration signal to... Amplitude and phase information at frequency location, calculated The compensation coefficient for frequency position is simple and easy to implement, enabling real-time (On the Fly) orthogonal imbalance compensation.

[0044] In this embodiment of the disclosure, the calibration signal can be obtained by upmixing the baseband single-tone signal using a superheterodyne structure (real number mixer) instead of using an orthogonal upmixer at the transmitting end. In this way, the calibration signal received by the receiving end does not have problems such as orthogonal imbalance (i.e., orthogonal mismatch).

[0045] In this embodiment of the disclosure, a real mixer refers to a non-orthogonal mixer, such as a single-sideband up-mixer or an RF mixer.

[0046] In this embodiment of the disclosure, the amplitude and phase information includes amplitude information and / or phase information.

[0047] In some exemplary embodiments, the signal to be processed may be an ultra-wideband signal; however, this disclosure does not limit this.

[0048] In some exemplary embodiments, receiving multiple calibration signals in step 401 specifically includes: Obtain one or more baseband signal frequencies ; For each baseband signal frequency The baseband single-tone signal receives at least two calibration signals.

[0049] In this embodiment of the disclosure, if the imbalance of the demodulator can be considered frequency-independent (e.g., in the narrowband case), then only one signal frequency needs to be considered. The baseband single-tone signal receives at least two calibration signals; if the demodulator imbalance is frequency-dependent, then multiple signal frequencies need to be selected. At each signal frequency The corresponding values ​​are calculated using the methods of the embodiments of this disclosure. The compensation coefficient for frequency position.

[0050] In some exemplary embodiments, the received at least two calibration signals include a first calibration signal and a second calibration signal, wherein there is a preset phase difference between the local oscillator signal corresponding to the first calibration signal and the local oscillator signal corresponding to the second calibration signal.

[0051] In this embodiment of the disclosure, the preset phase difference can be any angle value, such as 85°, 90°, 95°, etc., and this disclosure does not limit it.

[0052] It should be noted that the amplitude of the local oscillator signal corresponding to the first calibration signal and the amplitude of the local oscillator signal corresponding to the second calibration signal may be equal or unequal, and this disclosure does not impose any restrictions on this.

[0053] In some exemplary embodiments, the local oscillator signal of the real number mixer input corresponding to the first calibration signal is the same as the local oscillator signal of the I-channel mixer input of the quadrature mixer; The local oscillator signal corresponding to the real number mixer input of the second calibration signal is the same as the local oscillator signal input to the Q-channel mixer of the quadrature mixer.

[0054] In some exemplary embodiments, the calibration signal can be obtained by upmixing a baseband single-tone (or baseband single-frequency) signal using a real-number mixer.

[0055] In this embodiment of the present disclosure, there may be a preset phase difference between the local oscillator signal input to the real number mixer corresponding to the first calibration signal and the local oscillator signal input to the real number mixer corresponding to the second calibration signal.

[0056] In this embodiment of the present disclosure, for a communication device integrating transmission and reception, by making the local oscillator signal input to the real number mixer corresponding to the first calibration signal the same as the local oscillator signal input to the I-channel mixer of the quadrature mixer, and the local oscillator signal input to the real number mixer corresponding to the second calibration signal the same as the local oscillator signal input to the Q-channel mixer of the quadrature mixer, the transmitting end and the receiving end can share a single local oscillator signal generator. This eliminates the need for an additional local oscillator signal for calibration and eliminates the need for frequency misalignment between the local oscillator signal of the transmitting end and the local oscillator signal of the receiving end during calibration, thereby achieving image suppression and reducing resource overhead.

[0057] In some exemplary embodiments, the frequency is calculated according to the following formula. Position compensation coefficient: (18); (19); in, This indicates that the baseband calibration signal corresponding to the first calibration signal is in Amplitude and phase information obtained from frequency location. This indicates that the baseband calibration signal corresponding to the first calibration signal is in Amplitude and phase information obtained from frequency location. This indicates that the baseband calibration signal corresponding to the second calibration signal is in Amplitude and phase information obtained from frequency location. This indicates that the baseband calibration signal corresponding to the second calibration signal is in Amplitude and phase information obtained from frequency location.

[0058] The following is through Figure 5 The schematic diagram of the signal transceiver system shown is as follows: Figure 6A and Figure 6B The schematic diagram of the signal spectrum transformation relationship shown illustrates the principle of the orthogonal unbalance compensation method of this disclosure.

[0059] like Figure 5 As shown, the transmitter adopts a superheterodyne structure, and the receiver adopts a quadrature demodulation structure. In order to ensure that the local oscillator (LO) corresponding to the baseband single-tone signal of the transmitter can be accurately phase-shifted, the two LOs of the quadrature mixer of the receiver can be selected by a switch and connected to the LO signal input terminal of the real number mixer of the transmitter. This ensures that during the calibration process, the corresponding input LO of the real number mixer of the transmitter can be either the I-channel LO or the Q-channel LO of the receiver.

[0060] In this embodiment of the disclosure, the local oscillator signal input to the quadrature mixer is clock-synchronized with the local oscillator signal input to the real number mixer.

[0061] Figure 6A and Figure 6BThis is a schematic diagram illustrating the calibration principle of this disclosure. Figure 6A In this context, the LO at the transmitting end is the I-channel LO at the receiving end. Figure 6B In this context, the LO at the transmitting end is the Q-channel LO at the receiving end. Figure 6A For example, the baseband processor at the transmitting end transmits a baseband monotone signal (frequency: ),use This indicates that the phase is 0°. After mixing at the transmitter's LO (at this point, the transmitter's LO is the receiver's I-channel LO), the following can be obtained: Figure 6A shown and After transmitting it to the receiver and passing it through an orthogonal downmixer, the following equation can be obtained. (20); (twenty one); in, Desired signal frequency response, Desired signal frequency response, For mirror signal frequency response, For mirror signal frequency response, This indicates that when the LO at the transmitting end is the same as the I-channel LO at the receiving end, at the frequency point... Amplitude and phase information acquired from the location. This indicates that when the LO at the transmitting end is the same as the I-channel LO at the receiving end, at the frequency point... Amplitude and phase information acquired from the location.

[0062] Similarly, such as Figure 6B As shown, when the LO at the transmitter is the same as the Q-channel LO at the receiver, the following equation can be obtained: (twenty two); (twenty three).

[0063] in, This indicates that when the LO at the transmitting end is the Q-channel LO at the receiving end, at the frequency point... Amplitude and phase information acquired from the location. This indicates that when the LO at the transmitting end is the Q-channel LO at the receiving end, at the frequency point... Amplitude and phase information acquired from the location.

[0064] Multiply both sides of equations (20) and (21) by We can obtain: = (twenty four); (25).

[0065] Therefore, by combining equations (20), (21), (24), and (25), we can obtain: (26); (27); (28); (29).

[0066] This allows us to obtain the frequency. and frequency Position compensation coefficient: (30); (31).

[0067] In summary, by simply acquiring and calculating the amplitude and phase information for the required frequency, the corresponding compensation coefficient can be obtained. When the imbalance of the demodulator can be considered frequency-independent (in the narrowband case), the compensation coefficients corresponding to equations (30) and (31) can be used for frequency-independent inphase quadrature mismatch (FID-IQM) compensation.

[0068] When the imbalance of the demodulator is considered to be frequency-dependent, the compensation coefficients of multiple frequencies can be obtained and then fitted to obtain the coefficients of the compensation filter. The compensation filter is then used to compensate the conjugate signal of the demodulated baseband signal, thereby eliminating unwanted image signal components and achieving frequency-dependent inphase quadrature mismatch (FD-IQM) compensation.

[0069] This embodiment of the disclosure constructs four equations through two tests, and then obtains the result by solving the equations. Compensation coefficients at two frequency positions. During the two tests, the LO can be rotated at any angle, and the amplitude of the LO in each test can be arbitrary, as long as the entire circuit operates in the linear region and the ADC does not saturate.

[0070] For example, suppose the amplitude gain and phase delay of the Q-channel LO at the receiver are respectively and Then the equations (22) and (23) can be further expressed as: (32); (33).

[0071] Solving the system of equations (20), (21), (32), and (33) simultaneously, we can obtain the solution. Compensation coefficients at two frequency positions.

[0072] In some exemplary embodiments, for each baseband signal frequency The baseband single-tone signal receives at least three calibration signals and is calculated using the least squares criterion. The compensation coefficient for frequency position.

[0073] This disclosure applies to each signal frequency. This allows for more tests, the construction of more equations, and the use of the least squares approach to obtain more accurate estimates.

[0074] In some exemplary embodiments, in step 402, the baseband calibration signal is obtained by any of the following methods. Amplitude and phase information at frequency location: Perform a Fourier transform on the baseband calibration signal to obtain the amplitude and phase information at the corresponding frequency position; The corresponding frequencies of the two demodulated quadrature signals are shifted to zero frequency by a digital mixer, and the average value of the two quadrature signals at zero frequency is calculated to obtain the amplitude and phase information at the corresponding frequency position.

[0075] As mentioned above, , , and These represent the ADC data received by the receiver at specific frequency points during the two tests. Amplitude and phase information of the location. In this embodiment of the disclosure, this information can be obtained through the following two methods. , , and : 1) The amplitude and phase information of the corresponding frequency point can be obtained through Fourier transform (FFT); 2) The required frequency point can be shifted to zero frequency by using a digital mixer in the digital domain, and then the amplitude and phase information of the corresponding frequency point can be obtained by averaging the I / Q ADC data.

[0076] In some exemplary embodiments, in step 403, the calculated compensation coefficients are used to compensate the signal to be processed, including: Obtain the baseband signal to be processed after the quadrature mixer demodulates the signal to be processed, and calculate the conjugate signal of the baseband signal to be processed. Using the calculated The compensation coefficient at the frequency position is multiplied by the corresponding frequency component in the conjugate signal to obtain the compensated conjugate signal. The compensated conjugate signal is added to the baseband signal to be processed to obtain the compensated signal.

[0077] For FID-IQM compensation, the IQ compensation coefficients are almost identical at various frequencies. Therefore, to simplify the calculation, after obtaining a frequency... corresponding After applying the frequency position compensation coefficient, a compensation unit comprising a conjugate module, a multiplier, and an adder can be used to directly compensate the received signal to be processed. Optionally, the compensation unit may include a digital signal processor (DSP), etc. In some exemplary embodiments, the compensation unit may further include a delay module for delaying the baseband signal to be processed, so that the compensated conjugate signal is aligned with the time domain of the delayed baseband signal to be processed.

[0078] In some other exemplary embodiments, in step 403, the calculated compensation coefficients are used to compensate the signal to be processed, including: obtaining the calculated compensation coefficients for multiple baseband single-tone signals with signal frequencies within a preset bandwidth; fitting the obtained compensation coefficients to obtain the coefficients of the compensation filter; obtaining the baseband signal to be processed after demodulation processing of the signal to be processed by the quadrature mixer, and calculating the conjugate signal of the baseband signal to be processed; filtering the conjugate signal of the baseband signal to be processed using the compensation filter to obtain the filtered conjugate signal; and adding the filtered conjugate signal to the baseband signal to be processed to obtain the compensated signal.

[0079] In this embodiment, for FID-IQM compensation, the compensation coefficient is a single compensation coefficient; however, for FD-IQM compensation, since the IQ compensation coefficients may differ at different frequencies, multiple frequency compensation coefficients can be fitted into the coefficients of a compensation filter. This compensation filter is then used to filter the conjugate signal of the baseband signal to be processed, obtaining the filtered conjugate signal. The filtered conjugate signal is then added to the baseband signal to be processed to obtain the compensated signal. Similarly, a delay module can be provided to delay the baseband signal to be processed, ensuring that the time domain of the filtered conjugate signal is aligned with the time-delayed baseband signal to be processed.

[0080] The transmitter in this embodiment does not use an orthogonal upmixer, but instead uses a superheterodyne structure (real number mixer). This eliminates issues such as orthogonal mismatch at the transmitter. It eliminates the need for an additional local oscillator signal for calibration, and avoids frequency misalignment between the transmitter's and receiver's local oscillator signals during calibration, thus achieving image suppression and reducing resource overhead. The orthogonal imbalance compensation method in this embodiment is simple to implement and enables real-time calibration.

[0081] like Figure 7 or Figure 8 As shown, this disclosure also provides a signal receiving link, including a signal receiving channel 10, a calculation unit 20, and a compensation unit 30, wherein: Signal receiving channel 10 is configured to receive multiple calibration signals, which are non-orthogonal mismatched signals; to demodulate the received calibration signals using an orthogonal mixer (not shown in the figure) to obtain a baseband calibration signal; and to receive a signal to be processed, to demodulate the received signal to be processed to obtain a baseband signal to be processed. The computing unit 20 is configured to acquire the baseband calibration signal in Amplitude and phase information at frequency location, where, The baseband signal frequency corresponding to the calibration signal; based on the baseband calibration signal in Amplitude and phase information at frequency location, calculated Compensation coefficient for frequency position; The compensation unit 30 is configured to compensate the baseband signal to be processed using the calculated compensation coefficients.

[0082] In some exemplary embodiments, the compensation unit 30 includes a conjugate module 301, a multiplier 302, and an adder 303, wherein: The conjugate module 301 is configured to calculate the conjugate signal of the baseband signal to be processed; Multiplier 302 is configured to use the calculated The compensation coefficient at the frequency position is multiplied by the corresponding frequency component in the conjugate signal to obtain the compensated conjugate signal; and Adder 303 is configured to add the compensated conjugate signal to the baseband signal to be processed to obtain the compensated signal.

[0083] In some exemplary embodiments, the compensation unit 30 includes a delay module 304, wherein: The delay module 304 is configured to delay the baseband signal to be processed so that the compensated conjugate signal is aligned with the time domain of the delayed baseband signal to be processed.

[0084] In some exemplary embodiments, the compensation unit 30 includes a conjugate module 301, a compensation filter, and an adder 303, wherein: The calculation unit 20 is also configured to fit the compensation coefficients calculated for the multiple baseband single-tone signals received to obtain the coefficients of the compensation filter. The conjugate module 301 is configured to acquire the baseband signal to be processed obtained after the quadrature mixer demodulates the signal to be processed, and to calculate the conjugate signal of the baseband signal to be processed. The compensation filter is configured to filter the conjugate signal of the baseband signal to be processed, and obtain the filtered conjugate signal. Adder 303 is configured to add the filtered conjugate signal to the baseband signal to be processed to obtain the compensated signal.

[0085] In some exemplary embodiments, the compensation unit 30 includes a delay module 304, wherein: The delay module 304 is configured to delay the baseband signal to be processed so that the time domain of the filtered conjugate signal is aligned with the time domain of the delayed baseband signal to be processed.

[0086] In this embodiment of the disclosure, how specifically does the computing unit 20 perform calculations based on the baseband calibration signal? Amplitude and phase information at frequency location, calculated The compensation coefficient for frequency position can be referred to in the previous text, and will not be repeated here.

[0087] like Figure 9 As shown, this disclosure also provides a signal transceiver system, including a signal transmission link and a signal receiving link as described in any embodiment of this disclosure; The signal transmission link includes a baseband processor 501 and a real number mixer 502. The baseband processor 501 is configured to output a baseband single-tone signal, and the real number mixer 502 is configured to perform upmixing processing on the baseband single-tone signal. The signal transceiver system also includes a local oscillator signal generator 101 and a switch 102. The local oscillator signal generator 101 is used to output two orthogonal local oscillator signals to the quadrature mixer 103. The switch 102 is used to select one of the two orthogonal local oscillator signals according to the control signal of the control terminal of the switch 102 and output it to the real number mixer 502.

[0088] In this embodiment of the disclosure, the baseband processor 501 is a transmitting baseband processor, while the compensation unit 30 and the calculation unit 20 in the signal receiving link can be located within the receiving baseband processor.

[0089] In some exemplary embodiments, the signal transmission link further includes a real-to-analog converter (DAC) 503, configured to perform digital-to-analog conversion on the output baseband monotone signal to obtain an analog baseband monotone signal.

[0090] In some exemplary embodiments, the signal transmission link further includes a low-pass filter (LPF) 504 disposed between the real-to-digital converter 503 and the real-to-digital mixer 502, configured to perform low-pass filtering on the analog baseband single-tone signal output by the real-to-digital converter 503 and output it to the real-to-digital mixer 502.

[0091] In some exemplary embodiments, the signal transmission link further includes a power amplifier (PA) 505, wherein the power amplifier 505 is configured to amplify the upmixed baseband single-tone signal and output the amplified signal to the transmitting antenna.

[0092] In some exemplary embodiments, the signal transmission link further includes a transmitting antenna 506, wherein the transmitting antenna 506 is configured to radiate the amplified signal toward a preset spatial region.

[0093] In this embodiment, the signal amplified by the power amplifier 505 can be radiated into a predetermined spatial region through an integrated or external transmitting antenna 506. That is, the baseband processor 501, real-to-digital converter 503, low-pass filter 504, real-to-digital mixer 502, power amplifier 505, and transmitting antenna 506 can be integrated into a single device or are discrete components. For example, the baseband processor 501, real-to-digital converter 503, low-pass filter 504, real-to-digital mixer 502, and power amplifier 505 can be integrated into a package to form a SoC chip, while the transmitting antenna 506 can be connected to the chip's peripheral ports and formed on a carrier such as a PCB board. In some optional embodiments, the transmitting antenna 506 can also be integrated into the chip package to form an AiP or AoP, resulting in a chip structure with an encapsulated antenna.

[0094] In this embodiment, the baseband processor 501 generates a digital baseband single-tone signal and sends it to a real-to-digital converter 503 to obtain an analog baseband single-tone signal (without changing the phase information). This analog baseband single-tone signal is then input to a low-pass filter 504 to filter out out-of-band noise, and modulated by a real-to-digital mixer 502 to obtain a modulated radio frequency signal. Finally, the modulated radio frequency signal is radiated through a power amplifier 505 and a transmitting antenna 506.

[0095] In some exemplary embodiments, the signal transmission link may also include a Direct Digital Frequency Synthesizer (DDFS). Figure 9 (Not shown in the image), located between the baseband processor 501 and the real-to-digital converter 503, the direct digital frequency synthesizer can be configured to realize at least one of the following signal waveforms and transmission methods based on the received source signal: CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), and Digital IF (Digital Intermediate Frequency), so as to achieve flexible configuration of signal transmission form and transmission waveform.

[0096] In this embodiment of the disclosure, the local oscillator signal generator 101 can be an architecture including a phase-locked loop (PLL) that can provide electromagnetic wave (e.g., laser, microwave) signals.

[0097] In some optional embodiments, the signal transmission link may also include an error correction module for the DAC (TXDAC Board Error Correction) and an AWGN (additive white gaussian noise) module, etc., which are not shown in the figure. They can be added or removed according to actual needs.

[0098] In some optional embodiments, the signal receiving link may include an IQ mixer 103 and an IQ analog-to-digital converter (ADC) 105; wherein the IQ mixer 103 is configured to perform a mixing operation on the received calibration signal or signal to be processed based on the received local oscillator signal to obtain an analog baseband calibration signal or an analog baseband signal to be processed; and the IQ ADC 105 is configured to convert the analog baseband calibration signal or the analog baseband signal to be processed into a digital baseband calibration signal or a digital baseband signal to be processed.

[0099] In some exemplary embodiments, the signal receiving link may further include a receiving antenna 104, wherein the receiving antenna 104 is configured to receive a calibration signal or a signal to be processed, the calibration signal and / or the signal to be processed being an echo signal formed by the calibration signal and / or the signal to be processed transmitted by the signal transmitting link being reflected and / or scattered by a target object.

[0100] In some alternative embodiments, the receiving antenna 104 can be connected via the chip's peripheral port and formed on a carrier such as a PCB board. Meanwhile, in other alternative embodiments, the receiving antenna 104 can also be integrated into the chip package to form an AiP or AoP, i.e., a chip structure with an encapsulated antenna.

[0101] In some exemplary embodiments, the signal receiving link may also include a low noise amplifier (LNA) 107, which is disposed between the receiving antenna 104 and the IQ mixer 103 to amplify the signal received by the receiving antenna 104 with low noise before sending it to the IQ mixer 103.

[0102] In some exemplary embodiments, the signal receiving link may further include a low-pass filter (LPF) 108 and a high-pass filter (HPF) 109 connected in series, disposed between the IQ mixer 103 and the IQ analog-to-digital converter 105. The low-pass filter 307 and the high-pass filter 308 form a band-pass filter for filtering out out-of-band noise.

[0103] In some exemplary implementations, such as Figure 10 As shown, the signal transmission link also includes a TX HD3 compensation unit 507, which is used to solve the TX HD3 problem.

[0104] In some alternative embodiments, the transmit link can be calibrated by adding a receive link. Alternatively, the transmit link can be calibrated by multiplexing the receive link actually used for signal transmission and reception, with compensation performed based on the calibration data by the TX HD3 compensation unit in the transmit and / or receive links.

[0105] In some optional embodiments, the transmit link of the digital phase shifter architecture can be calibrated by reusing the receive link in the transmit-receive link; wherein, in other embodiments, the calibration operation involving the transmit link using the receive link and the calibration operation involving the receive link using the transmit link can be achieved by reusing the corresponding receive link or transmit link in the actual link for signal transmission and reception, or by adding a corresponding calibration receive link or calibration transmit link to achieve the calibration operation of the corresponding transmit link or receive link in the actual link for signal transmission and reception.

[0106] In some optional embodiments, a calibration module may be provided between the transmit link and the receive link. This calibration module can be configured to perform calibration operations on the transmit link by multiplexing the receive link. Simultaneously, the compensation unit can perform compensation operations on the transmitted signal at the transmit link end based on the parameters or coefficients obtained from the calibration operation of the calibration module. In other embodiments, a corresponding receive compensation unit may be provided simultaneously or separately in the receive link. In this case, the receive compensation unit can perform compensation on the echo signal at the receive link end based on the parameters or coefficients obtained from the aforementioned calibration operation.

[0107] In some alternative embodiments, such as Figure 11 As shown, the calibration module may include an internal self-test (BIST) unit. The BIST unit can be set between the signal output port of the transmit link and the signal input port of the receive link. That is, the transmit link directly sends the transmit signal to the receive link through the BIST unit, so as to realize the self-calibration operation of the receive link and / or transmit link without passing through the transmit antenna and the receive antenna.

[0108] In some alternative embodiments, such as Figure 12As shown, the calibration module described above may also include auxiliary circuit units, i.e., the signal output port of the transmit link is connected to any node between the IQ Mixer and the IQ ADC in the receive link through the BIST unit and the auxiliary circuit unit. For example, the Real mixer in the transmit link generates an RF signal of (z GHz ± x MHz) based on the x MHz digital phase-shifted baseband signal and the z GHz LO signal, and outputs it to the BIST unit through the output port. The BIST unit performs a y MHz frequency offset operation on the received RF signal to obtain an analog echo signal of (z GHz ± x MHz ± y MHz). After frequency reduction using the De-Modulator in the auxiliary circuit unit, a preset intermediate frequency signal (z GHz ± x MHz ± y MHz - z GHz = ± x MHz ± y MHz) is obtained. This intermediate frequency signal is then input to a preset node in the receive link to realize the calibration operation in the transmit link.

[0109] Optionally, the auxiliary circuit unit can be a demodulator circuit, the output of which can be connected to the node between the LPF and HPF in the receiving link, or between the HPF and VGA. Figure 12 (Not shown in the diagram, located between the HPF and IQ ADC) or any of the nodes between the VGA and IQ ADC. Additionally, to maximize the reuse of the transmit link structure, the output port of one transmit link, after passing through the BIST unit and auxiliary circuit unit, can be connected to the IQ branch of the receive link, as shown below. Figure 12 As shown. After completing the calibration of the transmission link, the HD3 compensation unit (TX HD3 compensation) and other compensation units in the aforementioned compensation unit (TXcompensation) can be used to perform compensation operations for HD3 and other problems in the transmission link based on the parameters obtained from the calibration.

[0110] In some optional embodiments, the BIST unit described above may include a phase-angle converter and a Real modulator connected in sequence. The auxiliary circuit unit may include an LNA, a Real De-Modulator, and a TIA (not shown in the figure) connected in sequence. Specifically, the phase-angle converter receives the RF signal output from the transmit link, while one input of the Real Modulator is connected to the output of the phase-angle converter, and the other input receives a y MHz BIST-LO signal to generate a preset echo signal. The LNA amplifies the received echo signal and sends it to one input of the Real De-Modulator. The other input of the Real De-Modulator receives a z GHz RX-LO signal. The output of the Real De-Modulator, after passing through the TIA, is connected to the corresponding node in its respective receive link to output the generated preset intermediate frequency signal to the two receive branches, thus achieving the calibration operation.

[0111] It should be noted that, for the calibration operation in this embodiment, if the transmit link transmits a swept frequency signal, in the actual calibration operation, the TX LO signal can be used as a single-tone signal for point-by-point calibration; at the same time, the TX LO signal can also be used as a swept frequency signal for large-bandwidth calibration operation, or even the swept frequency bandwidth calibration can be used to achieve the calibration operation of the swept frequency signal for the entire frequency band in one operation.

[0112] based on Figure 12 The structure shown can be further suppressed to a predetermined degree in the transmission link, such as HD3, by cascading at least two BIST units. By using two BIST units in series, the noise caused by the above-mentioned defects can be suppressed to -50dB, thereby effectively reducing the difficulty of developing and designing related link analog devices.

[0113] In some optional embodiments, in order to further improve the accuracy of the IQ Imbalance compensation coefficient, the ideal compensation coefficient can be approximated by iterative calibration and compensation, or the ideal compensation coefficient can be obtained by multi-observation calibration and compensation.

[0114] For example, regarding iterative calibration and compensation, the decision to stop the iteration operation can be based on the magnitude relationship between the compensation coefficients of two consecutive calibrations, or whether the difference between the compensation coefficients of two calibrations meets a preset iteration condition. The compensation coefficient obtained when the iteration stops is then used as the final compensation coefficient for the current scenario in subsequent operations. For multi-observation calibration and compensation, after multiple (e.g., three) calibration and compensation operations, the measurement data obtained from each operation can be subjected to FFT (Fast Fourier Transform) to obtain the corresponding amplitude and phase information. The measurements can then be subtracted and normalized to obtain the relevant data, and an observation matrix can be constructed. Subsequently, the corresponding compensation coefficients can be solved in reverse based on the data obtained by inverting this observation matrix.

[0115] In some alternative embodiments, based on the same idea as the above-described method for obtaining the IQ Imbalance compensation coefficient, methods such as iterative calibration and compensation, or multiple observation calibration and compensation, can also be used to obtain the HD3 compensation coefficient.

[0116] The examples of transmitters with compensation units mentioned above discussed the problem of solving harmonic distortion in transmitters. Research has found that some harmonic distortion may originate from components in the transmitter, such as mixers, that contain nonlinear characteristics.

[0117] by Figure 13 For example, Figure 13 This is a schematic diagram of the mixer structure in an embodiment of this disclosure. Figure 13 As shown, the mixer includes a voltage-to-current converter (V / I Converter), a current switch, and a current-to-voltage converter (I / V Converter). The voltage-to-current converter converts a received voltage signal into a current signal. The current switch is connected to the voltage-to-current converter and the second signal generator, and is used to process the current signal output by the voltage-to-current converter using a local oscillator signal. The current-to-voltage converter is connected to the current switch and is used to convert the current signal output by the current switch into a voltage signal.

[0118] In the above structure, because the voltage-to-current converter incorporates a transistor amplifier, the nonlinear characteristics of the transistor amplifier and the low frequency of the baseband signal result in harmonic signals in the output current signal of the voltage-to-current converter, corresponding to the baseband signal. For example, the harmonics (HD) caused by the third-order nonlinearity of the baseband can be simply referred to as HD3. Similarly, the harmonics caused by the fifth-order nonlinearity are called HD5. When the current switch processes the current signal output from the voltage-to-current converter, the harmonic frequencies are up-converted to the radio frequency (RF) band. Suppressing RF harmonic signals is complex and costly. If RF harmonic signals are not removed, it will affect the signal quality of transmission and reception, thus impacting measurement accuracy.

[0119] The compensation unit is used to input the generated cancellation signal to the signal transmission link to cancel the harmonic signals in the radio frequency signal. The compensation unit is independent of the first signal generator.

[0120] Therefore, the compensation unit may include a cancellation signal generator. The cancellation signal output by the compensation unit can suppress harmonic signals in the radio frequency signal, reduce harmonic components in the radio frequency signal, and thus improve the signal quality of the radio frequency signal output by the transmitter.

[0121] In this embodiment, for harmonic signals in the signal transmission link, the compensation unit uses feedback or, based on the characteristics of the transmitted wave, inputs a generated cancellation signal to the signal transmission link to cancel the harmonic signals in the radio frequency signal output by the signal transmission link. This cancellation signal has characteristics such as opposite phase and similar amplitude to the harmonic signals transmitted in the radio frequency transmission circuit, thereby achieving the purpose of suppressing harmonic signals.

[0122] In some examples, the compensation unit generates a compensation signal that includes cancellation based on parameters such as the phase, frequency, or amplitude of the baseband signal generated by the first signal generator, or even the path length of the LO signal.

[0123] For example, Figure 11 An example of a transmitter in which a compensation unit is connected to a signal transmission link is shown. Figure 11 In the structure shown, the compensation unit is a TX compensation unit. The TX compensation unit contains a cancellation signal generator (not shown) that can generate a cancellation signal based on the characteristics of the transmitted wave. For example, the cancellation signal generator can be a TX HD3 compensation unit.

[0124] The baseband processor controls the quadrature digital baseband signal generated by the TX DDFS. The TX compensation unit generates a quadrature compensation signal based on the parameters of the quadrature digital signal. The quadrature compensation signal and the quadrature digital signal are combined and sent to the RealDAC to be converted into an analog baseband signal. After LPF filtering, the signal is mixed by a mixer to obtain an RF signal based on the TX LO signal and the analog baseband signal. The PA amplifies the mixed signal and outputs it through the transmitting antenna. This compensation signal cancels at least part of the harmonic signals in the RF transmitting circuit, such as the HD3 harmonic signal. Therefore, the noise in the transmitted RF signal is greatly reduced. The RF signal can be an FMCW signal.

[0125] In other examples, the compensation unit generates a compensation signal based on harmonic information received from feedback via the radio frequency transmitting circuit. See also Figure 14 , Figure 14 for Figure 11 The diagram shows the structure of the compensation unit in the transmitter. Figure 14 As shown, the compensation unit includes a data acquisition circuit and a cancellation signal generator.

[0126] The acquisition circuit is coupled to the radio frequency transmitting circuit and is used to acquire signals from the radio frequency transmitting circuit to obtain an acquired signal. The acquired signal (or sampling signal) can reflect the waveform information (also known as harmonic parameters) in the harmonic signal, such as the phase of the dominant frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the dominant frequency signal, the power of the harmonic signal, and the power of the dominant frequency signal.

[0127] It should be noted that the harmonic parameters reflected in the acquired signal are related to the information carried by the signal that the acquisition circuit can acquire. For example, if the acquisition circuit is a power acquisition circuit, then the corresponding acquired signal includes the power of the main frequency. Alternatively, if the acquisition circuit utilizes at least part of the receiver's circuitry, then the acquired signal reflects the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, and the power of the main frequency signal, etc.

[0128] At least one of the aforementioned harmonic parameters can be extracted using analog circuitry. For example, the power of the output main frequency signal can be obtained through a coupler and a power detector. Alternatively, the advantages of frequency domain calculations in digital circuitry within a chip can be utilized to extract harmonic parameters. For instance, by coupling an RF transmitting circuit, a signal identical to the signal transmitted at the coupling point can be acquired as a sampling signal. This sampling signal carries the main frequency signal and harmonic signals. The sampling signal is converted into a digital signal by an ADC and then processed by digital circuitry in the frequency domain to obtain more harmonic parameters.

[0129] In one implementation, the input terminal of the acquisition circuit is connected to the output terminal or signal detection terminal of the mixer. This method can detect harmonic signals generated by the voltage-to-current converter and has a simplified acquisition circuit.

[0130] In another embodiment, the input terminal of the acquisition circuit is connected to the RF output terminal or the RF detection terminal of the RF transmitting circuit. The RF output terminal is, for example, the output terminal of the RF transmitting circuit. The RF detection terminal is, for example, the input or output terminal of at least one power amplifier (PA) stage in the RF transmitting circuit. This method can acquire more accurate harmonic parameters from the RF transmitting circuit, but it has a more complex circuit structure.

[0131] In some chips that include a BIST module, the acquisition circuit can obtain the acquired signal using some or all of the circuitry within the BIST module. For example, the input of the acquisition circuit is coupled to the RF output, and it includes a downconverter, a filter, etc., connected to a Real ADC to output a digital acquisition signal. The downconverter, filter, etc., can be multiplexed from the BIST module or a receiver.

[0132] The acquired signal is input to a cancellation signal generator. The cancellation signal generator is at least one circuit within the compensation unit. This cancellation signal generator is connected to the first signal generator, ensuring that the signal received by the radio frequency transmitting circuit simultaneously includes both the baseband signal and the cancellation signal.

[0133] For example, the cancellation signal generator includes the aforementioned cancellation signal generator and digital circuitry for extracting harmonic information. The digital circuitry for extracting harmonic information can be configured independently or at least partially shared with the digital circuitry within the chip.

[0134] One example of a digital circuit for extracting harmonic information is the use of a chip-based digital circuit for processing the baseband signal to extract harmonic frequency, main frequency, and main frequency power, which is then provided to a cancellation signal generator. The cancellation signal generator generates a cancellation signal based on the received parameters.

[0135] For example, a digital circuit for extracting harmonic information extracts the main frequency amplitude from the acquired signal and calculates the harmonic amplitude based on the difference between the preset main frequency amplitude and the harmonic amplitude. A cancellation signal generator then generates a harmonic compensation signal based on the calculated harmonic amplitude and other pre-configured harmonic parameters. These pre-configured harmonic parameters can be calculated based on the sweep range and phase of the main frequency signal to be transmitted by the chip.

[0136] The cancellation signal generator can be configured independently of the first signal generator, or at least partially shared. For example, the cancellation signal generated by the cancellation signal generator is input to the first signal generator, such that the baseband signal output by the first signal generator includes the cancellation signal. The cancellation signal generator may include a third harmonic generator and a fifth harmonic generator.

[0137] For example, the compensation unit further includes an adder coupled to the cancellation signal generator and the first signal generator to combine the baseband signal generated by the first signal generator and the cancellation signal generated by the cancellation signal generator. In this embodiment, the cancellation signal includes a cancellation signal Signal_HD3 generated by the third harmonic generator to cancel the third harmonic, and a cancellation signal Signal_HD5 generated by the fifth harmonic generator to cancel the fifth harmonic. The cancellation signals Signal_HD3 and Signal_HD5, along with the baseband signal generated by the first signal generator, are combined by the adder and output to the radio frequency transmission circuit.

[0138] In summary, the transmitter circuit examples provided in this disclosure, which employ a feedback method to pre-input cancellation signals into the RF transmitting circuit, can ensure that the RF signals transmitted by the chip contain sufficiently low harmonic signals under different environments.

[0139] This disclosure also provides a signal transceiver system, including a signal transmission link and a signal reception link, wherein: The signal transmission link includes a transmitter baseband processing module and a real number mixer. The transmitter baseband processing module is configured to output a calibration signal or a signal to be processed. The real number mixer is configured to perform up-mixing processing on the calibration signal or the signal to be processed. The signal receiving link includes a local oscillator signal generator, a switch, a quadrature mixer, and a receiving-end baseband processing module. The local oscillator signal generator is configured to output two quadrature local oscillator signals to the quadrature mixer. The switch is configured to select one of the two quadrature local oscillator signals according to the control signal from the control terminal and output it to the real number mixer. The quadrature mixer is configured to demodulate the received calibration signal or the signal to be processed to obtain a baseband calibration signal or a baseband signal to be processed. The receiving-end baseband processing module is configured to calculate the compensation coefficient based on the amplitude and phase information of the baseband calibration signal and use the compensation coefficient to compensate the baseband signal to be processed.

[0140] In some exemplary embodiments, the transmitting baseband processing module includes a transmitting baseband processor and a digital-to-analog converter, wherein the transmitting baseband processor is configured to output a digital calibration signal or a digital signal to be processed; and the digital-to-analog converter is configured to perform digital-to-analog conversion on the digital calibration signal or the digital signal to be processed.

[0141] In some exemplary embodiments, the receiving-end baseband processing module includes an analog-to-digital converter and a receiving-end baseband processor, wherein the analog-to-digital converter is configured to perform analog-to-digital conversion on the baseband calibration signal or the baseband signal to be processed; and the transmitting-end baseband processor is configured to calculate compensation coefficients based on the amplitude and phase information of the baseband calibration signal, and use the compensation coefficients to compensate the baseband signal to be processed.

[0142] In some exemplary embodiments, the signal transmission link further includes an amplifier configured to amplify the upmixed calibration signal or the signal to be processed.

[0143] In some exemplary embodiments, the signal receiving link further includes a low-noise amplifier, wherein the low-noise amplifier is configured to amplify the received calibration signal or signal to be processed in a low-noise manner.

[0144] This disclosure also provides an integrated circuit that may include any of the signal transceiver systems described above. Optionally, the integrated circuit may be an ultra-wideband chip (or die).

[0145] In some optional embodiments, the integrated circuit may be an AiP (Antenna-In-Package) chip structure, an AoP (Antenna-On-Package) chip structure, an AoC (Antenna-On-Chip) chip structure, or a RoP (Radiator-On-Package) chip structure, etc.

[0146] According to other embodiments of this disclosure, an electromagnetic wave device is also proposed. This electromagnetic wave device may include an antenna and an integrated circuit as described above. The integrated circuit is electrically connected to the antenna and is used to transmit and receive electromagnetic wave signals. For example, the electromagnetic wave device may include: a carrier, an integrated circuit as described in any of the above embodiments, and an antenna, etc. The integrated circuit may be disposed on the carrier; the antenna may be disposed on the carrier, or integrated with the integrated circuit as a single device disposed on the carrier (i.e., the antenna may be an antenna disposed in an AiP, AoP, AoC, or RoP structure); wherein the integrated circuit is connected to the antenna (i.e., the sensing chip or integrated circuit does not integrate an antenna, such as a conventional SoC), and is used to transmit and receive electromagnetic wave signals. The carrier may be a printed circuit board (PCB).

[0147] This disclosure provides an apparatus that may include: an apparatus body; and an electromagnetic wave device as described above disposed on the apparatus body; wherein the electromagnetic wave device is used for target detection and / or communication to provide reference information to the operation of the apparatus body.

[0148] This disclosure also provides a user terminal device, which can be manifested in the form of a general computing device. The components of the user terminal device may include, but are not limited to: at least one processing unit, at least one storage unit, a bus connecting different system components (including the storage unit and the processing unit), a display unit, etc. The storage unit stores program code, which can be executed by the processing unit to cause the processing unit to perform the methods described in this specification according to the various exemplary embodiments of this disclosure. The storage unit may include a readable medium in the form of volatile storage units, such as random access memory (RAM) and / or cache memory units, and may further include read-only memory units (ROM).

[0149] The storage unit may also include a program / utility having a set (at least one) of program modules, including but not limited to: an operating system, one or more applications, other program modules, and program data, each or some combination of these examples may include an implementation of a network environment.

[0150] A bus can represent one or more of several types of bus structures, including a memory cell bus or memory cell controller, a peripheral bus, a graphics acceleration port, a processing unit, or a local bus that uses any of the various bus structures.

[0151] The user terminal device can also communicate with one or more external devices (e.g., keyboards, pointing devices, Bluetooth devices, etc.), one or more devices that enable the user to interact with the user terminal device, and / or any device that enables the user terminal device to communicate with one or more other computing devices (e.g., routers, modems, etc.). This communication can be performed via input / output (I / O) interfaces. Furthermore, the user terminal device can also communicate with one or more networks (e.g., local area networks (LANs), wide area networks (WANs), and / or public networks, such as the Internet) via a network adapter. The network adapter can communicate with other modules of the user terminal device via a bus. It should be understood that, although not shown in the figures, other hardware and / or software modules can be used in conjunction with the user terminal device, including but not limited to: microcode, device drivers, redundant processing units, external disk drive arrays, RAID systems, tape drives, and data backup storage systems.

[0152] For example, the user terminal device in this embodiment may further include: a device body; and an electromagnetic wave device disposed on the device body as described in any of the above embodiments; wherein the electromagnetic wave device can be used to realize functions such as target detection and / or wireless communication.

[0153] According to some other embodiments of this application, an electromagnetic wave device is also proposed. This electromagnetic wave device may include an antenna and an integrated circuit as described above. The integrated circuit is connected to the antenna and is used to transmit and receive electromagnetic wave signals. For example, the electromagnetic wave sensor may include: a carrier, an integrated circuit as described in any of the above embodiments, and an antenna, etc. The integrated circuit may be disposed on the carrier; the antenna may be disposed on the carrier (i.e., the antenna may be an antenna disposed on a PCB board in a structure such as RoP), or integrated with the integrated circuit as a single device disposed on the carrier (i.e., the antenna may be an antenna disposed in a structure such as AiP, AoP, or AoC); wherein the integrated circuit is connected to the antenna (i.e., the sensing chip or integrated circuit does not integrate an antenna, such as a conventional SoC), and is used to transmit and receive electromagnetic wave signals. The carrier may be a printed circuit board (PCB).

[0154] In some optional embodiments, the aforementioned electromagnetic wave signal can be a centimeter wave band or a millimeter wave band (such as 3.1GHz, 24GHz, 60GHz, 77GHz, 94GHz, 120GHz, 140GHz, 220GHz, 250GHz, etc.). Specifically, the 3.1GHz centimeter wave signal can include 3.1GHz~10.6GHz, such as 3.1GHz, 5GHz, 5GHz, 6GHz, 8GHz, 10.6GHz, etc., or it can be such as 7.163-8.812GHz, etc.; the 77GHz millimeter wave signal can include signals from 76GHz to 81GHz, such as the frequency ranges of 76GHz~77GHz, 77GHz~79GHz, 79GHz~81GHz, etc., or fixed frequency points such as 76GHz, 77GHz, 78GHz, 79GHz, 80GHz, 81GHz, etc.

[0155] Specifically, based on the above embodiments, in one optional embodiment of this disclosure, the electromagnetic wave device can be disposed outside the device body or inside the device body. In other optional embodiments of this disclosure, the electromagnetic wave device can be partially disposed inside the device body and partially disposed outside the device body. This disclosure does not limit the specific implementation; it can be determined according to the circumstances.

[0156] In an optional embodiment, the aforementioned device body can be a component or product applied in fields such as smart cities, smart homes, transportation, smart homes, consumer electronics, security monitoring, industrial automation, digital keys, in-cabin detection (such as smart cockpits), indoor positioning, medical devices, and healthcare. For example, the device body can be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), liquid level / flow rate detection equipment, smart wearable devices (such as wristbands, glasses, etc.), smart home equipment (such as robot vacuum cleaners, door locks, televisions, air conditioners, smart lights, etc.), various communication devices (such as mobile phones, tablets, etc.), as well as devices such as barriers, intelligent traffic lights, intelligent signs, traffic cameras, and various industrial robotic arms (or robots). It can also be various instruments for detecting vital signs parameters and various devices equipped with such instruments, such as in-cabin vital sign detection in automobiles, indoor personnel monitoring, smart medical devices, and consumer user terminal devices.

[0157] From the above description of the embodiments, those skilled in the art will readily understand that the exemplary embodiments described herein can be implemented by software or by combining software with necessary hardware. The technical solutions according to the embodiments of this disclosure can be embodied in the form of a software product, which can be stored in a non-volatile storage medium (such as a CD-ROM, USB flash drive, external hard drive, etc.) or on a network, including several instructions to cause a computing device (such as a personal computer, server, or network device, etc.) to execute the methods described above according to the embodiments of this disclosure.

[0158] Software products may employ any combination of one or more readable media. A readable medium may be a readable signal medium or a readable storage medium. A readable storage medium may be, for example, but not limited to, an electrical, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, or device, or any combination thereof. More specific examples of readable storage media (a non-exhaustive list) include: electrical connections with one or more wires, portable disks, hard disks, random access memory (RAM), read-only memory (ROM), erasable programmable read-only memory (EPROM or flash memory), optical fiber, portable compact disk read-only memory (CD-ROM), optical storage devices, magnetic storage devices, or any suitable combination thereof.

[0159] Computer-readable storage media may include data signals propagated in baseband or as part of a carrier wave, carrying readable program code. Such propagated data signals may take various forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination thereof. A readable storage medium may also be any readable medium other than a readable storage medium that can transmit, propagate, or transfer a program for use by or in connection with an instruction execution system, apparatus, or device. The program code contained on the readable storage medium may be transmitted using any suitable medium, including but not limited to wireless, wired, optical fiber, RF, etc., or any suitable combination thereof.

[0160] Program code for performing the operations of this disclosure can be written in any combination of one or more programming languages, including object-oriented programming languages ​​such as Java and C++, and conventional procedural programming languages ​​such as C or similar languages. The program code can execute entirely on the user's computing device, partially on the user's computing device, as a standalone software package, partially on the user's computing device and partially on a remote computing device, or entirely on a remote computing device or server. In cases involving remote computing devices, the remote computing devices can be connected to the user's computing device via any type of network, including a local area network (LAN) or a wide area network (WAN), or can be connected to an external computing device (e.g., via the Internet using an Internet service provider).

[0161] Those skilled in the art will understand that the above modules can be distributed in the device as described in the embodiments, or they can be modified accordingly and placed in one or more devices that are unique to this embodiment. The modules in the above embodiments can be combined into one module, or they can be further divided into multiple sub-modules.

[0162] According to embodiments of this disclosure, a computer program is proposed, including computer programs or instructions, which, when executed by a processor, can perform the methods described above. In an optional embodiment, the integrated circuit described above can be an ultra-wideband chip. The types of digital functional modules in the integrated circuit can be determined according to actual needs.

[0163] It should be noted that wireless devices can transmit and receive ultra-wideband signals (such as ultra-wideband signals with a bandwidth greater than 500MHz) to achieve functions such as ultra-wideband communication, so as to provide detection and communication information to the device body, thereby assisting or even controlling the operation of the device body.

[0164] For example, when the aforementioned device is applied to an advanced driver assistance system (ADAS), the wireless device, as an on-board sensor, can assist the ADAS system in realizing application scenarios such as adaptive cruise control, automatic brake assist (AEB), blind spot detection warning (BSD), lane change assist warning (LCA), rear cross traffic alert (RCTA), parking assist, rear vehicle warning, collision avoidance, and pedestrian detection.

[0165] This disclosure also provides a user terminal device, including: a device body; and an electromagnetic wave device disposed on the device body as described in any embodiment of this disclosure; wherein the electromagnetic wave device is used for target detection and / or wireless communication to provide reference information to the operation of the device body, thereby assisting or even controlling the operation of the device body and / or other electronic devices disposed in the device body.

[0166] For example, when the aforementioned device is a vehicle, the electromagnetic wave device used as an in-vehicle sensor can be used to assist ADAS systems in realizing in-vehicle applications such as adaptive cruise control, automatic braking assist (AEB), blind spot detection warning (BSD), lane change assist warning (LCA), rear cross traffic alert (RCTA), assisted / automatic parking assist, rear vehicle warning, collision avoidance, pedestrian detection, as well as door collision avoidance, automatic opening and closing of the trunk door with foot kick, etc. It can also be used as a digital key for the vehicle.

[0167] The quadrature imbalance compensation method, signal receiving link, and signal transceiver system disclosed in this embodiment do not use orthogonal upmixers at the transmitting end, but instead use a superheterodyne structure (real number mixer), thus eliminating orthogonal imbalance issues in the received calibration signal. At the receiving end, an orthogonal mixer is used to demodulate the received calibration signal, and the signal is then processed according to the baseband calibration signal. Amplitude and phase information at frequency location, calculated The compensation coefficient for frequency position is simple and easy to implement, enabling real-time (on-the-fly) quadrature imbalance compensation. Furthermore, by making the local oscillator signal at the transmitting end one of the local oscillator signals input to the IQ mixer at the receiving end, image suppression can be achieved without the need for an additional local oscillator signal for calibration, and without the need for frequency misalignment between the local oscillator signals at the transmitting and receiving ends during calibration, thus reducing resource overhead.

[0168] The technical features of the above embodiments can be combined in any way. For the sake of brevity, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0169] The embodiments described above merely illustrate preferred embodiments of this disclosure and the technical principles employed. While the descriptions are specific and detailed, they should not be construed as limiting the scope of the invention patent. Various obvious changes, readjustments, and substitutions can be made by those skilled in the art without departing from the protection scope of this disclosure. Therefore, although this disclosure has been described in detail through the above embodiments, it is not limited to these embodiments. Many other equivalent embodiments may be included without departing from the concept of this disclosure, and the protection scope of this patent is determined by the appended claims.

Claims

1. An orthogonal unbalance compensation method, characterized in that, include: Receive multiple calibration signals, wherein the calibration signals are non-orthogonal mismatched signals; The received calibration signal is demodulated using an orthogonal mixer to obtain the baseband calibration signal; Obtain the baseband calibration signal in Amplitude and phase information at frequency location, where, The baseband signal frequency corresponding to the calibration signal; based on the baseband calibration signal in Amplitude and phase information at frequency location, calculated Compensation coefficient for frequency position; Receive the signal to be processed and compensate the signal using the calculated compensation coefficient.

2. The orthogonal unbalance compensation method according to claim 1, characterized in that, The receiving of multiple calibration signals specifically includes: Obtain one or more baseband signal frequencies ; For each baseband signal frequency The baseband single-tone signal is received at least twice as the calibration signal.

3. The orthogonal unbalance compensation method according to claim 2, characterized in that, The received at least two calibration signals include a first calibration signal and a second calibration signal, wherein: There is a preset phase difference between the local oscillator signal corresponding to the first calibration signal and the local oscillator signal corresponding to the second calibration signal.

4. The orthogonal unbalance compensation method according to claim 3, characterized in that, Calculate the frequency using the following formula. Position compensation coefficient: ； ； in, This indicates that the baseband calibration signal corresponding to the first calibration signal is in Amplitude and phase information obtained from frequency location. This indicates that the baseband calibration signal corresponding to the first calibration signal is in Amplitude and phase information obtained from frequency location. This indicates that the baseband calibration signal corresponding to the second calibration signal is in Amplitude and phase information obtained from frequency location. This indicates that the baseband calibration signal corresponding to the second calibration signal is in Amplitude and phase information obtained from frequency location.

5. The orthogonal unbalance compensation method according to claim 2, characterized in that, For each of the baseband signal frequencies The baseband single-tone signal, receiving at least three calibration signals, is calculated using the least squares criterion. The compensation coefficient for frequency position.

6. The orthogonal unbalance compensation method according to claim 1, characterized in that, The step of compensating the signal to be processed using the calculated compensation coefficient includes: Obtain the baseband signal to be processed after the quadrature mixer demodulates the signal to be processed, and calculate the conjugate signal of the baseband signal to be processed; Using the calculated The compensation coefficient for the frequency position is multiplied by the corresponding frequency component in the conjugate signal to obtain the compensated conjugate signal; The compensated conjugate signal is added to the baseband signal to be processed to obtain the compensated signal.

7. The orthogonal unbalance compensation method according to claim 1, characterized in that, The step of compensating the signal to be processed using the calculated compensation coefficient includes: For multiple baseband single-tone signals whose signal frequencies are within a preset bandwidth range, the calculated compensation coefficients are obtained respectively; The obtained compensation coefficients are fitted to obtain the coefficients of the compensation filter; Obtain the baseband signal to be processed after the quadrature mixer demodulates the signal to be processed, and calculate the conjugate signal of the baseband signal to be processed; The compensation filter is used to filter the conjugate signal of the baseband signal to be processed to obtain the filtered conjugate signal. The filtered conjugate signal is added to the baseband signal to be processed to obtain the compensated signal.

8. The orthogonal unbalance compensation method according to claim 1, characterized in that, The baseband calibration signal is obtained by any of the following methods. Amplitude and phase information at frequency location: Perform a Fourier transform on the baseband calibration signal to obtain the amplitude and phase information at the corresponding frequency position; The corresponding frequencies of the two demodulated quadrature signals are shifted to zero frequency by a digital mixer, and the average value of the two quadrature signals at zero frequency is calculated to obtain the amplitude and phase information at the corresponding frequency position.

9. The orthogonal unbalance compensation method according to claim 1, characterized in that, The signal to be processed is an ultra-wideband signal.

10. A signal receiving link, characterized in that, It includes a signal receiving channel, a computing unit, and a compensation unit, wherein: The signal receiving channel is configured to receive multiple calibration signals, which are non-orthogonal mismatched signals; to demodulate the received calibration signals using an orthogonal mixer to obtain a baseband calibration signal; and is also configured to receive a signal to be processed. The computing unit is configured to acquire the baseband calibration signal in Amplitude and phase information at frequency location, where, The baseband signal frequency corresponding to the calibration signal; based on the baseband calibration signal in Amplitude and phase information at frequency location, calculated Compensation coefficient for frequency position; The compensation unit is configured to compensate the signal to be processed using a calculated compensation coefficient.

11. A signal transceiver system, characterized in that, This includes the signal transmission link and the signal reception link, wherein: The signal transmission link includes a transmitting baseband processing module and a real number mixer, wherein the transmitting baseband processing module is configured to output a calibration signal or a signal to be processed; and the real number mixer is configured to perform up-mixing processing on the calibration signal or the signal to be processed. The signal receiving link includes a local oscillator signal generator, a switch, a quadrature mixer, and a receiving-end baseband processing module. The local oscillator signal generator is configured to output two quadrature local oscillator signals to the quadrature mixer. The switch is configured to select one of the two quadrature local oscillator signals according to a control signal from a control terminal and output it to the real-number mixer. The quadrature mixer is configured to demodulate the received calibration signal or signal to be processed to obtain a baseband calibration signal or a baseband signal to be processed. The receiving-end baseband processing module is configured to calculate a compensation coefficient based on the amplitude and / or phase information of the baseband calibration signal, and use the compensation coefficient to compensate the baseband signal to be processed.

12. The signal transceiver system according to claim 11, characterized in that, The transmitting baseband processing module includes a transmitting baseband processor and a digital-to-analog converter, wherein the transmitting baseband processor is configured to output a digital calibration signal or a digital signal to be processed; and the digital-to-analog converter is configured to perform digital-to-analog conversion on the digital calibration signal or the digital signal to be processed. The receiving-end baseband processing module includes an analog-to-digital converter and a receiving-end baseband processor. The analog-to-digital converter is configured to perform analog-to-digital conversion on the baseband calibration signal or the baseband signal to be processed. The transmitting-end baseband processor is configured to calculate a compensation coefficient based on the amplitude information and / or phase information of the baseband calibration signal, and use the compensation coefficient to compensate the baseband signal to be processed.

13. The signal transceiver system according to claim 11, characterized in that, The signal transmission link further includes an amplifier, wherein the amplifier is configured to amplify the upmixed calibration signal or the signal to be processed. The signal receiving link further includes a low-noise amplifier, wherein the low-noise amplifier is configured to amplify the received calibration signal or signal to be processed in a low-noise manner.

14. An integrated circuit, characterized in that, Includes the signal receiving link as described in claim 10, or the signal transceiver system as described in any one of claims 11 to 13.

15. The integrated circuit according to claim 14, characterized in that, The integrated circuit is a UWB chip.

16. An electromagnetic wave device, characterized in that, include: Carrier; The integrated circuit as described in any one of claims 14 to 15 is disposed on the carrier. An antenna is disposed on the carrier, or the antenna and the integrated circuit are integrated into a single device disposed on the carrier; the antenna includes a transmitting antenna and a receiving antenna, or includes a transmit / receive multiplexed antenna; The integrated circuit is connected to the antenna and is used to transmit and / or receive electromagnetic wave signals.

17. A user terminal device, characterized in that, include: Equipment body; as well as The electromagnetic wave device as described in claim 16 is disposed on the device body; The electromagnetic wave device is used for target detection and / or wireless communication to provide reference information for the operation of the device body.

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