Method for testing anti-interference suppression of receiving module signal line

Abstract

This invention relates to the field of anti-interference suppression testing technology, specifically to an anti-interference suppression testing method for signal lines of a receiving module. The method includes the following steps: injecting a signal into the signal line connected to the limiting diode at the front end of the receiving module; acquiring and generating a two-dimensional bispectral feature matrix; extracting the peak amplitude at a specific intermodulation frequency coordinate within the two-dimensional bispectral feature matrix; and calculating the peak amplitude with a benchmark test value. This invention extracts timing distortion evaluation parameters through the difference deviation ratio and refines them into two independent parameters: edge smoothing degree and phase shift time. This allows for the separate characterization of the influence of parasitic capacitance on signal edge morphology and timing delay. Finally, by comparing each parameter with the allowable timing delay limit, the distortion suppression capability is determined. This enables the overall evaluation process to have a closed-loop capability from frequency domain nonlinearity identification to time domain dynamic response and then to structured judgment, thereby improving the reliability of anti-interference performance evaluation.

Description

Technical Field

[0001] This invention relates to the field of anti-interference suppression testing technology, and in particular to a method for testing the anti-interference suppression of signal lines in a receiving module. Background Technology

[0002] Current technologies for testing the anti-interference capabilities of receiver module signal lines often rely on single-frequency interference or simple time-domain waveform observation, lacking methods for separating nonlinear intermodulation components. This makes it difficult to accurately identify intermodulation products under multi-frequency interference superposition conditions. In actual testing, high-order nonlinear distortion is easily misjudged as background noise, affecting the accuracy of anti-interference capability assessment. Furthermore, timing analysis typically relies solely on instantaneous amplitude changes or simple delay measurements, lacking a unified time and amplitude scale for normalization. This makes it difficult to consistently compare results under different test conditions. Under extreme disturbance conditions, residual waveform structures during the embedding process are often ignored, focusing only on peak reduction. This makes it difficult to effectively identify edge changes and phase shifts caused by parasitic parameters. For example, under transient high-voltage impacts, signal edge stretching may not be recorded, leading to timing misalignments in subsequent system integration that are difficult to trace. Therefore, improvements are needed. Summary of the Invention

[0003] The purpose of this invention is to address the shortcomings of existing technologies by proposing a test method for interference suppression of receiver module signal lines.

[0004] To achieve the above objectives, the present invention adopts the following technical solution: a method for testing the anti-interference suppression of receiver module signal lines, comprising the following steps: A signal is injected into the signal line of the limiting diode connection at the front end of the receiving module, and a two-dimensional bispectral feature matrix is ​​generated. The peak amplitude at a specific intermodulation frequency coordinate is extracted in the two-dimensional bispectral feature matrix. The peak amplitude is calculated with the reference test value to generate a nonlinear intermodulation suppression coefficient. Based on the nonlinear intermodulation suppression coefficient, the timing parameters of the output voltage of the receiving module signal line are extracted under interference-free conditions to generate a standard ideal waveform sequence. A transient stimulation signal is injected into the TVS tube connected in parallel to the receiving module signal line to collect and generate a residual distorted waveform sequence. The standard ideal waveform sequence and the residual distorted waveform sequence are called to construct a normalized reference parameter set, and the waveform morphology fidelity score is calculated and generated based on the normalized reference parameter set. The waveform morphology fidelity score is compared with a preset waveform reference threshold to generate timing distortion evaluation parameters. Based on the timing distortion evaluation parameters, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network and the phase shift time parameter are extracted and compared to obtain the deformation suppression capability judgment result.

[0005] Preferably, the step of obtaining the nonlinear intermodulation suppression coefficient is as follows: Two different frequency single-tone interference signals and a basic useful signal are synchronously injected into the signal line of the limiting diode connection of the front end of the receiving module. The output signal waveform is collected, and the sampled values ​​of the output signal waveform are combined according to the preset delay and the product of each combination is accumulated to obtain the third-order cumulative quantity. Based on the third-order cumulant, the cumulative response on the two frequency axes is expanded in the frequency domain to establish a two-dimensional bispectral feature matrix. Within the two-dimensional bispectral feature matrix, specific intermodulation frequency coordinates are locked, and the peak amplitude and adjacent amplitudes at the specific intermodulation frequency coordinates are extracted to obtain the intermodulation amplitude parameter set. Based on the intermodulation amplitude parameter set, the peak amplitude at a specific intermodulation frequency coordinate, the amplitude at the horizontal adjacent position, the amplitude at the vertical adjacent position, and the benchmark test value are retrieved to calculate the nonlinear intermodulation suppression coefficient.

[0006] Preferably, the steps for obtaining the standard ideal waveform sequence are as follows: Retrieve continuous output records of the receiving module signal line under interference-free conditions, exclude sudden jump records, repeated sampling records and sampling interval mismatch records according to the stability screening boundary corresponding to the nonlinear intermodulation suppression coefficient, and extract the sampling time, voltage value, adjacent sampling interval and adjacent voltage change point by point to obtain the output voltage timing parameters. The sampling times in the output voltage timing parameters are renumbered in chronological order, and the corresponding voltage values ​​are arranged sequentially according to their numbers. For missing numbers, voltage values ​​with the same trend of adjacent voltage changes are added, and for duplicate numbers, the voltage value in the middle position is retained. The components are then continuously spliced ​​along the time axis to generate a standard ideal waveform sequence.

[0007] Preferably, the step of obtaining the residual distorted waveform sequence is as follows: According to the standard ideal waveform sequence, a transient high voltage stimulation signal is injected into the TVS tube connected in parallel to the signal line of the receiving module. The voltage values ​​at each sampling time after the TVS tube performs the clamping action, the clamping start time, the clamping end time, and the voltage change in the recovery segment are recorded. The residual timing output voltage parameters are extracted and arranged and spliced ​​according to the sampling time sequence to generate a residual distorted waveform sequence.

[0008] Preferably, the steps for obtaining the normalized reference parameter set are as follows: Extract the time interval values ​​between all adjacent sampling times in the standard ideal waveform sequence, arrange the time interval values ​​according to the size of the values, and determine the time interval value corresponding to the middle position as the time interval reference. Then, extract the absolute value of the difference between all adjacent sampled voltage values ​​in the standard ideal waveform sequence, arrange the absolute value of the difference according to the size of the values, and determine the absolute value of the voltage difference corresponding to the middle position as the voltage difference reference, thus forming a normalized reference parameter group.

[0009] Preferably, the step of obtaining the waveform morphology fidelity score is as follows: Read the time node parameters and voltage node parameters corresponding to each sampling moment of the standard ideal waveform sequence one by one, and then read the time node parameters and voltage node parameters corresponding to each sampling moment of the residual distorted waveform sequence one by one. Calculate the absolute value of the time node difference and the absolute value of the voltage node difference for any set of cross-sampling positions. Then, perform normalization processing using the time interval reference and the voltage difference reference respectively. Write the Euclidean distance corresponding to the normalized time difference and the normalized voltage difference into the corresponding cell of the two-dimensional spatial difference matrix. Then, extend the matrix cell by cell according to the minimum value among the cumulative values ​​of the left matrix cell, the upper matrix cell, and the upper left matrix cell to form the minimum connected path feature group.

[0010] Calculate the waveform morphology fidelity score based on the minimum connected path feature set.

[0011] Preferably, the steps for obtaining the timing distortion assessment parameters are as follows: The waveform morphology fidelity score and the preset waveform reference threshold are placed in the same judgment interval. The positive and negative deviation directions, the absolute value of the difference, and the percentage position of the difference in the preset waveform reference threshold are recorded after subtracting the preset waveform reference threshold from the waveform morphology fidelity score. The deviation levels are then divided according to the percentage position and written into a unified judgment field to generate a time-series distortion evaluation parameter.

[0012] Preferably, the step of obtaining the deformation suppression capability determination result is as follows: Based on the timing distortion evaluation parameters, continuous sampling segments with consistent deviation directions are screened out along the time axis. The sampling span that the voltage across the continuous sampling segment reaches a stable amplitude is calculated. The edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network is extracted. Then, the lag time length of the peak deviation position relative to the reference position within the continuous sampling segment is calculated, and the phase shift time parameter is extracted. Thus, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network and the phase shift time parameter are obtained. Based on the edge smoothing parameter and phase shift time parameter caused by the parasitic capacitance of the receiving module's anti-interference network, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network is compared with the preset time delay tolerance limit value item by item. Then, the phase shift time parameter is compared with the preset time delay tolerance limit value item by item. The tolerance status flags corresponding to each comparison result are recorded, and the merging and discrimination are performed according to the tolerance status flags to obtain the deformation suppression capability judgment result.

[0013] Compared with the prior art, the advantages and positive effects of the present invention are as follows: In this invention, by introducing a dual-frequency single-tone interference signal superimposed on the basic useful signal at the front end of the receiving module signal line, and combining third-order cumulant calculation with two-dimensional frequency domain expansion to form a bispectral feature expression, the nonlinear intermodulation components exhibit clear aggregation characteristics in the frequency domain. This enables the extraction of amplitude at specific intermodulation frequency positions, avoiding the problem of nonlinear distortion being masked in traditional time-domain observation. Simultaneously, a nonlinear intermodulation suppression coefficient is constructed using the ratio of amplitude to benchmark test value, transforming anti-interference capability from a qualitative judgment into a quantifiable indicator. Under interference-free conditions, the output voltage timing sequence is rigorously screened, sorted, and reconstructed to form a standard ideal waveform sequence with time consistency and amplitude continuity. Furthermore, by introducing transient high-voltage stimulation and recording the residual voltage change trajectory during the clamping process, a residual distortion waveform sequence is constructed, enabling the dynamic response of the signal link under extreme disturbance conditions. The characteristics are fully characterized. Based on this, the time scale and amplitude scale are unified by a normalized reference parameter set. The difference between the two types of waveforms is mapped to a two-dimensional spatial structure and the minimum connected path is extracted. This makes the local misalignment error globally optimally constrained. The waveform morphology fidelity score is calculated by combining the cumulative path distance and path structure characteristics. This allows the evaluation result to reflect both amplitude offset and time stretching characteristics. Furthermore, the timing distortion evaluation parameter is extracted by the difference deviation ratio and refined into two independent parameters: edge slowing degree and phase shift time. This separates and characterizes the influence of parasitic capacitance on signal edge morphology and timing delay. Finally, the deformation suppression capability is judged by comparing it item by item with the timing delay tolerance limit. This gives the overall evaluation process a closed-loop capability from frequency domain nonlinearity identification to time domain dynamic response to structured judgment, thereby improving the reliability of anti-interference performance evaluation. Attached Figure Description

[0014] Figure 1 This is a schematic diagram of the steps of the present invention; Figure 2 This is a two-dimensional bispectral feature matrix amplitude distribution diagram; Figure 3 A comparison chart of the standard ideal waveform sequence and the residual distorted waveform sequence; Figure 4 It is a two-dimensional spatial difference matrix and a minimum connected path graph. Detailed Implementation

[0015] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and not intended to limit the invention.

[0016] Please see Figures 1-4 This invention provides a technical solution: a method for testing the anti-interference suppression of signal lines in a receiving module, comprising the following steps: A signal is injected into the signal line of the limiting diode connection at the front end of the receiving module, and a two-dimensional bispectral feature matrix is ​​generated. The peak amplitude at a specific intermodulation frequency coordinate is extracted in the two-dimensional bispectral feature matrix. The peak amplitude is calculated with the benchmark test value to generate a nonlinear intermodulation suppression coefficient. Based on the nonlinear intermodulation suppression coefficient, the timing parameters of the output voltage of the receiving module signal line are extracted under interference-free conditions to generate a standard ideal waveform sequence. A transient stimulation signal is injected into the TVS tube connected in parallel to the receiving module signal line to collect and generate a residual distorted waveform sequence. By calling the standard ideal waveform sequence and the residual distorted waveform sequence, a normalized reference parameter set is constructed, and the waveform morphology fidelity score is calculated based on the normalized reference parameter set. The waveform morphology fidelity score is compared with the preset waveform reference threshold to generate timing distortion evaluation parameters. Based on the timing distortion evaluation parameters, the edge smoothing parameters caused by the parasitic capacitance of the receiving module's anti-interference network and the phase shift time parameters are extracted and compared to obtain the deformation suppression capability judgment result.

[0017] The steps for obtaining the nonlinear intermodulation suppression coefficient are as follows: Two different frequency single-tone interference signals and a basic useful signal are synchronously injected into the signal line of the limiting diode connection of the front end of the receiving module. The output signal waveform is collected, and the sampled values ​​of the output signal waveform are combined according to the preset delay and the product of each combination is accumulated to obtain the third-order cumulative quantity. Based on the third-order cumulant, the cumulative response on the two frequency axes is expanded in the frequency domain to establish a two-dimensional bispectral feature matrix. Within the two-dimensional bispectral feature matrix, specific intermodulation frequency coordinates are locked, and the peak amplitude and adjacent amplitudes at the specific intermodulation frequency coordinates are extracted to obtain the intermodulation amplitude parameter set. Based on the intermodulation amplitude parameter set, the peak amplitude at a specific intermodulation frequency coordinate, the amplitude at the horizontally adjacent position, the amplitude at the vertically adjacent position, and the reference test value are retrieved to calculate the nonlinear intermodulation suppression coefficient. The calculation formula is as follows: ; in, The nonlinear intermodulation suppression coefficient is... The benchmark value is the amplitude of the two-dimensional bispectral feature matrix extracted at a preset reference frequency coordinate under conditions of no interference signal injection. The first in the two-dimensional bispectral characteristic matrix The first frequency position and the first The peak amplitude at a specific intermodulation frequency coordinate corresponding to a second frequency position. The first in the two-dimensional bispectral characteristic matrix The first frequency position and the first The amplitude of the horizontal adjacent position corresponding to the second frequency position. The first in the two-dimensional bispectral characteristic matrix The first frequency position and the first The amplitude of the horizontal preceding position corresponding to the second frequency position. The first in the two-dimensional bispectral characteristic matrix The first frequency position and the first The amplitude of the vertical next adjacent position corresponding to the second frequency position. The first in the two-dimensional bispectral characteristic matrix The first frequency position and the first The amplitude of the vertical preceding position corresponding to the second frequency position. The position number of a specific intermodulation frequency coordinate on the first frequency axis. The position number of a specific intermodulation frequency coordinate on the second frequency axis.

[0018] Specifically, two different frequency single-tone interference signals and a basic useful signal are synchronously injected into the signal line connected to the limiting diode at the front end of the receiving module. This is done via an RF signal generator, with the center frequency of the basic useful signal set to [value missing]. And the transmission power is fixed at The frequency of the first single-tone interference signal is set to And the power is set to The frequency of the second monotone interference signal is set to And the power is also set to A high-speed oscilloscope is used to acquire the output signal waveform, and the sampling rate is set to... And set the quantization precision to Continuous collection Millisecond data segments to obtain The data from each sampling point constitutes a discrete sequence of the output signal waveform. Discrete voltage amplitudes are extracted sequentially from this sequence in chronological order as each sample value. A first preset time delay and a second preset time delay are set for statistical calculation, and the search step size for the delay is set to... Each sampling period and the search range is set to [number]. to Each sampling period measures the first preset time delay. Each sampling period and the second preset time delay are measured. Given specific combinations of sampling periods, extract the sampled voltage value at the current moment in the sequence and shift it backward. The sampled voltage value and backward offset for each sampling period The sampled voltage values ​​for each sampling period are multiplied together to obtain the combined product at the current moment. This process is repeated by sliding the current moment position sequentially according to the time series, extracting the voltage values ​​at the three corresponding positions and multiplying them, until the shift to the next position is completed. When the position of each sampling period reaches the end of the sequence, all the combined products generated during the sliding process are summed. The sum is divided by the total number of actual sliding operations and averaged to obtain the cumulative response value under the set time delay. All possible combinations of the first and second preset time delays are traversed according to the set range. The sampling value delay extraction, multiplication and accumulation operations are repeatedly performed. The response values ​​calculated each time are arranged according to the two-dimensional grid coordinates of the delay to obtain the third-order cumulative amount.

[0019] Based on the third-order cumulant, the previously calculated third-order cumulant data is extracted. This data is presented as a two-dimensional discrete sequence based on the first and second time delays. The resolution of the frequency transformation is set to... After applying Hamming window functions to smooth the edges of the third-order cumulant data across two time delay dimensions, a two-dimensional discrete Fourier transform is performed to convert the time-domain response data into a two-dimensional frequency domain space composed of the first and second frequency axes. The magnitude of the complex operation result corresponding to each two-dimensional frequency coordinate point is calculated as the frequency domain amplitude at that frequency point. The frequency domain amplitudes of all frequency coordinate points are written into the corresponding matrix units according to the arrangement of the first frequency axis as the horizontal index and the second frequency axis as the vertical index, thus establishing a two-dimensional bispectral feature matrix based on the frequency of the first injected single-tone interference signal. and the frequency of the second single-tone interference signal By multiplying the first frequency by two and then subtracting the second frequency, the intermodulation frequency under third-order nonlinearity is determined as follows: In the two-dimensional bispectral feature matrix, the first frequency axis corresponds to And the second frequency axis corresponds to The grid intersection position is used as the specific intermodulation frequency coordinate. The value of the matrix unit corresponding to the coordinate in the two-dimensional bispectral feature matrix is ​​read as the amplitude peak at the specific intermodulation frequency coordinate. Taking the coordinate as the center starting point, move one resolution step in each of the positive and negative directions of the first horizontal frequency axis, and read the corresponding two grid values ​​as the amplitude of the adjacent horizontal position. Move one resolution step in each of the positive and negative directions of the second vertical frequency axis, and read the corresponding two grid values ​​as the amplitude of the adjacent vertical position. The extracted amplitude peak at the specific intermodulation frequency coordinate and the amplitude of the four adjacent positions are concatenated and recorded according to a predetermined format to obtain the intermodulation amplitude parameter group.

[0020] In the formula for calculating the nonlinear intermodulation suppression coefficient, the amplitude of a single peak is compensated for by introducing a two-dimensional peripheral gradient difference term of the intermodulation peak. In the two-dimensional bispectral feature matrix, the intermodulation distortion product is not only manifested as an isolated spectral peak, but also generates spectral diffusion at adjacent frequency points due to energy leakage. Conventional calculations that only extract the central peak for comparison are prone to underestimating the actual total scale of intermodulation interference energy. The steps to obtain the parameters are as follows: The benchmark value is the amplitude of the two-dimensional bispectral feature matrix extracted at a preset reference frequency coordinate under pure conditions without any interference signal injection. Its unit is millivolt (mV). The acquisition steps are as follows: cut off the interference source at the front end of the receiving module, and maintain only the input power at [value missing]. And the frequency is Useful basic signals, enable high-speed acquisition and obtain them. The output waveform sequence in milliseconds is processed using the same calculation steps as establishing a two-dimensional bispectral feature matrix to obtain a reference two-dimensional matrix in an interference-free state. The preset reference frequency coordinates are then positioned on the first frequency axis. With the second frequency axis The cross-grid is used to read the amplitude values ​​stored within each grid cell. For example, after actual testing and discrete Fourier transform extraction in a clean environment, the amplitude data at the reference frequency coordinate is read as follows. To obtain benchmark test values ; The steps to obtain the parameters are as follows: The peak amplitude at a specific intermodulation frequency coordinate in the two-dimensional bispectral feature matrix represents the voltage amplitude of the strongest nonlinear distortion component generated at a specific intermodulation frequency point when two-tone interference is input. The unit of measurement is millivolts (mV). For example, by consulting the stored data at the center coordinates of the intermodulation amplitude parameter group, the distortion amplitude at that coordinate position can be read. The peak amplitude at a specific intermodulation frequency coordinate is obtained. ; The steps to obtain the parameters are as follows: The amplitude at the corresponding lateral adjacent position in the two-dimensional bispectral characteristic matrix is ​​used to characterize the voltage amplitude level of intermodulation distortion energy propagating and leaking along the high-frequency direction of the second frequency axis. The unit of measurement is also millivolts. For example, retrieve the amplitude data of the second frequency axis delayed by one unit position from the intermodulation amplitude parameter group. The amplitude of the horizontal adjacent position is obtained. ; The steps to obtain the parameters are as follows: The amplitude at the corresponding lateral preceding position in the two-dimensional bispectral feature matrix represents the voltage amplitude of energy leakage of intermodulation distortion products along the low-frequency direction of the second frequency axis, with the unit being millivolts (mV). For example, the amplitude data extracted from the intermodulation amplitude parameter set by shifting the second frequency axis forward by one unit is: The amplitude of the horizontal preceding position is obtained. ; The steps to obtain the parameters are as follows: The amplitude at the corresponding longitudinal next-neighbor position in the two-dimensional bispectral feature matrix reflects the interference amplitude overflowing from the nonlinear distortion spike along the high-frequency direction of the first frequency axis. The unit of measurement is millivolts (mV). For example, the amplitude data read from the intermodulation amplitude parameter group, which is offset by one unit position in the positive direction of the first frequency axis, is: The amplitude of the vertical next neighbor position is obtained. ; The steps to obtain the parameters are as follows: The amplitude at the corresponding longitudinal preceding position in the two-dimensional bispectral characteristic matrix represents the amplitude of the interference leakage voltage radiated along the low-frequency region of the first frequency axis from the peak of nonlinear distortion. The unit of measurement is millivolts (mV). For example, extracting the amplitude data with a negative offset of one unit position on the first frequency axis from the intermodulation amplitude parameter set is: The amplitude of the vertical preceding position is obtained. ; Calculations based on parameters: Calculate the amplitude difference between adjacent horizontal positions, considering the index span of adjacent positions in the grid (i.e. minus () is dimensionless Therefore, we can directly substitute the voltage amplitude while keeping the dimension of the difference in millivolts. ; By squaring the horizontal difference, the dimension is increased to square millivolts. ; Calculate the amplitude difference between adjacent longitudinal positions. ; Squaring the vertical differences. ; Sum the squares of the differences between the horizontal and vertical directions and then divide by the dimensionless parameter. The mean of the two-dimensional peripheral gradient difference terms is obtained. ; Calculate the square of the peak amplitude at a specific intermodulation frequency coordinate at the center. ; The total squared energy of the intermodulation is obtained by adding the mean of the central squared term and the gradient difference term of the two-dimensional periphery. ; The equivalent total intermodulation distortion intensity is obtained by square rooting the summation result. ; The nonlinear intermodulation suppression coefficient is obtained by dividing the benchmark value by the equivalent total intermodulation distortion intensity. ; The results indicate the relative efficiency coefficient of the receiving circuit in resisting nonlinear interference and maintaining the complete transmission of the useful signal. A larger value indicates a smaller amplitude of the equivalent intermodulation distortion product excited by two-tone interference in the denominator, while the amplitude of the reference useful signal in the numerator remains at a high level. This indicates stronger filtering and clipping resistance of the receiving network against RF front-end interference, and better nonlinear intermodulation suppression capability in the signal line interference suppression test. When the value is smaller than... When the value is 1, it indicates that the intermodulation distortion energy has risen to a dangerous range that is close to or even threatens the demodulation threshold of the useful signal. This indicates that the limiting diode network in the hardware circuit has experienced extremely severe nonlinear cutoff distortion or avalanche conduction failure when resisting dual-tone synchronization interference. The nonlinear intermodulation suppression coefficient can provide a reliable quantitative decision benchmark for subsequent exclusion of sudden jump records, delineation of stable screening boundaries, and generation of standard ideal waveform sequences.

[0021] The steps for obtaining a standard ideal waveform sequence are as follows: Retrieve continuous output records of the receiving module signal line under interference-free conditions, exclude sudden jump records, repeated sampling records and sampling interval mismatch records according to the stability screening boundary corresponding to the nonlinear intermodulation suppression coefficient, and extract the sampling time, voltage value, adjacent sampling interval and adjacent voltage change amount point by point to obtain the output voltage timing parameters. The sampling times in the output voltage timing parameters are renumbered in chronological order, and the corresponding voltage values ​​are arranged sequentially according to their numbers. For missing numbers, voltage values ​​with the same trend of adjacent voltage changes are added, and for duplicate numbers, the voltage value in the middle position is retained. The data are then continuously spliced ​​along the time axis to generate a standard ideal waveform sequence.

[0022] Specifically, retrieve the continuous output records of the receiving module's signal line under interference-free conditions, and extract the previously calculated nonlinear intermodulation suppression coefficient, whose value is known to be... Based on this, and considering the thermal noise distribution characteristics of the receiving module under static conditions, a baseline voltage fluctuation value is set. The peak-to-peak value of the module's noise floor fluctuation when there is no signal input is measured, and half of this value is taken as the baseline. For example, the measured peak-to-peak value of the noise floor is... The baseline voltage fluctuation is then set to a value that represents the basic voltage fluctuation. The upper limit threshold of the stability screening boundary is obtained by multiplying the previously obtained nonlinear intermodulation suppression coefficient by the base voltage fluctuation reference value. The specific calculation process is as follows: Multiply The conclusion is This value will serve as the threshold for judging whether subsequent waveform voltage jumps are abnormal, based on the acquisition equipment. The ideal sampling interval benchmark value is calculated from the sampling rate. The allowable deviation of the time axis offset is set to ±5%, from which the acceptable range of the effective sampling time interval is calculated to be... to The system reads the absolute timestamp and voltage amplitude corresponding to each sampling node from the output record one by one. The absolute timestamp of the current node minus the absolute timestamp of the previous node is used as the adjacent sampling interval. The absolute value of the voltage amplitude of the current node minus the voltage amplitude of the previous node is used as the adjacent voltage change. The calculated adjacent sampling interval is compared with zero. If the difference equals... This is determined to be a redundant record from the same moment, and it is treated as a duplicate sample and removed. Adjacent sampling intervals are compared with the acceptable intervals; if the time difference is less than [a certain value], [the record is considered valid]. or greater than Records identified as sampling interval mismatch due to clock jitter are marked and excluded. The calculated adjacent voltage changes are compared with the previously calculated upper limit threshold of the stability screening boundary. Expand the size comparison; if the change is greater than... If the waveform is found to have distorted and jumped, it is classified as a sudden jump record and removed. All qualified data nodes that have been screened and retained are finally integrated. The sampling time, corresponding voltage value, calculated adjacent sampling interval and adjacent voltage change are captured one by one. These four indicators are bound into a complete data structure group to obtain the output voltage timing parameters.

[0023] The sampling times in the previously obtained output voltage timing parameters are renumbered chronologically. Starting from the first data entry of the output voltage timing parameters, the position number corresponding to the earliest sampling time is set to the number 1. The time information of the remaining entries is read one by one along the time axis. Each time a sampling time is advanced, the value of the current position number is incremented by one, thereby generating a set of integer sequence identifiers that increment from 1 to the total data length. The original voltage value of each record is bound to this newly generated set of integer sequence identifiers. All of them are placed in a one-dimensional array structure and arranged sequentially according to the rule of ascending sequence numbers. After initial arrangement, sequence continuity verification is initiated. The difference between the current item number and the previous item number is extracted, and it is determined whether this difference is greater than one. If the difference is greater than one, it is determined that there is a missing number position in the middle of the sequence. The voltage values ​​corresponding to the previous and next existing nodes in the missing interval are extracted. The difference between the subsequent voltage and the previous voltage is divided by the total span of the missing node plus one to obtain the single-step average voltage change trend. The previous voltage value is superimposed with this single-step average voltage change trend as the filling data for the first missing position. For example, the voltage at number 20 in the sequence is... And the voltage at position number twenty-three is Difference between the two ends Divide by the number span three to obtain the single-step trend quantity. Then fill in the missing number twenty-one. Fill in the missing number twenty-two. This process involves filling in voltage values ​​with consistent trends for adjacent voltage changes at all missing number positions. Then, it checks for stacking conflicts with the same sequence number within the array. If three sampling records are found under the same sequence number, they are arranged according to their original time sequence. The voltage value carried by the second record, which is in the middle, is extracted, while the first and third records are removed. This method retains the voltage value in the middle position for duplicate number positions. After clearing all redundancy and gaps, these voltage values ​​carrying absolutely continuous one-dimensional sequence numbers are pasted and spliced ​​one by one onto a unified discrete time axis template to generate a standard ideal waveform sequence.

[0024] The steps for obtaining the residual distorted waveform sequence are as follows: Based on the standard ideal waveform sequence, a transient high-voltage stimulation signal is injected into the TVS diode connected in parallel to the signal line of the receiving module. The voltage values ​​at each sampling time after the TVS diode performs clamping action, the clamping start time, the clamping end time, and the voltage change during the recovery segment are recorded. The residual timing output voltage parameters are extracted and then arranged and spliced ​​according to the sampling time sequence to generate a residual distorted waveform sequence.

[0025] Specifically, based on the previously generated standard ideal waveform sequence, a transient high-voltage stimulation signal to simulate the deterioration of the external electromagnetic environment is injected into the TVS transient voltage suppression diode connected in parallel to the signal line of the receiving module. This stimulation signal is set to have a rise time of... to Between and peak voltages up to A standard electrostatic discharge pulse is used to activate an oscilloscope to capture the real-time voltage fluctuations on the circuit after the electric shock. The datasheet of the parallel TVS diode is consulted to obtain its rated breakdown threshold; for example, the datasheet specifies a working withstand voltage of [value missing]. And the action embedding boundary is Then Set the clamping trigger determination threshold, extract the timing node voltages from the acquisition trajectory one by one from beginning to end, and compare them with the clamping trigger determination threshold. When comparing numerical values, if the read voltage value suddenly rises from a normal low level and is first greater than or equal to... At that time, it is confirmed that avalanche conduction is initiated inside the tube. This transient time point where the limit is exceeded is extracted as the clamping start time. The voltage values ​​of the trajectory sequence are scanned backward to find the inflection point of the voltage level drop after the pulse energy is discharged. When the voltage value drops from a high level and first breaks through the limit, the signal is detected. When restoring to the node below it, register its corresponding time node as the clamping end time, select all waveform points between the clamping start time and the clamping end time, and extract the voltage values ​​at each sampling time during this high-voltage suppression period in sequence. Continue sampling after the clamping end time. The time window is used as the recovery monitoring segment to extract this. The voltage at the initial moment of the window is subtracted from the voltage at the last moment, and the absolute value of the difference is extracted as the voltage change in the recovery period. The voltage values ​​of each node in the clamping period, the start and end timestamps, and the voltage drop difference index in the recovery period are packaged and recorded to form the residual timing output voltage parameters. All discrete voltage samples in this parameter set are extracted, and their respective sampling timestamps are read. According to the trend of the timestamp values ​​from small to large, these discrete voltage samples are repositioned and queued to generate the residual distorted waveform sequence.

[0026] The steps for obtaining the normalized reference parameter set are as follows: Extract the time interval values ​​between all adjacent sampling times in the standard ideal waveform sequence, arrange the time interval values ​​according to the size of the values, and determine the time interval value corresponding to the middle position as the time interval reference. Then extract the absolute value of the difference between all adjacent sampled voltage values ​​in the standard ideal waveform sequence, arrange the absolute value of the difference according to the size of the values, and determine the absolute value of the voltage difference corresponding to the middle position as the voltage difference reference, thus forming a normalized reference parameter group.

[0027] Specifically, the time interval values ​​between all adjacent sampling moments in the previously obtained standard ideal waveform sequence are extracted. For example, if the sequence contains two hundred sampling nodes, adjacent moments are subtracted pairwise according to their chronological order, resulting in one hundred and ninety-nine time interval values. These one hundred and ninety-nine time interval values ​​with picosecond dimensions are arranged in ascending order. The time interval value corresponding to the exact middle position in the arranged sequence is found and determined as the time interval benchmark. If the total number is odd, the absolute middle value is taken; if the total number is even, the average of the two middle values ​​is taken. For example, the time interval value at the 100th position after arrangement is... Then this The extracted values ​​are saved as a time interval reference. Then, the absolute values ​​of the voltage differences between all adjacent sampling nodes in the standard ideal waveform sequence are extracted. The voltage value of the next node is subtracted from the voltage value of the previous node, and the sign is removed before taking the absolute value. This process generates 199 absolute voltage difference values. These absolute voltage difference values, which have millivolt dimensions, are arranged in ascending order from smallest to largest. The absolute voltage difference value at the exact middle position is identified and determined as the voltage difference reference. For example, the voltage difference at the middle position after arrangement is... Then this The voltage difference benchmark is determined as the reference. The extracted time interval benchmark and the voltage difference benchmark are packaged and merged for storage to form a normalized reference parameter group.

[0028] The steps for obtaining the waveform morphology fidelity score are as follows: Read the time node parameters and voltage node parameters corresponding to each sampling moment of the standard ideal waveform sequence one by one, and then read the time node parameters and voltage node parameters corresponding to each sampling moment of the residual distorted waveform sequence one by one. Calculate the absolute value of the time node difference and the absolute value of the voltage node difference for any set of cross-sampling positions. Then, perform normalization processing using the time interval reference and the voltage difference reference respectively. Write the Euclidean distance corresponding to the normalized time difference and the normalized voltage difference into the corresponding cell of the two-dimensional spatial difference matrix. Then, extend the matrix cell by cell according to the minimum value among the cumulative values ​​of the left matrix cell, the upper matrix cell, and the upper left matrix cell to form the minimum connected path feature group. Based on the minimum connected path feature set, the waveform morphology fidelity score is calculated using the following formula: ; in, To preserve the fidelity of the waveform shape, This is the sum of the distances between all matrix elements along the least connected path. This represents the total number of path nodes contained in the minimum connected path. This represents the standard deviation of the slope values ​​of the movement of adjacent path nodes in the minimum connected path. The time interval reference in the normalized reference parameter set is the time interval value corresponding to the middle position after arranging all adjacent sampling times of the standard ideal waveform sequence according to their numerical magnitude. The voltage difference benchmark in the normalized reference parameter group is the absolute value of the voltage difference at the middle position of the absolute values ​​of the differences between all adjacent sampled voltage values ​​in the standard ideal waveform sequence, arranged according to their numerical magnitude. As a time unit reference constant, This is the voltage unit reference constant.

[0029] Specifically, the time and voltage node parameters corresponding to each sampling moment of the preceding standard ideal waveform sequence are read one by one from the data storage area as reference standards. Then, the time and voltage node parameters corresponding to each sampling moment of the preceding residual distorted waveform sequence are read one by one as comparison targets. A blank two-dimensional grid is constructed, with the nodes of the ideal waveform sequence as the vertical axis and the nodes of the distorted waveform sequence as the horizontal axis. For any set of cross-sampling positions within the grid, such as the 50th node of the ideal sequence and the 52nd node of the distorted sequence, the absolute values ​​of the actual time node difference and the absolute values ​​of the voltage node difference are calculated respectively. For example, the absolute value of the actual time node difference at this position is... And the absolute value of the voltage node difference is Then, call the previously generated normalized reference parameter set, and use the time interval reference respectively. and voltage difference reference Perform normalization processing, divide the actual time difference by the time base, and cancel out the physical units to obtain a dimensionless normalized time difference. Dividing the actual voltage difference by the voltage reference and canceling out the physical units yields a dimensionless normalized voltage difference. The dimensionless two-dimensional Euclidean distance corresponding to the intersection position is calculated by adding the square of the normalized time difference to the square of the normalized voltage difference and then taking the square root. The calculated Euclidean distance is written into the corresponding cell of the two-dimensional spatial difference matrix. After traversing all intersections and filling the matrix, the minimum cost path is searched from the starting alignment point in the upper left corner of the matrix to the lower right corner. For the current grid, the minimum value among the cumulative distance values ​​of the left matrix cell, the upper matrix cell, and the upper left diagonal matrix cell is selected. The minimum cumulative value is added to the Euclidean distance of the current grid as the new cumulative value of the current grid. This dynamic programming method is used to extend the path grid by grid towards the lower right corner endpoint. The grid coordinates of the grid with the minimum total cost are traced back from the endpoint. The coordinate index and distance data of each grid along the way are extracted to form the minimum connected path feature group.

[0030] In the waveform morphology fidelity score calculation formula, an exponential decay model based on minimum connected paths is introduced to quantify the morphological similarity between two waveforms. Traditional error calculation relies solely on the subtraction of amplitudes at the same moment, which can easily produce huge error artifacts due to small time shifts. In the core term of the exponential function, this formula first uses the sum of the total distances of the minimum connected paths divided by the total number of path nodes to obtain the average single-step alignment error, thereby evaluating the overall offset cost. Then, the ratio of the standard deviation of the path movement slope to the square root of the sum of squares of the dimensionless reference parameters is introduced as a morphological distortion penalty term. This penalty term can keenly capture the degree of local morphological distortion of the waveform during time stretching and voltage change. The steps to obtain the parameters are as follows: This is the sum of the distances between all matrix elements on the minimum connected path, representing the total morphological displacement cost incurred throughout the entire process from the starting alignment node to the ending node. For example, the minimum connected path obtained through backtracking traverses... Extract these grid cells sequentially. The distance values ​​within each unit are accumulated, and the final total sum of the accumulated distances is calculated as follows: This yields the sum of the distances between all matrix elements along the minimum connected path. ; The steps to obtain the parameters are as follows: The minimum connected path contains the total number of path nodes. For example, by iterating through the array sequence length of the previously extracted minimum connected path feature groups, the total number of nodes contained in the aligned path can be counted. The total number of path nodes contained in the minimum connected path is obtained from the grid nodes that have undergone cross-association comparison. ; The steps to obtain the parameters are as follows: This is the standard deviation of the moving slope values ​​of adjacent path nodes in the minimum connected path. It is used to measure the smoothness and consistency of the matching path during lateral time extension and vertical feature advancement. For example, it is calculated by extracting the coordinates of the feature group. The standard deviation of the moving slope values ​​of each adjacent path node is obtained by averaging and summing the squares of the deviations. ; The steps to obtain the parameters are as follows: For the time interval reference in the normalized reference parameter group, for example, retrieving the time interval reference data obtained from previous actual measurements and after median arrangement, the value at this position is read as... The time interval reference in the normalized reference parameter set is obtained. ; The steps to obtain the parameters are as follows: For the voltage difference reference in the normalized reference parameter group, for example, retrieving the previously sorted, identified, and established median voltage difference data object, and reading the specific value stored at that location. The voltage difference reference in the normalized reference parameter set is obtained. ; The steps to obtain the parameters are as follows: This serves as a time unit reference constant, for example, by pre-setting this constant based on the equipment's accuracy characteristics. The time unit reference constant is obtained. ; The steps to obtain the parameters are as follows: This is a voltage unit reference constant, used as a specific calibration reference value to eliminate the physical dimensions of the voltage parameter term in the denominator. For example, it is fixed as a constant for high-frequency level jitter in the millivolt range. The voltage unit reference constant is obtained. ; Calculations based on parameters: Calculate the dimensionless ratio of the time term. ; Calculate the dimensionless ratio of the voltage term. ; Square the dimensionless numbers individually to obtain... as well as ; Add the squares of the two values ​​together to get the result. ; The comprehensive reference dimensionless scale is obtained by performing a square root operation on the sum of the squares. ; Dividing the dimensionless moving slope standard deviation by the composite baseline dimensionless scale yields ; Adding the above division result to the constant 1 yields the coefficient of the morphological distortion penalty term. ; The average single-step alignment error is obtained by summing the total distances and dividing by the total number of nodes. ; The average single-step alignment error is multiplied by the morphological distortion penalty term coefficient to obtain the comprehensive attenuation index. ; Substituting the negative sign of the composite attenuation exponent into the natural exponential function yields the waveform morphology fidelity score. ; This result demonstrates the degree of fidelity in the reproduction of the residual distorted waveform relative to the standard ideal waveform in terms of time-domain and amplitude characteristics. The numerical result ranges from zero to one, and when this fidelity score is greater than... This indicates that the transient stimulus signal was rapidly and effectively clamped and suppressed by the anti-interference network connected in parallel with the receiving module. The distorted waveform recovered its timing trend and voltage step characteristics, which were highly consistent with the ideal waveform, within a very short time. This proves that the internal junction capacitance of the hardware network is small and its deformation suppression capability is excellent. When the numerical result is less than... When the waveform is significantly slowed down or even phase-shifted due to excessive parasitic capacitance in the anti-interference network or slow reverse recovery of the diode, the waveform edges are severely hampered, leading to a sharp increase in the dynamic alignment cost during the two-dimensional matching process. This fidelity score, as the core quantitative parameter for judging the severity of waveform damage, will be compared logically with the preset waveform benchmark threshold in the same judgment interval in subsequent steps to generate a timing distortion assessment parameter that reflects the hardware defect level.

[0031] The steps for obtaining the parameters for time series distortion assessment are as follows: The waveform morphology fidelity score and the preset waveform baseline threshold are placed in the same judgment interval. The positive and negative deviation directions, the absolute value of the difference, and the percentage position of the difference in the preset waveform baseline threshold are recorded after subtracting the preset waveform baseline threshold from the waveform morphology fidelity score. The deviation levels are then divided according to the percentage position and written into a unified judgment field to generate time series distortion evaluation parameters.

[0032] Specifically, retrieve the previously calculated waveform morphology fidelity score, for example, 0.6335. Review the fidelity score data of the module in 100 interference-free test records. Summate these 100 scores and divide by 100 to obtain the average of 0.9500. Set this average of 0.9500 as the preset waveform baseline threshold. Create a continuous value band from zero to one as the same judgment interval. Place the waveform morphology fidelity score of 0.6335 and the preset waveform baseline threshold of 0.9500 together within this value band for position observation. Subtract the preset waveform baseline threshold from the waveform morphology fidelity score, i.e., 0.6335 minus 0.9500, resulting in a negative 0.3165. Based on the characteristic of the negative sign in the calculation result, record the positive or negative deviation direction as negative deviation. After removing the negative sign, extract 0.3165 as the absolute value of the difference. Divide this absolute value of the difference by the preset waveform baseline threshold. The baseline threshold, 0.3165 divided by 0.9500, yields 0.3331. Converting 0.3331 to a percentage (33.31%), this is used as the percentage position of the difference within the preset waveform baseline threshold. A set of percentage level classification standards is established, defining a percentage position between 0% and 10% as a slight deviation level, between 10% and 30% as a moderate deviation level, and greater than 30% as a severe deviation level. Comparing 33.31% with this standard, since it is greater than 30%, it is classified as a severe deviation based on its percentage position. A unified judgment field containing multiple dimensions such as direction, value, and level is established. The recorded negative deviation direction, the absolute value of the difference of 0.3165, the percentage position of 33.31%, and the severe deviation level label are successively filled into this field for encapsulation, generating time series distortion assessment parameters.

[0033] The steps for obtaining the deformation suppression capability assessment result are as follows: Based on the timing distortion evaluation parameters, continuous sampling segments with consistent deviation directions are screened out along the time axis. The sampling span that the voltage across the continuous sampling segment reaches a stable amplitude is calculated. The edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network is extracted. Then, the lag time length of the peak deviation position relative to the reference position within the continuous sampling segment is calculated, and the phase shift time parameter is extracted. Thus, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network and the phase shift time parameter are obtained. Based on the edge smoothing parameter and phase shift time parameter caused by the parasitic capacitance of the receiving module's anti-interference network, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network is compared with the preset time delay tolerance limit value item by item. Then, the phase shift time parameter is compared with the preset time delay tolerance limit value item by item. The tolerance status flags corresponding to each comparison result are recorded, and the merging and discrimination are performed according to the tolerance status flags to obtain the deformation suppression capability judgment result.

[0034] Specifically, extract the previously generated timing distortion assessment parameters, read the negative deviation direction labels, and scan the local deviations point by point along the time axis of the waveform data from left to right. Find regions where multiple consecutive sampling points show negative deviations, and filter out continuous sampling segments with consistent deviation directions. For example, if a continuous negative deviation region containing fifty sampling points is extracted on the time axis, find the starting position of the voltage surge within this continuous sampling segment, and then find the ending position where the voltage fluctuation tends to level off. Set the condition for leveling off as the voltage change difference between three adjacent sampling points being less than 1.0 mV. Locate the first point that meets this condition as the stable point, extract the sampling point number of the starting position, for example, the 120th point, and then extract the sampling point number of the stable point, for example, the 165th point. Subtract the starting point number from the stable point number to obtain the span value of 45. Calculate the sampling span of 45 sampling points that the voltage reaches a stable amplitude within the continuous sampling segment, and extract the span of these 45 sampling points. As a parameter for edge slowing caused by parasitic capacitance in the receiver module's anti-interference network, since capacitor charging and discharging causes the step edge to lengthen and the larger the span value, the more severe the slowing effect, the more we continue to find the specific sampling time position where the actual distorted waveform voltage value reaches its highest peak value within the continuous sampling segment. For example, if the highest peak value is located at the 175th sampling time, we retrieve the reference time position where the peak value should have appeared in the previous standard ideal waveform sequence. For example, if the reference position is at the 140th sampling time, we subtract the reference position from the actual peak position to obtain a difference of 35 sampling intervals. For example, if each sampling interval is 416.67 picoseconds, multiplying 35 by 416.67 yields 14583.45 picoseconds. We calculate the lag time length of the peak deviation position relative to the reference position within the continuous sampling segment as 14583.45 picoseconds. We extract this time length value as the phase shift time parameter. By summing the obtained span value and lag time value, we obtain the parameter for edge slowing caused by parasitic capacitance in the receiver module's anti-interference network and the parameter for phase shift time.

[0035] Using the previously obtained parameters for edge smoothing caused by parasitic capacitance in the receiver module's anti-interference network and phase shift time, and referring to the receiver module's datasheet, we find that to ensure the normal operation of the subsequent decoding circuit, the waveform edge sampling span should not exceed fifty sampling points, and the total phase delay due to parasitic elements should not exceed 20,000 picoseconds. Based on these specifications, we set fifty sampling points as the preset allowable time delay limit for edge smoothing and 20,000 picoseconds as the preset allowable time delay limit for phase shift. We then compare the previously extracted parameter for edge smoothing caused by parasitic capacitance in the receiver module's anti-interference network (forty-five sampling points) with the preset time delay limit of fifty sampling points. Since forty-five is less than fifty and does not exceed the allowable boundary, we record this comparison. The permissible status flag corresponding to the result is marked as qualified. Then, the actual phase shift time parameter calculated earlier, i.e., 14583.45 picoseconds, is compared with the preset permissible limit value of 20000 picoseconds. Since 14583.45 is less than 20000 and is also within the safety limit, the permissible status flag corresponding to this comparison result is recorded as qualified. The rule for merging the judgment is set as follows: the overall acceptance is deemed to have passed only when all the items participating in the comparison are marked as qualified. If any item is marked as unqualified, the overall failure is deemed to have occurred. The merging judgment is performed according to this set rule. Since the previous two comparison results have each obtained qualified flags, the final conclusion after merging the two is that they have passed. This conclusion of passing is output as the review opinion in the test report, and the deformation suppression capability judgment result is obtained.

[0036] The above are merely preferred embodiments of the present invention and are not intended to limit the present invention in any other way. Any person skilled in the art may make changes or modifications to the above-disclosed technical content to create equivalent embodiments that can be applied to other fields. However, any simple modifications, equivalent changes, and modifications made to the above embodiments based on the technical essence of the present invention without departing from the scope of the present invention shall still fall within the protection scope of the present invention.

Claims

1. A method for testing the anti-interference suppression of signal lines in a receiving module, characterized in that, Includes the following steps: A signal is injected into the signal line of the limiting diode connection at the front end of the receiving module, and a two-dimensional bispectral feature matrix is ​​generated. The peak amplitude at a specific intermodulation frequency coordinate is extracted in the two-dimensional bispectral feature matrix. The peak amplitude is calculated with the reference test value to generate a nonlinear intermodulation suppression coefficient. Based on the nonlinear intermodulation suppression coefficient, the timing parameters of the output voltage of the receiving module signal line are extracted under interference-free conditions to generate a standard ideal waveform sequence. A transient stimulation signal is injected into the TVS tube connected in parallel to the receiving module signal line to collect and generate a residual distorted waveform sequence. The standard ideal waveform sequence and the residual distorted waveform sequence are called to construct a normalized reference parameter set, and the waveform morphology fidelity score is calculated and generated based on the normalized reference parameter set. The waveform morphology fidelity score is compared with a preset waveform reference threshold to generate timing distortion evaluation parameters. Based on the timing distortion evaluation parameters, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network and the phase shift time parameter are extracted and compared to obtain the deformation suppression capability judgment result.

2. The method for testing the anti-interference suppression of the receiving module signal line according to claim 1, characterized in that, The steps for obtaining the nonlinear intermodulation suppression coefficient are as follows: Two different frequency single-tone interference signals and a basic useful signal are synchronously injected into the signal line of the limiting diode connection of the front end of the receiving module. The output signal waveform is collected, and the sampled values ​​of the output signal waveform are combined according to the preset delay and the product of each combination is accumulated to obtain the third-order cumulative quantity. Based on the third-order cumulant, the cumulative response on the two frequency axes is expanded in the frequency domain to establish a two-dimensional bispectral feature matrix. Within the two-dimensional bispectral feature matrix, specific intermodulation frequency coordinates are locked, and the peak amplitude and adjacent amplitudes at the specific intermodulation frequency coordinates are extracted to obtain the intermodulation amplitude parameter set. Based on the intermodulation amplitude parameter set, the peak amplitude at a specific intermodulation frequency coordinate, the amplitude at the horizontal adjacent position, the amplitude at the vertical adjacent position, and the benchmark test value are retrieved to calculate the nonlinear intermodulation suppression coefficient.

3. The method for testing the anti-interference suppression of the receiving module signal line according to claim 1, characterized in that, The steps for obtaining the standard ideal waveform sequence are as follows: Retrieve continuous output records of the receiving module signal line under interference-free conditions, exclude sudden jump records, repeated sampling records and sampling interval mismatch records according to the stability screening boundary corresponding to the nonlinear intermodulation suppression coefficient, and extract the sampling time, voltage value, adjacent sampling interval and adjacent voltage change point by point to obtain the output voltage timing parameters. The sampling times in the output voltage timing parameters are renumbered in chronological order, and the corresponding voltage values ​​are arranged sequentially according to their numbers. For missing numbers, voltage values ​​with the same trend of adjacent voltage changes are added, and for duplicate numbers, the voltage value in the middle position is retained. The components are then continuously spliced ​​along the time axis to generate a standard ideal waveform sequence.

4. The method for testing the anti-interference suppression of the receiving module signal line according to claim 1, characterized in that, The steps for obtaining the residual distorted waveform sequence are as follows: According to the standard ideal waveform sequence, a transient high voltage stimulation signal is injected into the TVS tube connected in parallel to the signal line of the receiving module. The voltage values ​​at each sampling time after the TVS tube performs the clamping action, the clamping start time, the clamping end time, and the voltage change in the recovery segment are recorded. The residual timing output voltage parameters are extracted and arranged and spliced ​​according to the sampling time sequence to generate a residual distorted waveform sequence.

5. The method for testing the anti-interference suppression of the receiving module signal line according to claim 1, characterized in that, The steps for obtaining the normalized reference parameter set are as follows: Extract the time interval values ​​between all adjacent sampling times in the standard ideal waveform sequence, arrange the time interval values ​​according to the size of the values, and determine the time interval value corresponding to the middle position as the time interval reference. Then, extract the absolute value of the difference between all adjacent sampled voltage values ​​in the standard ideal waveform sequence, arrange the absolute value of the difference according to the size of the values, and determine the absolute value of the voltage difference corresponding to the middle position as the voltage difference reference, thus forming a normalized reference parameter group.

6. The method for testing the anti-interference suppression of the receiving module signal line according to claim 1, characterized in that, The steps for obtaining the waveform morphology fidelity score are as follows: Read the time node parameters and voltage node parameters corresponding to each sampling moment of the standard ideal waveform sequence one by one, and then read the time node parameters and voltage node parameters corresponding to each sampling moment of the residual distorted waveform sequence one by one. Calculate the absolute value of the time node difference and the absolute value of the voltage node difference for any set of cross-sampling positions. Then, perform normalization processing using the time interval reference and the voltage difference reference respectively. Write the Euclidean distance corresponding to the normalized time difference and the normalized voltage difference into the corresponding cell of the two-dimensional spatial difference matrix. Then, extend the matrix cell by cell according to the minimum value among the cumulative values ​​of the left matrix cell, the upper matrix cell, and the upper left matrix cell to form the minimum connected path feature group. Calculate the waveform morphology fidelity score based on the minimum connected path feature set.

7. The method for testing the anti-interference suppression of the receiving module signal line according to claim 1, characterized in that, The steps for obtaining the timing distortion assessment parameters are as follows: The waveform morphology fidelity score and the preset waveform reference threshold are placed in the same judgment interval. The positive and negative deviation directions, the absolute value of the difference, and the percentage position of the difference in the preset waveform reference threshold are recorded after subtracting the preset waveform reference threshold from the waveform morphology fidelity score. The deviation levels are then divided according to the percentage position and written into a unified judgment field to generate a time-series distortion evaluation parameter.

8. The method for testing the anti-interference suppression of the signal line of the receiving module according to claim 1, characterized in that, The steps for obtaining the deformation suppression capability determination result are as follows: Based on the timing distortion evaluation parameters, continuous sampling segments with consistent deviation directions are screened out along the time axis. The sampling span that the voltage across the continuous sampling segment reaches a stable amplitude is calculated. The edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network is extracted. Then, the lag time length of the peak deviation position relative to the reference position within the continuous sampling segment is calculated, and the phase shift time parameter is extracted. Thus, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network and the phase shift time parameter are obtained. Based on the edge smoothing parameter and phase shift time parameter caused by the parasitic capacitance of the receiving module's anti-interference network, the edge smoothing parameter caused by the parasitic capacitance of the receiving module's anti-interference network is compared with the preset time delay tolerance limit value item by item. Then, the phase shift time parameter is compared with the preset time delay tolerance limit value item by item. The tolerance status flags corresponding to each comparison result are recorded, and the merging and discrimination are performed according to the tolerance status flags to obtain the deformation suppression capability judgment result.

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