TUNNABLE FILTERS WITH MUTUALLY COUPLED INDUCTIVITIES

Abstract

A tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) with tunable suppression, comprising: a first inductor (L1); a second inductor (L2) mutually coupled to the first inductor (L1), wherein the first inductor (L1) is connected in series with the second inductor (L2); and a tunable capacitor circuit (42; 52; 62; 72; 74; 92) which is electrically connected in parallel to the first inductor (L1) which has a switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) and which is designed to match at least two notches in the frequency response of the tunable filter by changing a switching state of the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N), wherein the tunable filter is designed to filter a high-frequency signal.

Description

CROSS-REFERENCE TO PRIORITY ENTRANCES This application claims priority over US Provisional Patent Application No. 63 / 046,184, filed on June 30, 2020, entitled “TUNABLE FILTER WITH MUTUALLY COUPLED INDUCTORS”, and US Provisional Patent Application No. 63 / 071,261, filed on August 27, 2020, entitled “TUNABLE FILTER WITH MUTUALLY COUPLED INDUCTORS”. BACKGROUND Technical field Embodiments of this disclosure relate to filters designed to filter signals such as high-frequency signals. Description of related technology High-frequency communication systems (HF communication systems) can be used to transmit and / or receive signals over a wide frequency range. For example, an HF communication system can be used to wirelessly transmit HF signals in a frequency range of approximately 30 kilohertz (kHz) to 300 gigahertz (GHz), such as in the range of approximately 410 megahertz (MHz) to approximately 7.125 GHz for fifth-generation (5G) mobile communications in frequency band 1 (FR1). Examples of RF communication systems include, but are not limited to, mobile phones, tablets, base stations, network access points, customer-premises equipment (CPE), laptops and portable electronics. In certain applications, RF communication systems can process a large number of RF signals simultaneously. In such RF communication systems, filters with strong harmonic suppression may be desirable. Filters with strong harmonic suppression are desirable in a wide variety of applications. Document US 2019 / 0 109 575 A1 describes a linear, low-noise, high-quality, and wide-tunable filter circuit with one or more first reactive elements coupled between a first filter node and a first node. Publication US 10 951 190 B2 describes a filter and balancing circuit for an integrated circuit with a first capacitor connected in parallel to a first inductor, a second inductor coupled to the first inductor, and a variable second capacitor coupled between the first and second inductors. Document US 9,455,680 B2 describes RF filter structures that can have multiple filter paths provided by weakly coupled resonators. The filter paths can be interconnected, allowing for the implementation of additional filter paths between the input and output terminals of the RF filter structures. SUMMARY OF CERTAIN INVENTIVE ASPECTS The innovations described in the claims each have several aspects, none of which alone is responsible for its desirable properties. Without limiting the scope of the claims, some distinctive features of this disclosure are now briefly described. One aspect of this revelation is a tunable filter with tunable suppression. The tunable filter comprises a first inductor, a second inductor mutually coupled to the first inductor, and a tunable impedance circuit electrically connected to the first inductor. The tunable impedance circuit includes a switch and is designed to match at least two notches in the frequency response of the tunable filter by changing a switching state of the switch. The tunable filter is designed to filter a high-frequency signal. The switch can be designed to selectively couple one end of a capacitor to the first inductor. The tunable impedance circuit can include a second switch designed to selectively couple one end of a second capacitor to the first inductor. In certain applications, changing the switching state of the switch can adjust the position of at least three notches in the frequency response of the tunable filter. The position of a notch in the frequency response of the tunable filter can be based on at least one state of the tunable impedance circuit and the mutual coupling of the first inductor with the second inductor. The tunable impedance circuit can include a tunable capacitor circuit. This tunable capacitor circuit can provide shunt capacitance at a junction between the first and second inductors. Alternatively, the tunable capacitor circuit can be connected in parallel with the first inductor. The tunable capacitor circuit can also be connected in series with the second inductor. Furthermore, the tunable filter can include a second tunable capacitor circuit connected in parallel with the second inductor. The first inductor can be connected in series with the second inductor, a first capacitor can be connected in parallel with the first inductor, and a second capacitor can be connected in parallel with the second inductor. The tunable filter can also include a shunt capacitor coupled between the first and second inductors. The tunable impedance circuit can be designed to provide the first capacitance. The first inductor can be a series inductor, and the second inductor can be a shunt inductor. The tunable impedance circuit can include a tunable capacitor circuit connected in parallel with the first inductor. The tunable filter can further include an inductor-capacitor circuit in series with the first inductor. The inductor-capacitor circuit can include a third inductor, and the third inductor can be mutually coupled to at least one of the first and second inductors. The at least two notches can provide harmonic suppression. Another aspect of this disclosure relates to a method for filtering high-frequency signals. The method comprises filtering a first high-frequency signal with a tunable filter in a first state; after filtering the first high-frequency signal, changing the switching state of a switch of a tunable impedance circuit of the tunable filter from the first state to a second state in order to adjust the position of at least two harmonic corresponding notches in the frequency response of the tunable filter; and while the tunable filter is in the second state, filtering a second high-frequency signal with the tunable filter. In certain applications, changing the switching state of the switch can adjust the position of at least three notches in the frequency response of the tunable filter. Another aspect of this disclosure relates to a wireless communication device comprising a tunable filter and an antenna designed to transmit a high-frequency signal filtered by the tunable filter. The tunable filter includes a first inductor, a second inductor mutually coupled to the first inductor, and a tunable impedance circuit electrically connected to the first inductor. The tunable impedance circuit includes a switch and is designed to match at least two notches in the frequency response of the tunable filter by changing a switching state of the switch. The wireless communication device may include an antenna switch, and the tunable filter may be coupled between the antenna switch and the antenna. The wireless communication device may include a power amplifier and a band selector switch, and the tunable filter may be coupled between the power amplifier and the band selector switch. Another aspect of this disclosure concerns a tunable filter with harmonic rejection. The tunable filter comprises a first inductor, a second inductor mutually coupled to the first inductor, and a tunable capacitor circuit electrically connected to the first inductor. The tunable capacitor circuit includes N switches designed to adjust the effective capacitance of the tunable capacitor circuit to set the harmonic rejection of the tunable filter to at least 2 × 2N harmonics, where N is a positive integer greater than 1. The tunable filter is designed to filter a high-frequency signal. The N switches can be made adjustable to set the harmonic suppression of the tunable filter to at least 3 × 2N harmonics. The tunable capacitor circuit can be connected in parallel to the first inductor. The tunable filter can include a capacitor connected in parallel to the second inductor and a shunt capacitor connected in series between the first and second inductors. Changing the switching state of the first switch in a plurality of switches can adjust the position of at least two notches in the frequency response of the tunable filter. Changing the switching state of the first switch in a plurality of switches can adjust the position of at least three notches in the frequency response of the tunable filter. The second inductor can be a shunt inductor. The tunable capacitor circuit can be connected in parallel to the first inductor. The tunable filter can also include a shunt capacitor connected in series with the second inductor, which is electrically connected to the first inductor via the second inductor. The 2 × 2N harmonics can contain at least one second harmonic and at least one third harmonic. The 2 × 2N harmonics can contain at least one harmonic assigned to a fifth-generation New Radio operating band. The 2 × 2N harmonics can contain at least one harmonic assigned to a fifth-generation New Radio operating band, as well as at least one harmonic assigned to a fourth-generation Long Term Evolution operating band. Another aspect of this disclosure relates to a wireless communication device comprising a high-frequency front end with a tunable filter and an antenna communicating with the tunable filter. The tunable filter includes a first inductor, a second inductor mutually coupled to the first inductor, and a tunable capacitor circuit electrically connected to the first inductor. The tunable capacitor circuit includes N switches designed to adjust the effective capacitance of the tunable capacitor circuit to set the harmonic rejection of the tunable filter to at least 2 × 2N harmonics, where N is a positive integer greater than 1. The tunable filter is designed to filter a high-frequency signal. The antenna can be designed to filter a high-frequency signal filtered by the tunable filter. N can be at least 4. The high-frequency front end can include an antenna switch, and the tunable filter can be coupled between the antenna switch and the antenna. The high-frequency front end can include a power amplifier and a band selector switch, and the tunable filter can be coupled between the power amplifier and the band selector switch. The wireless communication device can be configured to implement dual-network operation, and the tunable filter can be configured to provide suppression during dual-network operation. The wireless communication device can also be configured to implement carrier bonding, and the tunable filter can be configured to provide suppression during carrier bonding. Another aspect of this disclosure relates to a high-frequency system comprising an antenna switch, an antenna connector, and a tunable filter coupled in a signal path between the antenna switch and the antenna connector. The tunable filter includes a first inductor, a second inductor mutually coupled to the first inductor, and a tunable capacitor circuit electrically connected to the first inductor. The tunable capacitor circuit includes N switches designed to adjust the effective capacitance of the tunable capacitor circuit to set the harmonic rejection of the tunable filter to at least 2 × 2N harmonics, where N is a positive integer greater than 1. The tunable filter is designed to filter a high-frequency signal. For the purpose of summarizing the disclosure, certain aspects, advantages, and novel features of the innovations are described herein. It should be clear that not all of these advantages can necessarily be achieved according to a particular embodiment. Therefore, the innovations may be implemented or designed in a way that achieves or optimizes one advantage or group of advantages as taught or indicated herein, without necessarily achieving other advantages taught or indicated herein. BRIEF DESCRIPTION OF THE DRAWINGS Embodiments of this disclosure are now described by means of non-limiting examples and with reference to the accompanying drawings. Fig. 1 shows a schematic representation of an example of a communication network. Fig. 2A shows a schematic block diagram of an example of a communication link that uses carrier bonding. Fig. 2B illustrates various examples of uplink carrier bonding in the communication link of Fig. 2A. Fig. 2C illustrates various examples of downlink carrier bonding in the communication link of Fig. 2A. Fig. 3 shows a diagram of an exemplary network topology with dual-network capability. Fig. 4 shows a schematic representation of a tunable filter according to one embodiment. Fig. 5 shows a schematic representation of a tunable filter according to another embodiment.Figure 6 shows a schematic representation of a tunable filter according to another embodiment. Figure 7 shows a schematic representation of a tunable filter according to another embodiment. Figure 8A shows a schematic representation of a tunable filter according to another embodiment. Figure 8B shows a schematic representation of a tunable filter according to another embodiment. Figure 8C shows a plot of the frequency response of the filter of Figure 8B. Figure 9 shows a schematic representation of a tunable filter according to one embodiment. Figure 10 shows a plot of the coupling coefficient as a function of the distance between the two mutually coupled inductors. Figure 11 shows a plot of a simulation illustrating the positions of the harmonic notches of the tunable filter of Figure 9 according to a first design example.Figure 12 shows a simulation plot illustrating the positions of the harmonic notches of the tunable filter in Figure 9 according to a second design example. Figure 13 shows a schematic representation of a tunable filter according to one embodiment. Figure 14 shows a simulation plot illustrating the positions of the harmonic notches of the tunable filter in Figure 13 according to a third design example. Figure 15A shows a schematic block diagram of a high-frequency system with a tunable filter according to one embodiment. Figure 15B shows a schematic block diagram of a high-frequency system with a tunable filter according to another embodiment. Figure 16 shows a schematic block diagram of an embodiment of a mobile device. DETAILED DESCRIPTION OF CERTAIN VERSIONS The following detailed description of certain embodiments presents various descriptions of specific embodiments. However, the innovations described here can be implemented in a variety of ways, for example, through the definition and scope of the claims. Reference is made in this description to the drawings, in which similar reference numerals may denote identical or functionally similar elements. It should be noted that the elements shown in the figures are not necessarily to scale. Furthermore, it should be noted that certain embodiments may include more elements than are shown in a drawing and / or a subset of the elements shown in a drawing. In addition, some embodiments may include any suitable combination of features from two or more drawings.The headings included here are for clarity only and do not necessarily affect the scope or significance of the claims. The International Telecommunication Union (ITU) is a specialized agency of the United Nations (UN) responsible for global issues relating to information and communication technologies, including the shared global use of the frequency spectrum. The 3rd Generation Partnership Project (3GPP) is a collaboration between groups of telecommunications standardisation bodies around the world, such as the Association of Radio Industries and Businesses (ARIB), the Telecommunications Technology Committee (TTC), the China Communications Standards Association (CCSA), the Alliance for Telecommunications Industry Solutions (ATIS), the Telecommunications Technology Association (TTA), the European Telecommunications Standards Institute (ETSI) and the Telecommunications Standards Development Society India (TSDSI). Within the framework of the ITU, 3GPP develops and maintains technical specifications for a wide range of mobile communication technologies, including, for example, second generation (2G) technology (e.g., Global System for Mobile Communications (GSM) and Enhanced Data Rates for GSM Evolution (EDGE)), third generation (3G) technology (e.g., Universal Mobile Telecommunications System (UMTS) and High Speed ​​Packet Access (HSPA)), and fourth generation (4G) technology (e.g., Long Term Evolution (LTE) and LTE-Advanced). The technical specifications controlled by 3GPP can be extended and revised through specification versions that span several years and can specify a variety of new features and developments. In one example, 3GPP introduced carrier aggregation (CA) for LTE in Release 10. Although initially launched with two downlink carriers, 3GPP expanded carrier aggregation in Release 14 to up to five downlink carriers and up to three uplink carriers. Other examples of new features and developments provided by 3GPP releases include License Assisted Access (LAA), Enhanced LAA (eLAA), Narrowband Internet of Things (NB-IoT), Vehicle-to-Everything (V2X), and High Power User Equipment (HPUE). 3GPP introduced Phase 1 of its fifth-generation (5G) technology in Release 15 and is currently developing Phase 2 of 5G technology in Release 16. Subsequent 3GPP releases will further develop and expand 5G technology. 5G technology is also referred to here as 5G New Radio (NR). 5G NR supports, or plans to support, a variety of features, such as millimeter-wave spectrum communication, beamformability, high spectral efficiency waveforms, low-latency communication, multiple radio numerology, and / or non-orthogonal multiple access (NOMA). While such RF functionalities offer network flexibility and increase user data rates, supporting them can present a number of technical challenges. The teachings contained herein apply to a wide variety of communication systems, including but not limited to communication systems that use advanced mobile communication technologies such as LTE-Advanced, LTE-Advanced Pro and / or 5G NR. Harmonic suppression In fifth-generation (5G) New Radio (NR) technology, harmonic suppression requirements can become increasingly challenging. With denser integration, signals can be generated simultaneously across more and more bands. Greater filter tunability to meet diverse 5G NR harmonic requirements is advantageous in a wide range of applications. Examples include carrier bundling, dual-band capability, coexistence of a 5G NR FR1 signal with a 5G NR FR2 signal, and other coexistence applications. Some solutions rely on harmonic suppression switches. In such solutions, one switch can control two harmonic states, usually by switching between two bands. Using this technique, N switches can control up to 2N harmonic states. When many bands are being served, a relatively large number of switches can be used to provide the desired harmonic suppression. This relatively large number of switches can be expensive to implement. Aspects of this disclosure relate to filters comprising mutually coupled inductors and a tunable capacitor circuit. The tunable capacitor circuit may include switches designed to selectively couple corresponding capacitors electrically to adjust the capacitance provided by the tunable capacitor circuit. When using N switches together with mutually coupled inductors, a filter can tune M × 2N harmonic states, where N is a positive integer greater than or equal to 2, and M is the number of harmonics of interest per band. In certain applications, M may be 1 greater than the number of mutual couplings of inductor pairs in a tunable filter. The filters disclosed herein can advantageously utilize inductive mutual coupling to increase the number of adjustable harmonic states. Compared to previous designs, the filters disclosed herein can reduce the space required and / or the cost of switches while maintaining the same number of adjustable harmonic states. Filters disclosed herein can filter any suitable harmonic. For example, filters disclosed herein can filter one or more of the following harmonics: second harmonic, third harmonic, fourth harmonic, fifth harmonic, sixth harmonic, etc. Such harmonics can be suppressed simultaneously by a filter designed according to the principles and advantages described herein. While embodiments related to harmonic suppression are discussed, all suitable principles and advantages of the filters described herein can also be used to provide any suitable out-of-band suppression and / or notch filtering effect. Communication network Fig. 1 is a schematic diagram of an example of a communication network 10. The communication network 10 includes a macrocell base station 1, a mobile device 2, a small cell base station 3 and a stationary wireless device 4. The communication network 10 shown in Fig. 1 supports communication using a variety of mobile communication technologies, including, for example, 4G LTE, 5G NR, and wireless local area networks (WLAN), such as WiFi. Dual networking capability with simultaneous 4G LTE and 5G NR communication with the mobile device 2 can be implemented in the communication network 10. Although various communication technology examples have been given, the communication network 20 can be adapted to support a wide variety of communication technologies. Figure 1 shows various communication links of communication network 10. These links can be duplexed in various ways, for example, by frequency division multiplexing (FDD) and / or time-division multiplexing (TDD). FDD is a type of high-frequency communication that uses different frequencies for transmitting and receiving signals. FDD can offer several advantages, such as high data rates and low latency. In contrast, TDD is a type of high-frequency communication that uses approximately the same frequency for transmitting and receiving signals, but with the transmission and reception being staggered. TDD can offer several advantages, such as efficient spectrum utilization and variable throughput allocation between transmit and receive directions. As shown in Fig. 2, the mobile device 2 communicates with the macrocell base station 1 via a communication link that uses a combination of 4G LTE and 5G NR technology. The mobile device 2 also communicates with the small cell base station 3. In the example shown, the mobile device 2 and the small cell base station 3 communicate via a communication link that uses 4G LTE, 5G NR, and Wi-Fi technology. In certain implementations, Enhanced License Assisted Access (eLAA) is used to combine one or more licensed frequency carriers (e.g., licensed 4G LTE and / or 5G NR frequencies) with one or more unlicensed carriers (e.g., unlicensed Wi-Fi frequencies). In certain implementations, the mobile device 2 communicates with the macrocell base station 1 and the smallcell base station 3 using 5G NR technology over one or more frequency bands within Frequency Range 1 (FR1) and / or over one or more frequency bands above Frequency Range 1 (FR1). These one or more frequency bands may be below 6 GHz. For example, wireless communication may use FR1, Frequency Range 2 (FR2), or a combination thereof. In one embodiment, the mobile device 2 supports an HPUE performance class specification. The small cell base station 3 shown also communicates with a stationary wireless device 4. The small cell base station 3 can be used, for example, to provide broadband operation using 5G NR technology. In certain implementations, the small cell base station 3 communicates with the stationary wireless device 4 via one or more millimeter wave frequency bands in the frequency range between 30 GHz and 300 GHz and / or upper centimeter wave frequency bands in the frequency range between 24 GHz and 30 GHz. In certain implementations, the small cell base station 3 communicates with the small cell base station 3 using beamforming. For example, beamforming can be used to focus the signal strength to overcome path losses, such as the high losses associated with communication over millimeter wave frequencies. The communication network 10 of Fig. 1 comprises the macrocell base station 1 and the smallcell base station 3. In certain implementations, the smallcell base station 3 can be operated with relatively lower power, shorter range, and / or fewer concurrent users compared to the macrocell base station 1. The smallcell base station 3 can also be referred to as a femtocell, picocell, or microcell. Although the communication network 10 is depicted as including two base stations, it can be implemented to include more or fewer base stations and / or base stations of other types. As shown in Fig. 1, base stations can communicate with each other wirelessly to provide a wireless return path. Additionally or alternatively, base stations can communicate with each other using wired and / or optical links. The communication network 10 of Fig. 1 is depicted as comprising a mobile device and a stationary wireless device. The mobile device 2 and the stationary wireless device illustrate two examples of user equipment (UE). Although the communication network 10 is depicted as comprising two user equipment, it can be used to communicate with more or fewer user equipment and / or other types of user equipment. For example, user equipment can include mobile phones, tablets, laptops, Internet of Things (IoT) devices, wearable electronics, and / or a wide variety of other communication devices. User terminal devices of the communication network 10 can utilize available network resources (e.g., available frequency spectra) in a variety of ways. In one example, Frequency Division Multiple Access (FDMA) is used to divide a frequency band into multiple carriers. Additionally, one or more carriers are assigned to a specific user. Examples of FDMA include Single Carrier FDMA (SC-FDMA) and Orthogonal FDMA (OFDMA). OFDMA is a multi-carrier technology that divides the available bandwidth into several mutually orthogonal narrowband subcarriers, which can be assigned separately to different users. Other examples of shared access include, but are not limited to, time-division multiple access (TDMA), where a user is assigned specific time slots to use a frequency resource; code-division multiple access (CDMA), where a frequency resource is shared by multiple users by assigning each user a unique code; space-division multiple access (SDMA), where beamforming is used to provide shared access through spatial division; and non-orthogonal multiple access (NOMA), where the power domain is used for multiple access. For example, NOMA can be used to serve multiple users with the same frequency, time, and / or code, but with different power levels. Enhanced Mobile Broadband (eMBB) refers to a technology for increasing the system capacity of LTE networks. For example, eMBB can refer to communications with a maximum data rate of at least 10 Gbps and a minimum of 100 Mbps for each user. Highly Reliable Low Latency Communication (uRLLC) refers to technologies for communication with very low latency, e.g., less than 2 milliseconds. uRLLC can be used for mission-critical communications, such as for autonomous driving and / or remote surgery applications. Massive Machine Communication (mMTC) refers to cost-effective, low-data-rate communications associated with wireless connections to everyday objects, such as those used in Internet of Things (IoT) applications. The communication network 10 of Fig. 1 can be used to support a variety of advanced communication functions, including, but not limited to, eMBB, uRLLC and / or mMTC. The peak data rate of a communication link (for example, between a base station and a user terminal) depends on a variety of factors. For example, the peak data rate can be affected by the channel bandwidth, modulation sequence, number of component carriers, and / or the number of antennas used for communication. In certain implementation forms, the data rate of a communication link can be, for example, M*B*log2(1+S / N), where M is the number of communication channels, B is the channel bandwidth, and S / N is the signal-to-noise ratio (SNR). Accordingly, the data rate of a communication link can be increased by increasing the number of communication channels (for example, by transmitting and receiving using multiple antennas), using a larger bandwidth (for example, by carrier bundling), and / or improving the SNR (for example, by increasing the transmission power and / or improving the receiver sensitivity). 5G NR communication systems can employ a wide variety of techniques to increase data rate and / or communication performance. Carrier bundling Figure 13A is a schematic diagram of an example of a carrier-bonded communication link. Carrier bonding can be used to extend the bandwidth of the communication link by supporting communications over multiple frequency carriers, thereby increasing user data rates and improving network capacity through the use of fragmented frequency allocations. Carrier bonding can present harmonic suppression challenges. Filters disclosed herein can be configured to provide harmonic suppression in carrier-bonded applications. High-frequency front-end architectures, as disclosed herein, can be used in carrier-bonded applications. In the illustrated example, a communication link is provided between a base station 21 and a mobile device 22. As shown in Fig. 2A, the communication link includes a downlink channel, which is used for RF communication from the base station 21 to the mobile device 22, and an uplink channel, which is used for RF communication from the mobile device 22 to the base station 21. Although Fig. 2A illustrates carrier bundling in the context of FDD communication, carrier bundling can also be used for TDD communication. In certain implementations, a communication link can provide asymmetric data rates for a downlink channel and an uplink channel. For example, a communication link can be used to support a relatively high downlink data rate to enable high-speed streaming of multimedia content to a mobile device, while a relatively slower data rate is provided for uploading data from the mobile device to the cloud. In the example shown, the base station 21 and the mobile device 22 communicate using carrier aggregation, which allows the bandwidth of the communication link to be selectively increased. Carrier aggregation includes contiguous aggregation (CA), in which contiguous carriers within the same operating frequency band are aggregated. Carrier aggregation can also be non-contiguous and include carriers that are separated in their frequency within a common band or in different bands. In the example shown in Fig. 2A, the uplink channel contains three aggregated component carriers fUL1, fUL2, and fUL3. Additionally, the downlink channel contains five aggregated component carriers fDL1, fDL2, fDL3, fDL4, and fDL5. Although this is an example of component carrier bundling or aggregation, more or fewer carriers can be aggregated for the uplink and / or downlink. Furthermore, the number of aggregated carriers can be varied over time to achieve the desired uplink and downlink data rates. For example, the number of aggregated carriers for uplink and / or downlink communication with respect to a particular mobile device can change over time. This could be as the device moves across the communication network and / or as network usage changes over time. Fig. 2B illustrates various examples of uplink carrier bundling for the communication link of Fig. 2A. Fig. 2B includes a first carrier bundling scenario 31, a second carrier bundling scenario 32 and a third carrier bundling scenario 33, which schematically represent three types of carrier bundling. Carrier bundling scenarios 31 to 33 illustrate different spectrum assignments for a first component carrier fUL1, a second component carrier fUL2, and a third component carrier fUL3. Although Fig. 2B is shown in the context of aggregating three component carriers, carrier bundling can be used to aggregate more or fewer carriers. Furthermore, although illustrated in the context of the uplink, the bundling scenarios are also applicable to the downlink. The first carrier bundling scenario 31 illustrates intraband coherent carrier bundling, in which component carriers that are adjacent in frequency and in a common frequency band are aggregated. For example, the first carrier bundling scenario 31 represents the aggregation of component carriers fUL1, fUL2, and fUL3, which are coherent and located within a first frequency band BAND1. With further reference to Fig. 2B, the second carrier bundling scenario 32 illustrates intraband non-continuous carrier bundling, in which two or more component carriers that are not adjacent in their frequency and within a common frequency band are aggregated. For example, the second carrier bundling scenario 32 represents the bundling of component carriers fUL1, fUL2, and fUL3, which are not contiguous but are located within a first frequency band BAND1. The third carrier bundling scenario 33 illustrates interband, non-contiguous carrier bundling, in which component carriers that are not adjacent in their frequency and in several frequency bands are combined. For example, the third carrier bundling scenario 33 represents the aggregation of the component carriers fUL1 and fUL2 of a first frequency band BAND1 with the component carrier fUL3 of a second frequency band BAND2. Figure 2C illustrates various examples of downlink carrier aggregation for the communication link of Figure 2A. The examples show different carrier aggregation scenarios 34 to 38 for different spectrum assignments of a first component carrier fDL1, a second component carrier fDL2, a third component carrier fDL3, a fourth component carrier fDL4, and a fifth component carrier fDL5. Although Figure 2C is shown in the context of aggregating five component carriers, carrier aggregation can be used to aggregate more or fewer carriers. Furthermore, although illustrated in the context of the downlink, the aggregation scenarios are also applicable to uplinks. The first carrier bundling scenario 34 represents the bundling of component carriers that are contiguous and located in the same frequency band. Additionally, the second carrier bundling scenario 35 and the third carrier bundling scenario 36 illustrate two examples of bundling that is not contiguous but is located in the same frequency band. Furthermore, the fourth carrier bundling scenario 37 and the fifth carrier bundling scenario 38 illustrate two examples of bundling in which component carriers that are not adjacent in their frequency and in several frequency bands are bundled. As the number of aggregated or bundled component carriers increases, so does the complexity of possible carrier bundling scenarios. Referring to Figures 2A to 2C, the individual component carriers used in carrier bonding can consist of a variety of frequencies, including, for example, frequency carriers in the same band or in multiple bands. Furthermore, carrier bonding is applicable both to implementations where the individual component carriers have approximately the same bandwidth and to implementations where the individual component carriers have different bandwidths. Certain communication networks assign a primary component carrier (PCC) or anchor carrier for uplink and a PCC for downlink to a specific user device. Additionally, if the mobile device communicates over a single frequency carrier for either uplink or downlink, the user device communicates via the PCC. To increase the bandwidth for uplink communication, the uplink PCC can be aggregated with one or more uplink secondary component carriers (SCCs). To increase the bandwidth for downlink communication, the downlink PCC can be aggregated with one or more downlink SCCs. In certain implementations, a communication network provides a network cell for each component carrier. Additionally, a primary cell can be operated with a PCC, while a secondary cell can be operated with an SCC. The primary and secondary cells can have different coverage areas, for example, due to different carrier frequencies and / or the network environment. License Assisted Access (LAA) refers to the aggregation of downlink carriers, where a licensed frequency carrier allocated to a mobile network operator is aggregated with a frequency carrier in unlicensed spectrum, such as Wi-Fi. LAA uses a downlink PCC in the licensed spectrum, which carries control and signaling information associated with the communication link, while unlicensed spectrum is aggregated for greater downlink bandwidth when available. LAA can operate with dynamic adjustment of secondary carriers to avoid and / or coexist with Wi-Fi users. Enhanced License Assisted Access (eLAA) refers to an evolution of LAA that aggregates licensed and unlicensed spectrum for both downlink and uplink. Dual network capability With the introduction of the 5G NR air interface standards, 3GPP enabled the simultaneous operation of 5G and 4G standards to facilitate the transition. This mode of operation can be referred to as "Non-Stand-Alone" (NSA) 5G mode or E-UTRAN New Radio Dual Networking (EN-DC) capability and involves the simultaneous transmission of both 4G and 5G carriers from a single user installation (UE). EN-DC can present harmonic suppression challenges. Filters disclosed herein can be configured to provide harmonic suppression in dual networking applications. Radio frequency front-end architectures, such as those disclosed herein, can be used in dual networking applications. In certain EN-DC applications, dual-network capability NSA involves overlaying 5G systems on top of an existing 4G core network. For dual-network capability in such applications, control and synchronization between the base station and the UE can be performed via the 4G network, while the 5G network is a complementary radio access network anchored within the 4G network. The 4G anchor can connect to the existing 4G network while simultaneously overlaying the 5G data and / or 5G control. Figure 3 shows a diagram of an exemplary network topology with dual-network capability. This architecture can leverage legacy LTE coverage to ensure continuity of link performance while 5G cells are gradually rolled out. A UE 30 can simultaneously transmit dual uplink carriers over LTE and NR. The UE 30 can transmit an LTE uplink carrier Tx1 to the eNodeB (eNB) 39a while simultaneously transmitting an LTE uplink carrier Tx2 to the gNodeB (gNB) 39b, thus achieving dual-network capability. Any suitable combination of uplink carriers Tx1, Tx2 and / or downlink carriers Rx1, Rx2 can be transmitted simultaneously over wireless links in the exemplary network topology shown in Figure 3. The eNB 39a can provide connectivity to a core network, such as an Evolved Packet Core (EPC). The gNB 39b can communicate with the core network via the eNB 39a.Control level data can be communicated wirelessly between the UE 30 and the eNB 39a. The eNB 39a can also exchange control level data with the gNB 39b. In the exemplary dual-network network topology shown in Fig. 3, all suitable combinations of standardized bands and radio access technologies (e.g., FDD, TDD, SUL, SDL) can be transmitted and received wirelessly. This can lead to technical challenges related to the many different radio channels and bands handled in UE 30. With a TDD-LTE anchor point, network operation can be synchronous, with operating modes limited to Tx1 / Tx2 and Rx1 / Rx2, or asynchronous, encompassing Tx1 / Tx2, Tx1 / Rx2, Rx1 / Tx2, and Rx1 / Rx2. If the LTE anchor is a frequency-shared duplex carrier (FDD carrier), TDD / FDD interband operation can simultaneously include Tx1 / Rx1 / Tx2 and Tx1 / Rx1 / Rx2. Tunable filters with harmonic suppression Harmonic suppression can be implemented in a variety of tunable filters. Exemplary embodiments of tunable filters are discussed in connection with Figures 4, 5, 6, 7, 8, 9, 10, 11, 12, 13 to 14. Any suitable combination of the features of these exemplary tunable filters can be implemented together. In 5G NR applications and others, meeting harmonic suppression requirements can be a challenge. Ensuring filters possess additional tunability can help meet these requirements. Embodiments disclosed herein relate to filters with at least one tunable impedance circuit and mutually coupled inductors to provide tunability across a variety of harmonics. Filters with mutually coupled inductors and at least one tunable impedance circuit with one or more switches can provide harmonic suppression across more harmonics than similar filters without mutually coupled inductors.This can lead to tunability for a greater number of harmonics with respect to the number of switches arranged in a tunable impedance circuit, compared to previous designs without mutually coupled inductors. In embodiments of the filters disclosed herein, two or three times as many harmonic states can be achieved in a filter due to the mutually coupled inductors compared to similar filters without mutual coupling. The principles and advantages disclosed herein can apply to a larger multiple of the number of harmonics relative to the number of switches in a tunable impedance circuit. The principles and advantages disclosed herein are not limited to harmonic suppression but can be applied to any other out-of-band suppression and / or notch filtering, as desired. Fig. 4 shows a schematic representation of a tunable filter 40 according to one embodiment. The tunable filter 40 is designed to filter a high-frequency signal propagating between a first terminal P1 and a second terminal P2. The tunable filter 40 can be a low-pass filter. As shown, the tunable filter 40 comprises a tunable impedance circuit 42, inductors, and capacitors. The inductance values ​​of the inductors and the capacitance values ​​of the capacitors can set a frequency response of the tunable filter 40, including the setting of which frequencies are passed and where notches appear in a frequency response for harmonic suppression. The inductors of the tunable filter 40 comprise a first inductor L1 and a second inductor L2. The first inductor L1 and the second inductor L2 are connected in series. The first inductor L1 and the second inductor L2 are mutually coupled. Mutual coupling (or negative feedback) can also be referred to as magnetic coupling or mutual inductive coupling. The first inductor L1 and the second inductor L2 have a coupling coefficient K. The inductors of the tunable filter 40 and / or one or more tunable filters described herein can incorporate any type of suitable inductor, such as surface-mount technology (SMT) inductors, one or more coils embedded on and / or in a substrate (e.g., a laminate substrate), or one or more inductors on a bare chip (e.g., a microcontroller).one or more inductors on the same raw chip as a switch in a tunable impedance circuit), one or more integrated passive inductors (“integrated passive devices”, IPD) or the like, or any suitable combination thereof. The capacitors of the tunable filter 40 comprise capacitors C11, C12, up to C1N of the tunable impedance circuit 42, as well as capacitors C2 and C3. The capacitors of the tunable filter 40 and / or one or more tunable filters described herein may include any type of suitable capacitors, such as surface-mount technology (SMT) capacitors, one or more capacitors on a bare chip (e.g., one or more inductors on the same bare chip as a switch in a tunable capacitor circuit), one or more integrated passive devices (IPDs), or the like, or any suitable combination thereof.In the tunable filter 40, the tunable impedance circuit 42 is a tunable capacitor circuit. The tunable capacitor circuit 42 is connected in parallel with the first inductor L1 in the tunable filter 40. The tunable capacitor circuit 42 is also connected in series with the second inductor L2 in the tunable filter 40. The tunable capacitor circuit 42 comprises a plurality of capacitors C11, C12, ..., C1N, each of which is connected in series with a corresponding switch S11, S12, ..., S1N. Each switch S11, S12, ..., S1N can selectively couple one end of a corresponding capacitor C11, C12, ..., C1N electrically to the first inductor L1. Accordingly, each switch S11, S12, ..., S1N can selectively switch a corresponding capacitor C11, C12, ..., C1N in parallel to the first inductor L1.The capacitor(s) connected electrically in parallel to the first inductor L1 provide an effective capacitance value for a specific state of the tunable capacitor circuit 42. Although embodiments disclosed herein may include tunable capacitor circuits, all the principles and advantages described herein can be applied to tunable impedance circuits. Such tunable impedance circuits may include tunable inductor circuits. Any suitable inductance of a filter comprising inductors and capacitors may be implemented by a tunable inductor circuit to enable harmonic suppression. Some tunable harmonic suppression filters may include one or more tunable capacitor circuits and one or more tunable inductor circuits. The tunable filter 40 is designed to provide harmonic rejection. The positions of harmonic notches in the frequency response of the tunable filter can be adjusted based on the state of the tunable impedance circuit 42. For example, closing a switch S11 can couple a capacitor C11 in parallel with the first inductor L1. This changes the effective capacitance of the tunable impedance circuit 42 and the positions of notches with respect to the frequency of the tunable filter 40 for harmonic rejection. With the mutually coupled inductors L1 and L2, switching a switch of the tunable impedance circuit 42 can adjust more positions of notches in the frequency response than a similar filter without mutually coupled inductors. The tunable filter 40 shown comprises a T-network. The T-network includes a first circuit with parallel inductors and capacitors, comprising the first inductor L1 and the tunable impedance circuit 42, a shunt capacitor C3, and a second circuit with parallel inductors and capacitors, comprising the second inductor L2 and the capacitor C2. Any suitable principles and advantages described herein may be applied to other suitable filter arrangements. Fig. 5 shows a schematic representation of a tunable filter 50 according to one embodiment. The tunable filter 50 corresponds to the tunable filter 40, except that an additional circuit with parallel inductors and capacitors and an additional shunt capacitor are provided. The additional circuit with parallel inductors and capacitors comprises a third inductor L3 and a capacitor C4. The additional shunt capacitor C5 is coupled between the additional circuit with parallel inductors and capacitors and the first inductor L1. As shown in Fig. 5, the first inductor L1 is reciprocally coupled to both the second inductor L2 and the third inductor L3. The first inductor L1 and the second inductor L2 have a coupling coefficient K12. The first inductor L1 and the third inductor L3 have a coupling coefficient K31.Changing the switching state of a switch of the tunable impedance circuit 42 in the tunable filter 50 can adjust the positions of three corresponding harmonics associated with notches in the frequency response of the tunable filter 50 through the mutual coupling of the tunable filter 50. Fig. 6 shows a schematic representation of a tunable filter 60 according to one embodiment. The tunable filter 60 corresponds to the tunable filter 50 of Fig. 5, except that each of the circuits with parallel inductors and capacitors has a corresponding tunable impedance circuit, and the inductors are mutually coupled. Although Fig. 6 shows three circuits with parallel inductors and capacitors, any suitable number of circuits with parallel inductors and capacitors can be provided. As shown in Fig. 6, the tunable impedance circuits 42, 52 and 62 are each arranged in parallel with a corresponding inductor L1, L2, L3. These tunable impedance circuits 42, 52 and 62 can each match an effective capacitance value in parallel with a corresponding inductor L1, L2, L3. The tunable impedance circuit 52 is connected in parallel to the second inductor L2 in the tunable filter 60. The tunable impedance circuit 52 comprises a plurality of capacitors C21, C22, ..., C2N, each arranged in series with a corresponding switch S21, S22, ..., S2N. Each switch S21, S22, ..., S2N can selectively couple a corresponding capacitor C21, C22, ..., C2N electrically to the second inductor L2. Accordingly, each switch S21, S22, ..., S2N can selectively connect a corresponding capacitor C21, C22, ..., C2N in parallel with the second inductor L2. The tunable impedance circuit 62 is connected in parallel to the third inductor L3 in the tunable filter 60. The tunable impedance circuit 62 comprises a plurality of capacitors C41, C42, ..., C4N, each arranged in series with a corresponding switch S41, S42, ..., S4N. Each switch S41, S42, ..., S4N can selectively couple a corresponding capacitor C41, C42, ..., C4N electrically to the second inductor L2. Similarly, each switch S21, S22, ..., S2N can selectively connect a corresponding capacitor C41, C42, ..., C4N in parallel with the third inductor L3. In the tunable filter 60, the first inductor L1 and the second inductor L2 have a coupling coefficient K12, the second inductor L2 and the third inductor L3 have a coupling coefficient K23, and the first inductor L1 and the third inductor L3 have a coupling coefficient K13. Because the inductors L1 and L3 are mutually coupled, the tunable filter 60 provides additional mutual coupling compared to the tunable filter 50. Fig. 7 shows a schematic representation of a tunable filter 70 according to one embodiment. The tunable filter 70 corresponds to the tunable filter 60 of Fig. 6, except that the shunt capacitors C3 and C5 are formed by tunable capacitor circuits 72 and 74, respectively. Fig. 7 illustrates the principles and advantages disclosed herein when applied to higher-order filters. The tunable impedance circuit 72 comprises a plurality of capacitors C31, C32, ..., C3N, each arranged in series with a corresponding switch S31, S32, ..., S3N. Each switch S31, S32, ..., S3N can selectively couple a corresponding capacitor C31, C32, ..., C3N electrically to a node between the first inductor L1 and the second inductor L2. Accordingly, each switch S31, S32, ..., S3N can selectively switch a corresponding capacitor C31, C32, ..., C3N electrically as a shunt capacitor. The tunable impedance circuit 74 comprises a plurality of capacitors C51, C52, ..., C5N, each arranged in series with a corresponding switch S51, S52, ..., S5N. Each switch S51, S52, ..., S5N can selectively couple a corresponding capacitor C51, C52, ..., C5N electrically to a node between the second inductor L2 and the third inductor L3. Similarly, each switch S51, S52, ..., S5N can selectively connect a corresponding capacitor C51, C52, ..., C5N electrically between the node and ground. Figure 7 illustrates that series and / or shunt capacitors can be formed by tunable impedance circuits. In certain applications, one or more of the capacitors of a filter can be formed by one or more tunable capacitor circuits, while one or more of the filter's capacitors can be formed by non-adjustable capacitors with a fixed capacitance value. Any suitable capacitor of a tunable filter can be formed by a tunable capacitor circuit designed according to the principles and advantages described herein. Additionally or alternatively, any suitable inductance of a tunable filter can be formed by a tunable inductor circuit designed according to the principles and advantages described herein.Adjusting capacitance and / or inductance values ​​through a tunable impedance circuit can adjust the harmonic rejection of a tunable filter. In certain applications, inductors can be placed in series with shunt capacitors to construct one or more harmonic traps. Such a shunt inductor can be coupled reciprocally with one or more series inductors. Fig. 8A illustrates an example of a shunt inductor coupled in series with a shunt capacitor, with the shunt inductor being reciprocally coupled with the series inductors. Fig. 8A shows a schematic representation of a tunable filter 80 according to one embodiment. The tunable filter 80 corresponds to the tunable filter 40 of Fig. 4, except that a shunt inductor LSin is connected in series with the shunt capacitor C3. The shunt inductor LSin is coupled to the first inductor L1 and the second inductor L2. The first inductor L1 and the shunt inductor LSin have a coupling coefficient K1S. The second inductor L2 and the shunt inductor LSin have a coupling coefficient K2S. Fig. 8B shows a schematic representation of a tunable filter 80' according to one embodiment. The tunable filter 80' corresponds to the tunable filter 80 of Fig. 8A, except that the second inductor L2 and the second capacitor C2 are not included. The tunable filter 80' can be considered equivalent to the tunable filter 80 in which the second inductor L2 has zero impedance and the second capacitor C2 has zero impedance. In the tunable filter 80, the second inductor L2 and the second capacitor C2 each have a non-zero impedance. Fig. 8C shows a plot of the frequency response of the tunable filter 80' of Fig. 8B. The plot corresponds to the tunable filter 80', which has a tunable capacitor circuit with four switches S11, S12, S13, and S14, each in series with a corresponding capacitor C11, C12, C13, and C14. As an example, the tunable filter 80' with the frequency response shown in Fig. 8C can provide the following harmonic suppression: second harmonic of band 8, third harmonic of band 8, third harmonic of band 12, second harmonic of band 13 / band 14, third harmonic of band 20 / band 26, second harmonic of band 28, third harmonic of band 28, and third harmonic of band 71. The tunable filter 80', which corresponds to the one in Fig.The frequency response shown in 8C is designed to shift two notches in opposite directions when the effective capacitance value C1 of the tunable capacitor circuit 42 is switched to meet the harmonic specification. Fig. 9 shows a schematic representation of a tunable filter 90 according to one embodiment. The tunable filter 90 corresponds to the tunable filter 40 of Fig. 4, except that the tunable filter 90 has a tunable impedance circuit 92 instead of the tunable impedance circuit 42. The tunable impedance circuit 92 provides an effective capacitance value C1. The tunable impedance circuit 92 can comprise any suitable tunable impedance circuit designed to match a capacitor coupled in parallel to the first inductor L1. As an example, the tunable impedance circuit 92 can be the tunable impedance circuit 42. As another example, the tunable impedance circuit 92 can be a tunable impedance circuit that includes a varactor whose capacitance value changes depending on the applied voltage.Exemplary technical explanations and embodiments will be explained in connection with Fig. 9. Without adhering to any particular theory, a theoretical explanation of tunable filter 90 is presented. For an intraband design, ωc is the central angular frequency. Neglecting the capacitance values ​​of the first capacitor C1 and the second capacitor C2, assuming that Z0 = 50 Ω for both terminals P1 and P2 of tunable filter 90, and assuming that the inductance values ​​of the first and second inductors L1 and L2 are both L, the capacitance value C3 of the third capacitor C3 corresponds to a simultaneous conjugate match according to Equation 1. Harmonic notches ω can be found according to equation 2. By selecting an appropriate coupling coefficient K for the first and second inductors L1 and L2, the capacitance value of the first capacitor C1 (and / or the second capacitor C2 and / or the third capacitor C3) can be adjusted so that both harmonic notches are tuned to desired frequency values. Additionally or alternatively, the inductance value of the first inductor L1 and / or the inductance value of the second inductor L2 can be adjusted so that both harmonic notches are tuned to desired frequency values. The inductor-capacitor tanks of the tunable filter 90 can also be designed to operate at any other resonant frequency, so that the notches can therefore be at frequency values ​​that are not multiples of the fundamental frequency. We define σ according to equation 3, where When δ = 0, the two notches are infinitely far apart. When δ = 1, the two notches overlap. The function σ(δ) is a monotonically increasing continuous function within the same domain and region of [0, 1]. From the equations for ω± it follows that: Assuming we have fixed capacitance values ​​for the second and third capacitors C2 and C3, as well as fixed inductance values ​​for the first and second inductors L1 and L2, then the capacitance value of the first capacitor C1 and the mutual coupling M between the first and second inductors L1 and L2 remain as system variables. If the capacitance value of the first capacitor C1 = 0, we have σ = δ = 0 from equation 4. Finding a solution for σ = 1 yields: This can be rewritten as: Assuming M2<< L1L2 for a second-order approximation, we can state the discriminant of equation 6: This implies that by appropriately choosing M, it can be designed to take on any value between 0 and 1. Accordingly, notches can theoretically be located at any position relative to each other in the frequency domain. Assuming that σ in equation 3 is constant for each capacitance value of the second capacitor C2, the mutual coupling M and the capacitance value of the first capacitor C1 can be determined. From equation 4 we obtain: Equation 8 can be rewritten as: If the capacitance value of the first capacitor C1 is unknown, equation 9 has a discriminant (assuming that M2<< L1L2) as follows: As long as M is chosen such that (1 - σ)L2C2 > -MC3, there will be two (positive) solutions for the capacitance value of the first capacitor C1 to satisfy the specified σ. The solutions can be found from Equation 11: Practically speaking, we can assume that the capacitance values ​​remain relatively constant if the capacitance value of the first capacitor C1 changes from C1- to C1+. This is consistent with the variational analysis described below. With α = L1C1+ L2C2- MC3, the following derivatives can be determined:•••••• Accordingly, if the capacitance of the first capacitor C1 is increased, the two notches would shift downwards together. However, if the capacitance is close to zero, we expect σ (or δ) to change very little when sampling the capacitance of the first capacitor C1 from C1- to C1+. Furthermore, increasing C1 should shift ω+ downwards, while ω- could shift upwards or downwards. The case of two notches shifting in opposite directions in the frequency domain could be used in a variety of applications to implement out-of-band rejection. The coupling coefficient of the first inductor L1 and the second inductor L2 of the tunable filter 90 was investigated. Fig. 10 shows a plot of the coupling coefficient against the distance between the first inductor L1 and the second inductor L2. Mutual coupling of inductors can be achieved by placing two SMT inductors relatively close together. As another example, mutual coupling of inductors can be achieved via two suitably aligned coils embedded in a laminate or on a chip. Fig. 10 shows a graph of the strength of the coupling coefficient of the first inductor L1 and the second inductor L2 of the tunable filter 90 in an embodiment where the first inductor L1 and the second inductor L2 each have an inductance value of 2.7 nanohenries (nH).Evidence 10 indicates that the physical layout of inductors influences the mutual coupling, and in particular the distance between inductors can affect the mutual coupling. The geometry of the mutually coupled inductors can also affect the mutual coupling. The coupling coefficient typically also depends on the polarity of the mutually coupled inductors. Reversing the polarity of one of the mutually coupled inductors can reverse the sign of the coupling coefficient (e.g., from positive to negative, or from negative to positive). The dots on the inductors shown in the diagrams indicate their polarity. A first design example of the tunable filter 90 of Fig. 9 will now be discussed. In this example, the tunable filter 90 has two operating frequencies (1 GHz and 0.9 GHz), which are achieved by switching to two different effective capacitance values ​​of the tunable impedance circuit 92. Each state has a 2fo rejection of >25 decibels (dB) and a 3fo rejection of >35 dB, where the 2fo rejection means a second harmonic rejection and the 3fo rejection means a third harmonic rejection. In this example, a constant quality factor (Q) of 20 is assumed for the first and second inductors L1 and L2. The tunable impedance circuit 92 is adjustable and / or switchable so that a first effective capacitance C1A is provided at an operating frequency of 1 GHz and a second effective capacitance C1B is provided at an operating frequency of 0.9 GHz.To meet these performance requirements, the tunable filter can have 90 components with the values ​​shown in Table 1 below. TABLE 1 Value3 nH1.31 pF1.99 pF3 nH1.41 pF1.87 pF-0.11 Fig. 11 shows a simulation plot illustrating the positions of the harmonic notches of the tunable filter 90 of Fig. 9, designed according to the first design example. A first curve in Fig. 11 shows the positions of the harmonic notches for the second and third harmonics at an operating frequency of 1 GHz. A second curve in Fig. 11 shows the positions of the harmonic notches for the second and third harmonics at an operating frequency of 0.9 GHz. These curves show the positions of two notches in the frequency response of the tunable filter 90, which shift with a change in the effective capacitance value of the tunable impedance circuit 92 at two different operating frequencies. Accordingly, changing the state of the switch can change the positions of the harmonic notches.The positions of two notches in the frequency response of the tunable filter 90 change when the tunable impedance circuit 92 has a switch designed to electrically connect one end of a capacitor to the first inductor L1 in order to change the state of the tunable impedance circuit 92. A second design example of the tunable filter 90 of Fig. 9 will now be discussed. In this example, the tunable filter 90 has three operating frequencies (1 GHz, 0.9 GHz, and 0.8 GHz), which are achieved by switching to three different effective capacitance values ​​of the tunable impedance circuit 92. Each state has a 2fO rejection of >25 dB and a 3fO rejection of >35 dB. In this example, a constant quality factor (Q) of 20 is assumed for the first and second inductors L1 and L2. The tunable impedance circuit 92 is adjustable and / or switchable such that a first effective capacitance C1A is provided at an operating frequency of 1 GHz, a second effective capacitance C1B is provided at an operating frequency of 0.9 GHz, and a third effective capacitance C1C is provided at an operating frequency of 0.8 GHz.To meet these performance requirements, the tunable filter can have 90 components with the values ​​shown in Table 2 below. TABLE 2 Value4 nH0.90 pF1.27 pF2.10 pF4 nH1.26 pF2.24 pF-0.084 Fig. 12 shows a simulation plot illustrating the positions of the harmonic notches of the tunable filter 90 of Fig. 9, designed according to the second design example. A first curve in Fig. 12 shows the positions of the harmonic notches for the second and third harmonics at an operating frequency of 1 GHz. A second curve in Fig. 12 shows the positions of the harmonic notches for the second and third harmonics at an operating frequency of 0.9 GHz. A third curve in Fig. 12 shows the positions of the harmonic notches for the second and third harmonics at an operating frequency of 0.8 GHz. These curves show the positions of two notches in the frequency response of the tunable filter 90, which shift with a change in the effective capacitance value of the tunable impedance circuit 92 at three different operating frequencies.Accordingly, changing the state of the switch can change the positions or locations of two notches in the frequency response of the tunable filter 90 if the tunable impedance circuit 92 has a switch which is designed to electrically connect one end of a capacitor to the first inductor L1 in order to change the state of the tunable impedance circuit 92. Fig. 13 shows a schematic representation of a tunable filter 130 according to one embodiment. The tunable filter 130 corresponds to the tunable filter 50 of Fig. 5, except that the tunable filter 130 has a tunable impedance circuit 92 instead of the tunable impedance circuit 42. A third design example relating to the tunable filter 130 will now be discussed. In this example, the tunable filter 130 has two operating frequencies (1 GHz and 0.9 GHz), which are achieved by switching to two different effective capacitance values ​​of the tunable impedance circuit 92. Each state has a 2fo rejection of >30 dB, a 3fo rejection of >50 dB, and a 4fo rejection of >60 dB, where 2fo rejection denotes second harmonic rejection, 3fo rejection denotes third harmonic rejection, and 4fo rejection denotes fourth harmonic rejection. In this example, the first and second inductors L1 and L2 are assumed to have a constant quality factor (Q) of 20.The tunable impedance circuit 92 is adjustable and / or switchable to provide a first effective capacitance C1A at an operating frequency of 1 GHz and a second effective capacitance C1B at an operating frequency of 0.9 GHz. To achieve these performance requirements, the tunable filter 130 can have components with the values ​​shown in Tables 3A and 3B below. TABLE 3A Value7 nH0.38 pF0.71 pF2.5 nH0.84 pF2.5 pF TABLE 3B TABLE 3B Value1.81 pF2.5 pF2.5 nH-0.017-0.086 Fig. 14 shows a simulation plot illustrating the positions of the harmonic notches of the tunable filter 130 of Fig. 13, designed according to the third design example. A first curve in Fig. 14 shows the positions of the harmonic notches for the second, third, and fourth harmonics at an operating frequency of 1 GHz. A second curve in Fig. 14 shows the positions of the harmonic notches for the second, third, and fourth harmonics at an operating frequency of 0.9 GHz. These curves show the positions of three notches in the frequency response of the tunable filter 130, which shift with a change in the effective capacitance value of the tunable impedance circuit 92 at two different operating frequencies. Filter designs with N switches providing M × 2N harmonic states are disclosed, where M is an integer greater than or equal to two. The filter designs disclosed herein utilize inductive mutual coupling to increase the number of adjustable harmonic states. Example designs with M = 2 (Design Examples 1 and 2) and M = 3 (Design Example 3) are described. Although the embodiments disclosed herein may relate to low-pass filters, all suitable principles and advantages disclosed herein can also be applied to other filter types, such as high-pass filters, band-pass filters, and / or band-stop filters. The principles and advantages disclosed herein can be implemented in a variety of filters. For example, filters disclosed herein may include or be incorporated within non-acoustic filters with passive impedance components. As another example, filters disclosed herein may be incorporated within a hybrid filter, which in certain applications includes inductive and capacitive components together with one or more acoustic wave resonators. For example, the inductive and capacitive components may establish a passband or a stopband of such a hybrid filter, and the one or more acoustic wave resonators may achieve one or more relatively sharp band edges for the hybrid filter. High-frequency systems with tunable filters The tunable filters disclosed herein may be used in high-frequency systems, such as a high-frequency front end. A tunable filter designed in accordance with any suitable principles and advantages may be used at any suitable point in a system that could benefit from the harmonic suppression provided by the filters disclosed herein. Fig. 15A shows a schematic block diagram of a high-frequency system 150 with a tunable filter 152. As shown in Fig. 15A, the tunable filter 152 is coupled between an antenna switch 154 and an antenna 155. The tunable filter 152 can be designed in accordance with any suitable principles and advantages described herein to provide harmonic suppression for high-frequency signals propagating between the antenna switch 154 and the antenna 155. Fig. 15B shows a schematic block diagram of a high-frequency system 156 with a tunable filter 157. As shown in Fig. 15B, the tunable filter 157 is coupled between a power amplifier 158 and a band selector switch 159. The band selector switch 159 can electrically connect an output of the power amplifier 158 to a high-frequency signal path for a specific operating band. Such a high-frequency signal path can include a bandpass filter with a passband corresponding to the operating band. The band selector switch 159 can, for example, be a multi-output switch that can selectively connect the tunable filter 157 electrically to a selected high-frequency signal path.The tunable filter 157 can be designed in accordance with any suitable principles and advantages described herein to provide harmonic suppression for high-frequency signals propagating between the output of the power amplifier 158 and the band selector switch 159. Wireless communication devices The tunable filters disclosed herein may be used in wireless communication devices, such as mobile communication devices. One or more tunable filters designed in accordance with any suitable principles and advantages may be used in any suitable wireless communication device. An example of such a wireless communication device is explained in connection with Fig. 16. Fig. 16 is a schematic diagram of an embodiment of a mobile device 800. The mobile device 800 includes a baseband system 801, a transceiver 802, a front-end system 803, antennas 804, a power control system 1505, a memory 806, a user interface 807 and a battery 808. The Mobile Device 800 can communicate using a variety of communication technologies, including but not limited to 2G, 3G, 4G (including LTE, LTE-Advanced and LTE-Advanced Pro), 5G NR, WLAN (e.g. Wi-Fi), WPAN (e.g. Bluetooth and ZigBee), WMAN (e.g. WiMAX) and / or GPS technologies. The transceiver 802 generates RF signals for transmission and processes incoming RF signals received by the antennas 804. It should be noted that various functionalities associated with transmitting and receiving RF signals can be achieved by one or more components, collectively represented as the transceiver 802 in Fig. 16. In one example, separate components (e.g., separate circuits or raw chips) can be provided for processing specific types of RF signals. The front-end system 803 assists in the processing of signals transmitted and / or received by the antennas 804. In the illustrated embodiment, the front-end system 803 comprises antenna tuning circuits 810, power amplifiers (PAs) 811, low-noise amplifiers (LNAs) 812, filters 813, switches 814, and signal-splitting / combining circuits 815. However, other implementations are also possible. The filters 813 can include one or more tunable harmonic-suppression filters, which incorporate one or more features of the embodiments disclosed herein. For example, the 803 front-end system can provide a number of functions, including, without limitation of generality, transmit signal amplification, receive signal amplification, signal filtering, switching between different bands, switching between different power modes, switching between transmit and receive modes, signal duplexing, signal multiplexing (for example, diplexing or triplexing), or any combination of these functions. In certain implementations, the Mobile 800 supports carrier bonding, providing flexibility to increase peak data rates. Carrier bonding can be used for both frequency division duplexing (FDD) and time division duplexing (TDD) and can be employed to bundle multiple carriers or channels. Carrier bonding includes contiguous bonding, where adjacent carriers within the same operating frequency band are bundled. Carrier bonding can also be non-contiguous and can include carriers that are frequency-separated within a common band or in different bands. The 804 antennas can include antennas used for a wide variety of different communication types. For example, 804 antennas can include antennas for transmitting and / or receiving signals associated with a wide variety of different frequencies and communication standards. In certain implementations, the antennas support 804 MIMO communication and / or switched diversity communication. For example, MIMO communication uses multiple antennas to transmit multiple data streams over a single radio frequency channel. MIMO communication benefits from a better signal-to-noise ratio, improved coding, and / or reduced signal interference due to spatial multiplexing differences in the radio environment. Switched diversity refers to communication where a specific antenna is selected for operation at specific times. For example, a switch can be used to select a particular antenna from a group of antennas based on a variety of factors, such as an observed bit error rate and / or a signal strength indicator. The Mobile Device 800 can be operated with beamforming in certain implementations. For example, the Front End System 803 can incorporate power amplifiers with controllable gain and phase shifters with variable phase to provide beamforming and directional characteristics for transmitting and / or receiving signals using the Antennas 804. For example, in the context of signal transmission, the amplitudes and phases of the transmit signals provided to the Antennas 804 can be controlled such that the signals emitted by the Antennas 804 are combined under constructive and destructive interference to obtain a focused transmit signal with beam-like characteristics, which exhibits higher signal strength in a predetermined direction of propagation.In the context of signal reception, the amplitudes and phases can be controlled so that more signal energy is received when the signal arrives at the 804 antennas from a particular direction. In certain implementations, the 804 antennas have one or more arrays of antenna elements to amplify the beamforming. The baseband system 801 is coupled to the user interface 807 to process various user inputs and outputs (I / O), such as voice and data signals. The baseband system 801 provides the transceiver 802 with digital representations of the transmitted signals, which the transceiver 802 processes to generate RF signals for transmission. The baseband system 801 also processes digital representations of received signals supplied by the transceiver 802. As shown in Fig. 16, the baseband system 801 is coupled to the memory 806 to enable operation of the mobile device 800. The 806 memory can be used for a wide variety of purposes, such as storing data and / or instructions to enable the operation of the 800 mobile device and / or providing storage for user information. The 805 power control system provides a number of power control functions for the 800 mobile device. In certain implementations, the 805 power control system includes a power amplifier supply control circuit that controls the supply voltages of the 811 power amplifiers. For example, the 805 power control system may be designed to modify the supply voltage(s) provided to one or more of the 811 power amplifiers to improve their efficiency, such as power added efficiency (PAE). As shown in Fig. 16, the power control system 805 receives a battery voltage from the battery 808. The battery 808 can be any suitable battery for use in the mobile device 800, including, for example, a lithium-ion battery. Applications, terminology and concluding remarks Each of the embodiments described above can be used in connection with mobile devices such as portable mobile phones. The principles and advantages of the embodiments can be used for any system or device, such as any wireless uplink communication device, that could benefit from any of the embodiments herein. The teachings herein are applicable to a variety of systems. Although this disclosure contains exemplary embodiments, the teachings described herein can be applied to a wide variety of structures. All the principles and advantages discussed herein can be implemented in connection with RF circuits designed to amplify and process signals with a frequency in the range of about 30 kHz to 300 GHz, such as in a frequency range of about 450 MHz to 8.5 GHz.The tunable filters disclosed herein can filter RF signals at frequencies up to and including millimeter wave frequencies, for example frequencies within FR2 of a 5G-NR specification. Aspects of this disclosure may be implemented in various electronic devices. Examples of such electronic devices may include, but are not limited to, consumer electronics products, components of consumer electronics products such as enclosed radio frequency modules, wireless uplink communication devices, wireless communication infrastructure, electronic test equipment, etc.Examples of electronic devices may include, but are not limited to, a mobile phone such as a smartphone, a portable computing device such as a smartwatch or earpiece, a telephone, a television, a computer monitor, a computer, a modem, a portable computer, a laptop, a tablet, a microwave oven, a refrigerator, an electronic system for a vehicle such as an automobile, a robot such as an industrial robot, an Internet of Things device, a stereo system, a digital music player, a radio, a camera such as a digital camera, a portable memory chip, a household appliance such as a washing machine or dryer, a peripheral device, a wristwatch, a clock, etc. Furthermore, electronic devices may also include unfinished products. Unless the context clearly requires otherwise, the words "comprise," "comprehensive," and the like in the description and claims are to be interpreted in an inclusive sense, as opposed to an exclusive or exhaustive one; that is, in the sense of "including but not limited to." The word "coupled," as used generally here, refers to two or more elements that are either directly connected or may be connected via one or more intermediate elements. Likewise, the word "connected," as used generally here, refers to two or more elements that are either directly connected or may be connected via one or more intermediate elements. Furthermore, the words "here," "above," "below," and words of similar meaning, when used in this application, refer to this application as a whole and not to a particular part thereof.Where the context allows, words in the detailed description above that refer to the singular or plural may also include the plural or singular form. Although certain embodiments of the inventions have been described, these embodiments are presented only as examples and are not intended to limit the scope of the disclosure. In fact, the novel filters, wireless communication devices, apparatus, and methods described herein can be implemented in a multitude of other configurations. Furthermore, various omissions, substitutions, and modifications can be made to the configuration of the filters, wireless communication devices, apparatus, and methods described herein without departing from the fundamental concept of the disclosure. For example, while processes or blocks are presented in a particular sequence, alternative embodiments can perform similar functions with different components and / or circuit arrangements, and some processes or blocks can be deleted, moved, added, subdivided, combined, and / or modified.Each of these processes or blocks can be implemented in different ways. Any suitable combination of elements and / or actions from the various embodiments described above can be combined to achieve further embodiments.

Claims

A tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) with tunable suppression, comprising: a first inductor (L1); a second inductor (L2) mutually coupled to the first inductor (L1), wherein the first inductor (L1) is connected in series with the second inductor (L2); and a tunable capacitor circuit (42; 52; 62; 72; 74; 92) which is electrically connected in parallel to the first inductor (L1) which has a switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) and which is designed to match at least two notches in the frequency response of the tunable filter by changing a switching state of the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N), wherein the tunable filter is designed to filter a high-frequency signal. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 1, wherein the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) is designed to selectively couple one end of a capacitor (C11, C12, ..., C1N) to the first inductor (L1). The tunable filter (60; 70; 80; 80'; 90; 130) according to claim 2, wherein the tunable capacitor circuit has a second switch (S11, S12, ..., S1N; S21, S22, ..., S2N) which is designed to selectively couple one end of a second capacitor (C11, C12, ..., C1N; C21, C22, ..., C2N) to the first inductor (L1). The tunable filter (50; 60; 70; 80; 80'; 90; 130) according to claim 1, further comprising a second tunable capacitor circuit connected in parallel to the second inductor (L2). The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 1, wherein the at least two notches provide harmonic suppression. A tunable filter (50; 60; 70; 80; 130) with tunable suppression, comprising a first inductor (L1); a second inductor (L2) mutually coupled to the first inductor (L1); and a tunable impedance circuit (42; 52; 62; 72; 74; 92) electrically connected to the first inductor (L1) having a switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) designed to adjust the position of at least three notches in the frequency response of the tunable filter by changing a switching state of the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N), wherein the tunable filter is designed to filter a high-frequency signal. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 6, wherein the tunable impedance circuit (92) comprises a tunable capacitor circuit. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 7, wherein the tunable capacitor circuit (92) is connected in parallel to the first inductor (L1). The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 8, wherein the first inductor (L1) is connected in series with the second inductor (L2). A tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) with tunable suppression, comprising a first inductor (L1); a second inductor (L2) mutually coupled to the first inductor (L1); and a tunable capacitor circuit (42; 52; 62; 72; 74; 92) electrically connected to the first inductor (L1), wherein the tunable capacitor circuit is designed to provide a shunt capacitance at a node between the first inductor (L1) and the second inductor (L2), wherein the tunable capacitor circuit (42; 52; 62; 72; 74; 92) has a switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) and is designed to produce at least two notches in the frequency response of the tunable filter by changing a switching state of the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) to adapt, the tunable filter being designed to filter a high-frequency signal. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 10, wherein the at least two notches provide harmonic suppression. A tunable filter (40; 50; 60; 70; 80; 80'; 90; 130)) with tunable suppression, comprising: a first inductor (L1); a second inductor (L2) coupled reciprocally to the first inductor (L1), wherein the first inductor (L1) is connected in series with the second inductor (L2), wherein a first capacitor is connected in parallel with the first inductor (L1) and a second capacitor is connected in parallel with the second inductor (L2); and a tunable impedance circuit (42; 52; 62; 72; 74; 92) electrically connected to the first inductor (L1), which has a switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) and which is designed to produce at least two notches in the frequency response of the tunable filter by changing a switching state of the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) to adapt, the tunable filter being designed to filter a high-frequency signal . The tunable filter (50; 60; 70; 80; 80'; 90; 130) according to claim 12, further comprising a shunt capacitor (C3) coupled between the first inductor (L1) and the second inductor (L2). The tunable filter (50; 60; 70; 80; 80'; 90; 130) according to claim 12, wherein the tunable impedance circuit is designed to provide for the first capacitor. A tunable filter (80; 80'; 90) with tunable suppression, comprising: a first inductance (L1), wherein the first inductance (L1) is a series inductance; a second inductance (L2) mutually coupled to the first inductance (L1), wherein the second inductance (L2) is a shunt inductance; and a tunable impedance circuit (42; 52; 62; 72; 74; 92) electrically connected to the first inductor (L1) having a switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N) designed to match at least two notches in the frequency response of the tunable filter by changing a switching state of the switch (S11, S12, ..., S1N; S21, S22, ..., S2N; S31, S32, ..., S3N; S41, S42, ..., S4N; S51, S52, ..., S5N), wherein the tunable filter is designed to filter a high-frequency signal. The tunable filter (80; 80'; 90) according to claim 15, wherein the tunable impedance circuit comprises a tunable capacitor circuit connected in parallel to the first inductor (L1). The tunable filter (80; 80'; 90) according to claim 16, further comprising an inductor-capacitor circuit in series with the first inductor (L1). The tunable filter (80; 80') according to claim 17, wherein the inductor-capacitor circuit has a third inductor (L3) which is mutually coupled to at least one of the first inductor (L1) and the second inductor (L2). A method for filtering high-frequency signals, comprising: filtering a first high-frequency signal with a tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) in a first state; after filtering the first high-frequency signal, changing the switching state of a switch of a tunable impedance circuit (42; 52; 62; 72; 74; 92) of the tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) from the first state to a second state in order to adjust the position of at least three harmonic corresponding notches in the frequency response of the tunable filter (40; 50; 60; 70; 80; 80'; 90; 130), which incorporates mutually coupled inductors and the tunable impedance circuit. (42; 52; 62; 72; 74; 92) has which is electrically connected to the mutually coupled inductors; and while the tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) is in the second state, filtering a second high-frequency signal with the tunable filter (40; 50; 60; 70; 80; 80';90; 130).; A wireless communication device (800) comprising: a tunable filter (813) according to claim 1; and an antenna (804) designed to transmit a high-frequency signal filtered by the tunable filter (813). A tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) with harmonic rejection, comprising: a first inductor (L1); a second inductor (L2) mutually coupled to the first inductor (L1); and a tunable capacitor circuit (42; 52; 62; 72; 74; 92) electrically connected to the first inductor (L1), and having N switches designed to adjust the effective capacitance of the tunable capacitor circuit to set the harmonic rejection of the tunable filter to at least 2 × 2N harmonics, where N is a positive integer greater than 1 and where the tunable filter is designed to filter a high-frequency signal. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 21, wherein the N switches are designed to set a harmonic suppression of the tunable filter to at least 3 × 2N harmonics. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 21 or 22, wherein the tunable capacitor circuit is connected in parallel to the first inductor (L1). The tunable filter (50; 60; 70; 80; 80'; 90; 130) according to claim 22, further comprising a capacitor in parallel to the second inductor (L2) and a shunt capacitor between the first inductor (L1) and the second inductor (L2) connected in series. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to one of claims 21 to 24, wherein a change in the switching state of a first switch of the N switches adjusts the position of at least two notches in the frequency response of the tunable filter. The tunable filter (50; 60; 70; 80; 130) according to claim 25, wherein a change in the switching state of a first switch of the N switches adjusts the position of at least three notches in the frequency response of the tunable filter. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to one of claims 21 to 26, wherein the second inductance (L2) is a shunt inductance. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to claim 27, wherein the tunable capacitor circuit is connected in parallel to the first inductor (L1). The tunable filter (50; 60; 70; 80; 80'; 90; 130) according to claim 28, further comprising a shunt capacitor (C3; C5) connected in series with the second inductor (L2), which is electrically connected to the first inductor (L1) via the second inductor (L2). The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to one of claims 21 to 29, wherein the 2 × 2N harmonics have at least one second harmonic and at least one third harmonic. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to any one of claims 21 to 30, wherein the 2 × 2N harmonics have at least one harmonic that is assigned to a fifth generation New Radio operating band. The tunable filter (40; 50; 60; 70; 80; 80'; 90; 130) according to any one of claims 21 to 31, wherein the 2 × 2N harmonics comprise at least one harmonic associated with a fifth generation New Radio operating band and at least one harmonic associated with a fourth generation Long Term Evolution operating band. A wireless communication device (803) comprising: a high-frequency front end (803) with a tunable filter (813) comprising a first inductor, a second inductor mutually coupled to the first inductor, and a tunable capacitor circuit electrically connected to the first inductor and comprising N switches designed to adjust the effective capacitance of the tunable capacitor circuit to set the harmonic rejection of the tunable filter to at least 2 × 2N harmonics, where N is a positive integer greater than 1, wherein the tunable filter is designed to filter a high-frequency signal; and an antenna (804) communicating with the high-frequency front end (803). The wireless communication device (800) according to claim 33, wherein the antenna (804) is designed to transmit a high-frequency signal filtered by the tunable filter (813). The wireless communication device (800) according to claim 33 or 34, wherein the high-frequency front end (803) has an antenna switch (154) and the tunable filter (152; 813) is coupled between the antenna switch (154) and the antenna (155; 804). The wireless communication device (800) according to one of claims 33 to 35, wherein the high-frequency front end (803) comprises a power amplifier (811; 158) and a band selector switch (159), and the tunable filter (157; 813) is coupled between the power amplifier (811; 158) and the band selector switch (159). The wireless communication device (800) according to one of claims 33 to 36, wherein the wireless communication device is designed to implement dual network operation, and the tunable filter (813) is designed to provide suppression during dual network operation. The wireless communication device (800) according to one of claims 33 to 36, wherein the wireless communication device is designed to implement a carrier bundling operation, and the tunable filter (813) is designed to provide suppression during carrier bundling. The wireless communication device (800) according to one of claims 33 to 38 wherein N is at least 4. A high-frequency system (150) comprising: an antenna switch (154); a tunable filter (152) comprising a first inductor, a second inductor mutually coupled to the first inductor, and a tunable capacitor circuit electrically connected to the first inductor and comprising N switches designed to adjust the effective capacitance of the tunable capacitor circuit to set the harmonic rejection of the tunable filter to at least 2 × 2N harmonics, where N is a positive integer greater than 1, wherein the tunable filter is designed to filter a high-frequency signal; and an antenna connector (155), wherein the tunable filter (152) is coupled into a signal path between the antenna switch (154) and the antenna connector (155).

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