Synchronous motor control device, electric vehicle and synchronous motor control method

Abstract

A synchronous motor control device comprising a torque command conversion unit that converts a torque command for a synchronous motor into a second torque command that gradually increases in a range including a peak of the torque of the synchronous motor, a difference calculation unit that calculates a difference between the torque of the synchronous motor and the second torque command, a voltage phase control unit that controls a voltage phase angle to reduce the difference, and a power conversion unit that converts DC power into AC power based on the voltage phase angle and outputs the converted AC power to the synchronous motor, wherein the difference calculation unit corrects the difference in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor,by calculating the second torque setpoint based on a predetermined correction value.

Description

Technical area

[0001] The present invention relates to a synchronous motor control device, an electric vehicle and a synchronous motor control method. background

[0002] DC power is converted into AC power by a power conversion unit, and a square-wave voltage is applied to a synchronous motor through voltage phase control to rotate the synchronous motor. Through this voltage phase control, the output power can be improved over a high speed range of the synchronous motor, and the number of switching operations in the power conversion unit can be reduced, thus reducing switching loss. To improve the voltage utilization efficiency of the synchronous motor, single-pulse control is used, in which the number of pulses in one cycle of the square-wave voltage is reduced. In single-pulse control, torque control is performed by operating a voltage phase based on the torque of the synchronous motor and a torque command.

[0003] PTL 1 discloses a motor control device including a voltage command calculation unit that calculates a voltage command used for voltage phase control according to a torque command, a phase calculation unit that calculates a phase command of the applied voltage according to the torque command, a first estimation unit that estimates the torque generated in a motor as a first torque based on a current command, a second estimation unit that estimates the torque generated in the motor as a second torque based on a current flowing through the motor, and a correction unit that corrects the phase command to reduce the deviation between the first torque and the second torque to calculate the voltage command. Citation listPatent literature

[0004] PTL 1: JP 2017-229127 A Overview of the inventionTechnical problem

[0005] Fig. Figure 17 illustrates a relationship between the voltage phase angle θv and the output torque. Here, Ov is set to 0 degrees with respect to the q-axis. Up to the maximum (minimum) torque, the torque increases (decreases) monotonically with respect to the voltage phase angle θv. However, when the maximum (minimum) torque is exceeded, the torque decreases (increases) again. For this reason, it is necessary to perform control within a range where the voltage phase angle θv does not exceed the maximum (minimum) torque, and it is necessary to limit a torque command (or the voltage phase angle θv) with a margin from the maximum (minimum) torque.Therefore, in the method of operating the voltage phase angle θv based on an estimated torque calculated from a current value or the like as in PTL 1, the torque that can actually be output is limited relative to the torque that can actually be output by a motor. In view of the above, this problem can be solved by controlling the voltage phase angle θv by using a physical quantity that monotonically increases even near the maximum (minimum) torque. For example, it is conceivable to use the d-axis current id or the d-axis magnetic flux Φd. Fig. 18 illustrates a relationship between the voltage phase angle θv and the d-axis magnetic flux Φd, and Fig. Figure 19 illustrates a relationship between the voltage phase angle θv and the d-axis current id. In a case of the d-axis magnetic flux Φd or the d-axis current id, the operation is stable even near the maximum (minimum) torque because the voltage phase angle θv increases monotonically until it reaches 180 degrees (-180 degrees).

[0006] A problem to be solved by the present invention is that in a case where a voltage phase angle is operated based on a torque command such as the d-axis magnetic flux Φd or the d-axis current id, when a sign of a torque command changes, the deviation cannot be calculated correctly, overshoot occurs, and a first-order delay response as designed cannot be realized. Solution to the problem

[0007] A synchronous motor control device according to the invention includes a torque command conversion unit that converts a torque command for a synchronous motor into a second torque command that gradually increases in a range including a peak of the torque of the synchronous motor, a difference calculation unit that calculates a difference between the torque of the synchronous motor and the second torque command, a voltage phase control unit that controls a voltage phase angle to reduce the difference, and a power conversion unit that converts DC power into AC power based on the voltage phase angle and outputs the converted AC power to the synchronous motor, wherein the difference calculation unit corrects the difference by calculating the second torque command in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor.calculated based on a predetermined correction value.

[0008] A synchronous motor control method according to the present invention is a synchronous motor control method in a synchronous motor control device having a power conversion unit that converts DC power into AC power based on a voltage phase angle and outputs the converted AC power to a synchronous motor, the synchronous motor control method comprising converting a torque command for the synchronous motor into a second torque command that gradually increases in a range including a peak of the torque of the synchronous motor, calculating a difference between the torque of the synchronous motor and the second torque command, controlling the voltage phase angle to reduce the difference, and correcting the difference by calculating the second torque command in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor.based on a predetermined correction value. Advantageous effects of the invention

[0009] According to the present invention, even if a sign of a torque command changes, no overshoot occurs, and a first-order deceleration response can be realized as designed. Short description of the drawings [ Fig. 1] Fig. 1 is a block configuration diagram of a synchronous motor control device according to a first embodiment. [ Fig. 2] Fig. 2 is a block configuration diagram of a control unit according to the first embodiment. [ Fig. 3] Fig. 3 is a block configuration diagram of a difference calculation unit according to the first embodiment. [ Fig. 4] Fig. 4 is a diagram illustrating an operating point of torque expressed in coordinates represented by a d-axis magnetic flux and a q-axis magnetic flux in the first embodiment. [ Fig. 5] Fig. 5 is a block configuration diagram of a second difference calculation unit according to the first embodiment. [ Fig. 6] The Fig. 6(A), Fig. 6(B), Fig. 6(C), Fig. 6(D) and Fig. 6(E) are graphs showing response waveforms in a case where the first embodiment is applied. [ Fig. 7] Fig. 7 is a block configuration diagram showing a variant of the second difference calculation unit in the first embodiment. [ Fig. 8] Fig. 8 is a diagram illustrating an example of a look-up table according to a variant of the first embodiment. [ Fig. 9] Fig. 9 is a diagram illustrating an operating point of torque in coordinates represented by a d-axis magnetic flux and a q-axis magnetic flux in a comparative example. [ Fig. 10] Fig. 10 is a block configuration diagram illustrating the second difference calculation unit in the comparative example. [ Fig. 11] The Fig. 11(A), Fig. 11(B), Fig. 11(C), Fig. 11(D) and Fig. 11(E) are graphs illustrating the response waveforms in the comparative example. [ Fig. 12] Fig. 12 is a block configuration diagram of a difference calculation unit according to a second embodiment. [ Fig. 13] Fig. 13 is a diagram illustrating an operating point of torque in coordinates represented by the d-axis current and the q-axis current in the second embodiment. [ Fig. 14] Fig. 14 is a block configuration diagram of a second difference calculation unit 302 according to the second embodiment. [ Fig. 15] Fig. 15 is a block configuration diagram illustrating a variant of the second difference calculation unit in the second embodiment. [ Fig. 16] Fig. 16 is a diagram illustrating an example of a lookup table according to the second embodiment. [ Fig. 17] Fig. Figure 17 is a graph illustrating a relationship between a voltage phase angle and the torque output from a synchronous motor. [ Fig. 18] Fig. Figure 18 is a graph illustrating a relationship between a voltage phase angle and a d-axis magnetic flux. [ Fig. 19] Fig. Figure 19 is a graph illustrating a relationship between a voltage phase angle and the d-axis current. [ Fig. 20] Fig. 20 is a configuration diagram of an electric vehicle in a third embodiment. Description of the embodiments

[0010] An embodiment of the present invention will be described below with reference to the drawings. The description below and the drawings are an example for describing the present invention, and for the sake of clarity, omissions and simplifications are appropriately made. The present invention can be embodied in various other forms. Unless otherwise specified, the number of each of the individual components may be one or more than one.

[0011] The positions, sizes, shapes, areas, and the like of each of the individual components illustrated in the drawings may not represent the actual positions, sizes, shapes, areas, and the like in some cases to facilitate understanding of the invention. Therefore, the present invention is not necessarily limited to any position, size, shape, area, and the like disclosed in the drawings. [First embodiment]

[0012] Fig. 1 is a block configuration diagram of a synchronous motor control device 1 according to the present embodiment.

[0013] The synchronous motor control device 1 converts DC power from a DC power supply 2 into AC power and drives a synchronous motor 3 with the converted AC power.

[0014] For example, a synchronous motor (permanent magnet synchronous motor, PMSM) 1 has a rotating field type configuration in which a permanent magnet is provided in a rotor and an armature winding is provided in a stator. The synchronous motor 3 operates by an interaction between a magnetic flux generated by a permanent magnet provided in a rotor (not shown) and a magnetic field generated by the current of each phase flowing through a three-phase winding attached to an armature. Note that the synchronous motor 3 is not limited to a permanent magnet synchronous motor, but may be a synchronous motor such as a wound synchronous motor. The synchronous motor 3 is a drive source of an electric vehicle and rotates to drive the electric vehicle.

[0015] The synchronous motor 3 includes a magnetic pole position sensor 31. The magnetic pole position sensor 31 functions to detect a magnetic pole position of a rotor of the synchronous motor 3 as a phase relative to a stator. The magnetic pole position sensor 31 outputs a detected rotor phase θ, which is input to a control unit 100 of the synchronous motor control device 1. The magnetic pole position sensor 31 can be implemented by a resolver, a rotary encoder, an absolute encoder, and the like.

[0016] The DC power supply 2 supplies DC power to a power conversion unit 400 of the synchronous motor control device 1. The DC power supply 2 is, for example, a chargeable and dischargeable battery.

[0017] The synchronous motor control device 1 includes the control unit 100, a voltage detector 200, a gate drive circuit 300, the power conversion unit 400, and a current detector 500.

[0018] The control unit 100 generates a gate signal GS based on a torque command value T* or the like provided by a host controller (not shown) of an electric vehicle, and performs torque control of the synchronous motor 3. Details of the generation of the gate signal GS will be described later.

[0019] The voltage detector 200 is connected to a wiring 133 on the high potential side and a wiring 134 on the low potential side of the DC power supply 2 and detects a potential difference between them. Since this potential difference is typically a high voltage of, for example, 100 V or more, the voltage detector 200 converts the potential difference into a low voltage that can be detected by the control unit 100 and outputs a value of this converted DC voltage Vdc to the control unit 100.

[0020] The gate drive circuit 300 converts the gate signal GS generated by the control unit 100 into a potential required to switch the switching elements 114, 115, 116, 117, 118, and 119 in the power conversion unit 400 between on and off, and outputs the potential to the power conversion unit 400.

[0021] The power conversion unit 400 includes a smoothing capacitor 138 and switching elements 114 to 119, which form an inverter with upper and lower arms for three phases. The smoothing capacitor 138 suppresses the pulsation of the DC voltage generated by the switching operation of the switching elements 114 to 119. The power conversion unit 400 operates the switching elements 114 to 119 according to the gate signal GS described later, converts the DC power supplied from the DC power supply 2 into AC power, and drives the synchronous motor 3.

[0022] The inverter includes switching elements 114, 115, 116, 117, 118, and 119, as well as freewheeling diodes 120, 121, 122, 123, 124, and 125. The switching elements 114 to 119 are, for example, Si-IGBTs and include a gate terminal, a collector terminal, and an emitter terminal. The freewheeling diodes 120 to 125 are connected between the collector terminals and emitter terminals of the switching elements 114 to 119. In a case where the collector terminals of the switching elements 114 to 119 have a higher potential than the emitter terminals, the freewheeling diodes 120 to 125 cause current to flow through the freewheeling diodes 120 to 125 to prevent a high reverse voltage from being applied to the switching elements 114 to 119. However, the inverter can also be formed by another semiconductor element, without being limited to a combination of a Si IGBT and a freewheeling diode.

[0023] The switching of the switching elements 114 to 119 between on and off is performed by the gate signal GS, which is connected to a gate terminal of each of the switching elements 114 to 119.

[0024] The emitter terminal of switching element 114 and the collector terminal of switching element 115 are connected to each other, and a connection point between them is connected to a U-phase wiring 105 so that a U-phase current iu flows. The emitter terminal of switching element 116 and the collector terminal of switching element 117 are connected to each other, and a connection point between them is connected to a V-phase wiring 106 so that a V-phase current iv flows. The emitter terminal of switching element 118 and the collector terminal of switching element 119 are connected to each other, and a connection point between them is connected to a W-phase wiring 107 so that a W-phase current iw flows. The collector terminals of switching elements 114, 116, and 118 are connected to each other and connected to the wiring 133 on the high potential side.In addition, the emitter terminals of the switching elements 115, 117 and 119 are connected to each other and connected to the wiring 134 on the low potential side.

[0025] The current detector 500 detects current flowing through the phase wirings 105, 106 and 107 and outputs the values ​​of the detected phase currents iu, iv and iw to the control unit 100.

[0026] The torque command T* from a host controller, a value of the DC voltage Vdc from the voltage detector 200, values ​​of the phase currents iu, iv, and iw from the current detector 500, and the rotor phase θ from the magnetic pole position sensor 31 are input to the control unit 100. The controller 100 refers to the DC voltage Vdc, the phase currents iu, iv, and iw, and the rotor phase θ, and generates the gate signal GS for turning on and off the switching elements 114, 115, 116, 117, 118, and 119 at an appropriate timing according to the torque command T*. By the gate signal GS, the phase currents iu, iv, and iw flowing through a winding of the synchronous motor 3 are controlled to realize torque control of the synchronous motor 3. The gate signal GS is a PWM (pulse width modulation) signal in which the phase currents iu, iv and iw become sinusoidal signals with phases 120 degrees different from each other.Although described in detail later, the PWM signal is generated based on square wave three-phase voltage commands Vu*, Vv* and Vw*, which are based on the voltage phase angle θv and the rotor phase θ.

[0027] In the present embodiment, the control unit 100 uses one-pulse control in which the number of pulses in one cycle of the rectangular-wave three-phase voltage commands Vu*, Vv*, and Vw* is one to improve the voltage utilization efficiency of the synchronous motor 3. Since the voltage amplitude is fixed in the one-pulse control, voltage phase control is performed. In the voltage phase control, the voltage phase angle θv is controlled based on the torque of the synchronous motor 3 and a torque command, thus performing torque control.

[0028] Fig. 2 is a block configuration diagram of the control unit 100 according to the first embodiment.

[0029] The control unit 100 includes a torque command conversion unit 201, a three-phase and two-phase conversion unit 202, a magnetic flux calculation unit 203, a difference calculation unit 204, a speed calculation unit 205, a voltage phase control unit 206, a square wave generation unit 207, and a PWM gate signal generation unit 208.

[0030] The control unit 100 performs torque control by allowing the three-phase currents iu, iv, and iw to flow through the synchronous motor 3. In this case, the control unit 100 employs what is called a current vector control system, in which fixed coordinates of the three phases are converted and control is performed with two-phase rotation coordinates synchronized with a rotor phase represented by the d-axis and the q-axis.

[0031] The torque command conversion unit 201 receives the input of the torque command T*, a DC voltage value Vdc, and the rotational angular velocity ω, and calculates a d-axis current command id* and a q-axis current command iq* from their input. Furthermore, the torque command conversion unit 201 calculates a d-axis magnetic flux command Φd* and a q-axis magnetic flux command Φq* based on the calculated d-axis current command id* and the q-axis current command iq* by referring to a look-up table or the like.

[0032] Furthermore, a magnetic flux limit Φlim is calculated based on Equation (1) below. The magnetic flux limit Φlim represents the amplitude of a d-axis magnetic flux and a q-axis magnetic flux and is calculated by Equation (1) below, ignoring a voltage drop across the resistance R of the motor winding. Φlim=sqrt(Φd2+Φq2)=Vdc×Kv / ω

[0033] Here, Kv is a voltage utilization coefficient and is 2 / π in the case of single-pulse control. The rotational angular velocity ω is calculated by the velocity calculation unit 205 based on the rotor phase θ. The abbreviation "sqrt" stands for the square root.

[0034] The torque command conversion unit 201 converts the torque command T*, which is a first torque command for the synchronous motor 3, into a second torque command that gradually increases in a region A near a torque peak of the synchronous motor 3. An example of the second torque command is the d-axis magnetic flux command Φd* or the d-axis current command id*. The reason for using the second torque command that gradually increases in a region including a torque peak of the synchronous motor 3 as the torque command T* will be described later. The converted d-axis magnetic flux command value Φd*, the q-axis magnetic flux command value Φq*, the d-axis current command value id*, the q-axis current command value iq*, and the calculated magnetic flux limit value Φlim are output to the difference calculation unit 204.

[0035] The three-phase and two-phase conversion unit 202 performs coordinate conversion of the values ​​of the phase currents iu, iv, and iw from the detector 500 into the d-axis and the q-axis based on information about the rotor phase θ, and outputs the d-axis current id and the q-axis current iq to the difference calculation unit 204 and the magnetic flux calculation unit 203.

[0036] The magnetic flux calculation unit 203 calculates the d-axis magnetic flux Φd and the q-axis magnetic flux Φq by using a look-up table or the like based on the d-axis current id and the q-axis current iq, and outputs the d-axis magnetic flux Φd and the q-axis magnetic flux Φq to the difference calculation unit 204.

[0037] The difference calculation unit 204 receives the inputs of the d-axis current command id*, the q-axis current command iq*, the d-axis current id, and the q-axis current iq, and also receives the inputs of the d-axis magnetic flux command Φd*, the q-axis magnetic flux command Φq*, the d-axis magnetic flux Φd, the q-axis magnetic flux Φq, and the magnetic flux limit Φlim. Based on these inputs, a first difference ΔT1, which is a difference between the torque of the synchronous motor 3 and the first torque command, and a second difference ΔT2, which is a difference between the torque of the synchronous motor 3 and the second torque command, are calculated.In a case where the sign of the torque command is inverted from a current operating point of the synchronous motor 3, a predetermined correction value, for example, the magnetic flux limit value Φlim, is added to the second torque command to calculate the second difference ΔT2. Details will be described later. The calculated first difference ΔT1 and second difference ΔT2 are output to the voltage phase control unit 206.

[0038] Based on the first difference ΔT1 or the second difference ΔT2, the voltage phase control unit 206 outputs the voltage phase angle θv at which the difference becomes small, that is, the difference becomes zero. Based on the rotational angular velocity ω, the voltage phase control unit 206 determines the area A near a peak of the first torque, which is the torque command value T*, and an area B that is not near the peak of the first torque, for example, by using a weighted average. Then, feedback control is performed by using the second difference ΔT2 in the area A and the first difference ΔT1 in the area B so that each difference becomes zero, and the voltage phase angle θv is output. The feedback control uses, for example, PI control.Note that the voltage phase control unit 206 may perform feedback control using only the second difference ΔT2. In this case, the difference calculation unit 204 calculates only the second difference ΔT2.

[0039] The square wave generation unit 207 calculates and outputs the square wave three-phase voltage command values ​​Vu*, Vv*, and Vw* by using a SIGN function or the like based on the voltage phase angle θv and the rotor phase θ.

[0040] The PWM gate signal generation unit 208 calculates the PWM duty cycle based on the three-phase voltage command values ​​Vu*, Vv*, and Vw*. Then, six gate signals GS, which are PWM signals, are generated from the PWM duty cycle, taking dead time and the like into account, and output to the gate drive circuit 300.

[0041] Fig. 3 is a block configuration diagram of the difference calculation unit 204 according to the first embodiment.

[0042] The difference calculation unit 204 includes a first difference calculation unit 301, a second difference calculation unit 302 and a low-pass filter 303.

[0043] The first difference calculation unit 301 receives the input of the d-axis current command value id*, the q-axis current command value iq*, the d-axis magnetic flux command value Φd*, the q-axis magnetic flux command value Φq*, the d-axis current iq, the q-axis current iq, the d-axis magnetic flux Φd, and the q-axis magnetic flux Φq, and outputs the first torque T1, a first torque command value T1*, and the first difference ΔT1. The first torque T1, the first torque command value T1*, and the first difference ΔT1 are calculated by equations (2), (3), and (4) below, respectively. T1=Φd×iq−Φq×id T1*=Φd*×iq*−Φq*×id* ΔT1=T1*−T1

[0044] The low-pass filter 303 is a low-pass filter that simulates a current control response and outputs a q-axis magnetic flux command value ΦqLPF* after filtering with the q-axis magnetic flux command value Φq* as an input signal. However, the output to the second difference calculation unit 302 may be the q-axis magnetic flux command value Φq* before filtering.

[0045] The second difference calculation unit 302 receives the input of the first torque T1, the first torque command value T1*, the d-axis magnetic flux Φd, the d-axis magnetic flux command value Φd*, and the q-axis magnetic flux command value ΦqLPF* after filtering, and calculates the second difference ΔT2. Details will be explained later with reference to Fig. 5 described.

[0046] Here, a change of a torque sign is described with reference to Fig. 4 described.

[0047] Fig. 4 is a diagram illustrating an operating point of torque in coordinates represented by a d-axis magnetic flux and a q-axis magnetic flux in the present embodiment.

[0048] Fig. Figure 4 illustrates the transition of the operating point when a torque sign changes from positive to negative. The actual torque represented by the d-axis magnetic flux Φd is an operating point C, and the target torque represented by the d-axis magnetic flux command value Φd* is an operating point D. In addition, a d-axis magnetic flux Φd0 represents the d-axis magnetic flux Φd at a point where the q-axis magnetic flux Φq is 0.

[0049] During single-pulse control, the voltage amplitude is fixed and the magnetic flux amplitude is also fixed. Therefore, the operating point moves, as shown in Fig. 4, on a circumference defined by a magnetic flux-limiting circle Φ (= sqrt(Φd^2 + Φq^2)). This means that since the voltage amplitude is fixed during single-pulse control, the amplitude of a magnetic flux is also fixed. Therefore, a magnetic flux moves around a circumference with a constant magnetic flux. If a path running along this circumference with a constant magnetic flux can be simulated, a travel distance ΔΦd of the operating point can be correctly calculated.

[0050] In a case where a torque command changes gradually from positive to negative, as in Fig. 4, the current operating point C indicated by the d-axis magnetic flux Φd moves on a circumference of the magnetic flux limiting circle Φ and moves via the d-axis magnetic flux Φd0 to the operating point D indicated by the d-axis magnetic flux command value Φd*. Therefore, the moving distance ΔΦd of the operating point is expressed by Equation (5) below. ΔΦd=Φd0−Φd+Φd0−Φd*=2Φd0−Φd*−Φd

[0051] The second difference ΔT2 is expressed by equation (6) below using ΔΦd. ΔT2=ΔΦd×ΦqLPF*

[0052] By applying equation (5) to equation (6), the second difference ΔT2 can be calculated using equation (7) below when a torque sign change occurs. ΔT2=(2Φd0−Φd*−Φd)ΦqLPF*

[0053] Since the d-axis magnetic flux Φd0 when the q-axis magnetic flux Φq = 0 is the d-axis magnetic flux Φd, the magnetic flux limit Φlim can be calculated using equation (8) below when Φd = Φd0 and Φq = 0 are substituted into equation (1). Φlim=sqrt(Φd02+02)=Φd0

[0054] Therefore, in the present embodiment, in a case where the torque sign changes from positive to negative or from negative to positive (T1 × T1* < 0), a difference is corrected by calculating the second difference ΔT2 using Equation (9) below. In a case where the torque sign does not change (T1 × T1* >= 0), the second difference ΔT2 is calculated using Equation (10) below. ΔT2=(2Φlim−Φd*−Φd)×ΦqLPF* ΔT2=(Φd*−Φd)×ΦqLPF*

[0055] Fig. 5 is a block configuration diagram of the second difference calculation unit 302 according to the first embodiment.

[0056] The second difference calculation unit 302 is configured to calculate the second difference ΔT2 in a case where T1 × T1* < 0 based on equation (9) or in a case where T1 × T1* >= 0 based on equation (10).

[0057] A doubler 312 doubles the magnetic flux limit value Φlim, and an adder / subtractor 322 subtracts the d-axis magnetic flux command value Φd* from the doubled magnetic flux limit value Φlim and outputs a d-axis correction magnetic flux command value Φd** to one input of a switch 332. The d-axis magnetic flux command value Φd* is fed to another input of the switch 332.

[0058] A multiplier 342 multiplies the first torque T1 by the first torque command value T1*. A sign determiner 352 determines a sign of this multiplication result and outputs a result to the switch 332. Note that the sign determiner 352 may employ another determination method, such as receiving the first torque T1 and the first torque command value T1* to determine whether their signs match, regardless of a multiplication result by the multiplier 342. Upon receiving the determination of the sign determiner 352, the switch 332 outputs the d-axis correction magnetic flux command value Φd** in a case where the sign of the torque command value is inverted. In a case where the sign of the torque command value is not inverted, the d-axis magnetic flux command value Φd* is output.That is, the switch 332 outputs the d-axis correction magnetic flux command value Φd** instead of the d-axis magnetic flux command value Φd* during a period in which the sign of the torque command value is inverted.

[0059] An adder / subtractor 362 subtracts the d-axis magnetic flux Φd from the d-axis magnetic flux command value Φd* or the d-axis correction magnetic flux command value Φd** output from the switch 332 to calculate the travel distance ΔΦd of the operating point. The travel distance ΔΦd is multiplied by the q-axis magnetic flux command value ΦqLPF* after filtering with a multiplier 372 and is output as the second difference ΔT2.

[0060] That is, it is determined whether the sign of the torque command is inverted with respect to a current operating point of the synchronous motor 3, and the second difference ΔT2 is calculated by Equation (9) in a case where the sign changes, and by Equation (10) otherwise. This allows a first-order deceleration response to be realized even in a case where the sign of the torque command changes.

[0061] The Fig. 6(A), Fig. 6(B), Fig. 6(C), Fig. 6(D) and Fig. 6(E) are graphs showing response waveforms in a case where the present embodiment is applied. Fig. 6(A) shows the d-axis current id and the d-axis current setpoint id*, Fig. 6(B) shows the q-axis current iq and the q-axis current setpoint iq*, Fig. 6(C) shows the first torque T1 and the first torque setpoint T1*, Fig. 6(D) shows the d-axis magnetic flux Φd and the d-axis magnetic flux command value Φd*, and Fig. Figure 6(E) shows the phase current. In each graph, the horizontal axis represents time and the vertical axis represents a value normalized with a maximum value set to 1. In addition, in the Fig. In the graphs shown in Figures 6(A) to 6(D), the setpoint is shown by a dashed line and a response waveform of a setpoint is shown by a solid line. Fig. In the graph shown in Fig. 6(E), the U-phase current iu is shown by a solid line, the V-phase current iv is shown by a dashed line, and the W-phase current iw is shown by a dotted line with a short dot pitch.

[0062] As in Fig. As shown in Figure 6(C), each response waveform is an example of a case where the first torque command value T1* changes gradually from positive to negative in a situation where the first torque T1, which is a current operating point, is negative. In this case, as shown in Fig. 6(D), during a period Tn in which the sign of the torque command is inverted, the d-axis correction magnetic flux command Φd** (see Fig. 5) instead of the d-axis magnetic flux command value Φd*. Note that in Fig. 6(D) the d-axis magnetic flux command value Φd* is shown by a dashed line, but a dashed line in the period Tn in which the sign of the torque command value is inverted shows the d-axis correction magnetic flux command value Φd**.

[0063] As in Fig. As shown in Figure 6(C), the first torque T1 responding to the first torque command T1* is a first-order delay response. As shown in Fig. 6(B), the q-axis current iq is also a first-order delay response. As shown in Fig. As shown in Figure 6(A), the d-axis current id once approaches 0 A but does not oscillate, and then responds to the d-axis current command value id*, which is the command current. That is, it can be seen that the operation is performed on the circumference of the magnetic flux limiting circle Φ. Furthermore, as shown in Fig. 6(E), the amplitude of the phase current at the time of the response does not change. As described above, even in a case where the sign of the torque command changes, no oscillating response is obtained, and a first-order delay response can be realized. A case where the present embodiment is not applied will be described later with reference to Fig. 11 described.

[0064] Fig. 7 is a block configuration diagram showing a variant of the second difference calculation unit 302 in the first embodiment.

[0065] A difference to the Fig. 5 is that the d-axis magnetic flux Φd0 is used for feeding into the doubler 312. The same section as in Fig. 5 is designated by the same reference numeral, and the description of the section is simplified.

[0066] Equation (8) shows a method for calculating Φd0 as the magnetic flux limit Φlim calculated by Equation (1). This is because it is assumed that a voltage drop across the motor winding resistance in a high-speed region is sufficiently small with respect to the speed electromotive force and can be approximated as Φd0 ≈ Φlim as in Equation (8).

[0067] In a case where the voltage drop across the motor winding resistance R cannot be ignored, the magnetic flux limit Φlim in equation (1) is expressed by equation (11) below. Φlim=sqrt((R×id / ω−Φq)2+(R×iq / ω+Φd)2)

[0068] Since the d-axis magnetic flux Φd0 is the d-axis magnetic flux Φd when the q-axis magnetic flux Φq = 0 (iq = 0), substituting Φd = Φd0, Φq = 0, and iq = 0 into Equation (11), we obtain Equation (12) below. The d-axis current id0 represents the d-axis current id at the point of the d-axis magnetic flux Φd0. ϕd0=sqrt(Φlim2−(R×id0 / ω)2)

[0069] The second difference ΔT2 can be calculated by substituting the d-axis magnetic flux Φd0 expressed by Equation (12) into Equation (7). However, to directly implement Equation (12), it is necessary to calculate the d-axis current id0, and it is difficult to perform the calculation in real time considering the nonlinearity of the motor inductance and the like. Therefore, the d-axis magnetic flux Φd0 is calculated taking into account the motor winding resistance R by referencing the d-axis magnetic flux Φd0 in a lookup table 382 based on the magnetic flux limit value Φlim and the DC voltage Vdc calculated by Equation (1).

[0070] As in Fig. As shown in Figure 7, the lookup table 382 outputs the d-axis magnetic flux Φd0 corresponding to the input magnetic flux limit value Φlim and the DC voltage Vdc. Details of the lookup table 382 will be described later. The d-axis magnetic flux Φd0 is input to the doubler 312, and the second difference ΔT2 is calculated in the same manner as described below with reference to Fig. 5 described.

[0071] Fig. 8 is a diagram showing an example of the lookup table 382 according to a variation of the first embodiment. The x-axis represents the magnetic flux limit value Φlim, the y-axis represents the DC voltage Vdc, and the z-axis represents the d-axis magnetic flux Φd0.

[0072] Assuming that the magnetic flux limit Φlim in equation (12) is a constant value, the rotational angular velocity ω is large when the DC voltage Vdc is high, and the influence of the voltage drop of the second term on the right is small, so the d-axis magnetic flux Φd0 is basically the same as the magnetic flux limit Φlim. Conversely, when the DC voltage Vdc is small, the rotational angular velocity ω decreases and the influence of the voltage drop increases, so the d-axis magnetic flux Φd0 is small with respect to the magnetic flux limit Φlim. For this reason, as shown in Fig. 8, a table in which, when the DC voltage Vdc is low, a change in the d-axis magnetic flux Φd0 with respect to the DC voltage Vdc is large, and when the DC voltage Vdc is high, the d-axis magnetic flux Φd0 becomes substantially constant.

[0073] Fig. 9 is a diagram illustrating a torque operating point in coordinates represented by a d-axis magnetic flux and a q-axis magnetic flux in a comparative example. This comparative example is a diagram illustrating an example to which the present embodiment is not applied, and it is a diagram to enhance understanding of the present embodiment.

[0074] Similar to Fig. 4 illustrates Fig. 9 shows the transition of the operating point when a torque sign changes from positive to negative. The actual torque represented by the d-axis magnetic flux Φd is an operating point C, and the target torque represented by the d-axis magnetic flux command value Φd* is an operating point D. In addition, a d-axis magnetic flux Φd0 represents the d-axis magnetic flux Φd at a point where the q-axis magnetic flux Φq is 0.

[0075] As mentioned with reference to Fig. As described in Figure 4, the voltage amplitude is fixed during single-pulse control, and a magnetic flux moves on a circumference shown by the magnetic flux limiting circle Φ (= sqrt(Φd^2 + Φq^2)) with a constant magnetic flux. Therefore, if a path running on the circumference of the same magnetic flux can be simulated, the moving distance ΔΦd of the operating point can be correctly calculated.

[0076] In a case where a torque setpoint changes gradually from positive to negative, such as in the case of Fig. 4, the current operating point C indicated by the d-axis magnetic flux command value Φd passes through a circumference of the magnetic flux limiting circle Φ and moves via the d-axis magnetic flux Φd0 to the operating point D indicated by the d-axis magnetic flux command value Φd*. In order to correctly calculate the moving distance ΔΦd, it is necessary to use Equation (6) as described above.

[0077] However, in the comparative example to which the present embodiment is not applied, the moving distance ΔΦd is simply calculated as in equation (13) below regardless of a change in the torque sign. ΔΦd=Φd*−Φd

[0078] This means that in the comparison example, the movement distance ΔΦd of the operating point, which runs on the circumference of the same magnetic flux, cannot be calculated correctly.

[0079] Then the second difference ΔT2 is expressed by equation (14) below from equation (6) above. ΔT2=(Φd*−Φd)ΦqLPF*

[0080] Fig. 10 is a block configuration diagram illustrating the second difference calculation unit 302 in the comparative example. This comparative example is a block configuration diagram in the case of the embodiment described with reference to Fig. 9 described comparative example.

[0081] Fig. Figure 10 shows a configuration in which the second difference ΔT2 is calculated based on equation (14). The d-axis magnetic flux Φd is subtracted from the d-axis magnetic flux command value Φd* by the adder / subtractor 362 to calculate the moving distance ΔΦd of the operating point. The moving distance ΔΦd is multiplied by the q-axis magnetic flux command value ΦqLPF* after filtering by a multiplier 372 and is output as the second difference ΔT2.

[0082] In the Fig. 9 and Fig. In the comparative example shown in Figure 10, a difference between the d-axis magnetic flux Φd and the d-axis magnetic flux command value Φd* is simply calculated as shown in Equation (13), regardless of a change in the torque sign, so the second difference ΔT2 is calculated to be small. As a result, in a case where the sign of the torque command value changes, an expected step response cannot be realized.

[0083] The Fig. 11(A), Fig. 11(B), Fig. 11(C), Fig. 11(D) and Fig. 11(E) are graphs showing the response waveforms in the comparative example. The case of the Fig. 9 and Fig. 10 shown comparative example.

[0084] Fig. 11(A) shows the d-axis current id and the d-axis current setpoint id*, Fig. 11(B) shows the q-axis current iq and the q-axis current setpoint iq*, Fig. 11(C) shows the first torque T1 and the first torque setpoint T1*, Fig. 11(D) shows the d-axis magnetic flux Φd and the d-axis magnetic flux command value Φd*, and Fig. Figure 11(E) shows the phase current. In each graph, the horizontal axis represents time and the vertical axis represents a value normalized with a maximum value set to 1. In addition, the graphs in the Fig. 11(A) to 11(D), the setpoint is shown by a dashed line and a response waveform of a setpoint is shown by a solid line. Fig. In the graph shown in Fig. 11(E), the U-phase current iu is shown by a solid line, the V-phase current iv is shown by a dashed line, and the W-phase current iw is shown by a dotted line with a short dot pitch.

[0085] As in Fig. As shown in Fig. 11(C), each response waveform is an example of a case where the first torque command value T1* changes gradually from positive to negative in a situation where the first torque T1, which is a current operating point, is negative.

[0086] In the present embodiment, as shown in Fig. 6(D), the d-axis magnetic flux command value Φd* calculated based on a predetermined correction value is input for a predetermined period of time by changing the torque sign. On the other hand, in this comparative example, such control is not performed.

[0087] As in Fig. As shown in Figure 11(C), the first torque T1, which responds to the first torque command T1*, has an oscillating response. As shown in Fig. 11(B), the q-axis current iq also exhibits an oscillatory response. As shown in Fig. 11(A), the oscillation of the d-axis current id increases and then responds to the d-axis current command value id*, which is the target current. In addition, as shown in Fig. 11(E), the amplitude of the phase current also increases at the time of response. This indicates that there is a possibility of overcurrent. As described above, in a case where the sign of the torque command changes, an undershoot (overshoot) of the first torque T1 and the q-axis current iq occurs, and a first-order delay response cannot be realized.

[0088] According to the first embodiment, in a case where the sign of the torque command value is inverted from the current operating point of the synchronous motor 3, by correctly calculating a difference between the torque of the synchronous motor 3 and a command value (d-axis magnetic flux command value Φd*), overcurrent and undershoot (overshoot) can be suppressed to realize a first-order deceleration response, and as a result, stable control of the synchronous motor becomes possible. [Second embodiment]

[0089] In the first embodiment, the example in which the d-axis magnetic flux command value Φd* is used as the second torque command value will be described. In the second embodiment, an example in which the d-axis current command value id* is used as the second torque command value will be described.

[0090] The block configuration diagram of the synchronous motor control device 1 of Fig. 1 and the block configuration diagram of the control unit 100 of Fig. 2 shown in the first embodiment have the same configuration in the second embodiment.

[0091] Fig. 12 is a block configuration diagram of the difference calculation unit 204 according to the present embodiment. The difference calculation unit 204 includes a first difference calculation unit 301, a second difference calculation unit 302, and a low-pass filter 303. The first difference calculation unit 301 and the low-pass filter 303 are similar to those of the first embodiment. The second difference calculation unit 302 is different from that of the first embodiment.

[0092] The second difference calculation unit 302 uses the first torque T1, the first torque command value T1*, the d-axis current command value id*, the q-axis current command value iq*, and the q-axis magnetic flux command value ΦqLPF* after filtering as input to calculate the second difference ΔT2. Details will be described later with reference to Fig. 14 described.

[0093] Fig. Figure 13 is a diagram showing an operating point of torque in coordinates represented by the d-axis current and q-axis current.

[0094] Fig. Figure 13 illustrates the transition of the operating point when a torque sign changes from positive to negative. The actual torque, represented by the d-axis current id, is the operating point C, and the target torque, represented by the d-axis current command value id*, is the operating point D. In addition, the d-axis current id0 represents the d-axis current id when the q-axis current iq is 0.

[0095] As described above, the operating point moves on a magnetic flux limiting ellipse Φidq because the voltage amplitude is fixed during single-pulse control. The magnetic flux limiting ellipse Φidq specifies the magnetic flux limiting circle Φ (= Φlim) through coordinates represented by the d-axis current and the q-axis current, and is expressed by Equation (15) below when a voltage drop across the winding resistance R of the synchronous motor 3 is sufficiently small with respect to the speed electromotive force. Φidq=Φlim=sqrt((−Lq×iq)2+(Ld×id+Φm)2)

[0096] Where Ld is the d-axis inductance, Lq is the q-axis inductance and Φm is a magnetic flux coefficient.

[0097] In a case where the torque command changes gradually from positive to negative, the current operating point C runs on the magnetic flux limiting ellipse and moves via the d-axis current command id0 to the operating point D indicated by the d-axis current command id*. Taking this process into account, in a case where the torque sign changes from positive to negative or from negative to positive (T1 × T1* < 0), a difference is corrected by calculating the second difference ΔT2 using Equation (16) below. In a case where the torque sign does not change (T1 × T1* >= 0), the second difference ΔT2 is calculated using Equation (17) below. However, instead of ΦqLPF, the q-axis current iq can also be used, and a value obtained by applying a low-pass filter that simulates a current control response to the q-axis current iq can be used. ΔT2=(2id0−id*−id)×ΦqLPF* ΔT2=(id*−id)×ΦqLPF*

[0098] Here, the d-axis current id0 is expressed by Equation (18) below by substituting id = id0 and iq = 0 in Equation (15). id0=(Φlim−Φm) / Ld

[0099] In general, the synchronous motor 3 used in an electric vehicle is greatly affected by magnetic saturation, and Φm and Ld change depending on the current. Considering the above, in the present embodiment, the d-axis current id0 is referenced in a lookup table based on Φlim calculated by Equation (1).

[0100] Fig. 14 is a block configuration diagram of the second difference calculation unit 302 according to the second embodiment.

[0101] The second difference calculation unit 302 is configured to calculate the second difference ΔT2 in a case where T1 × T1* < 0 based on equation (16) or in a case where T1 × T1* >= 0 based on equation (17).

[0102] A lookup table 392 outputs the d-axis current id0 corresponding to the applied magnetic flux limit value Φlim. The d-axis current id0 is fed into the doubler 312.

[0103] The doubler 312 doubles the d-axis current id0, and the adder / subtractor 322 subtracts the d-axis current command id* from the doubled d-axis current id0 and outputs a d-axis correction current command id** to one input of the switch 332. The d-axis current command id* is input to another input of the switch 332.

[0104] A multiplier 342 multiplies the first torque T1 by the first torque command value T1*. The sign determiner 352 determines this multiplication result and outputs a result to the switch 332. Upon receiving the determination of the sign determiner 352, the switch 332 outputs the d-axis correction current command value id** in a case where the sign of the torque command value is inverted. If the sign of the torque command value is not inverted, the d-axis current command value id* is output. That is, during a period in which the sign of the torque command value is inverted, the switch 332 outputs the d-axis correction current command value id** instead of the d-axis current command value id*.

[0105] The adder / subtractor 362 subtracts the d-axis current from the d-axis current command value id* or the d-axis correction current command value id** output from the switch 332 to calculate a moving distance Δid of the operating point. The moving distance Δid is multiplied by the q-axis magnetic flux command value ΦqLPF* after filtering by the multiplier 372 and is output as the second difference ΔT2.

[0106] That is, it is determined whether the sign of the torque command is inverted with respect to a current operating point of the synchronous motor 3, and the second difference ΔT2 is calculated by Equation (16) in a case where the sign changes, and by Equation (17) otherwise. This allows a first-order deceleration response to be realized even in a case where the sign of the torque command changes.

[0107] Fig. 15 is a block configuration diagram illustrating a variant of the second difference calculation unit 302 in the second embodiment.

[0108] At the Fig. In the second difference calculation unit 302 shown in Figure 14, only Φlim is used to reference the d-axis current id0. Fig. 15 is referenced to the d-axis current id0 using the magnetic flux limit Φlim and the DC voltage Vdc taking into account a voltage drop across the winding resistance R of the synchronous motor 3. The same section as in Fig. 14 is designated by the same reference numeral, and the description of the section is simplified.

[0109] If the calculation is performed taking into account a voltage drop across the motor winding resistance R in equation (15), equation (19) below is obtained. Φidq=Φlim=sqrt((R×id / ω−Lq×iq)2+((R×iq / ω−Ld×id+Φm)2)

[0110] Substituting id = id0 and iq = 0 into equation (19), we obtain equation (20) below. Φlim=sqrt((R×id0 / ω)2+(Ld×id0+Φm)2)

[0111] Assuming that the magnetic flux limit Φlim in Equation (20) is a constant value, when the DC voltage Vdc is high, the rotational angular velocity ω is large, and the influence of the voltage drop of the first term on the right is small, so the d-axis current id0 is basically consistent with Equation (18). Conversely, when the DC voltage Vdc is small, the rotational angular velocity ω decreases and the influence of the voltage drop increases.

[0112] Fig. 16 is a diagram illustrating an example of a lookup table 392' according to the second embodiment. The x-axis represents the magnetic flux limit value Φlim, the y-axis represents the DC voltage Vdc, and the z-axis represents the d-axis current id0.

[0113] As in Fig. 16, in a case where the DC voltage Vdc is low, a change in the d-axis current id0 with respect to the DC voltage Vdc is large, and in a case where the DC voltage Vdc is high, the d-axis current id0 is substantially constant.

[0114] According to the second embodiment, in a case where the sign of the torque command value is inverted from the current operating point of the synchronous motor 3, by correctly calculating a difference between the torque of the synchronous motor 3 and a command value (d-axis current command value id*), overcurrent and undershoot (overshoot) can be suppressed to realize a first-order deceleration response, and as a result, stable control of the synchronous motor becomes possible. [Reason for using the second torque setpoint]

[0115] The following describes the reason for using the second torque command value, which gradually increases in a range including a peak of the torque of the synchronous motor 3, as the torque command value T*.

[0116] Fig. Figure 17 is a graph illustrating a relationship between the voltage phase angle θv and the torque output by the synchronous motor 3. The horizontal axis represents the voltage phase angle θv, and the voltage phase angle θv is 0 degrees with respect to the q-axis. The vertical axis represents the torque.

[0117] The torque increases monotonically with respect to the voltage phase angle θv up to the maximum torque. However, when the maximum torque is exceeded, the torque decreases. Further, the torque decreases monotonically with respect to the voltage phase angle θv up to the minimum torque. However, when the minimum torque is exceeded, the torque increases again. For this reason, it is necessary to perform control in a range where the voltage phase angle θv does not exceed the maximum torque or the minimum torque, and it is necessary to limit a torque command (or the voltage phase angle θv) with a margin vo of the maximum torque or the minimum torque.For this reason, in a case where the voltage phase angle θv is operated based on the actual torque, the estimated torque calculated from a current value, or the like, the torque that can actually be used is limited with respect to the torque that can actually be output from the synchronous motor 3.

[0118] To solve this, it is conceivable to control the voltage phase angle θv by using a physical quantity that increases monotonically even near the maximum (minimum) torque, that is, the second torque command. For example, the d-axis magnetic flux command Φd* and the d-axis current command id* are used.

[0119] Fig. Figure 18 is a graph illustrating a relationship between the voltage phase angle θv and the d-axis magnetic flux Φd. Fig. Figure 19 is a graph illustrating a relationship between the voltage phase angle θv and the d-axis current id.

[0120] As in the Fig. 18 and Fig. As shown in Figure 19, in the case of the d-axis magnetic flux Φd or the d-axis current id, the voltage phase angle θv increases monotonically from 0 degrees to 180 degrees (-180) degrees, so that the torque can be stably controlled even near the maximum (minimum) torque.

[0121] However, in a case where the voltage phase angle θv is controlled based on the d-axis magnetic flux Φd and the d-axis current id, and in a case where the sign of the torque command is inverted from the current operating point of the synchronous motor 3, a difference between the torque of the synchronous motor 3 and the command value is not correctly calculated, and the control of the synchronous motor 3 becomes unstable. In the present embodiment, as described above, in a case where the sign of the torque command is inverted from the current operating point of the synchronous motor 3, a difference between the torque of the synchronous motor 3 and the command value can be correctly calculated and corrected, no overshoot occurs, a first-order deceleration response as designed can be realized, and the synchronous motor 3 can be stably controlled.

[0122] In the first embodiment and the second embodiment, the d-axis magnetic flux command value Φd* and the d-axis current command value id* are described as examples of the second torque command value. Without being limited to the d-axis magnetic flux command value Φd* and the d-axis current command value id*, the second torque command value may be a value within a range that includes the maximum torque or the minimum torque shown in the graph of Fig. 17, gradually increases or decreases, or a value obtained by combining these values ​​may be used. Furthermore, in the first embodiment and the second embodiment, the example in which a difference is corrected by calculating the second torque command value based on a magnetic flux limit value will be described. However, the present invention is not limited to the magnetic flux limit value, and another correction value may also be used. The other correction value only needs to be a parameter that represents the moving distance ΔΦd of the torque command value calculated with reference to Fig. 4, and may in some cases be a constant corresponding to a characteristic of the synchronous motor 3, and, in short, the other correction value need only be a value for correcting the difference to a large value in a case where the sign of the torque command is inverted. [Third Embodiment]

[0123] Fig. 20 is a configuration diagram of an electric vehicle 1000 in a third embodiment.

[0124] The Fig. The synchronous motor control device 1 shown in Fig. 20 is the synchronous motor control device 1 described in the first and second embodiments. The synchronous motor control device 1 converts DC power from the DC power supply 2 into AC power to drive the synchronous motor 3.

[0125] The synchronous motor 3 is connected to a transmission 601. The transmission 601 is connected to a drive shaft 603 via a differential gear 602 and supplies power to a wheel 604. Note that a configuration in which the transmission 601 is not provided and the connection is made directly to the differential gear 602, or a configuration in which the synchronous motor 3 and the synchronous motor controller 1 are attached to each of a front wheel and a rear wheel, may be used.

[0126] In automotive applications, for example, high-speed torque is related to acceleration performance on a highway. In particular, when high-speed torque becomes insufficient on an uphill road, the speed cannot be increased, and therefore, there is a strong demand for improving high-speed torque. Furthermore, there is a need to reduce the size and weight of the synchronous motor 3 from the perspective of improving power consumption, and the synchronous motor control device 1 capable of realizing the maximum torque that can be output by the synchronous motor 3 is important.Similarly, a moving body in a railway is the same as in an automobile, and the acceleration performance during high-speed travel affects the travel time between stations and is therefore similarly important. Even in a case where the sign of the torque command is inverted from the current operating point of the synchronous motor 3, a difference between the torque of the synchronous motor 3 and a command value is correctly corrected to stabilize the control of the synchronous motor 3. In the electric vehicle 1000, in a scene where the sign of the torque command changes, vibration suppression control, gear shifting, and the like can be considered.By applying the synchronous motor control device 1 described in the first and second embodiments as the synchronous motor control device 1, it is possible to improve the acceleration performance in a high-speed range and the stability of control in an automobile and a railway.

[0127] In the first to second embodiments, the synchronous motor control device 1 will be described with reference to the block configuration diagram. Some or all of the blocks except the power conversion unit 400 in these block configuration diagrams may be realized by a processor (for example, CPU or GPU) and a program executed by the processor. In this case, the program is executed by the processor to perform predetermined processing using a storage resource (for example, a memory) and / or an interface device (for example, a communication port) and the like accordingly, and thus the subject of the processing may be the processor. Similarly, the subject of the processing performed by executing the program may be a controller, a device, a system, a computer, or a node including a processor.The object of the processing performed by executing the program need only be a computing unit and may include a special circuit (for example, FPGA or ASIC) that performs a specific processing.

[0128] The program can be installed on a device such as a computer from a program source. The program source can be, for example, a program distribution server or a computer-readable storage medium. In a case where the program source is a program distribution server, the program distribution server can include a processor and a memory resource that stores a distribution target program, and the processor of the program distribution server can distribute the distribution target program to another computer. Further, in the description below, two or more programs can be implemented as one program, or one program can be implemented as two or more programs.

[0129] According to the embodiment described above, an operating effect can be achieved as follows. (1) The synchronous motor control device 1 includes the torque command conversion unit 201 that converts the torque command T* for the synchronous motor 3 into the second torque command (d-axis magnetic flux command Φd* or d-axis current command id*) that gradually increases in a range including a peak of the torque of the synchronous motor 3, the difference calculation unit 204 that calculates a difference (the second difference ΔT2) between the torque of the synchronous motor 3 and the second torque command, the voltage phase control unit 206 that controls the voltage phase angle θv to reduce the difference, and the power conversion unit 400 that converts DC power into AC power based on the voltage phase angle θv and outputs the converted AC power to the synchronous motor 3, and the difference calculation unit 204 corrects the difference (second difference ΔT2) by adjusting the second torque setpoint in a casein which a sign of the torque command T* is inverted from a current operating point of the synchronous motor 3, based on a predetermined correction value (the magnetic flux limit Φlim). As a result, even in a case where a sign of a torque command changes, no overshoot occurs, and a first-order deceleration response as designed can be realized. (2) The synchronous motor control method is a synchronous motor control method in the synchronous motor control device 1 including the power conversion unit 400 that converts DC power into AC power based on the voltage phase angle θv and outputs the converted AC power to the synchronous motor 3. The synchronous motor control method includes converting the torque command T* for the synchronous motor 3 into the second torque command (d-axis magnetic flux command Φd* or d-axis current command id*) that gradually increases in a range including a peak of the torque of the synchronous motor 3, calculating a difference (the second difference ΔT2) between the torque of the synchronous motor 3 and the second torque command, controlling the voltage phase angle θv to reduce the difference, and the difference (second difference ΔT2) in a casein which a sign of the torque command is inverted from a current operating point of the synchronous motor 3, is corrected by calculating the second torque command based on a predetermined correction value (the magnetic flux limit value Φlim). As a result, even in a case where a sign of a torque command changes, no overshoot occurs, and a first-order deceleration response as designed can be realized.

[0130] The present invention is not limited to the above-described embodiment, and other forms conceivable within the scope of the technical idea of ​​the present invention are also included within the scope of the present invention, as long as the features of the present invention are not impaired. Furthermore, the above-described embodiments and variants can be combined with each other. List of reference symbols 1 synchronous motor control device 2 DC power supply 3 Synchronous motor 31 Magnetic pole position sensor 100 control unit 114 to 119 switching element 120 to 125 freewheeling diode 133 Wiring on the high potential side 134 Wiring on the low potential side 105, 106, 107 Phase wiring 200 voltage detector 201 Torque setpoint conversion unit 202 Three-phase and two-phase conversion unit 203 Magnetic flux calculation unit 204 Difference calculation unit 205 Speed ​​calculation unit 206 Voltage phase control unit 207 Square wave generating unit 208 PWM gate signal generation unit 300 Gate control circuit 382, 392, 392' Lookup table 400 power conversion unit 500 current detector 601 gearbox 602 differential gear 603 drive shaft 604 wheels 1000 electric vehicles T* Torque setpoint ω rotational angular velocity iu, iv, iw phase current GS gate signal Vdc DC voltage θv voltage phase angle id* d-axis current setpoint iq* q-axis current setpoint Φd* d-axis magnetic flux command value Φq* q-axis magnetic flux command value id d-axis current iq q-axis current Φd d-axis magnetic flux Φq q-axis magnetic flux Φlim magnetic flux limit ΦqLPF* q-axis magnetic flux command value after filtering ΔT1 first difference ΔT2 second difference Vu*, Vv*, Vw* Three-phase voltage setpoint QUOTES CONTAINED IN THE DESCRIPTION

[0000] This list of documents submitted by the applicant was generated automatically and is included solely for the convenience of the reader. This list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions. Cited patent literature

[0000] JP 2017-229127 A

[0004]

Claims

[1] Synchronous motor control device comprising: a torque command conversion unit that converts a torque command for a synchronous motor into a second torque command that gradually increases in a range including a peak of the torque of the synchronous motor; a difference calculation unit that calculates a difference between the torque of the synchronous motor and the second torque command value; a voltage phase control unit that controls a voltage phase angle to reduce the difference; and a power conversion unit that converts DC power into AC power based on the voltage phase angle and outputs the converted AC power to the synchronous motor, wherein the difference calculation unit corrects the difference in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor by calculating the second torque command based on a predetermined correction value. [2] Synchronous motor control device according to claim 1, wherein the torque command conversion unit generates a d-axis magnetic flux command and a q-axis magnetic flux command as the second torque command, the difference calculation unit calculates the difference based on a d-axis magnetic flux and a q-axis magnetic flux based on the current of the synchronous motor and based on the d-axis magnetic flux command value and the q-axis magnetic flux command value, and the difference calculation unit further corrects the difference in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor by calculating the d-axis magnetic flux command based on the correction value. [3] Synchronous motor control device according to claim 1, wherein the torque command conversion unit generates a d-axis current command and a q-axis current command as the second torque command, the difference calculation unit calculates the difference based on the d-axis current and the q-axis current based on the current of the synchronous motor and based on the d-axis current command value and the q-axis current command value, and the difference calculation unit further corrects the difference in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor by calculating the d-axis current command based on the correction value. [4] The synchronous motor control device according to claim 2 or 3, wherein the correction value is a magnetic flux limit value calculated based on the DC voltage supplied to the power conversion unit and the rotational angular velocity of the synchronous motor. [5] The synchronous motor control device according to claim 1, wherein the difference calculation unit determines that a sign of the torque command value is inverted from a current operating point of the synchronous motor based on a sign of a multiplication result of the torque of the synchronous motor and the second torque command value. [6] Electric vehicle that has: the synchronous motor control device according to one of claims 1 to 3; and the synchronous motor controlled by the synchronous motor control device, The electric vehicle is supplied with energy by the synchronous motor. [7] A synchronous motor control method in a synchronous motor control device including a power conversion unit that converts DC power into AC power based on a voltage phase angle and outputs the converted AC power to a synchronous motor, the synchronous motor control method comprising: Converting a torque command for the synchronous motor into a second torque command that gradually increases in a range including a peak of the torque of the synchronous motor; Calculating a difference between the torque of the synchronous motor and the second torque setpoint; Controlling the voltage phase angle to reduce the difference; and Correcting the difference by calculating the second torque command based on a predetermined correction value in a case where a sign of the torque command is inverted from a current operating point of the synchronous motor.

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