TEMPERATURE SENSOR

Description

Domaine technique

[0001] This description relates generally to electronic circuits, and, more specifically, to temperature sensors. Technique antérieure

[0002] Many integrated circuits, for example those based on silicon, include a temperature sensor. This temperature sensor, or temperature measurement circuit, allows the temperature of the integrated circuit to be measured, for example to allow adjustment of the integrated circuit's parameters, such as its supply voltage, based on the measured temperature.

[0003] A known type of temperature sensor uses a differential measurement of a time-domain quantity, either between a constant time-domain quantity that is a function of temperature and a time-domain quantity that varies with temperature, or between two time-domain quantities that vary with temperature but have very different temperature coefficients. An example of a time-domain quantity is the frequency of an oscillator. Such temperature sensors are disclosed, for example, by US patent applications 2012 / 170616 A1 and CN 104 568 208 A.

[0004] However, known temperature sensors based on differential frequency measurement have drawbacks. Résumé de l'invention

[0005] There is a need to overcome all or part of the drawbacks of known integrated temperature sensors based on differential frequency measurement.

[0006] An embodiment corresponding to the invention as claimed overcomes all or part of the disadvantages of known integrated temperature sensors based on differential frequency measurement.

[0007] The embodiment corresponding to the invention as claimed provides for a temperature sensor comprising: The sensor consists of a first oscillator and a second oscillator; a counter configured to count a number of periods of the second oscillator during a duration determined by one period of the first oscillator; and a calculation circuit configured to determine a temperature value from said counted number (P). The sensor is characterized in that: one of the first and second oscillators is configured so that its frequency depends linearly on the temperature or the inverse of the temperature by a coefficient that is a multiple of the inverse of a resistance value of a resistive component of said first and second oscillators; said resistive component is implemented by a switched capacitor; and a control frequency of the switched capacitor is determined by the frequency of the other first and second oscillator.

[0008] According to one embodiment, said first and second oscillators comprise: an oscillating circuit; and a current source configured to provide a bias current to the oscillating circuit, the current source being of the absolute temperature proportional type or of the absolute temperature complementary type.

[0009] According to one embodiment: The current source includes the resistive component; the resistive component is configured to convert a voltage into a first current; and the current source is configured so that the bias current is a copy of the first current up to a multiplicative factor.

[0010] According to one embodiment, the oscillating circuit is a ring oscillator.

[0011] According to one embodiment, the oscillating circuit is a relaxation oscillator comprising an RS flip-flop.

[0012] According to one embodiment, the oscillating circuit includes the resistive component.

[0013] According to one embodiment: The oscillating circuit comprises at least one first inverter having its threshold voltage determining the frequency of said one of the first and second oscillators; and said one of the first and second oscillators further comprises: a second inverter identical to the first oscillator and having its output connected to its input, an error amplifier configured to provide a signal indicating a difference between an output voltage of the second inverter and a reference threshold voltage, and a circuit configured to control a threshold voltage of the first and second inverters from the signal provided by the error amplifier so that the threshold voltage of the first and second inverters is equal to the reference voltage.

[0014] According to one embodiment, the sensor includes a calibration circuit configured for: receive from the counter the number P of periods of the second oscillator counted during the duration determined by the period of the first oscillator; determine, when the calibration circuit receives a calibration request, and from a known calibration temperature and the number P, a constant coefficient J such that: a) P is equal to J times the calibration temperature if said one of the first and second oscillators is the first oscillator and has its frequency depending linearly on the inverse of the temperature or if said one of the first and second oscillators is the second oscillator and has its frequency depending linearly on the temperature;or b) P is equal to J times the inverse of the temperature if said one of the first and second oscillators is the second oscillator and has its frequency depending linearly on the inverse of the temperature or if said one of the first and second oscillators is the first oscillator and has its frequency depending linearly on the temperature. ;

[0015] According to one embodiment, the sensor is configured so that the duration determined by the frequency of the first oscillator is at least 100 times greater than one period of the second oscillator.

[0016] According to one embodiment, the sensor is configured so that the control frequency of the switched capacitance is at least 10 times greater than the frequency of said one of the first and second oscillators.

[0017] According to one embodiment, each of the first and second oscillators is quartz-free. Brève description des dessins

[0018] These features and advantages, as well as others, will be described in detail in the following description of particular embodiments, given by way of non-limiting example, in relation to the attached figures, among which: there figure 1 represents, in block form, an example of a temperature sensor; the figure 2 represents, in block form, one embodiment of a temperature sensor; the figure 3 represents, in block form, an embodiment of an oscillator for the sensor of the figure 2 ; there figure 4 represents an example of an embodiment of a current source for the oscillator of the figure 3 ; there figure 5 represents an example of an embodiment of an oscillating circuit of the oscillator of the figure 3 ; and the figure 6 represents an example of an oscillator circuit of the figure 3 . Description des modes de réalisation

[0019] The same elements have been designated by the same reference numerals in the different figures. In particular, structural and / or functional elements common to the different embodiments may have the same reference numerals and may have identical structural, dimensional and material properties.

[0020] For the sake of clarity, only the steps and elements necessary for understanding the described embodiments have been shown and detailed. In particular, the various known integrated circuits in which a temperature sensor is integrated have not been detailed, as the described embodiments and variants are compatible with these known integrated circuits.

[0021] Unless otherwise specified, when referring to two connected elements, this means directly connected without any intermediate elements other than conductors, and when referring to two coupled elements, this means that these two elements can be connected or linked through one or more other elements.

[0022] In the description that follows, when referring to absolute positional qualifiers, such as the terms "front", "back", "top", "bottom", "left", "right", etc., or relative positional qualifiers, such as the terms "above", "below", "superior", "inferior", etc., or to orientational qualifiers, such as the terms "horizontal", "vertical", etc., unless otherwise specified, it refers to the orientation of the figures.

[0023] Unless otherwise specified, the expressions "approximately", "roughly", "about", and "on the order of" mean within 10%, preferably within 5%.

[0024] There figure 1 represents, in block form, an example of a temperature sensor 1, and more particularly of a temperature sensor 1 based on a differential frequency measurement.

[0025] Sensor 1 includes a first oscillator LO1 and a second oscillator LO2, as well as a counter COUNTER.

[0026] The COUNTER is configured to count, over a duration D determined by the period of the first oscillator LO1 (i.e., by the period of an output signal from oscillator LO1), a number P of periods of the second oscillator. The duration D corresponds to a multiple, preferably an integer, of the period of oscillator LO1.

[0027] For example, the COUNTER includes an input I1 configured to receive a signal sig1 with a period determined by the period of the oscillator LO1; that is, a signal sig1 whose period is a multiple, preferably an integer, of the period of the output signal of the oscillator LO1. For example, the signal sig1 is the output signal of the oscillator LO1.

[0028] For example, the COUNTER also includes an input I2 configured to receive a number N, preferably an integer. The number N, together with the period of the signal sig1, determines the counting time D. Indeed, the COUNTER is configured so that the duration D is equal to N times T1, where T1 is the period of the signal sig1.

[0029] For example, the COUNTER also includes a CK input configured to receive the output signal sig2 from the LO2 oscillator.

[0030] For example, the COUNTER includes an output O configured to provide the number P.

[0031] At the end of a counting time D, the number P is equal to N.F2 / F1 with F1 the frequency of the signal sig1, and F2 the frequency of the signal sig2.

[0032] At least one of the oscillators LO1 and LO2 has its frequency which depends linearly on the temperature T or the inverse of the temperature (i.e. 1 / T).

[0033] For example, when the frequency F1 of the signal sig1, and therefore the frequency of the oscillator LO1, is independent of temperature, and the frequency F2 of the signal sig2 depends linearly on the temperature T according to the relation F2 = AT, with A being a constant coefficient, then P = NAT / F1. Since the frequency F1 is constant with temperature, the number N is known, and the coefficient A is constant, P = BT with B = NA / F1. By determining B, for example during a calibration step, the measured temperature T is directly equal to the ratio P / B.

[0034] For example, when the frequency F1 of the signal sig1, and therefore the frequency of the oscillator LO1, is independent of temperature, and the frequency F2 of the signal sig2 depends linearly on the inverse of the temperature T according to the relation F2 = A / T, with A a constant coefficient, then P = NA / (F1.T). Since the frequency F1 is constant with temperature, the number N is known, and the coefficient A is constant, P = B / T, with B = NA / F1. By determining B, for example during a calibration step, the measured temperature T is directly equal to the ratio B / P.

[0035] For example, when the frequency F2 of the signal sig2 is independent of temperature and the frequency F1 of the signal sig1, and therefore the frequency of the oscillator LO1, depends linearly on the temperature T according to the relation F1 = ET, with E a constant coefficient, then P = N.F2 / (ET). Since the frequency F2 is constant with temperature, the number N is known, and the coefficient E is constant, P = B / T with B = N.F2 / E. By determining B, for example during a calibration step, the measured temperature T is directly equal to the ratio B / P.

[0036] For example, when the frequency F2 of the signal sig2 is independent of temperature and the frequency F1 of the signal sig1, and therefore the frequency of the oscillator LO1, depends linearly on the inverse of the temperature T according to the relation F1 = E / T, with E a constant coefficient, then P = N.F2.T / E. Since the frequency F2 is constant with temperature, the number N is known, and the coefficient E is constant, P = BT with B = N.F2 / E. By determining B, for example during a calibration step, the measured temperature T is directly equal to the ratio P / B.

[0037] In the examples above, one of the oscillators must have a temperature-independent frequency. Such an oscillator can be implemented using a quartz crystal, but this is undesirable due to the complexity, size, and cost of sensor 1. An oscillator providing a temperature-independent frequency can also be implemented using a method other than a quartz crystal, but at the cost of even greater complexity for sensor 1.

[0038] Furthermore, even when using an oscillator whose intrinsic frequency is independent of temperature, in practice this oscillator must be biased with a biasing circuit that generally includes one or more resistors. These resistors have resistance values ​​that depend on temperature, meaning that, in practice, the oscillator's frequency will not be completely independent of temperature. It is therefore necessary to take at least two calibration points to determine the relationship between the temperature T (or the inverse of the temperature, 1 / T) and the number P counted.

[0039] In other examples, both frequencies F1 and F2 may depend on the temperature. At least two calibration points are then required to determine the measured temperature from the number P, which is undesirable.

[0040] There figure 2 represents, in block form, an embodiment of a temperature sensor 2 based on a differential measurement of the frequency of an oscillator.

[0041] Sensor 2, like sensor 1, comprises an oscillator LO1', an oscillator LO2', and the counter COUNTER. The counter COUNTER is, for example, identical to the one described in relation to the figure 1 In particular, the COUNTER is configured to count the number P of periods of the LO2' oscillator during a duration D determined by the period of the LO1' oscillator.

[0042] The LO1' oscillator provides a periodic signal sig3 with frequency F3. The frequency F3 determines the counting time D. More specifically, the time D is equal to N*T1, where T1 is the period of a signal sig1 with a frequency F1 determined by the frequency F3 of the signal sig3, and therefore by the frequency F3 of the LO1' oscillator. In other words, the frequency F1 of the signal sig1 is equal to 1 / Q times the frequency F3 of the signal sig3, where Q is a constant number greater than or equal to 1, preferably an integer. Thus, the time D is equal to NQ / F3 and is therefore determined by the period T3 of the signal sig3. The signal sig3 and the number N are supplied to the counter COUNTER.

[0043] For example, when the coefficient Q is strictly greater than 1, sensor 2 includes a DIVQ circuit configured to receive the signal sig3 and to provide the signal sig1 with a frequency F1 equal to 1 / Q times the frequency F3 of the signal sig3. In other words, the DIVQ circuit is a frequency divider by Q configured to provide the signal sig1 from the signal sig3.

[0044] As an alternative example, when the coefficient Q is equal to 1, the DIVQ circuit can be omitted, the signal sig1 then being directly the signal sig3.

[0045] As an example, the numbers N and Q are constant numbers fixed once for the entire operating time of sensor 2. As an alternative example, at least one of the numbers N and Q is a parameter that is set by the user, so that the user can change the value of this number, for example during the lifetime of sensor 2 and / or depending on the application using sensor 2.

[0046] The LO2' oscillator provides a periodic sig2 signal at frequency F2. The sig2 signal is supplied to the COUNTER.

[0047] For example, as in figure 1 , the signal sig1 is supplied to the input I1 of the COUNTER, the number N is supplied to the input I2 of the COUNTER, the signal sig2 is supplied to the input CK of the COUNTER and the output O of the counter supplied the number P.

[0048] In a manner similar to what has been indicated in relation to the figure 1 At the end of each counting period D, the number P is equal to N.F2 / F1, therefore to N'.F2 / F3 with N' equal to NQ

[0049] One of the oscillators LO1' and LO2' is configured so that its frequency depends linearly on the temperature T, or the inverse of the temperature (1 / T). Furthermore, this oscillator LO1' or LO2' is configured so that its frequency is partially determined by the inverse of a resistance value Rv of a resistive component R in this oscillator. In other words, this oscillator is configured so that the coefficient linearly relating the temperature T (or the inverse of the temperature T) to its frequency is a multiple of 1 / Rv, where Rv is the resistance value of the resistive component R.

[0050] In the example of the implementation method of the figure 2 , the LO2' oscillator is configured so that its frequency F2 depends linearly on the temperature T, according to a constant coefficient multiple of 1 / Rv, the component R then being part of this LO1' oscillator.

[0051] Furthermore, the resistive component R is implemented by a switched capacitor. In other words, component R is a switched capacitor. The switched capacitor R comprises a capacitive element C, a switch IT1 connected in series with component C between terminals 200 and 202 of the resistive component R, and a switch IT2 connected in parallel with element C. Switches IT1 and IT2 are controlled in opposite phase. The control frequency Fc of the switched capacitor R is determined by the frequency of the oscillator LO1' or LO2' that does not include element R, that is, by the frequency F3 of the oscillator LO1' in the example of the figure 2 In other words, the frequency Fc is equal to H times the frequency of the oscillator LO1' or LO2' not including the component R, that is to say, H times the frequency F3 in the example of the figure 1 , H being a constant positive number. Preferably, H is chosen so that the frequency Fc is at least ten times greater than the frequency of the oscillator LO1' or LO2' in which the component R is implemented, i.e., so that H.F3 > 10.F2 in the example of the figure 2 .

[0052] Thus, in the example of the figure 2 , where F2 is equal to GT / Rv with G a constant coefficient, as the value Rv of resistance of the resistive component R is equal to 1 / (Cv.Fc), with Cv the value of capacitance of the capacitive element C, then F2 = GTCv.Fc = GTCv.F3.H.

[0053] Substituting F2 with its expression above in the equation P = N'.F2 / F3, it follows that P is equal to N'.GTCv.H. Since N', G, Cv and H are constants, P is equal to JT, with J a constant coefficient and equal to N'.G.Cv.H.

[0054] Advantageously, the coefficient J does not depend on the frequency F3 of the LO1' oscillator. The frequency F3 of the LO1' oscillator can therefore depend on the temperature without changing the value of the coefficient J, and thus without distorting the measurement of T. Indeed, to determine T from P, it is then sufficient to know the coefficient J.

[0055] In one embodiment, the LO1' or LO2' oscillator that does not include component R also does not include a crystal, and, more generally, neither the LO1' nor LO2' oscillators include a crystal. This reduces the complexity, size, and cost of sensor 2 compared to a sensor 1 in which either the LO1 or LO2 oscillator includes a crystal.

[0056] In one embodiment, the oscillators LO1' and LO2' have the same structure or topology, which reduces the dependence of sensor 2 on the supply voltage and / or common-mode noise. For example, both oscillators LO1' and LO2' each include a ring oscillator, or each include a relaxation oscillator, for example, a relaxation oscillator including an RS flip-flop.

[0057] According to one embodiment, the value of the coefficient J is determined during a calibration phase. For example, when calibration is required, a number P is counted over a corresponding duration D, knowing the calibration temperature T1 at which the sensor 2 is located. The value of the coefficient J is then equal to P / T1 in the example of the figure 2 Advantageously, only one calibration point is needed.

[0058] According to one embodiment, the calibration phase is implemented by a calibration circuit for sensor 2 (not shown in figure 2 This calibration circuit is configured to receive the number P counted by the counter COUNTER for a duration D, and to determine, when the calibration circuit receives a calibration request corresponding to this counting duration D, the coefficient J relating the number P to the known calibration temperature T1. This assumes that the calibration request received by the calibration circuit is issued while sensor 2 is at the calibration temperature T1. For example, the calibration request is provided to sensor 2 and its calibration circuit by an operator who has previously placed sensor 2 at the calibration temperature T1. The value of the calibration temperature T1 can be known beforehand and stored in the calibration circuit's memory, or it can be provided to sensor 2 and its calibration circuit along with the calibration request.

[0059] Sensor 2 includes a calculation circuit (not shown in figure 2 The calculation circuit is configured to determine the value of the temperature T from the number P. More precisely, the calculation circuit is configured to determine the value of the temperature from the number P and the constant coefficient J relating P to the temperature T.

[0060] Preferably, to obtain sufficient measurement accuracy, the duration D is chosen to be at least 100 times greater than the duration of one period of the LO2' oscillator.

[0061] Preferably, the control frequency Fc of the switched capacitance R is at least 10 times greater than the frequency of the oscillator comprising this switched capacitance R, namely the frequency F2 of the LO2' oscillator in the example of the figure 2 Thus, in the example of the figure 2 Preferably, Fc > 10.F2.

[0062] There figure 3 represents, in block form, an example of an embodiment of a sensor oscillator. figure 2 More specifically, the figure 3 represents, in block form, an example of an embodiment of the oscillator of sensor 2 which includes the switched capacitance R, namely the LO2' oscillator, using the example of the figure 2 However, the LO1' oscillator can be implemented identically in the case where the switched capacitance R is part of the LO1' oscillator rather than the LO2' oscillator.

[0063] The LO2' oscillator includes an oscillating circuit (OC), for example, a relaxation oscillator which may include an RS flip-flop, or for example, a ring oscillator. The OC circuit provides the output signal of the oscillator, that is, the output signal sig2 of the LO2' oscillator in this example.

[0064] The LO2' oscillator further includes a 300 current source configured to provide a bias (or supply) Ibias current to the oscillating circuit OC.

[0065] The 300 current source is of the CTAT (Complementary To Absolute Temperature) or PTAT (Proportional To Absolute Temperature) type. Thus, the Ibias current varies linearly with temperature.

[0066] As an example, the 300 current source has one terminal connected to a VDD supply potential and another terminal connected to a OC circuit supply terminal.

[0067] According to one embodiment, although this is not illustrated by the figure 3 , the switched capacitance R ( figure 2 ) is part of the oscillating circuit OC.

[0068] As an alternative, although this is not illustrated by the figure 3 , the switched capacitance R ( figure 2 ) is part of the current source 300. Preferably, in such a variant, the resistive element R is configured to convert a voltage across terminals 200, 202 ( figure 2 The resistive element R converts a first current, and the current source 300 is configured so that its bias current varies proportionally, for example linearly, with this first current. More precisely, when the source 300 is of type PTAT or CTAT, respectively, the component R is configured to convert a PTAT voltage or CTAT voltage into a first PTAT current or CTAT current, respectively, and the source 300 is configured so that the bias current varies proportionally, for example linearly, with the first current.

[0069] There figure 4 represents an example of an embodiment of the oscillator's 300 current source figure 3 . In this example embodiment, the current source 300 is of type PTAT.

[0070] The 300 current source comprises two MOS (Metal Oxide Silicon) transistors, T1 and T2, connected in series between a supply potential VDD and a reference potential GND. Transistor T1 is connected to VDD, and transistor T2 is connected to GND. In this example, where VDD is positive relative to GND, transistor T1 is a P-channel MOS transistor, and transistor T2 is an N-channel MOS transistor. Specifically, the source of transistor T1 is connected to VDD, the drain of T1 is connected to the drain of T2, and the source of T2 is connected to GND. Furthermore, the drain of transistor T2 is connected to its gate.

[0071] The current source 300 further comprises a resistive element and two MOSFETs, T3 and T4, connected in series between the supply voltage VDD and the reference voltage GND. Transistor T3 is connected to VDD, and transistor T4 is coupled to GND via the resistive element. The series combination of transistors T1 and T2 is connected in parallel with the series combination of transistors T3 and T4 and the resistive element. In this example, where VDD is positive relative to GND, transistor T3 is a P-channel MOSFET, and transistor T4 is an N-channel MOSFET. For example, the source of transistor T3 is connected to VDD, the drain of transistor T3 is connected to the drain of transistor T4, and the source of transistor T4 is coupled to GND via the resistive element. Furthermore, transistor T3 has its drain connected to its gate.

[0072] In addition, transistors T1 and T3 have their gates connected to each other, and transistors T2 and T4 have their gates connected to each other.

[0073] The resistive element coupling transistor T4 to the GND potential is configured to convert a voltage across its terminals into a first current, and the current source 300 is configured so that its Ibias current copies this first current, that is to say so that the Ibias current is equal to the first current up to a multiplicative factor, this multiplicative factor being constant.

[0074] In this example embodiment, the resistive component R is part of the current source 300, and corresponds to the resistive element coupling transistor T4 to the GND potential.

[0075] In another unshown example where the resistive component R is part of the oscillating circuit OC, the resistive element coupling transistor T4 to the GND potential is implemented by a resistor.

[0076] The 300 current source further includes a MOS transistor T5, with a channel of the same type as that of transistor T3. The gate of transistor T5 is connected to the gate of transistor T3, its source is connected to the VDD potential, and its drain is configured to supply the Ibias current.

[0077] In the example of the figure 4 The bias current is equal to ((kT) / (Rv.q)).ln(n), with k the Boltzmann constant, T the temperature in Kelvin, q the charge of an electron, n the ratio between the widths of transistors T2 and T4 and Rv the resistance value of component R in ohms.

[0078] Thus, the Ibias current varies linearly with temperature.

[0079] Although an example of a PTAT-type current source has been described above, a person skilled in the art will be able to foresee other examples of PTAT-type current sources, and, more generally, other PTAT or CTAT-type current sources. Preferably, in these other implementation examples of the current source, where the current source includes the resistive component R, this component R is configured to convert a voltage across its terminals into a first current, and the source is configured so that the bias current is a copy of the first current, that is, equal to the first current up to a multiplicative factor, this multiplicative factor being constant.

[0080] There figure 5 represents an example of an embodiment of the OC circuit of the oscillator of the figure 3 .

[0081] In this example, the OC circuit is a relaxation oscillator, and, more precisely, a relaxation oscillator including an RS flip-flop (referenced as 500 in figure 5 ).

[0082] In this example, the R component ( figure 2 ) is not part of the OC circuit, and is part of the 300 current source, the latter being implemented, for example, as illustrated in figure 4 .

[0083] The OC circuit includes a 502 power supply terminal configured to receive Ibias current.

[0084] The OC circuit includes two MOS transistors T6 and T7 in series between node 502 and a node 504 configured to receive the reference potential GND, and two MOS transistors T8 and T9 connected in series between nodes 502 and 504, with transistors T6 and T8 connected to node 502 and transistors T7 and T9 connected to node 504.

[0085] In this example, where the potential VDD to which source 300 is connected, supplying the Ibias current ( figure 2 ) is positive with respect to the GND potential, transistors T6 and T8 are P-channel and transistors T7 and T9 are N-channel.

[0086] As an example, transistors T6 and T8 have their sources connected to node 502 and their drains connected to the drains of transistors T7 and T9 respectively, the latter having their sources connected to node 504.

[0087] The gates of transistors T6 and T7 are connected to a Qb output of the 500 flip-flop, and the gates of transistors T8 and T9 are connected to a Q output of the 500 flip-flop.

[0088] A capacitive element C1 couples node 504 to a connection node 506 of transistor T6 to transistor T7, and a capacitive element C2 couples node 504 to a connection node 508 of transistor T8 to transistor T9. Preferably, although not essential, capacitive elements C1 and C2 are identical, that is, they have the same capacitance value.

[0089] The OC circuit further includes an even number of series inverters coupling node 506 to an input R of the 500 flip-flop and the same even number of series inverters coupling node 506 to an input S of the RS flip-flop.

[0090] In the example of the figure 5 The OC circuit includes two inverters INV1 and INV2 in series coupling node 506 to input R, and two inverters INV3 and INV4 coupling node 508 to input S. Inverter INV1 has its input connected to node 506, and inverter INV3 has its input connected to node 508. As an example, at least inverters INV1 and INV3 are identical, and preferably, inverters INV1 to INV4 are all identical.

[0091] The output signal sig2 of the oscillating circuit OC, therefore of the oscillator LO2' in this example, has a frequency F2 equal to Ibias / (2.C1v.Vth), with C1v the capacitance value of each of the capacitive elements C1 and C2, and Vth the switching threshold of the inverters INV1 and INV3.

[0092] Thus, in this example, since the bias current Ibias depends linearly on the temperature T and is partly determined by the value 1 / Rv, the frequency F2 will also depend linearly on the temperature T according to a constant coefficient that is a multiple of 1 / Rv. In other words, F2 is equal to GT / Rv, where G is a constant coefficient. Since the value of Rv is equal to 1 / (Cv.Fc), then F2 = GTCv.Fc, from which it follows that F2 = GTCv.F3.H, since Fc = H.F3. Substituting the expression for F2 above into the equation P = N'.F2 / F3, we find that P is equal to N'.GTCv.H, therefore JT, where J is a constant coefficient and equal to N'.G.Cv.H.

[0093] More specifically, in the case where source 300 is implemented in the manner illustrated in relation to the figure 4 and where the oscillator is implemented in the manner described in relation to the figure 5 The bias current Ibias is equal to ((kT) / (Rv.q)).ln(n), therefore to ((kTCv.Fc) / q).ln(n) = ((kTCv.F3.H) / q).ln(n), and the frequency F2 is equal to Ibias / (2.C1v.Vth). It follows that F2 = (kTCv.F3.H.ln(n)) / (q.2.C1v.Vth). Substituting F2 with its expression above into the equation P = N'.F2 / F3, we obtain: P = (N'.kTCv.F3.H.ln(n)) / (q.2.C1v.Vth.F3), therefore P = (N'.kTCv.H.ln(n)) / (q.2.C1v.Vth), or P= JT with J = (N'.k.Cv.H.ln(n)) / (q.2.C1v.Vth), the coefficient J being constant and independent of temperature, at least to the first order.

[0094] One advantage of the particular example above is that the coefficient J does not depend on any resistance value that might vary with temperature.

[0095] Another advantage of this particular example is that the capacitance value ratio Cv / C1v is temperature independent to first order, because it is a ratio between values ​​which, if they vary with temperature T, have identical coefficients of variation with temperature.

[0096] Although the threshold Vth may depend on the temperature T, this dependence of the threshold Vth on temperature is negligible compared to the dependence of the bias current Ibias on temperature, and the threshold Vth can therefore be considered constant.

[0097] To determine the coefficient J, a calibration phase as described previously in relation to the figure 2 can be implemented, in particular, in the specific example above, to determine the value of the Vth threshold which may depend on the manufacturing process.

[0098] The particular example described above illustrates an embodiment in which the LO2' oscillator has its frequency F2 which depends linearly on T according to a constant coefficient multiple of 1 / Rv, and where the switched capacitance R of value Rv is implemented in the current source 300.

[0099] As an alternative example, the LO2' oscillator can be implemented in the manner described in relation to the figure 3 and its OC circuit is a ring oscillator.

[0100] In other embodiments, the switched capacitance R is implemented in the OC circuit of the LO2' oscillator, and the switched capacitance R of the current source 300 is replaced by a resistive component.

[0101] As has been indicated in relation to the particular example of the figure 5 The threshold Vth of an inverter depends on the temperature T, but this dependence can be neglected.

[0102] More generally, in an oscillator of the type described in relation to the figure 3 , when the frequency of the oscillating circuit OC of this oscillator is at least partly determined by the threshold voltage Vth of one or more inverters of the oscillating circuit OC, the dependence of the threshold Vth on temperature is considered negligible.

[0103] However, to improve the measurement accuracy of a sensor 2 in which the oscillator comprising the switched capacitance R is of the type described in relation to the figure 3 and its frequency at least partly determined by the threshold voltage Vth of one or more inverters of the oscillating circuit OC of this oscillator, an embodiment provides for controlling the threshold voltage Vth of these inverters to make it constant and independent of temperature.

[0104] In such an embodiment, this oscillator has a circuit configured to maintain the voltage Vth constant and independent of temperature, for example by changing the power supplied to these inverters and / or by modulating the back gate voltage of the transistors implementing these inverters in SOI technology (from the English "Silicon On Insulator").

[0105] There figure 6 illustrates an example of such a voltage control circuit Vth of inverters of the oscillating circuit OC of an oscillator, for example inverters INV1 and INV3 of the oscillating circuit OC of the figure 5 .

[0106] The circuit, referenced as 700 in figure 6 , includes an INVref inverter identical to the inverter or inverters whose threshold Vth is controlled by circuit 700.

[0107] The INVref inverter has its output connected to its input, hence its output is set to the threshold voltage Vth of the INVref inverter.

[0108] The 700 circuit further includes an error amplifier ERR configured to provide a Vth_CTRL signal representative of the error, or deviation, between the inverter output voltage INVref, equal to Vth, and a reference or setpoint voltage Vref.

[0109] The 700 circuit further includes a CTRL circuit configured to provide a signal fb from the Vth_CTRL signal. The ERR circuit is configured to control the threshold voltage of the INVref inverter and the inverters, for example INV1 and INV3, of the OC circuit, so that the output voltage of the INVref inverter, and therefore the threshold voltage Vth of this INVref inverter and of the inverters of the OC circuit having their Vth voltage controlled by the 700 circuit, is equal to the Vref voltage.

[0110] Thus, the 700 circuit allows the threshold voltage of the inverters it controls to be controlled by the reference voltage Vref.

[0111] For example, when inverters are implemented by MOS transistors on SOI, the fb signal is a back-gate control signal of these transistors.

[0112] Examples of embodiments in which the LO2' oscillator has its frequency F2 which depends linearly on the temperature and includes the switched capacitance R controlled from the F3 frequency of the LO1' oscillator.

[0113] A person skilled in the art can foresee, from the description above, embodiments in which the oscillator LO2' has its frequency F2 which depends linearly on the inverse of the temperature, i.e., on 1 / T, and includes the switched capacitance R controlled from the frequency F3 of the oscillator LO1'. In this case, the frequency F2 is equal to G / (Rv.T) with G a constant coefficient and the value Rv of the resistive component R is equal to 1 / (Cv.Fc), therefore to 1 / (Cv.H.F3) with H a constant coefficient, from which it follows that F2 = (G.Cv.F3.H) / T. By replacing F2 with its expression above in the equation P = N'.F2 / F3, it follows that P is equal to (N'.G.Cv.H) / T, therefore to J / T, with J a constant coefficient equal to N'.G.Cv.H. Advantageously, the coefficient J then does not depend on the frequency F3 of the LO2' oscillator.To determine T from P, it is then sufficient to know the coefficient J, for example by determining it during a calibration phase such as described in relation to the . figure 2 This calibration phase is, for example, implemented by the calibration circuit described in relation to the figure 2 .

[0114] In these embodiments, when the frequency of the LO2' oscillator, and therefore the coefficient J, depends on the threshold voltage Vth of one or more inverters of the LO2' oscillator, a person skilled in the art will be able to implement a control of the threshold voltage Vth of this or these inverters, for example by means of circuit 700 described in relation to the figure 6 , so that this voltage Vth is constant and independent of temperature.

[0115] Examples of embodiments have been described above in which the LO2' oscillator has its frequency F2 which depends linearly on the temperature or the inverse of the temperature, and includes the switched capacitance R controlled from the frequency F3 of the LOI1' oscillator.

[0116] A person skilled in the art can foresee, from the above description, embodiments in which the LO1' oscillator has a frequency F3 that depends linearly on the temperature or inversely on the temperature, and includes the switched capacitance R, this switched capacitance being controlled from the frequency F2 of the LO2' oscillator. In this case, the LO1' oscillator is, for example, implemented in the manner described in relation to the figure 3 .

[0117] For example, a person skilled in the art can foresee, from the description above, embodiments in which the oscillator LO1' has a frequency F3 that depends linearly on the temperature T and includes the switched capacitance R controlled by the frequency F2 of the oscillator LO2'. In this case, the frequency F3 is equal to GT / Rv, where G is a constant coefficient, and the value Rv of the switched capacitance R is equal to 1 / Cv.Fc, where Fc = H.F2 and H is a constant coefficient. It follows that F3 = G.Cv.H.F2.T. Substituting F3 with its expression above into the equation P = N'.F2 / F3, it follows that P is equal to (N'.F2) / (G.Cv.H.F2.T), therefore to J / T, where J is a constant coefficient and equal to N' / (G.Cv.H). Advantageously, the coefficient J then does not depend on the frequency F2 of the LO2' oscillator, which can vary with temperature.To determine T from P, it is then sufficient to know the coefficient J, for example by determining it during a calibration phase such as described in relation to the . figure 2 This calibration phase is, for example, implemented by the calibration circuit described in relation to the figure 2 .

[0118] As an alternative example, a person skilled in the art can foresee, from the above description, embodiments in which the oscillator LO2' has a frequency F3 that depends linearly on the inverse of the temperature, 1 / T, and includes the switched capacitance R controlled by the frequency F2 of the oscillator LO2'. In this case, the frequency F3 is equal to G / (Rv.T), where G is a constant coefficient, and the value Rv of the switched capacitance R is equal to 1 / Cv.Fc, where Fc = H.F2 and H is a constant coefficient. It follows that F3 = (G.Cv.H.F2) / T. Substituting F3 with its expression above in the equation P = N'.F2 / F3, we find that P is equal to (N'.F2.T) / (G.Cv.H.F2), and therefore to JT, with J being a constant coefficient equal to N' / (G.Cv.H). Advantageously, the coefficient J then does not depend on the frequency F2 of the LO2' oscillator, which can vary with temperature.To determine T from P, it is then sufficient to know the coefficient J, for example by determining it during a calibration phase such as described in relation to the . figure 2 This calibration phase is, for example, implemented by the calibration circuit described in relation to the figure 2 .

[0119] In these embodiments, when the frequency of the LO2' oscillator, and therefore the coefficient J, depends on the threshold voltage Vth of one or more inverters of the LO2' oscillator, a person skilled in the art will be able to implement a control of the threshold voltage Vth of this or these inverters, for example by means of circuit 700 described in relation to the figure 6 , so that this voltage Vth is constant and independent of temperature.

[0120] Thus, in all the examples of embodiments and variants described, the LO1' or LO2' oscillator, which does not include the switched capacitance R and whose frequency depends linearly on T / Rv or 1 / (T.Rv), can have its frequency depend on the temperature without distorting the temperature measurement, because its frequency determines the control frequency Fc of the switched capacitance R, which allows the influence of the frequency of this oscillator to be eliminated in the linear relationship linking P and the temperature T or the inverse of the temperature 1 / T.

[0121] Various embodiments and variants have been described. Those skilled in the art will understand that certain features of these various embodiments and variants could be combined, and other variants will become apparent to them. In particular, the implementation of the LO1' or LO2' oscillator including the switched capacitance R is not limited to the structure described in relation to the figure 3 , and even less so to the examples of 300 current sources and OC oscillating circuits described in relation to the figures 3 à 5 Furthermore, when the LO1' or LO2' oscillator, including the switched capacitance R, has a structure of the type of that of the figure 3 The person skilled in the art will be able to predict a 300 source of type CTAT rather than PTAT as described in relation to the figure 4In particular, the person skilled in the art will be able, in embodiments, to implement the source 300 so that it includes the switched capacitance R to convert a voltage into a current determining the bias current.

[0122] Finally, the practical implementation of the described methods and variants is within the reach of the person in the trade, based on the functional indications given above.

Claims

1. Temperature sensor (2) comprising: a first oscillator (LO1') and a second oscillator (LO2'); a counter (COUNTER) configured to count a number (P) of periods of the second oscillator (LO2') over a duration determined by a period of the first oscillator (LO1'); and a calculation circuit configured to determine a temperature value based on said counted number (P), the sensor being characterized in that: one (LO1'; LO2') of the first and second oscillators (LO1', LO2') is configured so that its frequency linearly depends on temperature or on the inverse of temperature according to a coefficient multiple of the inverse of a resistance value of a resistive component (R) of said one (LO1'; LO2') of the first and second oscillators; said resistive component (R) is implemented by a switched capacitive element; and a control frequency of the switched capacitive element (R) is determined by the frequency of the other one (LO2'; LO1') of the first and second oscillators.

2. Sensor according to claim 1, wherein said one (LO1'; LO2') of the first and second oscillators (LO1', LO2') comprises: an oscillating circuit (OC); and a current source (300) configured to deliver a bias current (Ibias) to the oscillating circuit (C), the current source (300) being of the type proportional to absolute temperature or of the type complementary to absolute temperature.

3. Sensor according to claim 2, wherein : the current source (300) comprises the resistive component (R) ; the resistive component (R) is configured to convert a voltage into a first current; and the current source (300) is configured so that the bias current (Ibias) is a copy of the first current to within a multiplication factor.

4. Sensor according to claim 3, wherein the oscillating circuit (OC) is a ring oscillator.

5. Sensor according to claim 3, wherein the oscillating circuit (OC) is a relaxation oscillator comprising an RS flip-flop (500).

6. Sensor according to claim 2, wherein the oscillating circuit (OC) comprises the resistive component (R).

7. Sensor according to any one of claims 2 to 6, wherein : the oscillating circuit (OC) comprises at least one first inverter (INV1, INV3) having its threshold voltage determining the frequency of said one (LO1'; LO2') of the first and second oscillators (LO1', LO2'); and said one (LO1'; LO2') of the first and second oscillators further comprises: a second inverter (INVref) identical to the first oscillator and having its output connected to its input, an error amplifier (ERR) configured to deliver a signal (Vth_CTRL) indicating a deviation between an output voltage of the second inverter (INVref) and a reference threshold voltage (Vref), and a circuit (CTRL) configured to control a threshold voltage of the first and second inverters (INVref; INV1, INV3) based on the signal (Vth_CTRL) delivered by the error amplifier (ERR) so that the threshold voltage of the first and second inverters is equal to the reference voltage (Vref) .

8. Sensor according to any one of claims 1 to 7, wherein the sensor comprises a calibration circuit configured to: - receive from the counter (COUNTER) the number P of periods of the second oscillator (LO2') counted over the duration determined by the period of the first oscillator (LO1'); - determine, when the calibration circuit receives a calibration request, and based on a known calibration temperature and on number P, a constant coefficient J such that: a) P is equal to J times the calibration temperature if said one of the first and second oscillators (LO1', LO2') is the first oscillator (LO1') and has its frequency linearly depending on the inverse of temperature or if said one of the first and second oscillators is the second oscillator (LO2') and has its frequency linearly depending on temperature; or b) P is equal to J times the inverse of temperature if said one of the first and second oscillators is the second oscillator (LO2') and has its frequency linearly depending on the inverse of temperature or if said one of the first and second oscillators is the first oscillator (LO1') and has its frequency linearly depending on temperature.

9. Sensor according to any one of claims 1 to 8, wherein the sensor is configured so that the duration determined by the frequency of the first oscillator is at least 100 times greater than a period of the second oscillator (LO2').

10. Sensor according to any one of claims 1 to 9, wherein the sensor is configured so that the control frequency of the switched capacitive element (R) is at least 10 times greater than the frequency of said one of the first and second oscillators.

11. Sensor according to any one of claims 1 to 10, wherein each of the first and second oscillators comprises no quartz.