Differential input, single-ended output quadrature hybrid coupler

Abstract

A Differential Input, Single-ended Output Quadrature Hybrid Coupler (DISO-QHC) is based on baluns, one or two phase shifters, and a Wilkinson combiner. The DISO-QHC is configured to receive two differential, quadrature RF signals from differential PAs and output a combined, single-ended RF signal to drive an antenna element or subarray. Impedance matching transformers configured as baluns perform the differential-to-single-ended conversion and PA output impedance matching, without any requirement on coupling factor. One or two phase shifting circuits align the single-ended, quadrature RF signals in phase. A Wilkinson combiner circuit combines the power of the outputs of the phase shifting circuits. Area and losses are reduced by combining components from the phase shifting circuits with the baluns, and with the Wilkinson combiner circuit (which is implemented with lumped-reactance components rather than transmission lines). The DISO-QHC does not have a specific isolation port, but the isolation resistor within the Wilkinson combiner dissipates the differential and reflected signal from the PAs resulting from the incoming wave reflected from the antenna. The DISO-QHC is compatible with a LMBA functionality. By injecting a current at the center point of the isolation resistor in the Wilkinson combiner, the impedance at PA ports can be modulated.

Description

[0001] DIFFERENTIAL INPUT, SINGLE-ENDED OUTPUT QUADRATURE HYBRID COUPLER

[0002] TECHNICAL FIELD

[0003] The present disclosure relates generally to wireless communications, and in particular to a Quadrature Hybrid Coupler conjured to convert a differential Radio Frequency (RF) input to a single-ended RF output having substantially constant current under dynamically varying load impedance.

[0004] BACKGROUND

[0005] Wireless communication networks are ubiquitous in many parts of the world. These networks continue to grow in capacity and sophistication. To accommodate more users, different types of devices, and different use cases, the technical standards governing the operation of wireless communication networks continue to evolve. The fourth generation (4G) of network standards has been deployed, the fifth generation (5G) is in development and early deployment, and the sixth generation (6G) is in design. With each generation, technological advances improve the capacity and spectral efficiency of the wireless communication system. For example, 5G added new frequency bands, and applied beamforming. This trend is expected to continue in 6G by exploiting additional frequency bands, and applying more advanced beamforming.

[0006] 5G added a second frequency range, FR2. This provided significant new available spectrum in the range 24.25-52.6 GHz. At these high frequencies, wavelengths are small. This is advantageous, as antenna elements are also small, allowing for antenna arrays with hundreds, or even thousands, of antenna elements. However, carriers at these high frequencies suffer higher path loss, and hence have limited range, compared to conventional wireless telecom operating frequencies. Beamforming is one technique featured in 5G and 6G, to improve both coverage and capacity.

[0007] Beamforming refers to the use of antennas having increased and controllable directionality, whereby an RF transmission (or reception sensitivity) is narrow, and is “aimed” in a specific direction. This is enabled by transmitting or receiving signals with controlled relative phase and gain in the antenna elements (or subarrays of antenna elements). The relative phases of, e.g., transmit signals sent to each antenna element are controlled to create constructive or destructive interference, thus amplifying the signal in some directions, and attenuating it in others, and hence controlling the direction in which the beam is transmitted. Similar phase manipulation of signals from antenna elements (or subarrays) in a receive antenna can also result in beamforming the sensitivity of an antenna array in receiving signals. Also, multiple orthogonal beams can be formed and aimed in different directions, thus simultaneously addressing multiple wireless devices, also known as User Equipment (UE). To form robust beams, antenna elements are normally placed tightly together. For example, a distance of A / 2 is commonly used (where A is the RF wavelength), to form arbitrary beams without folding. However, the tight antenna spacing causes high electromagnetic coupling between the antennas, and additionally signals leak in between the antennas. The beamsteering, combined with the antenna coupling, makes the impedance seen by each power amplifier (PA) driving the antenna elements (or subarrays) deviate from a designed impedance.

[0008] The PA is designed assuming a nominal load impedance for optimal output power, linearity, and efficiency. The PA amplifies and delivers electrical power to the antenna element / subarray, which converts it to an electromagnetic signal. However, if the load impedance seen by the PA diverges from its designed (optimum) value, there is an impedance mismatch, which degrades PA performance.

[0009] To direct a beam to a desired direction, a phase shift is required between signals sent to different antenna elements (or subarrays). The same signal, except for the phase shift, is present at all antenna elements, and electromagnetic energy of the signal leaks between them. This is seen by the PAs as a mismatch from an optimal (matched) impedance, which is not present when no phase shifts are introduced to steer the beam. The designed impedance seen by the PA is referred to as the impedance in the boresight direction ( / .e., where the RF signal is radiated normal to the plane of the antenna element). When coupling is present between the antenna elements (due to spacing), and the same signal is sent on all antennas, but with different phases, this is experienced by the PA as load impedance variation and mismatch, even though it originates from antenna leakage and to the delay introduced by the phase shifter ( / .e., the mismatch typically grows higher as the beam-angle increases, since the relative phase shift between antennas increases). Because the impedance mismatch causes a partial reflection of the RF signal from the antenna element (or subarray) back toward the PA, a standing wave is generated along the transmission line connecting the two. This is quantified in the art as an antenna impedance Voltage Standing Wave Ratio (VSWR), which is calculated in terms of the reflection coefficient or return loss (also known as the s11 parameter). Assuming the antenna and PA are impedance-matched for signals transmitted in the boresight direction, the active impedance load, or VSWR, typically grows higher as the beam-angle increases, since the relative phase shift in between antenna elements (or subarrays) increases.

[0010] At low frequencies, an isolator can be inserted in between a PA and its antenna element / subarray, to ensure that the VSWR is not transferred to the PA. In high frequency Advanced Antenna System (AAS), there is no room to fit an isolator at each PA output.

[0011] The Background section of this document is provided to place aspects of the present disclosure in technological and operational context, to assist those of skill in the art in understanding their scope and utility. Approaches described in the Background section could be pursued, but are not necessarily approaches that have been previously conceived or pursued. Unless explicitly identified as such, no statement herein is admitted to be prior art merely by its inclusion in the Background section.

[0012] SUMMARY

[0013] The following presents a simplified summary of the disclosure in order to provide a basic understanding to those of skill in the art. This summary is not an extensive overview of the disclosure and is not intended to identify key / critical elements of aspects of the disclosure or to delineate the scope of the disclosure. The sole purpose of this summary is to present some concepts disclosed herein in a simplified form as a prelude to the more detailed description that is presented later.

[0014] According to one or more aspects of the present disclosure described and claimed herein, a Differential Input, Single-ended Output Quadrature Hybrid Coupler (DISO-QHC) is based on baluns, one or two phase shifters, and a Wilkinson combiner. The DISO-QHC is configured to receive differential, quadrature RF signals from differential PAs and output a combined, single-ended RF signal to drive an antenna element or subarray. Impedance matching transformers configured as baluns perform the differential-to-single-ended conversion and PA output impedance matching, without any requirement on coupling factor. One or two phase shifting circuits align quadrature RF signals in phase by imparting a total of 90° of phase difference to the balun outputs. A Wilkinson combiner circuit combines the power of two phase- aligned, single-ended RF signals. Area and losses are reduced by combining components from the phase shifting circuits with the baluns, and with the Wilkinson combiner circuit (which is implemented with lumped-reactance components rather than transmission lines). The DISO- QHC does not have a specific isolation port, but the isolation resistor within the Wilkinson combiner dissipates the differential and reflected signal from the PAs resulting from the incoming wave reflected from the antenna. The DISO-QHC is compatible with a LMBA functionality. By injecting a current at the center point of the isolation resistor in the Wilkinson combiner, the impedance at PA ports can be modulated.

[0015] One aspect relates to a Differential Input, Single-ended Output (DISO) Quadrature Hybrid Coupler (QHC), configured to receive two differential, quadrature Radio Frequency (RF) signals and output a combined, single-ended RF signal having substantially constant current under dynamically varying load impedance. The DISO-QHC includes two output impedance matching transformers, each receiving one of the differential, quadrature RF signals, and each configured as a balun outputting a single-ended RF signal; at least one output phase shifting circuit connected to a balun output and configured to impart a total of 90° of phase difference between the single-ended RF signals; and an output power combiner circuit configured to combine the outputs of the at least one phase shifting circuit and output the combined, single- ended RF signal. Another aspect relates to a method of converting two differential, quadrature Radio Frequency (RF) signals into a combined, single-ended RF signal having substantially constant current under dynamically varying load impedance. The two differential, quadrature RF signals are converted into two single-ended, quadrature RF signals in two output impedance matching transformers configured as baluns. A total of 90° of phase difference is imparted between the two single-ended, quadrature RF signals by at least one output phase shifting circuit connected to a balun output. The outputs of the at least one phase shifting circuit are combined into one single-ended RF signal in an output power combiner circuit.

[0016] Yet another aspect relates to User Equipment (UE) operative in a wireless communication network. The UE includes processing circuitry and communication circuitry operatively connected to the processing circuitry. The communication circuitry includes differential RF Power Amplifiers (PAs) arranged in pairs and a Differential Input, Single-ended Output (DISO) Quadrature Hybrid Coupler (QHC) associated with each pair of PAs. The DISO- QHC is configured to receive two differential, quadrature Radio Frequency (RF) signals from the PAs and output a combined, single-ended RF signal having substantially constant current under dynamically varying load impedance. The DISO-QHC includes two output impedance matching transformers, each receiving one of the differential, quadrature RF signals, and each configured as a balun outputting a single-ended RF signal; at least one output phase shifting circuit connected to a balun output and configured to impart a total of 90° of phase difference between the single-ended RF signals; and an output power combiner circuit configured to combine the outputs of the at least one phase shifting circuit and output the combined, single-ended RF signal.

[0017] Yet another aspect relates to a base station operative in a wireless communication network. The base station includes processing circuitry and communication circuitry operatively connected to the processing circuitry. The communication circuitry includes differential RF Power Amplifiers (PAs) arranged in pairs and a Differential Input, Single-ended Output (DISO) Quadrature Hybrid Coupler (QHC) associated with each pair of PAs. The DISO-QHC is configured to receive two differential, quadrature Radio Frequency (RF) signals from the PAs and output a combined, single-ended RF signal having substantially constant current under dynamically varying load impedance. The DISO-QHC includes two output impedance matching transformers, each receiving one of the differential, quadrature RF signals, and each configured as a balun outputting a single-ended RF signal; at least one output phase shifting circuit connected to a balun output and configured to impart a total of 90° of phase difference between the single-ended RF signals; and an output power combiner circuit configured to combine the outputs of the at least one phase shifting circuit and output the combined, single-ended RF signal. BRIEF DESCRIPTION OF THE DRAWINGS

[0018] The present disclosure will now be described more fully hereinafter with reference to the accompanying drawings, in which aspects of the disclosure are shown. However, this disclosure should not be construed as limited to the aspects set forth herein. Rather, these aspects are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the disclosure to those skilled in the art. Like numbers refer to like elements throughout.

[0019] FIG. 1 depicts an exemplary array of 64 antenna elements, wired into 32 2x1 subarrays.

[0020] FIG. 2 is a hardware block diagram of a Radio Frequency Integrated Circuit (RFIC) transceiver for connection to 32 subarrays of antenna elements.

[0021] FIG. 3 is a schematic diagram of a Quadrature Hybrid Coupler.

[0022] FIG. 4A is a block diagram of a balanced Power Amplifier (PA).

[0023] FIG. 4B is a graph of the output power of each PA, and the balanced PA, of FIG. 4A, over a full VSWR circle.

[0024] FIG. 5A is a schematic diagram of a Load-Modulated Balanced Amplifier (LMBA).

[0025] FIG. 5B is a plot of real and imaginary components of impedance in the LMBA of FIG. 5A.

[0026] FIG. 5C is a Smith chart plot of load impedances as modulated in the LMBA of FIG. 5A.

[0027] FIG. 6 is a schematic diagram of a Differential Balanced PA.

[0028] FIG. 7A is a schematic diagram of an 180° Wilkinson splitter circuit implemented with transmission lines.

[0029] FIG. 7B is a schematic diagram of a 180° Wilkinson splitter circuit implemented with lumped-reactive components.

[0030] FIG. 8 is a block and schematic diagram of a Distributed Transformer circuit to merge phase shifters and baluns to reduce the overall loss on a signal path.

[0031] FIG. 9 is a schematic diagram of a Differential Input, Single-ended Output Quadrature Hybrid Coupler (DISO-QHC) according to aspects of the present disclosure.

[0032] FIG. 10A is a diagram of a Wilkinson combiner circuit implemented with transmission lines.

[0033] FIG. 10B is a diagram of a Wilkinson combiner circuit implemented with lumped- reactance components.

[0034] FIG. 11 is a schematic diagram showing DISO-QHC at the output of PAs, and a mirror circuit at the PA inputs.

[0035] FIG. 12 is a block and schematic diagram of an LCL type phase shifting circuit and its partial combination with the secondary winding of a transformer.

[0036] FIG. 13 is a block and schematic diagram of a CLC type phase shifting circuit.

[0037] FIG. 14 is a schematic diagram a DISO-QHC with an LCL type phase shifting circuit, showing the reduction of component count by merging reactive components. FIG. 15 is a schematic diagram of a DISO-QHC with CLC type phase shifting circuits, showing the reduction of component count by merging reactive components.

[0038] FIG. 16 is schematic diagram of a DISO-QHC showing the elimination of reflected signal from the antenna element(s).

[0039] FIG. 17 is a schematic diagram of a DISO-QHC type input circuit with LCL type phase shifting circuits, showing the reduction of component count by merging reactive components.

[0040] FIG. 18 is a schematic diagram of a DISO-QHC type input circuit with CLC type phase shifting circuits, showing the reduction of component count by merging reactive components.

[0041] FIG. 19 is a schematic diagram of a DISO-QHC implemented as a LMBA by injecting current between the resistors of the Wilkinson combiner circuit.

[0042] FIG. 20 is a Smith chart plot showing modulation of the load impedance by controlling the injected current in FIG. 19.

[0043] FIG. 21 is a flow diagram of a method of converting two differential, quadrature RF signals into a single combined RF signal having substantially constant current under dynamically varying load impedance.

[0044] FIG. 22 is a hardware block diagram of a wireless device in a wireless communication network.

[0045] FIG. 23 is a hardware block diagram of a base station in a wireless communication network.

[0046] DETAILED DESCRIPTION

[0047] For simplicity and illustrative purposes, the present disclosure is described by referring mainly to an exemplary aspect thereof. In the following description, numerous specific details are set forth in order to provide a thorough understanding of the present disclosure. However, it will be readily apparent to one of ordinary skill in the art that the present disclosure may be practiced without limitation to these specific details. In this description, well known methods and structures have not been described in detail so as not to unnecessarily obscure the present disclosure.

[0048] FIG. 1 depicts an example antenna array according to aspects disclosed herein. In this example, the array consists of 8x8 dual-polarized antenna elements. The antenna elements are pairwise interconnected to form 2x1 subarrays, and thus reduce the number of active radio chains required to connect to the antenna and apply beamforming. The subarrays are numbered SAO to SA31 (moving left to right and top to bottom). In general, antenna elements of an array may be grouped into any number of subarrays, each comprising any number of individual antenna elements. Subarrays in an antenna array need not be the same - that is, some antennal elements may be grouped into one or more m x n subarrays, and other antennal elements may be grouped into one or more i x j subarrays, where m i and / or n j. As used herein, a subarray may include from one antenna element up to all of the antenna elements in an antenna array.

[0049] FIG. 2 depicts relevant portions of a Radio Frequency Integrated Circuit (RFIC) according to one aspect. This RFIC has 8 bidirectional IQ baseband ports, an internal port expansion by four, and thus 32 antenna connections. Each antenna branch, also referred to as an RF tile, has its own Phase Locked Loop (PLL), to enable beamforming by controlling the relative phase between antenna elements or subarrays. In transmit mode, the IQ baseband signal is split to four branches, upconverted to RF using an IQ-modulator, and amplified by a PA. An antenna switch connects either the transmitter or the receiver branch to the antenna element subarray. As depicted in FIG. 2, the tile connects to the antenna element subarrays in column 1 of the antenna system depicted in FIG. 1 (that is, subarrays SAO, SA8, SA16, and SA24). In receive mode, each antenna element subarray signal is amplified by a Low Noise Amplifier (LNA), downconverted to baseband, and added to the other three branches sharing the IQ-interface. The receiver can also be reused as a transmit observation receiver (TOR), to sense the PA output signal, such as for use in closed-loop Digital Pre-Distortion (DPD) operation (alternatively, the RFIC may include separate TOR circuitry).

[0050] The transceiver of FIG. 2 can be used together with the exemplary antenna array of FIG. 1. The four RF tiles sharing a baseband IQ-interface are here each connected to antenna element subarrays in one column of the antenna array. For example, the antenna connections depicted in FIG. 2 may connect to the antenna element subarrays of column 1 (SAO, SA8, SA16, and SA24). As discussed above, when coupled to antennas or subarrays as shown in FIG. 2, the PAs and transmitters interact with each other through the antenna coupling. Accordingly, the PAs effectively see a time-varying impedance mismatch due to beamforming, which can be expressed in terms of VSWR. The VSWR impacts PA output power, efficiency, and linearity, and consequently the phased-array beam and its direction.

[0051] Reducing the sensitivity of PAs to the varying load impedance would thus improve RF system performance. Various means of reducing PA load sensitivity are known in the art.

[0052] One approach to reducing the sensitivity of PAs to impedance mismatches is a balanced PA circuit, implemented for example with Quadrature Hybrid Couplers (QHC). A QHC is special case of the general class of directional couplers for which the coupling is 3dB. FIG. 3 depicts one implementation of a QHC using transmission lines or microstrips. The transmission lines are of electrical length A / 4, where A is the wavelength of the fundamental frequency. In other aspects, the QHC may be built from lumped reactive elements, such as inductive and capacitive devices. Such an implementation may be attractive for integration, as such QHCs may be fabricated in less area than a transmission line or microstrip implementation. For example, Robert C. Frye, et al. describe a CMOS implementation of a QHC suitable for integration on a high frequency RFIC, in the paper A 2GHz Quadrature Hybrid Implemented in CMOS Technology, published at the IEEE 2002 Custom Integrated Circuits Conference p. 287, the disclosure of which is incorporated herein by reference in its entirety.

[0053] An ideal QHC is a symmetric, lossless, passive, four-port network, which imparts a 90° phase shift. Because it is symmetric, a QHC can split an input signal into two output signals (having 90° phase offsets) or combine two input signals (having 90° phase offsets) into one output signal. If a signal is incident on only one port of a QHC, there is a port from which no power will exit, referred to as the isolation port. When configured as a splitter or combiner, the isolation port is often terminated to ground through a characteristic impedance (e.g., 50Q).

[0054] As the Frye paper cited above discloses, one known application of a QHC is to implement a balanced PA, which exhibits reduced sensitivity to impedance mismatches by the load. This application is further described by Guiseppe Berretta, et al. in the paper A Balanced CDMA2000 SIGe HBT Load Insensitive Power Amplifier, published in the IEEE Radio and Wireless Symposium, 2006, pp. 523-526, the disclosure of which is incorporated herein by reference in its entirety. FIGs. 4A and 4B, copied from the Berretta paper, depict a schematic diagram of a balanced amplifier implemented using two QHCs, and a graph of output powers, respectively. As depicted in FIG. 4A, an RF signal is input to port 1 of a first QHC configured as a splitter, with a standard 50Q impedance connected to ground at port 4. The QHC outputs the signal on both output ports 2 and 3, with a 90° phase offset. These are amplified by PAs, which output the amplified quadrature RF signals, through impedance matching circuits, to a second QHC configured as a combiner, at input ports 2 and 3. The combined RF signal is output at port 1 , with a standard 50Q impedance connected to ground at port 4.

[0055] FIG. 4B shows the output power for each PA as the load impedance varies over a full VSWR=4 circle ( / .e., 0-360°). Because of the quadrature operation, the two PAs compensate each other, and the output power of the balanced PA circuit - the top curve in FIG. 4B - is nearly flat. A balanced PA architecture thus exhibits a high insensitivity to impedance mismatch, such as that caused by coupling between antenna array elements in beamforming operations Another known approach to reducing the sensitivity of PAs to impedance mismatches is a Load-Modulated Balanced Amplifier (LMBA). The LMBA architecture relies on an output QHC where a current source is connected to the isolation port. Depending on amplitude and phase of the injected current, the load presented to the PAs ports is modulated both reactively and resistively. One implementation of this architecture is described by Valdrin Qunaj, et al. in the paper A K a-Band Doherty-Like LMBA for High-Speed Wireless Communication in 28-nm CMOS, published in the IEEE Journal of Solid-state Circuits, Vol. 56, No. 12, Dec. 2001 , the disclosure of which is incorporated herein by reference in its entirety. Figures 5A-5C are reproduced from this paper. In this design, the LMBA technique is used to obtain a Doherty amplifier behavior. For the largest modulation ratio, the amplitude of the current injected into the combiner must be equal to the current of a PA. Yet another known approach to reducing the sensitivity of PAs to impedance mismatches is a Differential Balanced PA. European Patent Specification 1 209 756 B1 , the disclosure of which is incorporated herein by reference in its entirety, discloses the use of differential PAs in a balanced architecture. By using two QHCs at the input and output, a combination of 0 / 90° (balanced) and 0 / 180°(differential) phases is obtained. A differential signal can be applied directly to the circuit as depicted in FIG 6 (reproducing Fig. 16 from the patent). This approach combines the benefit of differential and balanced PAs. However, this implementation requires many passive elements: four QHCs in total, and additional baluns are required to perform a conversion to single-ended.

[0056] Still another known approach to reducing the sensitivity of PAs to impedance mismatches is the use of N° Wilkinson splitters. H.S. Nagi, in Miniature lumped element 180° Wilkinson divider, published in the IEEE MTT-S International Microwave Symposium Digest, 2003, pp. 55-58, vol.1 and Dan Kuylenstierna, et al., in Lumped-Element Quadrature Power Splitters Using Mixed Right / Left-Handed Transmission Lines, published in the IEEE Transactions on Microwave Theory and Techniques, Vol. 53, No. 8, Aug. 2005, the disclosures of both of which are incorporated herein by reference in their entireties, disclose lumped Wilkinson splitters with 90° and 180° phase, respectively.

[0057] N° Wilkinson splitters are formed by a classic in-phase Wilkinson splitter followed by phase shifters, as shown in FIGs. 7A (transmission line) and 7B (lumped element). Kuylenstierna et al. propose to reduce the number of passive components by merging inductors from the Wilkinson splitter and capacitors from the phase shifter that are connected in parallel. The implementation Nagi proposed only introduces phase shifts and does not perform a conversion from single-ended to differential, and therefore operates with single-ended PAs.

[0058] In a phased array system, the number of phase shifters and signal splitters / combiners is proportional to the number of branches in the system. According to one aspect of the present disclosure, components of phase shifters and baluns are merged to reduce the overall loss on the signal path. The split / combine is realized by a Distributed Transformer, in which impedance scales with the number of branches in the system. This architecture is depicted in FIG. 8. The circuit performs both differential to single-ended conversion and signal combining.

[0059] Balanced PAs are considered a primary solution to antenna mismatch issues. As illustrated in FIG. 4A, the PAs are followed by matching components, like transformers and inductors, to achieve the greatest output power and efficiency possible. Transformers included in the matching components can also perform a differential to single-ended conversion, for differential PAs. These matching components are followed by a QHC, which can be implemented as inductor- and / or transformer-based couplers, Lange couplers, or quarter wavelength microstrips / striplines. The passive matching and QHC components both consume area and generate losses, reducing both output power and efficiency. Thus, reducing and merging the components’ functionality into a smaller number of elements may improve the overall RF performance.

[0060] FIG. 9 depicts a Differential Input, Single-ended Output (DISO) Quadrature Hybrid Coupler (QHC), referred to herein as a DISO-QHC 10. The DISO-QHC 10 is connected to the outputs of two differential, quadrature PAs (PA1 , PA2) and receives two differential RF signals from the PAs. As well known in the art, the RF signals output by each PA have a 180° phase difference (differential), and the two RF signals are further offset from each other by 90° (quadrature). The DISO-QHC 10 comprises two impedance matching transformers 12, 14, each configured as a balun, to perform differential to single-ended signal conversion. The DISO- QHC 10 further comprises at least one phase shifting circuit 18, 20 each connected to a balun 12, 14 output and configured to impart a total of 90° of phase difference between the single-ended RF signals, and a power combiner circuit 22 configured to combine the outputs of the phase shifting circuits and output a combined, single-ended RF signal.

[0061] The phase shifters 18, 20 impart a total of 90° phase difference between the two single- ended RF signals, thus realigning them in phase from their quadrature relationship at the outputs of the PAs. This is indicated in FIG. 9 by the notations 1 and Qi+90°. For example, in one embodiment, the DISO-QHC 10 may comprise only one phase shifting circuit 20 imparting the full 90° of phase shift. In this case, the output of balun 12 has no phase shifting circuit (i.e. , (Di=0). In other embodiments, a phase shifting circuit 18, 20 is provided for each RF signal, with the combined phase shifts resulting in a 90° difference in phase. As another example, where Qi=-45°, the phase shifting circuit 18 imparts a -45° phase shift, while the phase shifting circuit 20 imparts a (-45+90)=+45° phase shift, yielding a total phase difference of 90°. Those of skill in the art will readily appreciate that the phase shifting circuits 18, 20 may impart any amount of phase shift, so long as the combined phase shift totals to a phase difference of 90° between the two RF signals.

[0062] The power combiner circuit 22 is preferably a Wilkinson combiner circuit. As known in the art, and as depicted in FIG. 10A, a Wilkinson combiner uses quarter wavelength transformers or transmission lines to connect each of dual ports (2, 3) to a single port (1), with a resistor (R) between the dual ports. When configured as a power splitter, the circuit divides power from a signal input at the single port (1) equally between the dual port outputs (2, 3). The resistor (R) between the dual ports (2, 3) enables the two outputs to be matched, while also providing isolation. The resistor does not dissipate any power, and as a result the Wilkinson power divider can theoretically be lossless, although in practice there are always some losses. When configured as a power combiner, the circuit combines the power of matched signals input to the dual ports (2, 3), and outputs one signal at the single port (1). The resistor (R) absorbs any mismatch in phase and amplitude between signals at the dual ports (2, 3), allowing only the common mode of the signals to pass to the single port (1). Accordingly, the Wilkinson combiner is typically only used to combine the outputs of matched transistors, where the amplitudes and phases will be closely matched. FIG. 10B depicts a lumped element implementation of a Wilkinson combiner, where the A / 4 transformers are implemented with capacitors and inductors.

[0063] FIG. 11 depicts the RF signal inputs to the PAs (PA1 , PA2) as being differential, quadrature signals. In one embodiment, these inputs are generated from an RF signal by a near-mirror image of the DISO-QHC 10 connected at the input. This configuration is depicted in FIG. 11. A Wilkinson combiner circuit configured as a splitter 24 divides the RF signal into two, and at least one phase shifting circuit 26, 28 imparts a total of 90° of phase difference between the two signals output by the splitter 24 (quadrature). These are converted to differential signals at impedance matching transformers 30, 32 configured as baluns, each generating a differential output from a single-ended input signal. The differential, quadrature RF signals are then input to the PAs (PA1, PA2). Those of skill in the art will appreciate that the differential, quadrature RF signals input to the PAs could be generated otherwise, such as in a differential quadrature frequency conversion mixer circuit.

[0064] FIG. 12 shows one of the phase shifting circuits 18, 20, and its implementation in lumped element form using an LCL filter circuit. To the right, the secondary winding of the associated impedance matching transformer (balun) 12, 14 is depicted, showing that the two inductances are in parallel. As such, the inductances may be combined, such as by altering the inductance of the balun secondary winding to account for both inductances. In this manner, component count may be reduced, both saving area and reducing losses.

[0065] For completeness, FIG. 13 shows a phase shifting circuit 18, 20 implemented as a CLC filter circuit. In this implementation, combination of the inductor with part of the balun 12, 14 is not possible.

[0066] FIG. 14 shows how a compact DISO-QHC 10 may be formed, according to one aspect, by combining components. In the upper representation, the signal from PA1 has no phase shifter circuit 18, and the phase shifter circuit 20 (of 90°) on the signal from PA2 is of the LCL form, and it has been combined with the secondary winding of the balun 14, as depicted in FIG. 12. In the lower representation of FIG. 14, the left-side shunt capacitors in the A / 4 portion of the Wilkinson combiner 22, and the remaining shunt inductor of the phase shifting circuit 20, are combined and replaced with two equivalent shunt capacitors, of values Ceq,i and Ceq,2. These are given by:

[0067] Note that,

[0068] The two right-side shunt capacitors in the A / 4 portion of the Wilkinson combiner 22 combine to a single shut capacitor of value 2C. This places the inductors of the A / 4 portion of the Wilkinson combiner 22 in series. The DISO-QHC 10 in this case is reduced to only two baluns, one series capacitor, two equivalent shunt capacitors, one shunt 2C capacitor, and one inductor - a reduction from eight discrete components to five.

[0069] FIG. 15 shows the formation of another compact DISO-QHC 10, according to a second aspect. In the upper representation, the phase shifter circuit 18 is of the CLC form, and the phase shifter circuit 20 is of the LCL form. One inductance of the latter has been combined with the secondary winding of the balun 14, as depicted in FIG. 12. In the lower representation of FIG. 15, several additional components are combined. One capacitor of the phase shifter circuit 18 is combined with the output capacitance of the balun 12, by adjusting the balun 12. The other capacitor of the phase shifter circuit 18 is combined with the left-side shunt capacitor in the A / 4 portion of the Wilkinson combiner 22, resulting in one shunt capacitor of value Ceq,3 . As above, the two right-side shunt capacitors in the A / 4 portions of the Wilkinson combiner 22 are combined, yielding a second shunt capacitor of value 20. Finally, also as above, the left-side shunt capacitor in the lower A / 4 portion of the Wilkinson combiner 22, and the remaining shunt inductor of the phase shifting circuit 20, are combined and replaced with the equivalent shunt capacitor of value Ceq,2. In this aspect, eleven discrete components are reduced to six. This both saves area and reduces losses.

[0070] The DISO-QHC 10 does not contain a specific isolation port with a termination resistor. However, FIG. 16 illustrates the propagation of an incoming / forward wave from the antenna ( / .e., one reflected by an impedance mismatch). Within the Wilkinson combiner, the signal is split and appears in phase at ports 2 and 3 (shown above the line). One of the forward waves travels through the 0-degree phase shifter 18 ( / .e., there is no phase shifter 18), is reflected at the PA1 drain terminal, and then once again passes through the 0-degree phase shifter 18 before reaching port 2 (shown below the line). The second forward wave travels through the 90- degree phase shifter 20 (above the line), is reflected at the PA2 drain terminal, and then once again passes through the 90-degree phase shifter 20 before reaching port 3 (below the line). Effectively, the reflected signals appear differentially ( / .e., 180° out of phase) across the isolation resistor in the Wilkinson combiner 22, where the signal is dissipated. Hence, the remaining reflected signal, directed at the antenna port, is 0. The same result is obtained for any values of the phase shifting circuits 18, 20 that sum to 90°. For example, if the phase shifting circuit 18 imparts -45° and the phase shifting circuit 20 imparts +45°, each signal passes through the respective circuits 18, 20 twice, resulting in one having a phase shift of -90° and the other having +90°, for a total of 180°, or differential, phase relationship.

[0071] As discussed above, a mirror of the DISO-QHC 10 can be used to generate the differential, quadrature RF signals input to the PAs. Components in these circuits can similarly be combined, reducing area and losses. FIG. 17 depicts two Wilkinson splitters 40, 42, each receiving one leg of a differential RF signal (e.g., following a balun, not shown in FIG. 17). Phase shifting circuits 44 and 46 on one output of each Wilkinson splitter 40, 42 impart a 90° phase shift. The phase shifting circuits 44 and 46 are of the LCL form. The Wilkinson splitter 40, 42 outputs are then cross-coupled to the PAs, such that the signals input to each PA are 180° out of phase (differential), and the differential signal input to PA2 is 90° out of phase (quadrature) with the signal input to PA1. The lower portion of FIG. 17 depicts how components can be combined, to reduce component count and hence area and losses. The right-side inductors of the LCL phase shifting circuits 44, 46 are combined to one inductor between the inputs to PA2. The left-side inductors of these phase shifting circuits 44, 46 are combined with components in the A / 4 portions of the Wilkinson splitters 40, 42, yielding the two shunt Ceqcapacitances.

[0072] FIG. 18 shows a similar input arrangement, wherein the phase shifting circuits 44 and 46 are of the CLC form. The lower portion of FIG. 17 depicts how components can be combined, to reduce component count and hence area and losses. The right-side capacitors of the CLC phase shifting circuits 44, 46 are combined to one capacitor between the inputs to PA2. The leftside capacitors of these phase shifting circuits 44, 46 are combined with components in the A / 4 portions of the Wilkinson splitters 40, 42, yielding the two shunt Ceqcapacitances.

[0073] In both the input circuits of FIGs. 17 and 18, only two Wilkinson splitters with adjusted capacitances (in the lumped-element A / 4 portions), two series elements (capacitors / inductors) and one shunt element (inductor / capacitor), are needed for input matching and to generate the phases of the signals driving the PA stages. If better amplitude matching is required, each pair of differential RF drive signals must have either LCL or CLC phase shifters. However, the maximum space savings will be achieved for the solution in Figure 17, as only a single inductor is required.

[0074] FIGs. 12 and 13 show the phase shifting circuits 18, 20 implementations with two inductors and one capacitor, LCL, and two capacitors and one inductor, CLC, respectively. One of the shunt inductors can be combined with a transformer. Thus, at one side of the phase shifting circuit 18 it should have the characteristic impedance of the inventive QHC; on the other side it transforms the characteristic impedance to the desired load impedance of the PA. In the inventive circuit, a differential PA could be connected between the two terminals of the transformer. The shunt inductance connected to the Wilkinson combiner terminal is parallel with the capacitance needed for the combiner but can be merged and replaced by a single capacitor as in FIG. 14.

[0075] Similarly, for the “0-degree” phase shifting circuit 18, which is just a short circuit, a transformer can be used directly to connect another pair of PA stages. For better amplitude balancing between the different PAs, in one aspect, one -45- and one +45-degree phase shifting circuits are inserted, as in FIG. 15. The CLC elements can be absorbed by any tuning capacitance needed by the transformers 12, 14 and the Wilkinson combiner 22. Effectively, the two PAs have either a series capacitance or series inductance before connecting to the Wilkinson combiner 22. Conventional inductor- and transformer-based QHCs require a large coupling factor, or turns ratio, such as in the range -0.7-0.8, to achieve combination / splitting of signals with 0- and 90-degrees of phase shift.

[0076] According to aspects of the present disclosure, there is no transformer which requires a certain coupling factor for the combination of the phase-shifted signals. The only transformers used are configured as baluns 12, 14 for combining the outputs of the differential PAs, and here the coupling can be set freely to achieve the required bandwidth and impedance transformation. Thus, the transformers 12, 14 are natural and obvious parts of the PA’s required output matching network.

[0077] DISO-QHCs 10 according to aspects of the present disclosure allow a reduction in the number of components between the PAs and the antenna, therefore reducing the overall losses. Table 1 below compares the area required for DISO-QHCs 10 according to aspects of the present disclosure, and for different known approaches to PA load impedance variation in beamforming. As explained above, the DISO-QHCs 10 merges components, and in particular reduces the required number of inductors. Accordingly, the area metric in Table 1 is the number of inductors per Transmit / Receive Function (TRF). Because the required input phases can be generated in various ways, the input network is not included in this comparison.

[0078] Table 1 : Comparison of DISO-QHC area to prior art solutions

[0079] DISO-QHCs according to aspects of the present disclosure may have two input signals with 0- and 90-degrees of phase shift, or two phase shifts of any amount that sum to 90° (e.g., -45° and +45°). The DISO-QHC is useful for differential signals; and combines differential input signals into a single-ended output signal. The DISO-QHC dissipates the reflected waves, due to the incoming forward wave from the antenna, despite not having a conventional isolated port.

[0080] In summary, referring to FIGs. 14 and 15:

[0081] • The DISO-QHCs are based on two transformers acting as baluns, one or two phase shifters, and a Wilkinson combiner.

[0082] • The QHC elements are absorbed and merged with the remaining circuit elements to reduce component count.

[0083] • When a single LCL phase shifter is used, a single series capacitance occurs prior to the connection of one the PAs to the combiner, and the second PA connects directly to the combiner. When one LCL and one CLC phase shifter are used, in the second PA path, there will be a series inductance prior to the connection of the combiner.

[0084] • The transformers are natural parts of the PAs’ matching networks, and do not have any specific requirements on their coupling factors. The transformers are used to transform the characteristic impedance of the QHC to the required load impedance of the PA, and perform differential-to-single-ended conversion.

[0085] • The DISO-QHC does not have any specific isolation port, but the isolation resistor within the Wilkinson combiner dissipates the differential and reflected signal from the PAs resulting from the incoming wave reflected from the antenna.

[0086] • In total, the following components are needed to combine two differential PA stages: one (or two) inductors, one resistor, four capacitors, as well as two transformers (which are part of the impedance transformation networks needed at the PA outputs) and corresponding tuning capacitors.

[0087] • The same approach can be used for input matching for the two differential PAs. Only two Wilkinson splitters with adjusted capacitances, two series elements (capacitors / inductors) and one shunt element (inductor / capacitor), are needed.

[0088] The DISO-QHC is also compatible with the load-modulating balanced amplifier (LMBA) approach, as shown in FIG. 19. The LMBA DISO-QHC circuit 50 includes a modified Wilkinson combiner 52, which includes an auxiliary current source 54 at the center point of the resistor. The current source 54 modulates the impedance at each input of the Wilkinson combiner 52. Note that the LMBA DISO-QHC 50 of FIG. 20 has no phase shifting circuit 18 (0° phase shift) at the output of PA1 , and a 90° phase shifting circuit 20 at the output of PA2, which has been combined with other components, as discussed above with reference to FIG. 14.

[0089] Both amplitude and phase of the injected current are used to modulate the impedance seen by the PA ports. FIG. 20 shows this graphically in a Smith chart. With no injected current (ILMBA = 0), VSWR = 1 , defining a point on the Smith chart. When a current equal to that of a PA, and in phase with it, is injected (ILMBA = IRA), then VSWR = 1.7. When that injected current is doubled, VSWR = 3. Altering the phase of the injected current IPA controls the position on the VSWR circle. For example, FIG. 20 shows the “current phase” point on the VSWR circle for 2*1 PA at 90°.

[0090] Table 2 provides a few examples, where the impedances are also expressed in terms of VSWR for clarity. Table 2: Examples of impact of injected current on PA impedances

[0091] FIG. 21 depicts the steps in a method 100 of converting two differential, quadrature RF signals into a single combined RF signal having substantially constant current under dynamically varying load impedance. The two differential, quadrature RF signals are converted into two single-ended, quadrature RF signals in two output impedance matching transformers configured as baluns (block 102). A total of 90° of phase shift is imparted between the two RF signals by at least one output phase shifting circuit connected to a balun output (block 104). The phase-shifted RF signals are combined into a single RF signal an output power combiner circuit (block 106). The method 100 may be performed by a DISO-QHC 10 in any device or node driving antenna arrays that implement beamforming. FIG. 22 for example illustrates a hardware block diagram of a wireless device 80 as implemented in accordance with one or more embodiments. A wireless device 80 is any type of device capable of communicating with a network node and / or access point using radio signals. A wireless device 80 may therefore refer to a machine- to-machine (M2M) device, a machine-type communications (MTC) device, a Narrowband Internet of Things (NB loT) device, etc. The wireless device 80 may also be referred to as a User Equipment (UE), such as a cellular telephone or “smartphone,” however, the term UE should be understood to encompass any wireless device 80. A wireless device 80 may also be referred to as a radio device, a radio communication device, a wireless device, a wireless terminal, or simply a terminal - unless the context indicates otherwise, the use of any of these terms is intended to include device-to-device UEs or devices, machine-type devices, or devices capable of machine-to-machine communication, sensors equipped with a wireless device, wireless-enabled table computers, mobile terminals, smart phones, laptop-embedded equipped (LEE), laptop-mounted equipment (LME), USB dongles, wireless customer-premises equipment (CPE), etc. In the discussion herein, the terms machine-to-machine (M2M) device, machinetype communication (MTC) device, wireless sensor, and sensor may also be used. It should be understood that these devices, although referred to as UEs, but may be configured to transmit and / or receive data without direct human interaction.

[0092] In some embodiments, the wireless device 80 includes a user interface 82 (display, touchscreen, keyboard or keypad, microphone, speaker, and the like); in other embodiments, such as in many M2M, MTC, or NB loT scenarios, the wireless device 80 may include only a minimal, or no, user interface 82 (as indicated by the dashed lines of block 82 in FIG. 22). The wireless device 80 also includes processing circuitry 84; memory 86; and communication circuitry 88 to effect wireless communication across an air interface to one or more radio network nodes, such as a base station, and / or access points. The communication circuitry 88 includes a DISO-QHC 10 associated with each pair of PAs, as described herein. The DISO- QHCs are connected to antenna elements, or subarrays of antenna elements, of an antenna array 89, such as an Advanced Antenna System (AAS), which implements beamforming by phase control. As indicated by the dashed lines, the antenna array 89 may protrude externally from the wireless device 80, or the antenna array 89 may be internal. In some embodiments, a wireless device 80 may include a sophisticated user interface 82, and may additionally include features such as a camera, accelerometer, satellite navigation signal receiver circuitry, vibrating motor, and the like (not depicted in FIG. 22).

[0093] Figure 23 depicts a hardware block diagram of a base station 90 operative in a wireless communication network. The base station 90 includes processing circuitry 92; memory 94; and communication circuitry 96 to effect wireless communication across an air interface to one or more wireless devices 80. The communication circuitry 96 includes a DISO-QHC 10 associated with each pair of PAs, as described herein. The DISO-QHCs are connected to antenna elements, or subarrays of antenna elements, of an antenna array 98, such as an Advanced Antenna System (AAS), which implements beamforming by phase control. As indicated by the broken connection to the antenna array 98, the antenna array 98 may be physically located separately from the base station 90, such as mounted on a tower, building, or the like. Although the memory 96 is depicted as being internal to the processing circuitry 94, those of skill in the art understand that the memory 96 may also be external. Those of skill in the art additionally understand that virtualization techniques allow some functions nominally executed by the processing circuitry 94 to actually be executed by other hardware, perhaps remotely located (e.g., in the so-called “cloud”). The base station 90 is known in LTE as an eNodeB or eNB, and in New Radio (NR) as gNB. In general, in other wireless communication networks, the base station 90 may be known as a Radio Base Station, Base Transceiver Station, Access Point, or the like.

[0094] Note that apparatuses described herein may perform the method 100 herein and any other processing by implementing any functional means, modules, units, or circuitry. In one embodiment, for example, the apparatuses comprise respective circuits or circuitry configured to perform the steps shown in the method figures. The circuits or circuitry in this regard may comprise circuits dedicated to performing certain functional processing and / or one or more microprocessors in conjunction with memory. For instance, the circuitry may include one or more microprocessor or microcontrollers, as well as other digital hardware, which may include digital signal processors (DSPs), special-purpose digital logic, and the like. The processing circuitry may be configured to execute program code stored in memory, which may include one or several types of memory such as read-only memory (ROM), random-access memory, cache memory, flash memory devices, optical storage devices, etc. Program code stored in memory may include program instructions for executing one or more telecommunications and / or data communications protocols as well as instructions for carrying out one or more of the techniques described herein, in several embodiments. In embodiments that employ memory, the memory stores program code that, when executed by the one or more processors, carries out the techniques described herein.

[0095] Those skilled in the art will also appreciate that aspects herein further include corresponding computer programs.

[0096] A computer program comprises instructions which, when executed on at least one processor of an apparatus, cause the apparatus to carry out any of the respective processing described above. A computer program in this regard may comprise one or more code modules corresponding to the means or units described above.

[0097] Aspects further include a carrier containing such a computer program. This carrier may comprise one of an electronic signal, optical signal, radio signal, or computer readable storage medium.

[0098] In this regard, aspects herein also include a computer program product stored on a non- transitory computer readable (storage or recording) medium and comprising instructions that, when executed by a processor of an apparatus, cause the apparatus to perform as described above. Aspects further include a computer program product comprising program code portions for performing the steps of any of the embodiments herein when the computer program product is executed by a computing device. This computer program product may be stored on a computer readable recording medium.

[0099] DISO-QHCs according to aspects of the present disclosure present numerous advantages over prior art solutions to mitigating performance degradation caused by dynamic PA load impedance variations, such as due to beamforming.

[0100] Conventional transformer-based QHCs require a large coupling factor, such as -0.7-0.8, to achieve combination / splitting of signals with 0- and 90-degrees of phase shift. The DISO- QHC eliminates the requirement on a large coupling factor in any transformer, and allows the transformer be used for differential-to-single-ended conversion and impedance transformation. The coupling factor can be set freely and used to achieve the required targeted bandwidth and impedance transformation.

[0101] In a lumped representation, an LCL phase shifter can be represented by two shunt inductors and one in-series capacitor. One of the shunt inductors is merged into a transformer, which in turn can be used for differential-to-single-ended conversion of a differential PA. The second shunt inductor is merged with the capacitance needed for the Wilkinson combiner, resulting in a single equivalent shunt capacitance. This approach reduces the number of passive components inserted between the PA and the antenna port.

[0102] To reduce an amplitude imbalance between the PA outputs, two phase shifters can be used, one LCL as described above, and one CLC phase shifter. The CLC elements can be absorbed by any tuning capacitance needed by the transformers and the Wilkinson combiner. Effectively, the two PAs have either a series capacitance or series inductance before connecting to the Wilkinson combiner.

[0103] The DISO-QHC does not have any specific isolation port, but the isolation resistor within the Wilkinson combiner dissipates the differential and reflected signal from the PAs resulting from the incoming wave reflected from the antenna.

[0104] The DISO-QHC is compatible with an LMBA functionality. By injecting a current at the center point of the isolation resistor in the Wilkinson combiner, the impedance at PA ports can be modulated in both amplitude and phase.

[0105] In total the following additional components are needed to combine two differential PA stages: one (or two) inductor(s), one resistor, four capacitors, and two transformers with corresponding tuning capacitors

[0106] The same approach is applicable to input matching and generating the phases of the signals driving the two differential PA stages. One Wilkinson splitter with adjusted capacitances, two series elements (capacitors / inductors) and one shunt element (inductor / capacitor), are needed for input matching and to generate the phases of the signals driving the PA stages. The present invention may, of course, be carried out in other ways than those specifically set forth herein without departing from essential characteristics of the invention. The present embodiments are to be considered in all respects as illustrative and not restrictive, and all changes coming within the meaning and equivalency range of the appended claims are intended to be embraced therein.

Claims

CLAIMSWhat is claimed is:

1. A Differential Input, Single-ended Output, DISO, Quadrature Hybrid Coupler, QHC (10), configured to receive two differential, quadrature Radio Frequency, RF, signals and output a combined, single-ended RF signal having substantially constant current under dynamically varying load impedance, characterized by: two output impedance matching transformers (12, 14), each receiving one of the differential, quadrature RF signals, and each configured as a balun outputting a single-ended RF signal; at least one output phase shifting circuit (18, 20) connected to a balun (12, 14) output and configured to impart a total of 90° of phase difference between the single- ended RF signals; an output power combiner circuit (22) configured to combine the outputs of the at least one phase shifting circuit (18, 20) and output the combined, single-ended RF signal.

2. The DISO-QHC (10) of claim 1 wherein the at least one output phase shifting circuit (18, 20) comprises one phase shifting circuit of + / - 90°.

3. The DISO-QHC (10) of claim 1 wherein the at least one output phase shifting circuit (18, 20) comprises two phase shifting circuits, one connected to each balun output, that sum to + / - 90° of phase difference.

4. The DISO-QHC (10) of any preceding claim wherein an inductance of a secondary winding of a balun (12, 14) is combined with a phase shifting circuit (18, 20) to reduce component count.

5. The DISO-QHC (10) of any preceding claim wherein the output power combiner (22) is a Wilkinson Combiner.

6. The DISO-QHC (10) of claim 5 wherein the Wilkinson Combiner (22) comprises lumped reactive components.

7. The DISO-QHC (10) of claim 6 wherein inductances or capacitances in the Wilkinson Combiner (22) are combined with inductances or capacitances of one or more output phase shifting circuits (18, 20) to reduce component count.

8. The DISO-QHC (10) of any preceding claim wherein the DISO-QHC (10) consists of two transformers (12, 14) and corresponding tuning capacitors, one or two inductors, one resistor and four capacitors.

9. The DISO-QHC (10) of any preceding claim wherein RF signal components reflected from a load into the output are substantially cancelled.

10. The DISO-QHC (10) of any preceding claim wherein the two differential, quadrature RF signals are generated by two differential Power Amplifiers (PA1 , PA2), and further comprising: an input power combiner circuit (24), configured as a splitter operative to receive a single-ended RF input signal and output two single-ended RF input signals; at least one input phase shifting circuit (26, 28) connected to an output of the input power splitter circuit (24) and configured to impart a total of 90° of phase difference between the two single-ended RF input signals; and two input impedance matching circuits (30, 32), each receiving a single-ended RF signal, and each configured as a balun generating a differential RF signal, wherein the differential balun outputs are connected to the inputs of the two differential Power Amplifiers (PA1 , PA2).

11. The DISO-QHC (50) of any of claims 5-10, wherein a resistor of the Wilkinson combiner circuit (52) comprises two resistors connected in series, and further comprising: a current source (54) connected between the two resistors of the Wilkinson combiner circuit (52), whereby the load impedances seen by Power Amplifiers (PA1 , PA2) generating the two differential, quadrature RF signals are modulated by controlling the frequency, phase, and amplitude of current output by the current source (54).

12. A method (100) of converting two differential, quadrature Radio Frequency, RF, signals into a combined single-ended RF signal having substantially constant current under dynamically varying load impedance, characterized by: converting the two differential, quadrature RF signals into two single-ended, quadrature RF signals in two output impedance matching transformers (12, 14) configured as baluns; imparting a total of 90° of phase difference between the two single-ended, quadrature RF signals by at least one output phase shifting circuit (18, 20) connected to a balun (12, 14) output; and combining the outputs of the at least one phase shifting circuit (18, 20) into one single- ended RF signal in an output power combiner circuit (22).

13. The method (100) of claim 12 wherein imparting a total of 90° of phase difference between the two single-ended, quadrature RF signals comprises imparting Qi° of phase shift to one RF signal and Qi+90° of phase shift to the other RF signal.

14. The method (100) of claim 13 wherein Qi=0°.

15. The method (100) of any of claims 12-14 further comprising reducing component count by combining an inductance of a secondary winding of a balun (12, 14) with a phase shifting circuit (18, 20).

16. The method (100) of any of claims 12-15 wherein the output power combiner (22) is a Wilkinson Combiner comprising lumped reactive components.

17. The method (100) of claim 16 further comprising reducing component count by combining inductances or capacitances in the Wilkinson Combiner (22) with inductances or capacitances of one or more output phase shifting circuits (18, 20).

18. The method (100) of any of claims 12-17 further comprising substantially cancelling RF signal components reflected from a load into the output.

19. The method (100) of any of claims 12-18 wherein the two differential, quadrature RF signals are generated by two differential Power Amplifiers (PA1 , PA2), and further comprising: converting a single-ended RF input signal into two single-ended RF input signals in an input power combiner circuit (24) configured as a splitter; imparting a total of 90° of phase difference between the two single-ended RF input signals in at least one input phase shifting circuit (26, 28) connected to an output of the input power splitter circuit (24); and converting the two single-ended, quadrature RF signals into differential quadrature RF signals in two input impedance matching circuits (30, 32); and inputting the differential quadrature RF signals to the two differential Power Amplifiers (PA1 , PA2).

20. The method (100) of claim 19 wherein the two input impedance matching circuits(30, 32) comprise impedance matching transformers, each configured as a balun generating a differential output signal from a single-ended input signal.

21. The method (100) of any of claims 16-20, wherein a resistor of the Wilkinson combiner circuit (22) comprises two resistors connected in series, and further comprising: modulating the load impedances seen by Power Amplifiers (PA1 , PA2) generating the two differential, quadrature RF signals by injecting a current between the two resistors of the Wilkinson combiner circuit (22), and by controlling the frequency, phase, and amplitude of the injected current.

22. User Equipment, UE (80), operative in a wireless communication network, comprising: processing circuitry (84); and communication circuitry (88) operatively connected to the processing circuitry (84) and comprising differential RF Power Amplifiers, PAs (PA1 , PA2), arranged in pairs; and a Differential Input, Single-ended Output, DISO, Quadrature Hybrid Coupler, QHC (10), associated with each pair of PAs (PA1, PA2), the DISO-QHC (10) configured to receive two differential, quadrature Radio Frequency, RF, signals from the PAs (PA1 , PA2) and output a combined, single- ended RF signal having substantially constant current under dynamically varying load impedance, the DISO-QHC (10) characterized by: two output impedance matching transformers (12, 14), each receiving one of the differential, quadrature RF signals, and each configured as a balun outputting a single-ended RF signal; at least one output phase shifting circuit (18, 20) connected to a balun (12, 14) output and configured to impart a total of 90° of phase difference between the single-ended RF signals; and an output power combiner circuit (22) configured to combine the outputs of the at least one phase shifting circuit (18, 20) and output the combined, single-ended RF signal.

23. A base station (90) operative in a wireless communication network, comprising: processing circuitry (92); and communication circuitry (96) operatively connected to the processing circuitry (92) and comprising differential RF Power Amplifiers, PAs (PA1 , PA2), arranged in pairs; and a Differential Input, Single-ended Output, DISO, Quadrature Hybrid Coupler, QHC (10), associated with each pair of PAs (PA1, PA2), the DISO-QHC (10) configured to receive two differential, quadrature Radio Frequency, RF, signals from the PAs (PA1, PA2) and output a combined, single-ended RF signal having substantially constant current under dynamically varying load impedance, the DISO-QHC (10) characterized by: two output impedance matching transformers (12, 14), each receiving one of the differential, quadrature RF signals, and each configured as a balun outputting a single-ended RF signal; at least one output phase shifting circuit (18, 20) connected to a balun (12, 14) output and configured to impart a total of 90° of phase difference between the single-ended RF signals; and an output power combiner circuit (22) configured to combine the outputs of the at least one phase shifting circuit (18, 20) and output the combined, single-ended RF signal.

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