Regulation system for a converter with multiple active bridges with hybrid power supply, associated method and device

Abstract

Regulation system for a converter with multiple active bridges with hybrid power supply, associated method and device The present invention is a regulation system (28) for a converter (10) with multiple active bridges with hybrid power supply comprising: - an input current port (20) comprising a bridge of H switches (30) of which a switch (T1) is controlled by a first law having a first duty cycle, - at least one output voltage port (22, 24) comprising a bridge of H switches (38, 40) of which a switch (T21, T31) is controlled by a second law out of phase with respect to the first law, the regulation system (28) regulating the converter (10) to a setpoint comprising a setpoint current value for the current of the current port (20) and a setpoint voltage value for a voltage port (22, 24). . Figure for abstract: figure 3

Description

Title of the invention: System for regulating a converter with multiple active bridges with hybrid power supply, associated method and device FIELD OF THE INVENTION

[0001] The present invention relates to a system for regulating a converter with multiple active bridges with hybrid power supply. The invention also relates to an associated regulation method and converter. TECHNOLOGICAL BACKGROUND OF THE INVENTION

[0002] The integration of renewable energy sources and energy storage systems into power electronic applications has generated interest in multi-port converters.

[0003] A specific topology of such converters is that of MAB converters, the acronym MAB referring to the English term “Multi-Active Bridge” literally meaning “Multiple active bridges”.

[0004] An example of using such a converter is shown in [Fig.l] where the converter interacts with the network, a battery, a load (resistance in this figure) and a solar panel.

[0005] This shows that such a structure has the advantage that the production, consumption and storage of electrical energy can be carried out in a single location.

[0006] Furthermore, MAB converters have intrinsic galvanic isolation due to the fact that the transformer connects the ports via respective windings. This aspect is important to enable the converters to be used for energy and load sources with large differences.

[0007] It is known to use MAB converters with voltage ports.

[0008] However, for certain uses and in particular for cases of photovoltaic panels, it is desirable to use bridges supplied with current but this leads to a hybrid power supply converter whose overvoltage control is delicate. Summary of the invention

[0009] There is therefore a need for a system for regulating a converter with multiple active bridges with hybrid power supply making it possible to ensure proper operation of the converter.

[0010] For this purpose, the description describes a system for regulating a converter with multiple active bridges with hybrid power supply, the converter with multiple active bridges comprising:

[0011] - an input current port comprising an H-switch bridge for which a reference switch is defined, the reference switch being called the first reference switch and being controlled by a first control law, the first control law having a first duty cycle,

[0012] - at least one output voltage port comprising an H-switch bridge for which a reference switch is defined, each reference switch being called a second reference switch and being controlled by a respective second control law, each second control law having a second predefined duty cycle and a phase shift relative to the first control law,

[0013] - a transformer having windings, each winding being connected to a port by a respective isolation interface,

[0014] the regulation system being capable of regulating the converter with multiple active bridges to a setpoint, the setpoint comprising a setpoint current value for the current of the current port and at least one setpoint voltage value for a voltage port, the regulation system comprising:

[0015] - a unit of measurement of measured values, the measured values ​​comprising the current of the current port and the voltage of each voltage port,

[0016] - a first control unit of the first reference switch, the first control unit comprising:

[0017] - a first sub-unit for calculating a first desired initial value, the first desired initial value being a desired value for the first duty cycle equal to the sum of the result of a first calculation function applied to the setpoint values ​​and a safety margin,

[0018] - a first subunit for determining the difference between the current value measured and the set current value, to obtain a determined current deviation,

[0019] - a first correction subunit suitable for converting the current deviation determined a first corrective value for the first duty cycle by applying a first conversion function,

[0020] - a first subunit for adding the first desired initial value and the first corrective value, to obtain a first candidate value,

[0021] - a first adjustment subunit suitable for adjusting the first candidate value to obtain a first value to be applied between two first extreme values, the lowest first extreme value being the result of the first calculation function applied to the measured values, - a first sub-unit for applying the first control law having as its duty cycle the first value to be applied, and

[0022] - for each voltage port having a set voltage value, a second control unit of the second reference switch of the voltage port considered, each second control unit comprising:

[0023] - a second subunit for calculating a second desired initial value, the second desired initial value being a desired value for the phase shift equal to the result of a second calculation function applied to the first desired initial value,

[0024] - a second subunit for determining the difference between the voltage value measured for the voltage port considered and the set voltage value for the voltage port considered, to obtain a determined voltage difference,

[0025] - a second correction subunit suitable for converting the voltage difference determined a second correction value for the phase shift by applying a second conversion function,

[0026] - a second subunit for adding the second desired initial value and the second corrective value, to obtain a second candidate value,

[0027] - a second adjustment subunit suitable for adjusting the second value candidate to obtain a second value to be applied between two second extreme values, the second lowest extreme value being the result of the second calculation function applied to the first desired initial value, and

[0028] - a second sub-unit for applying the second control law having as phase shift the second value to apply.

[0029] According to particular embodiments, the regulation system has one or more of the following characteristics, taken in isolation or in all technically possible combinations:

[0030] - each adjustment subunit is capable of giving as the value to be applied the candidate value unchanged when the candidate value is between the two extreme values ​​and otherwise the extreme value closest to the candidate value.

[0031] - the second highest extreme value is the result of a third function of calculation applied to the first desired initial value.

[0032] - at least one calculation function among the second calculation function and the third calculation function is an affine function.

[0033] - each correction subunit is a proportional integral corrector.

[0034] - each conversion function is a first-order function.

[0035] - each H-bridge has two midpoints each isolation interface comprising two lines connecting a respective midpoint with one end of the associated winding, one of the two lines comprising a resistance in series with an inductance.

[0036] - each winding has turns, the first conversion function having a gain depending on the measured values ​​for the voltages, the desired second initial value, the number of turns of each winding and the inductances of the insulation interfaces.

[0037] - each winding has turns, the second conversion function having a gain depending on the measured current value, the first desired initial value, the second desired initial value, the number of turns of each winding and the inductances of the insulation interfaces.

[0038] - each winding has turns, the first calculation function (dependent, in addition, the number of turns of each winding and the inductances of the insulation interfaces.

[0039] - an output voltage port is a bidirectional port.

[0040] - each determination subunit is a subtractor.

[0041] - the safety margin is a predefined value.

[0042] The description also describes a hybrid-powered multi-active bridge converter, the multi-active bridge converter comprising:

[0043] - an input current port operating at a current and comprising a bridge of H-switches for which a reference switch is defined and operating at a current, the reference switch being called the first reference switch and being controlled by a first control law, the first control law having a first duty cycle,

[0044] - at least one output voltage port operating at a voltage, each output voltage port voltage comprising an H-switch bridge for which a reference switch is defined, each reference switch being called a second reference switch and being controlled by a respective second control law, each second control law having a second predefined duty cycle and a phase shift relative to the first control law,

[0045] - a transformer having windings, each winding being connected to a port by a respective isolation interface, and

[0046] - a regulation system as previously described.

[0047] The description also provides a method for regulating a hybrid-powered multi-active bridge converter, the multi-active bridge converter comprising:

[0048] - an input current port comprising an H-switch bridge for which a reference switch is defined, the reference switch being called the first reference switch and being controlled by a first control law, the first control law having a first duty cycle,

[0049] - at least one output voltage port comprising an H-switch bridge for which a reference switch is defined, each reference switch being called a second reference switch and being controlled by a respective second control law, each second control law having a second predefined duty cycle and a phase shift relative to the first control law,

[0050] - a transformer having windings, each winding being connected to a port by a respective isolation interface,

[0051] the method being suitable for regulating the converter with multiple active bridges to a setpoint, the setpoint comprising a setpoint current value for the current of the current port and at least one setpoint voltage value for a voltage port, the method comprising the steps of:

[0052] - measurement of measured values, the measured values ​​including the current of the port of current and voltage of each voltage port,

[0053] - control of the first reference switch, the control step including:

[0054] - a first calculation of a first desired initial value, the first value initial desired being a desired value for the first duty cycle equal to the sum of the result of a first calculation function applied to the setpoint values ​​and a safety margin,

[0055] - a first determination of the difference between the measured current value and the setpoint current value, to obtain a determined current deviation,

[0056] - a first correction converting the determined current deviation into a first corrective value for the first duty cycle by applying a first conversion function,

[0057] - a first addition of the first desired initial value and the first corrective value, to obtain a first candidate value,

[0058] - a first adjustment suitable for adjusting the first candidate value to obtain a first value to be applied between two first extreme values, the lowest first extreme value being the result of the first calculation function applied to the measured values,

[0059] - a first application of the first control law having as ratio cyclic the first value to be applied, and

[0060] - for each voltage port having a set voltage value, command of the second reference switch of the voltage port considered, each control step of the second switch comprising:

[0061] - a second calculation of a second desired initial value, the second value desired initial being a desired value for the phase shift equal to the result of a second calculation function applied to the first desired initial value,

[0062] - a second determination of the difference between the voltage value measured for the voltage port considered and the set voltage value for the voltage port considered, to obtain a determined voltage difference,

[0063] - a second correction converting the determined voltage deviation into a second corrective value for the phase shift by application of a second conversion function,

[0064] - a second addition of the second desired initial value and the second corrective value, to obtain a second candidate value,

[0065] - a second adjustment suitable for adjusting the second candidate value, for obtaining a second value to be applied between two second extreme values, the second lowest extreme value being the result of the second calculation function applied to the first desired initial value, and

[0066] - a second application of the second control law having as phase shift the second value to apply.

[0067] In the present description, the expression “suitable for” means indifferently “adapted for”, “adapted to” or “configured for”. Brief description of the drawings

[0068] Characteristics and advantages of the invention will appear on reading the description which follows, given solely by way of non-limiting example, and made with reference to the appended drawings, in which:

[0069] - [Fig.l] [Fig.l] is a schematic representation of an example of use of a converter with multiple active bridges with hybrid power supply,

[0070] - [Fig.2] [Fig.2] represents an electrical diagram of an example of a converter to multiple active bridges with hybrid power supply, the converter including a regulation system,

[0071] - [Fig.3] [Fig.3] represents a diagram of a part of the regulation system,

[0072] - [Fig.4] [Fig.4] represents a diagram of another part of the regulation system,

[0073] - [Fig.5] [Fig.5] represents the evolution of the input and output signals of the converter of [Fig.2] in ideal operation,

[0074] -1[Fig.6]] [Fig.6] schematically illustrates the star-delta conversion, and

[0075] - [Fig.7]l[Fig.8]][Fig.9][Fig.lO] [Fig.7] to 10 present simulation results obtained by the applicant.

[0076] DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

[0077] According to the example described, the converter 10 with multiple active bridges with hybrid power supply comprises three ports 20, 22 and 24, a transformer 26 and a regulation system 28.

[0078] In this sense, the converter 10 is a triple active bridge converter.

[0079] Such a converter 10 is sometimes called a TAB converter, the acronym “TAB” referring to the corresponding English name of “Triple Active Bridge”.

[0080] According to the example of [Fig.2], the converter 10 comprises an input current port 20, a first voltage port 22 and a second output voltage port 24.

[0081] The first voltage port 22 is here both an input port and an output port (bidirectional character).

[0082] Such a converter 10 may, for example, be used to connect a photovoltaic module to the input current port 20, a battery to the first voltage port 22 and a direct current load to the second output voltage port 24.

[0083] In such a case, the input current reference of the input current port 20 would correspond to the tracking of the maximum power point of the photovoltaic module. This tracking is more often designated by the English abbreviation MPPT which refers to the corresponding English name of “Maximum Power Point Tracker”.

[0084] The current port 20 comprises a current source 30 and a first H-switch bridge 32.

[0085] The current source 30 is capable of delivering a current denoted l\ for the following.

[0086] The current source 30 comprises a voltage generator 34, a resistor Rf and an inductance Lf in series.

[0087] In each of the following notations, to facilitate readability, the reference sign of a component may correspond to its value. Typically, the resistance of the current source 30 is denoted by the reference sign Rf and the value of the resistance is denoted Rf.

[0088] The voltage generator 34 produces a voltage Vp

[0089] The voltage generator 34 has a positive pole and a negative pole.

[0090] The current source 30 is capable of operating at a current determining the operating current of the current port 20.

[0091] Current port 20 is a unidirectional port.

[0092] This means that the current port 20 is only suitable for sending power to the transformer 26.

[0093] To simplify the notations, each H-switch bridge will be called an H-bridge.

[0094] The first H-bridge 32 comprises four branches 32B1 to 32B4, a first end 32E1 and a second end 32E2 and two midpoints 32M1 and 32M2.

[0095] The first branch 32B1 extends between the first end 32E1 and the first midpoint 32M1, the second branch 32B2 extends between the first end 32E1 and the second midpoint 32M2, the third branch 32B3 extends between the first midpoint 32M1 and the second end 32E2 and the fourth branch 32B4 extends between the second midpoint 32M2 and the second end 32E2.

[0096] The first end 32E1 is connected to the inductance Lf of the current source 30 and the second end 32E2 to the negative pole of the voltage generator 34.

[0097] As its name “switch bridge” indicates, each branch 32B1 to 32B4 of the first H-bridge 32 comprises a switch Ti to T4 respectively.

[0098] Each switch here is a transistor.

[0099] Furthermore, since each switch of an H-bridge has in the example described an all-or-nothing operation, the term switch will be used in the remainder of the description to designate the transistors. Nevertheless, what will be described is valid more generally for a switch instead of a switch.

[0100] Furthermore, each branch 32B1 to 32B4 of the first H-bridge 32 comprises two diodes, a diode DA1 to DA4 in parallel with the switch Ti to T4 and a diode DI to D4 in series with the switch Ti to T4.

[0101] Furthermore, the first switch Ti is defined as the reference switch of the current port 20 and will be referred to as the first reference switch Ti in the following.

[0102] The first reference switch Ti of the current port 20 is controlled by a first control law LCp. The first control law LCi has a first duty cycle noted Di.

[0103] The fourth switch T4 is controlled by the same first control law LCi while the second switch T2 and the third switch T3 are controlled by the same law phase-shifted by 180° relative to the first control law LCp. The phase shift of 180° corresponds to a time shift of half the switching period Ts / 2.

[0104] In this respect, it would be possible to apply what will be described by choosing as the first reference switch one of the switches T2, T3 or T4.

[0105] The first voltage port 22 comprises a second voltage source 36 and a second H-bridge 38.

[0106] The first voltage port 22 operates at a voltage, called the second voltage V2. This second voltage V2 corresponds here to the voltage delivered by the second voltage source 36.

[0107] The first voltage port 22 is a bidirectional port.

[0108] This means that the first voltage port 22 receives power from the transformer 26 and delivers power through the transformer 26.

[0109] The second H-bridge 38 has a structure similar to that of the first H-bridge 30 (that of the current port 20) with four branches 38B1 to 38B4, ends 38E1 and 38E2 and two midpoints 38M1 and 38M2.

[0110] In the case shown, each branch 38B1 to 38B4 of the second H-bridge 38 also comprises a transistor and a diode in parallel similarly to the branches of the first H-bridge 30 but does not comprise a diode in series.

[0111] For the rest, the transistor of a branch 38Bi of the second H bridge 38 is noted T2i.

[0112] The first end 38E1 of the second H-bridge 38 is connected to the positive pole of the second voltage source 36 and the second end 38E2 of the second H-bridge 38 is connected to the negative pole of the second voltage source 36.

[0113] Furthermore, the first switch T2i is defined as the reference switch of the first voltage port 22, which will hereinafter be the second reference switch T2i to avoid any confusion with the first reference switch Tb

[0114] The second reference switch T2[ of the first voltage port 22 is controlled by a second control law LC2.

[0115] The second control law LC2 presents a second duty cycle noted D2 and a phase shift, noted second ^2, relative to the first control law LCp

[0116] For the remainder, the second duty cycle noted D2 is set to a predefined value, chosen in a non-limiting manner here as equal to 0.5.

[0117] The second voltage port 24 comprises a load 39 and a third H-bridge 40.

[0118] Load 39 comprises a capacitor C3 parallel with a resistor RL3.

[0119] The second voltage port 24 operates at a voltage, called third voltage V3. This third voltage V3 corresponds here to the voltage across the resistor RL3.

[0120] The second voltage port 24 is a unidirectional port.

[0121] This means that the second voltage port 24 receives power from the transformer 26.

[0122] The third H-bridge 40 has a structure similar to that of the second H-bridge 38 (that of the first voltage port 22) with four branches 40B1 to 40B4, ends 40E1 and 40E2 and two midpoints 40M1 and 40M2.

[0123] In the case shown, each branch 40B1 to 40B4 comprises the same components as branches 38B1 to 38B4 of the third H-bridge 38, namely a transistor and a diode in parallel.

[0124] For the rest, the transistor of a branch 40B1 to 40B4 of the third H-bridge 40 is noted T3i.

[0125] The first end 40E1 of the third H-bridge 40 is connected to one terminal of the capacitor C3 and one terminal of the resistor RL3 so that the second end 40E2 of the third H-bridge 40 is connected to another terminal of the capacitor C3 and another terminal of the resistor RL3.

[0126] Furthermore, the first switch T3i is defined as the reference switch of the second voltage port 24, which will subsequently be the third reference switch T3i to avoid any confusion with the other reference switches Ti and T21.

[0127] The third reference switch T3[ of the second voltage port 24 is controlled by a third control law LC3.

[0128] The third control law LC3 presents a third duty cycle noted D3 and a phase shift, noted third phase shift ^3, relative to the first control law LCi.

[0129] For the remainder, the third duty cycle noted D3 is set to a predefined value, chosen in a non-limiting manner here as equal to 0.5.

[0130] The transformer 26 has windings 42, 44 and 46 connected by a core 47

[0131] Each winding 42, 44 and 46 is connected to a port 20 to 24 by a respective isolation interface 48 to 52.

[0132] More specifically, the transformer 26 has a first winding 42 connected to the current port 20 by a first insulation interface 48, a second winding 44 connected to the first voltage port 22 by a second insulation interface 50 and a third winding 46 connected to the second voltage port 24 by a third insulation interface 52.

[0133] Each winding is a set of turns extending between a first end denoted XE1 (where X is the reference sign of the winding considered) and a second end XE2.

[0134] The first isolation interface 48 comprises a first inductance Li in series with a first resistance Rb

[0135] The first midpoint 32M1 of the first H-bridge 32 is connected to the first resistor Ri and the second midpoint 32M2 of the first H-bridge 32 is connected to the second end 42E2 of the first winding 42.

[0136] The second isolation interface 50 comprises a second inductor L2 in series with a second resistor R2.

[0137] The first midpoint 38M1 of the second H-bridge 38 is connected to the inductance L2 and the second midpoint 38M2 of the second H-bridge 38 is connected to the second end 44E2 of the second winding 44.

[0138] The third isolation interface 52 comprises a third inductor L3 in series with a third resistor R3.

[0139] The first midpoint 40M1 of the third H-bridge 40 is connected to the third inductance L3 and the second midpoint 40M2 of the third H-bridge 40 is connected to the second end 46E2 of the third winding 46.

[0140] In each of the insulation interfaces 48, 50 or 52, the resistance Rb R2 or R3 and the inductance Lb L2 or L3 are those of the associated windings 42, 44 or 46. As such, the resistance RB R2 or R3 corresponds to a parasitic resistance.

[0141] In some embodiments, these resistors or inductors come from external components.

[0142] The regulation system 28 is capable of regulating the converter 10 with multiple active bridges to a setpoint.

[0143] In this case, the setpoint is a setpoint current value for the current of the current port 20 and a setpoint voltage value for the second voltage port 24.

[0144] In fact, in the example described, the first voltage port 22 has no setpoint, the voltage of the first voltage port 22 is imposed by the voltage value V2 delivered by the second voltage source 36 and its current is imposed by the fact that the algebraic sum of the incoming and outgoing powers of converter 10 is zero.

[0145] The setpoint current value is denoted lïyef, and the setpoint voltage value for the second voltage port 24 is denoted Vjref.

[0146] The regulation system 28 ensures that at all times, the operation of the converter 10 does not damage any component of the converter 10.

[0147] The various conditions are detailed in the Appendix section.

[0148] The regulation system 28 comprises a measuring unit 54, a first control unit 56, a second control unit 58 and a third control unit 60.

[0149] The measuring unit 54 is capable of measuring several values ​​of the converter 10.

[0150] According to the example described, the measured values ​​include the current Ii of the current port 20 and the voltage V2 and of each voltage port 22 and 24.

[0151] The measuring unit 54 is connected to the other control units 56 to 60 to communicate the measured values ​​to them.

[0152] The first control unit 56 is capable of controlling the first reference transistor Ti using a first adapted control law LCi.

[0153] As visible in Figures 2 and 3, the first control unit 56 comprises a first calculation sub-unit 62, a first determination sub-unit 64, a first correction sub-unit 66, a first addition sub-unit 68, a first adjustment sub-unit 70 and a first application sub-unit 72.

[0154] The first calculation sub-unit 62 is capable of calculating a first desired initial value noted D^.

[0155] For this, the first calculation sub-unit 62 applies a first calculation function FC1 to the setpoint values ​​then adds a safety margin £ to the result.

[0156] The first calculation function FC1 further depends on the number of turns of each winding 42 to 46 and the inductance of the insulation interfaces 48 to 52.

[0157] The result of the first calculation function FC1 is noted

[0158] The first calculation function FC1 is such that:

[0159] n__w i + 2

[0160] where: • is the number of turns of the first winding 42, • w2 is the number of turns of the second winding 44, • n3 is the number of turns of the third winding 46, • vV, with r ' _ f ù2 r for k= 1,2 or 3 r _ ^k ~ l nk / ^k b~ • Ts is the switching period corresponding to the inverse of the conversion frequency fs chosen 1.

[0161] Once this value Dj mjn_rey is obtained, it comes:

[0162] £

[0163] According to the example described, the safety margin £ is a predefined value, for example set at 0.01.

[0164] According to a more elaborate variant, the safety margin £ takes into account other elements such as variations in voltage sources or uncertainty in the value of the inductance.

[0165] The first determination sub-unit 64 is capable of determining the difference between the measured current value I j and the setpoint current value 1\ rej\

[0166] The first determination subunit 64 is, here, a subtractor.

[0167] The first determination subunit 64 thus obtains the current deviation determined, noted and which therefore verifies:

[0168] M 1 =

[0169] The first correction sub-unit 66 is capable of converting the determined current deviation into a first corrective value for the first duty cycle.

[0170] The first corrective value is noted ADh

[0171] To obtain the first correction value AZ)b the first correction sub-unit 66 applies a first conversion function to the determined current difference A / j.

[0172] The first conversion function is denoted G^.

[0173] The first conversion function here is a first-order function.

[0174] More specifically, the first conversion function G^ has a gain depending on the measured values ​​for the voltages, the desired second initial value, the number of turns of each winding 42 to 46 and the inductance of the insulation interfaces 48 to 52.

[0175] According to the example described, the first conversion function Gflr is such that:

[0176] / Ad AT)

[0177] Where: • s denotes the Laplace variable, ■.........., a - Lsa-Ltl)-fh-V 2+ Lfi-La)-n}-v 3jef • the notation A means small variation around the equilibrium point, • , And . 4Lj C=L~f +ÂÜXÎ7)'

[0178] The first correction sub-unit 66 thus forms a proportional-integral corrector.

[0179] Alternatively, it is also possible that the first correction subunit 66 uses a more elaborate first correction function as proposed in the appendix.

[0180] The first addition subunit 68 is capable of obtaining a first candidate value by adding the first desired initial value Duq with the first corrective value ADb

[0181] It comes like this:

[0182] Dl£-Dyq + ADl

[0183] The first adjustment sub-unit 70 is capable of obtaining a first value to be applied DUtpp.

[0184] For this, the first adjustment sub-unit 70 adjusts the first candidate value D\e to obtain the first value to be applied Duipp.

[0185] This adjustment aims to ensure that the first value to be applied D\ppp is between two first extreme values.

[0186] The first two extreme values ​​are noted and

[0187]

[0188]

[0189]

[0190]

[0191]

[0192]

[0193]

[0194]

[0195]

[0196]

[0197]

[0198]

[0199]

[0200] The first extreme value I>imm is the result of the first calculation function FC1 applied to the measured values ​​according to the formula described above, that is to say it comes: D1'min = + 2 According to the example described, the first extreme value is set to the maximum value for the first duty cycle, namely 1. The first adjustment sub-unit 70 is capable of giving as the value to be applied the unchanged candidate value when the candidate value is between the two extreme values ​​and otherwise the extreme value closest to the candidate value. In other words, it comes: ^iupp “ [ ^l.miw D}jnax ] 1 lapp — l,c < , &\ptpp~ ^l,max^ > ^Ljnax This gives a first control law to apply, this first control law having as its duty cycle the first value to apply. The first application subunit 72 is capable of applying the first control law to be applied. The second control unit 58 and the third control unit 60 respectively control the second reference switch T2i and the third reference switch T3[. The third control unit 60 being the control unit comprising the most sub-units because a setpoint is fixed for the third reference switch T3i, it is first described before the second control unit 58. As seen in [Fig.4], the third control unit 60 comprises a third calculation sub-unit 82, a third determination sub-unit 84, a third correction sub-unit 86, a third addition sub-unit 88, a third adjustment sub-unit 90 and a third application sub-unit 92. The third calculation sub-unit 82 is suitable for calculating a third desired initial value denoted cA<^. For this, the third calculation sub-unit 82 applies a second calculation function FC2 to the first desired initial value. The second calculation function FC2 is here a function calculated from the mathematical model presented in the following part, at a certain operating point.

[0201] As will appear, the value is obtained by solving a system of two equations with two unknowns once the value of the first duty cycle is fixed.

[0202] The third determination sub-unit 84 is capable of determining the difference between the voltage value measured for the second output voltage port 24 and the set voltage value V^f for the second output voltage port 24.

[0203] The third determination subunit 84 is, here, a subtractor.

[0204] The third determination subunit 84 thus obtains the voltage difference determined, noted AV3 and which therefore verifies:

[0205] AV3=Vw-y3

[0206] The third correction sub-unit 86 is capable of converting the determined voltage difference AV3 into a third corrective value for the third phase shift.

[0207] The third corrective value is noted A(pv

[0208] To obtain the third corrective value A <p3, la troisième sous-unité de correction 88 applique une deuxième fonction de conversion sur l’écart de tension déterminé AV3.

[0209] The second conversion function is denoted Gv^r.

[0210] The second conversion function Gv^r is here a first-order function.

[0211] More specifically, the second conversion function Gv^ has a gain depending on the measured current value of the first desired initial value, the second desired initial value, the number of turns of each winding 42 to 46 and the inductance of the insulation interfaces 48 to 52.

[0212] According to the example described, the second transfer function Gv^ is such that:

[0213] r ^3.1 M | - (s+d)

[0214] Where: • j--L_, and a - C3RL}

[0215] & ~ + + ) • R in subscript designates the real value of the first harmonic (of current here), • I in subscript designates the imaginary value, • is the current flowing in the inductance Ln of [Fig.6] (see details with reference to this figure), and • is the current flowing in the inductance L23 of [Fig.6] (see details with reference to this figure).

[0216] The third correction subunit 86 thus forms a proportional-integral corrector.

[0217] The third addition subunit 88 is capable of obtaining a third candidate value by adding the third desired initial value ^^q with the third corrective value

[0218] It comes like this:

[0220] The third adjustment sub-unit 90 is capable of obtaining a third value to be applied (P^PP.

[0221] For this, the third adjustment sub-unit 90 adjusts the third candidate value to obtain the third value to be applied

[0222] This adjustment aims to ensure that the third value to be applied ^app is between two third extreme values.

[0223] The two third extreme values ​​are noted ^min and ^max.

[0224] Each of the third extreme values ​​and ^max is the result of the application of a respective affine function applied to the first desired initial value D^q.

[0225] More specifically, the third extreme value ^^nin is the result of the second calculation function FC2 applied to the first desired initial value so that in this case:

[0226] m — tt( Ds

[0227] For the third extreme value is the result of a third calculation function FC3 applied to the first desired initial value D\£q. The third calculation function FC3 is such that:

[0228] (p

[0229] The third adjustment sub-unit 90 is then capable of giving as the value to be applied the unchanged candidate value when the candidate value is between the two extreme values ​​and otherwise the extreme value closest to the candidate value. In other words, it comes: if e [ 1 ' 3,upp r 3.c ' 3p [ ' ijmn ' 3,max J ». — . if (D- < (D- . T .\app r \imn r 3fi r 3,mm (p^ = (p. If (pQ > 0, T3#pp r ^nax ' 3fi r 3puix This gives a third control law to apply, this third law

[0230]

[0231] ■

[0232] control having as duty cycle the third predefined value and as phase shift the third value to be applied ^3PPP.

[0233] The third application sub-unit 92 is capable of applying the third control law to be applied.

[0234] As regards the second control unit 58, compared to the other two control units 54 and 60, it operates in open loop.

[0235] This implies that the second control unit 58 comprises only a second calculation sub-unit 94, a second adjustment sub-unit 96 and an application sub-unit 98.

[0236] The operations of these second subunits 94, 96 and 98 are similar to those described for the corresponding third subunits (the subunits 82, 90 and 92).

[0237] Due to its open-loop operation, the second control unit 58 does not include second sub-units having the role of the third sub-units 84 to 88.

[0238] As explained in the demonstration section, the regulation system 28 thus makes it possible to ensure stable operation of the converter 10, for any operating point.

[0239] Furthermore, the regulation system 28 can be applied to other configurations of the converter 10.

[0240] Examples of such configurations are given in Part 2 of the Annex. ANNEX 1 - GENERAL CASE 1.1 - Converter topology

[0241] The converter 10 is a hybrid DC-DC multi-port converter having a DC-AC current switch on the current port side and two voltage inverters on the voltage port side.

[0242] The abbreviation "DC" refers to the corresponding English name "Direct Current" which literally means "direct current" while the abbreviation "AC" refers to the corresponding English name "Alternative Current" which literally means "alternating current"

[0243] [Fig. 2] shows the presence of a temporary current source 26 at current port 20 (port 1) and H-bridges 38 and 40 at the first voltage port 22 (port 2) and the second voltage port 24 (port 3). Current port 20 represents a power source (e.g., a photovoltaic module), first voltage port 22 (port 2) is bidirectional (e.g., battery system and electrical grid), and second voltage port 2 (port 3) is a DC load.

[0244] The ports 20, 22 and 24 of this converter 10 are coupled due to the presence of an inductance on each isolation interface 48 to 52, which can be either a leakage inductance of the transformer 26 alone or in series with an external inductance. Consequently, this converter 10 behaves like a multi-system variable having multiple inputs and multiple outputs (MIMO). In other words, changing a control parameter has an effect on ports 20, 22 and 24 of the converter 10.

[0245] Controlling the power flow of this converter 10 can be achieved by regulating the input current h of the current port 20 and the output voltage V3 of the second voltage port 24.

[0246] Because the system is a coupled multiport 10 converter, the algebraic sum of all system input and output powers should approximately be 0 (or equal to the system losses).

[0247] Thus, the power flowing from and into the first voltage port 22 is imposed according to the following formula (neglecting the power stored in the magnetic core 47 of the transformer 26):

[0248] P2= prP3

[0249] Where: • P\ is the power flow from and into current port 20, and • P^ is the power flow from and into the second port of voltage 24.

[0250] The waveform of the AC signals of the converter 10 circulating in the windings 42, 44, 46 of the transformer 26 are visible in [Fig.5] as well as the control signals of the reference switches Tp T2p and T3p

[0251] The trapezoidal shape of the evolution of the current iu shown in Figure 5 corresponds to the current flowing in the first winding. The parameters r2 and are respectively the AC voltages on the transformer 26 side of the voltage ports 22 and 24. is the duty cycle of the control of the switches of the first current port 20 (0.5 <Dt< 1). et ^3 sont respectivement les déphasages (en radians) des signaux de commandes des interrupteurs de référence T2i et T3[ par rapport au interrupteur de référence Tp Les rapports cycliques des commandes des interrupteurs des ports de tension 22 et 24 sont fixés h = A = 50% sur le mode de fonctionnement, de sorte que les tensions et sont les tensions visibles sur la figure 5. P s désigne la période de commutation ( / 5 est la fréquence de commutation).

[0252] The shapes of the waveforms in Figure 5 are an approximation of the actual shape of the signals. In reality, the current iu is not perfectly trapezoidal, as it is slightly affected by the switching of the voltage ports 22 and 24 at times 6 and for the first voltage port 22 on one side and ^2 and C> for the second voltage port 24 on the other side.

[0253] Figure 6 shows the star-delta delta equivalence of the circuits of the windings of transformer 26 at port 1, with v2 = S2V2 and v3 = SyV3. S2 and 53 respectively are the switching functions of ports 2 and 3. n\ and respectively designate the number of turns of each winding 42, 44 and 46 of the transformer 26.

[0254]

[0255]

[0256] In these equivalent circuits, the current port 20 is replaced by an equivalent voltage source. The expression of the voltage vi on this port at the transformer 26 is detailed later. The voltage at point A- can be obtained by the following relation RI: ^ = £^ + £^ / +£^3' Or: The inductances of the equivalent delta circuit are calculated as follows:

[0257] N / A, ¥i = j 'k

[0258]

[0259]

[0260]

[0261]

[0262]

[0263]

[0264]

[0265]

[0266] 1.2 - Operating principle For a given operating point, the operating cycle of this converter 10 is divided into four successive time intervals. The first time interval gathers the instants f such that 0 < t < t0. At time t = 0, switches T1 and T4 are turned on. Switches T2 and T3 are already turned on (usually before t = 0 due to the previous implementation of the cycle). In this time interval, all switches of current port 20 are turned on, which implies that V] = 0. So, using the RI relation, it comes: vx = Llrv2'+Lc.v3' At the voltage ports, switches T22, T24, T32 and T34 should be in the on state, so that: yh,v <- y — and < 0 This allows the current (li) to increase from -7½ to A in the interval. Neglecting the resistance R\ in series with the leakage inductance at port 1, the current in is expressed according to the following relation R2: 4.1(0 y* LifV^+LcV^ Lt

[0267]

[0268]

[0269]

[0270]

[0271]

[0272]

[0273]

[0274]

[0275]

[0276]

[0277]

[0278]

[0279]

[0280]

[0281]

[0282] Or: • is the value of the input current i[ from current port 20 to the point of chosen operation. This current is considered constant at each operating point since it is limited by the inductance Lf, which is a value much larger than the value of the inductance As in approaches the value / j, the current begins to flow more through diodes Di and D4 and less through diodes D2 and D3. At — ?o, diodes D2 and D3 go into the blocked state. The second time interval gathers the instants 1 such that < L. . During this time interval, the input current L flows through T4, Di and D4 and in~L- The switches T2 and T3 go into the blocked state between A) and 6 at zero current (also designated by the acronym ZCS referring to the English term for "Zero Current Switching" literally meaning "switching at zero current"). The switching of the voltage bridges 22 and 24 is then implemented to ensure that the current in of the transformer 26 switches in the next time interval. At t — t], the H-bridge 38 of voltage port 2 switches and _ y . At t — the H-bridge 40 of voltage port 3 switches and — y^ Ideally, this does not impact the value of the current in since it is imposed by the current source 26 (Lf > L^). In this model, the ideal case is considered. Consequently, the approximation is made that the current in is perfectly trapezoidal and that the resistance Ri is negligible, which leads to the following expression for the AC voltage at port 1: ^1=^ =.....OÏf...... >0 The third time interval gathers the instants z such that <t<t2 At time = , switches T2 and T3 are put into the zero current on state. As a result, all switches at current port 20 are again in the on state and the voltage = 0. The voltage at the star point becomes = L^' + > 0, with r and The current decreases from L to - / j and verifies the relationship Next R3: At time t — diodes Di and D4 are in the off state. The fourth time interval gathers the instants z such that t3 < t < Ts.

[0283] The input current flows entirely through T2, T3, D2 and D3 in this time interval and in = - / {.

[0284] At time t = the switches Ti and T4 are put in the zero current passing state.

[0285] At times t = and t = the voltage ports 2 and 3 are respectively switched and „ ' _ v — and v _ v — •

[0286] The AC voltage at port 1 is written:

[0287] _ V1 - Kv - (1-£J < 0

[0288] This cycle is then repeated for each switching period Ts. 1.3 - Operating conditions

[0289] In order for the power transfer between the ports to be done with smooth switching over the entire operating range for all bridges, while preventing overvoltages at port 1, three main conditions must be met.

[0290] To avoid overvoltages and have smooth switching at port 1, the switches should not be in the off state until the transformer current has completely reversed and the series diodes have blocked. Otherwise, these switches would block a significant portion of the current in the inductive portion, causing a sudden overvoltage at port 1.

[0291] Such a condition can therefore be written as follows:

[0293] Where .-Wi with reference to the relation R2. 0 “

[0294] This results in a relation R4:

[0295] D> J.

[0296] The AC voltages at voltage ports v2' and v3' should be reversed after the current Îli is completely reversed and the associated diodes and switches for port 1 are blocked.

[0297] If L and / or U are reversed before, the current in may not be able to reverse, so no power can be exchanged between the ports. Furthermore, if the voltages U and / or v3 are reversed between the blocking of the diodes and the associated switch, the diodes may switch on again, thus causing overvoltages at port 1. This condition also ensures smooth switching on the voltage ports and the associated conditions can be written:

[0298] f >f »5 —M 2 +^72æ+ 4

[0299] and

[0300] , >t _ i T.

[0301] This results in the two conditions corresponding to the following relations R5 and R6:

[0302] di _ 1}

[0303] and

[0304] ^>„.(0^1)

[0305] It can be deduced from relations R2 and R3 that the slope of the current in during its reversal is proportional to +Le.v^ ),

[0306] Thus, the reversal of this current is guaranteed if both the AC voltages of ports 2 and 3 have the same sign (both negative for a positive slope and positive for a negative slope).

[0307] Furthermore, the slope of the current in will be maximized in this way. Thus, conditions R5 and R6 will always be met simultaneously.

[0308] It thus appears that these conditions depend on the operating point of the converter 10 due to the fact that ^i, ^3, ^t, ^2 and ^3 vary according to the desired power, making the control of this topology complex.

[0309] 1.4 - Generalized mean model of the converter 10

[0310] The equations of state of the system studied are the system corresponding to the following relation R7:

[0311] ^=^1-^.1-^1 ■Æ _ 21 . ç : 01 .ci il z3 dt ~ ~ RL3 + ^13-^ + ^23-^ r £12 _ V » * ^12 dt -V1" ^2^2-^2^^12^12 ^}3~3F -^^3-^3^13-^13 ,^2 Ç UJ? i '23 dt “ «2 ô2-F 2 ' « ^3-^3-

[0312] Where: • f v, for0 <t<- / ’ = T, [ - V] for ~ <t<Ts • 1 for you <t<t5 , et ~ -1 for 0 <t<tiett5<t<Ts • 1 for t^ <t<tf, SMt} = 7 -1 for 0 <t <t2 et t6<t< Ts

[0313] The average model usually used for power electronic circuits takes into account the average values ​​of the state variables to transform this discrete model into a continuous model. This averaging model cannot be implemented for

[0314]

[0315]

[0316]

[0317]

[0318]

[0319] a 10 converter as it results in zero transformer current (since it is an AC type variable). Therefore, the generalized average model of this system is developed to study it as a continuous model while representing the AC signals with better accuracy than the classical average model. Thus, DC signals are represented by average values ​​(coefficient of order 0 of the Fourier series) and AC signals are represented by their fundamentals (coefficient of order 1 of the Fourier series). The k-th coefficient of the Fourier series of a variable x is denoted ( x) , and it is complex and verified the following relation R8: The large signal model of the system can thus be derived from relation 7 to obtain the system corresponding to the following relation R9: x dt Lf ~ Lf Lt • \ $1 / q , 2 »1 y ç K ​​z . x 2 «i, ç v.. x - = -^+7).7:.(53)^((13)^+7:.7:.(53),,.((13),, + ^(0^+7)-^-(53) „-(('23) u ~dT~ = -1^12)^+^12)5,+ --^-^2 / ■ X R I2 / ; x ,2211 T7 dt - ÏR- Ln-\h2l Li2 Ln -V2

[0320] / . X ^23 / . X «1^1 / / Tz i dt - -^5-(^3)^-123^^3)5,+ ^ L23 -^2" «3 L23 '(^3)0 Or: the index R denotes the real part of a complex number, the index I denotes the imaginary part of a complex number, ( O 0=( S,) 0 = 0, < $2) ïr=^C0^D^ + (P2}' ( $2) 5,= -hin^Dp + tp^ (Ss) = ( $3) -isin(D}n + <p3y • 0^,,=1(^4^( / ),^)-^.(^^(01^) + (¼^^ and . (VdC:lK=-^L,,y£+LJY^^

[0321] The large signal model can be represented in matrix form with:

[0322] X = AX + BU

[0323] where: [ (i^ (V3) 0'12)^ (*12) u O13) O13) ]j (^23) iÆ (hi) ' and MriV2f

[0324] The input parameters of the system control are the duty cycle D{ and the phase shifts ^3-

[0325] The output parameters of the system control are (i| ) $ and ( V3) .

[0326] As a result, the average equations obtained are non-linear.

[0327] Linearization must be performed at the operating point to be able to use conventional linear controllers.

[0328] The small-signal model of this system is obtained by introducing small perturbations in the system variables at the operating point and using a Taylor series expansion, so that:

[0329] (x) = xeq + (x)

[0330] Where: • the variables surmounted by the symbol “A” represent the associated small signals (disturbances around the operating point), and • Xeii denotes the value of (x} at the operating point which is sometimes also called the equilibrium point.

[0331] The disturbances of the voltage sources around the mean values ​​can be neglected in this study (y1 _ _ q).

[0332] This is mainly due to their slow variation compared to the fast dynamics of the control (e.g., the voltage of the photovoltaic panel, the battery system or the electrical network). The obtained linearized mathematical model has an order of 8. 1.5 - Reduced order model of the converter 10

[0333] The reduced-order model of a system is a simplified model that can more easily be used and utilized in simulations. In addition, it makes the design of the system's controllers much simpler.

[0334]

[0335]

[0336]

[0337]

[0338]

[0339]

[0340]

[0341]

[0342]

[0343]

[0344]

[0345]

[0346] As a result, it will be possible to recalculate the system controllers in real time when a change in the operating point occurs. However, the main disadvantage of order reduction is lower accuracy for the mathematical model. Reduced-order mean modeling relies on the separation of the system dynamics in the frequency domain into two parts: low-frequency dynamics (slow variables) and high-frequency dynamics (fast variables). After this separation, only the dominant dynamics of the system is taken into account for the study of the system's behavior. For converter 10, the DC variables can be considered as the slow variables and the AC variables as the fast variables. In this case, the slow variables represent the input and output parameters of the system, while the fast variables represent the internal operation of converter 10. Since the purpose of converter 10 control is to regulate the input and output parameters, the dominant low-frequency dynamics are retained and the fast dynamics are ignored in the reduced-order model. The two subsystems can thus be represented as follows, by separating the state vector X of the unreduced system into two parts: Or : • the index “s” designates the slow dynamic subsystem, • the index “f” designates the fast dynamic subsystem, Xf = [ {1,2)<'12)('b)^ <>'b}„ feL]' Thus, the system of relation R9 becomes: Xs = AssX, + AsfXf + Bs.U X = AX + BU^ . Xf = AfsXs + AffX f + Bf.U Or: • The matrices Ass, Asf, Afs and Aff are parts of the matrix A designated before, Ass being the gain matrix between the variables Xs and Xs Asf the gain matrix between the variables Xs and Xf, Afs the gain matrix between the variables Xf and Xs and Aff the gain matrix between the variables Xf and Xfj and • Bs and Bf are parts of the matrix B linking Xs and Xf to the input matrix U. The matrices Ass, Asf, Aj^ Af^Bs and Bf are obtained using a rearrangement of the matrices A and B.

[0347]

[0348]

[0349]

[0350]

[0351]

[0352]

[0353]

[0354]

[0355]

[0356]

[0357] To obtain the reduced-order model, the fast dynamic subsystem is first solved at a chosen operating point (equilibrium point) (X= 0), considering that the slow variables are constant and equal to their average values, that is to say that: Xs = Xs^ so that (^)= / and (V3) = The average response Xfeq of the fast dynamic subsystem is thus obtained. Then, for the slow dynamic subsystem, the fast variables are replaced by the average response calculated previously. This gives Xf — X The slow dynamic subsystem is then linearized around the chosen operating point while ignoring the dynamics of the fast variables. This gives a reduced-order linearized model of the converter 10, whose expression is given by an RIO relation corresponding to the following two-equation system: Vi / 0 / R, dt - \ ' Lf ' / ^\n,(nor ^3^ • \11 / Q + Lf< !-£„)x^teq 1 + w ZLh nl ■A 2LC n. 4 «1 ~dt~ = - + + " / tC - -n3 • [ 3J?,^Sill ( \£q^ 4- ) + / 1 SJ^-COS ( D + (Pj.eq ) ++ (p3^ ) + + / 23j^cos ( D ieqjr + j ,^3 It thus appears that the order of the mathematical model has been reduced from 8 to 2 using the reduced-order modeling technique. This makes it simpler to analyze the dynamic behavior and design the control of the converter 10. The application of the Laplace transform to the RIO relation model leads to a reduced transfer function linking on the one hand the input current I] of port 1 to the duty cycle by a first transfer function Gnr and on the other hand the output DC voltage of port 3 to the phase shift ^3 by a second transfer function The first transfer function G^r is expressed according to the following relation RI 1: . _ as+b ^2~ — (s+cXs+4) Or:

[0358]

[0359]

[0360]

[0361]

[0362]

[0363]

[0364]

[0365]

[0366]

[0367]

[0368] • _ 2Ll> V 2G ni T / , a - -¾ ■V 2 + Lf(l-La) -¾•V 3,eq • = ' CVJ -(^1 / - / - 1 1 Ip* / 1 / / , / -3111 ) + + 723j / / ) / SÎn( Di / 7 / r+ j ^COSf D), And • ... £4i C - L, + L^-L^T, This expression of the first transfer function Gnr can be further simplified by ignoring the dynamics of the voltage V3, i.e. by considering that this value C3 does not vary much around its nominal value. This gives a first reduced transfer function of order 1 corresponding to the following relation R12: G^s _ has -A (S+C) The second transfer function Gv$r is expressed according to the following relation R13: / ' / _ ^'9 I „ _ e \ $ 7 — (p^S) I — (s+d) Or: • / 7--i—, and a “ c^L3 ¢ = + / .1^^^8( / ^1 / ^ + ^-,^) + + / .2iy^-COS(..D]^?j^ + ) The order of the first and second reduced transfer functions Gvt and Gn is independent of the number of ports of the converter 10. Thus, the reduced-order model just described for a 10 hybrid-powered converter can be generalized to an n-port 10 MAB converter, 1 port being a current-powered port while the (n-1) others are voltage-powered ports. These first and second reduced transfer functions are then still of the first order regardless of the number of ports of the 10 MAB converter, so that the voltages of the voltage supply ports do not vary much around the average values ​​at a certain operating point. These reduced order models can be used for converter analysis 10 and for designing closed-loop controllers. 1.6 - Control strategy There are two parameters to be controlled in this system (the output control parameters), namely the input DC current of port 1 and the output DC voltage of port 3.

[0369] However, the input control parameters are three in number and are the duty cycle D and the phase shifts ^2 and ^3.

[0370] Thus, at a chosen operating point, there is an infinite combination of values ​​for the input control parameters which can give the desired values ​​while respecting the relationships R4 to R6. The elements of such a combination will be named and for the input control parameters and and for the output control parameters.

[0371] Furthermore, from the system R9, it can be noted that the output control parameters I\eq and V^q depend on all the input control parameters and This shows that the converter structure 10 studied is coupled and that a change in one of the control variables has an effect on all ports.

[0372] One way to control a converter 10 is to make Dj,eq equal to a minimum allowed value D\min in the relation R4 with a safety margin £ at each operating point. It comes thus:

[0373] D^ = D]jni„+£

[0374] The value of £ can be chosen arbitrarily, for example set to 0.01.

[0375] The corresponding values ​​of ^2^ and ^.eq are then calculated from the system of relation R9 at the chosen operating point.

[0376] Then, using the relations R5 and R6, a minimum allowed value for ^2 and ^3 is obtained.

[0377] The calculated values ​​of the three input control parameters are then sent to the system.

[0378] Adding a feedback loop to the converter control 10 allows the ports to be decoupled.

[0379] The approximations made to develop the mathematical model will result in an error in the steady state if only open loop control is applied.

[0380] A PI controller is therefore used for each of the control loops to remove this error (see Figures 3 and 4). These controllers are calculated based on the reduced order model presented previously. Adjustment subunits 70 and 90 are added to ensure that conditions R4 to R6 are always satisfied.

[0381] In the case of figures 3 and 4, the maximum authorized for the value of is noted ^Ijnax and is equal to 1.

[0382] Similarly, the maximum allowed for the values ​​of ^2 and ^3 is noted ^max and is calculated as follows:

[0383] ls < Tl and t6 < T, - = .r, ( | - D, )

[0384] The transfer function of the conversion subunit 66 is expressed as follows: F03851 x MW

[0386] Using the first transfer function Gu of relation R12, a first-order transfer function is obtained.

[0387] It is chosen y — 1.

[0388] The closed-loop transfer function CLTF^ thus becomes:

[0389] crrr ra 1 ÇLlF^S) s+Kp]ar^+l

[0390] Where: • — —1— is the time constant of the closed-loop system.

[0391] Now, the response time of this closed-loop transfer function to obtain 95% of the reference current is such that tr । 95% = 3.T (.

[0392] The gain Kp\ is then chosen for the desired value of 6-1.95%, so that this gain verifies:

[0393] K ,__1_ "Ai.95%

[0394] As for the transfer function of the conversion subunit 86, it is expressed as follows:

[0395]

[0396] Using the second transfer function Gr3r of relation R13, a first-order transfer function is obtained.

[0397] It is chosen t — 1. / 3 d

[0398] The closed-loop transfer function CLT F?, thus becomes:

[0399] rTTr ( . _ _ 1

[0400] Where: • is the time constant of the closed-loop system.

[0401] Now, the response time of this closed-loop transfer function to obtain 95% of the reference voltage is such that / ,-3 95% — 3.T^.

[0402] The gain Kp\ is then chosen for the desired value of 6-3.95%, so that this gain verifies:

[0403] is ,--2__ ~ edri.^ 1.6 - Experimental results

[0404] Figure 7 shows the results of simulations of a closed-loop control of current 71.

[0405] The values ​​of the parameters of the simulated converter 10 are given in Table 1 below.

[0406] [Tables 1] Parameters Value in simulation V] 200 V Vi 400 V fs 20 KHz Lf 0.016 H Rf 10 mQ 83 pH R} 10 mQ L2 83 pH r2 10 mQ l3 230 pH R3 10 mQ c3 100 pH Re3 1 mQ Lm 8.3mH «1 100 tums «2 83 tums «3 124 tums P, T x ^nominal 3 kW (received) P * ïnominal 3.5 kW (delivered) Pmax (between two ports) 4 kW RL3 120 Q

[0407] The response time was chosen so that the values ​​of the parameters of the PII controller of the R14 relations are such that = 3ms.

[0408] Observation of [Fig.7] shows that the response time of the open loop is indeed equal to the chosen value of 3 ms, which validates the development of the mathematical model of the controller.

[0409] Figure 8 shows the results of the same simulation for closed-loop control of the output voltage V3.

[0410] The response time was chosen so that the values ​​of the parameters of the PI3 controller of the RI5 relations are such that ^3 95% = 10ms.

[0411] Observation of [Fig.8] shows that the response time of the open loop is indeed equal to the chosen value of 10 ms, which validates the development of the mathematical model of the controller.

[0412] Figures 9 and 10 show the evolution of each of the voltage values ​​ri, F3 and ILi respectively obtained by simulation and theoretically.

[0413] Comparison of the two figures clearly shows that the signals obtained by simulation are consistent with the theoretical forms.

[0414] The minimal difference between the characteristics chosen for the calculation of the controller parameters and the characteristics of the simulated responses is due to simplification assumptions that were made, in particular the first harmonic approximation of the AC signals and the order reduction of the model.

[0415] To improve the accuracy of the model, higher order harmonics could be considered. However, this leads to model complexity.

[0416] The model described makes it possible to obtain the best compromise between precision and complexity. 2 - OTHER CASES

[0417] In the structure of the converter 10, the coupling between the ports 20, 22 and 24 is generated by the presence of the inductances of the isolation interfaces 48, 50 and 52.

[0418] However, the absence of an inductance on one of the ports is a special case that can be regulated with the same regulation strategy.

[0419] In such a case, the conversion functions without order reduction would be written:

[0420] / \ m I i

[0421] and:

[0422] / \ nC.P+t, bWSJ 1 r J —...................................•.....y........................................v............................................... rV; IV, l^RA - TR' SI / “«2 I • 2 ,, / Y • » 2 ■ l Y and 0 = • ^23 being the phase shift between voltages V2 and V3.

[0423] It may be noted here that in this case, only the voltage port that does not have an inductance (called the master port) is subject to the control constraints imposed by the current port. Therefore, the external phase shift of this port has minimum and maximum permissible values. The external phase shifts of the other voltage ports may have any calculated value to obtain a desired value of their DC output voltage.

[0424] Additionally, the sources / loads connected to the ports can be changed. For example, load 39 can be replaced with a voltage source and port 3 can be made bidirectional.

[0425] Similarly, a load 39 can replace the voltage source of port 2.

[0426] In each of these cases, at least one voltage port connected to a stable voltage source, the average value of which does not vary too much over time.

[0427] An electrical network or a battery are examples of such a stable voltage source.

[0428] Finally, the mathematical model and control strategy proposed in this patent can be generalized to a MAB converter consisting of a total of n ports, n being any integer and one port being powered by current.

[0429] In such a case, the multi-active bridge converter 10 comprises the input current port 20 and at least one output voltage port.

[0430] Each output voltage port is associated with a control unit conforming to the control unit described with reference to [Fig.4], so that the regulation system 28 comprises (n-1) control unit(s).

[0431] In each case, the proposed converter can in particular be used in domestic applications or in electric vehicles.

Claims

33

1. Claims A regulation system (28) for a hybrid-powered multi-active bridge converter (10), the multi-active bridge converter (10) comprising: - an input current port (20) comprising an H-switch bridge (30) for which a reference switch (TJ) is defined, the reference switch (TJ) being called the first reference switch (TJ) and being controlled by a first control law (LCi), the first control law (LCi) having a first duty cycle (Dj), - at least one output voltage port (22, 24) comprising an H-switch bridge (38, 40) for which a reference switch (T2b T31) is defined, the reference switch being called a second reference switch (T2b T3i) and being controlled by a respective second control law (LC2, LC3), each second control law (LC2, LC3) having a second predefined duty cycle (D2, Dÿ and a phase shift (¾ ^3) with respect to the first control law (LCi), - a transformer (26) having windings (42, 44, 46), each winding (42, 44, 46) being connected to a port (20, 22, 24) by a respective isolation interface (48, 50, 52), the regulation system (28) being capable of regulating the converter (10) with multiple active bridges to a setpoint, the setpoint comprising a setpoint current value (Aref) for the current of the current port (20) and at least one setpoint voltage value (Vyref) for said at least one voltage port (22, 24), the regulation system (28) comprising: - a unit for measuring measured values, the measured values ​​comprising the current ( / Q) of the current port (20) and the voltage (V2, V3) of each voltage port (22, 24), - a first control unit (56) for controlling the first reference switch (Ti), the first control unit (56) comprising: - a first sub-unit (62) for calculating a first desired initial value ( / ^uq), the first desired initial value (i^Uq) being a desired value for the first duty cycle equal to the sum of the result of a first calculation function (FC1) applied to the setpoint values ​​V1ref) and a safety margin (f), - a first sub-unit (64) for determining the difference between the measured current value (I0) and the set current value (^uef), to obtain a determined current difference (A / j), - a first correction sub-unit (66) capable of converting the determined current deviation^ / ]) into a first corrective value (ADj for the first duty cycle (D^ by applying a first conversion function (G / Q, - a first addition subunit (68) of the first desired initial value (^uq) and the first corrective value (ADj), to obtain a first candidate value (^u), - a first adjustment sub-unit (70) capable of adjusting the first candidate value (^k) to obtain a first value to be applied (¾) between two first extreme values ​​(■^tmin, ^imax), the lowest first extreme value (^unin) being the result of the first calculation function (FC1) applied to the measured values ​​(I\, V2, V3), - a first application sub-unit (72) of the first control law having as its duty cycle the first value to be applied (^u), and - for each voltage port (22, 24) having a set voltage value (V31.ef), a second control unit (58, 60) of the second reference switch (T2i, T3i) of the voltage port (22, 24) considered, each second control unit (58, 60) comprising: - a second calculation sub-unit (82) of a second desired initial value (^3,eq), the second desired initial value (^3,eq) being a desired value for the phase shift (¾ ^3) equal to the result of a second calculation function (FC2) applied to the first desired initial value (Di,eq), - a second sub-unit (84) for determining the difference between the measured voltage value (V2, V3) for the voltage port (22, 24) considered and the set voltage value (V3 ref) for the voltage port (22, 24) considered, to obtain a determined voltage difference (AV3), - a second correction sub-unit (86) capable of converting the determined voltage difference (AV3) into a second corrective value (Aç> ) for the phase shift (^2, ^3) by applying a second conversion function (G^), - a second sub-unit for adding the second desired initial value (^q) and the second corrective value to obtain a second candidate value (^3.0), - a second adjustment sub-unit (90) capable of adjusting the second candidate value (^3,c) to obtain a second value to be applied (^3,app) between two second extreme values ​​(^3,min, ^3jnax), the lowest second extreme value (^3.min) being the result of the second calculation function (FC2) applied to the first desired initial value (^i^q), and - a second application sub-unit (92) of the second control law having as phase shift (^2, ^3) the second value to be applied (¾a).

2. Regulation system according to claim 1, in which each adjustment sub-unit (70, 90) is capable of giving as the value to be applied CDupp, ^tapp) the candidate value (^u-, ^3,c) unchanged when the candidate value peak, ^3,c) is between the two extreme values ​​(^imin, ^tmax, ^3,min, ^3max) and otherwise the extreme value (^.min, ^.max) closest to the candidate value Pic, ^.c).

3. A control system according to claim 1 or 2, wherein the second highest extreme value (^3,max) is the result of a third calculation function (FC3) applied to the first desired initial value (^i^q).

4. A control system according to claim 3, wherein at least one calculation function among the second calculation function (FC2) and the third calculation function (FC3) is an affine function.

5. A control system according to any one of claims 1 to 4, wherein each correction sub-unit (66, 86) is a proportional-integral corrector.

6. A control system according to any one of claims 1 to 5, wherein each conversion function (Giif, G^ is a first-order function.

7. A control system according to any one of claims 1 to 6, wherein each H-bridge (32, 38, 40) comprises two midpoints (32M1, 32M2, 38M1, 38M2, 40M1, 40M2), each insulation interface (48, 50, 52) comprising two lines connecting a respective midpoint (32M1, 32M2, 38M1, 38M2, 40M1, 40M2) with an end (42E1, 42E2, 44E1, 44E2, 46E1, 46E2) of the associated winding (42, 44, 46), one of the two lines comprising a resistor (Rb R2, R3) in series with an inductance (Lb L2, L3).

8. Regulation system according to claim 7, in which each winding (42, 44, 46) comprises turns, the first conversion function (G^) having a gain depending on the measured values ​​for the voltages (V2, V3\ of the second desired initial value (^3,eq), the number of turns of each winding (42, 44, 46) and the inductances (Lb L2, L3) of the insulation interfaces (48, 50, 52).

9. A regulation system according to claim 7 or 8, wherein each winding (42, 44, 46) comprises turns, the second conversion function (Gv^r) having a gain depending on the measured current value ( / }), the first desired initial value (^i^q), the second desired initial value (^3,eq), the number of turns of each winding (42, 44, 46) and the inductances (Lb L2, L3) of the insulation interfaces (48, 50, 52).

10. Regulation system according to any one of claims 7 to 9, in which each winding (42, 44, 46) comprises turns, the first calculation function (FC1) further depending on the number of turns of each winding (42, 44, 46) and the inductances (Lb L2, L3) of the insulation interfaces (48, 50, 52).

11. A control system according to any one of claims 1 to 10, wherein at least one output voltage port (22) is a bidirectional port.

12. A control system according to any one of claims 1 to 11, wherein each determination subunit (64, 84) is a subtractor.

13. A control system according to any one of claims 1 to 12, wherein the safety margin (f) is a predefined value.

14. A hybrid-powered multi-active bridge converter (10), the multi-active bridge converter (10) comprising: - an input current port (20) comprising an H-switch bridge (30) for which a reference switch (TJ is defined and operating at a current, the reference switch (Ti) being called the first reference switch (Ti) and being controlled by a first control law (LCi), the first control law (LCi) having a first duty cycle (£>1), - at least one output voltage port (22, 24), each voltage port (22, 24) comprising an H-switch bridge (38, 40) for which a reference switch (T2b T31) is defined, each reference switch being called the second reference switch (T21, T31) and being controlled by a respective second control law (LC2, LC3), each second control law (LC2, LC3) having a second predefined duty cycle (^2, ^3) and a phase shift (^2, ^3) with respect to the first control law (LCi), - a transformer (26) having windings (48, 50, 52),each winding (48, 50, 52) being connected to a port (20, 22, 24) by a respective isolation interface (48, 50, 52), and - a regulation system (28) according to any one of claims 1 to 13.,

15. A method for regulating a hybrid-powered multi-active bridge converter (10), the multi-active bridge converter (10) comprising: - an input current port (20) comprising an H-switch bridge (30) for which a reference switch (TJ) is defined, the reference switch (TJ being called a first reference switch (Ti) and being controlled by a first control law (LCi), the first control law (LCi) having a first duty cycle (U0, - at least one output voltage port (22, 24) comprising an H-switch bridge (38, 40) for which a reference switch (T21, T31) is defined, the reference switch being called a second reference switch (T2b T3i) and being controlled by a respective second control law (LC2, LC3), each second control law (LC2, LC3) having a second ... predefined duty cycle (D2, and a phase shift (^2, ^3) with respect to the first control law (LCJ, - a transformer (26) having windings (42, 44, 46), each winding (42, 44, 46) being connected to a port (20, 22, 24) by a respective isolation interface (48, 50, 52), the method being suitable for regulating the converter (10) with multiple active bridges to a setpoint, the setpoint comprising a setpoint current value (Z^ef) for the current of the current port (20) and at least one setpoint voltage value (V^æf) for the at least one output voltage port (22, 24), the method comprising the steps of: - measuring measured values, the measured values ​​comprising the current (Zj) of the current port (20) and the voltage (V 2, V3) of each voltage port (22, 24), - control of the first reference switch (Ti), the control step comprising: - a first calculation of a first desired initial value (Z)^), the first desired initial value (D|,eq) being a desired value for the first duty cycle (Due) equal to the sum of the result of a first calculation function (FC1) applied to the setpoint values ​​(Zi.ref, V^f and a safety margin (£), - an initial determination of the difference between the measured current value (Zi) and the set current value (Z 1.^), to obtain a determined current difference (AZ3), - a first correction converting the determined current deviation (AZ|) into a first corrective value (ADj) for the first duty cycle (ZZt) by applying a first conversion function (Gnr), - a first addition of the first desired initial value (ZZj^q) and the first corrective value (AD]), to obtain a first candidate value (D!c), - a first adjustment suitable for adjusting the first candidate value (Dlc) to obtain a first value to be applied (ZZi,app) between two first extreme values ​​(DJjnjn, Dlm.;xy the lowest first extreme value (D]^) being the result of the first calculation function (FC1) applied to the measured values ​​(Zb V2, V3), - a first application (72) of the first control law having as duty cycle the first value to be applied (A,a), and - for each voltage port (22, 24) having a set voltage value (V^f), control of the second reference switch (T2b T3i) of the voltage port (22, 24) considered, each step of control of the second switch (T2b T3i) comprising: - a second calculation of a second desired initial value (^3,eq), the second desired initial value (^3,eq) being a desired value for the phase shift (^2, ^3) equal to the result of a second calculation function (FC2) applied to the first desired initial value (^teq), - a second determination of the difference between the measured voltage value (V2, V3) for the voltage port (22, 24) considered and the set voltage value (V3ref) for the voltage port (22, 24) considered, to obtain a determined voltage difference (AV3), - a second correction converting the determined voltage difference (AV3) into a second corrective value (A#> ) for the phase shift (^2, ^3) by applying a second conversion function (Gv3r), - a second addition of the second desired initial value (^3.eq) and the second corrective value (A^3), to obtain a second candidate value (^3,c), - a second adjustment suitable for adjusting the second candidate value (^ac) to obtain a second value to be applied (¾^ between two second extreme values ​​(^3.min, ^3,max), the second lowest extreme value (^3,min) being the result of the second calculation function (FC2) applied to the first desired initial value (Dteq), and - a second application of the second control law having as phase shift (^2, ^3) the second value to be applied (^a).

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