Linear frequency modulation pulsed radar

Abstract

A radar signal generation device comprising: a generator (110) of periodically repeated series of trigger pulses (sPRFs) having a linearly variable time spacing; and an oscillator (120) configured to receive said trigger pulses as input and to generate a train of periodic oscillations corresponding to each said trigger pulse; characterized in that said oscillator (120) has a variable frequency depending on a control signal (Vtune); and in that the device also comprises a generator (112) of said control signal, adapted to linearly vary the frequency of the oscillator (120) from one train of periodic oscillations to another.Radar device comprising a transmit channel (1) including such a device for generating a radar signal; and a receive channel (2) configured to receive echoes of said radar signal and to demodulate them synchronously with their generation so as to extract time-of-flight information. Figure for the abbreviation: Fig. 4.

Description

Title of the invention: Linear frequency modulation pulsed radar

[0001] The invention relates to the field of radar, and more particularly to radar-on-a-chip, which can be used, in particular, for measuring a patient's vital signs or as a presence detector, see for example (Antide 2020). These radars must exhibit both high spatial resolution (on the order of a few centimeters) and low power consumption (a few tens of mW).

[0002] A commonly used technique for these applications is that of frequency-modulated continuous-wave, duty-cycled (FMCW-DC), see for example (Liu 2019) and (Siligaris 2023). Indeed, this technique exploits the principle of compressing a wave emitted with a wide bandwidth to a narrow bandwidth in the intermediate frequency, which makes it possible to use analog-to-digital converters (ADCs) with a relatively low acquisition rate, of a few MSps or tens of MSps (IMSps = 10⁶ samples per second - acronym for "Mega Samples per second").However, when dealing with targets exhibiting highly diverse radar cross sections (RCS), it will be necessary to use ADCs with high dynamic range, for example 9 to 12 bits or more, and therefore relatively high power consumption.

[0003] An alternative is to use the ultra-wideband radio pulse (IR-UWB) technique, in which a short pulse is emitted and the time of flight of its echo is measured to determine the distance to the target. See, for example, (Andersen 2017). One advantage of this technique is that, since the echo signals are separated in time, it is possible to apply automatic gain control (AGC) to enable the acquisition of a highly contrasting environment in the presence of very diverse RCS objects while limiting the dynamic range of the ADCs. On the other hand, obtaining good spatial resolution relies on the use of a high acquisition rate, with several GSpS (IGSps = 10⁹ samples per second).

[0004] The invention aims to overcome, in whole or in part, the aforementioned drawbacks of the prior art. More particularly, it aims to enable the use of ADCs with lower dynamic range than FMCW-DC radars and a lower acquisition rate than IR-UWB radars, without sacrificing either the spatial resolution or the RCS dynamic range of detectable targets.

[0005] According to the invention, this objective is achieved by generating a radar signal consisting of periodically repeated series of periodic oscillation trains. The oscillation trains within the same series have a linearly variable time spacing, while within a series, the oscillation frequency varies linearly from one periodic oscillation train to the next. Since the radar signal is pulsed (it is indeed composed of periodic oscillation trains spaced apart), the echo signals are separated in time, as in the case of the IR-UWB technique, allowing for gain adjustment that permits the use of relatively low dynamic range ADCs even in the presence of a high-contrast environment.Furthermore, the combination of variable spacing of oscillation trains and a modification of the oscillation frequency from one train to another allows - as will be demonstrated later - a compression of the echo bandwidth, permitting the use of ADCs with a relatively low acquisition rate without sacrificing spatial resolution, as in the case of the FMCW-DC technique.

[0006] Also, an object of the invention is a radar signal generation device comprising: - a generator of periodically repeated series of trigger pulses, the pulses of the same series having a linearly variable time spacing; and - an oscillator configured to receive said trigger pulses as input and to generate a train of periodic oscillations corresponding to each said trigger pulse; characterized in that - said oscillator exhibits a variable frequency depending on a control signal; and in that - the device also includes a generator of said control signal, adapted to vary linearly in time the frequency of the oscillator from one periodic oscillation train to another for periodic oscillation trains triggered by trigger pulses belonging to the same series.

[0007] According to particular embodiments of such a device:

[0008] - The oscillator and the generator of said control signal can be configured to that the average frequency of the oscillations of said periodic oscillation trains is a multiple, by a factor K>1, of an average repetition frequency of the trigger pulses.

[0009] - More particularly, said trigger pulse series generator includes a first signal generator and a trigger pulse generator, the first signal generator being configured to generate a first signal consisting of periodic repetitions of modulated oscillation trains linearly in frequency, the trigger signal generator being configured to receive said first signal as input and to generate said trigger pulses synchronously with the oscillations of said first signal.

[0010] - More particularly, the oscillator and the generator of said control signal can be configured so that the frequency of the periodic oscillation train generated in response to the nth trigger pulse of said series is given by: ft - v / f j- ttn \ i °ù fpRFO is the frequency of the oscillations of the first U UU - JM rPRF0 * Tcflirp [s V njj signal at the beginning of each frequency-modulated oscillation train, bprf the TChirp rate frequency modulation of said oscillations and ts ( n ) an instant of generation of said nth trigger pulse, with ts(n — 1) ~ 0.

[0011] - Said first signal generator can be configured so that the modulation in frequency of oscillations of said first signal is controlled by said control signal.

[0012] - The K factor can advantageously be greater than or equal to 10 and preferably between 100 and 1000.

[0013] - The device may also include a pulse shaping device for modify the time profile of said periodic oscillation trains.

[0014] - Said oscillator may be a microwave oscillator.

[0015] - The device may also include a device for applying jamming of phase to said periodic oscillation trains.

[0016] Another object of the invention is a radar device comprising: - a transmission channel comprising a device for generating a radar signal as described above; and - a receiving channel configured to receive echoes of said radar signal and to demodulate them synchronously with their generation in order to extract time-of-flight information.

[0017] According to particular embodiments of such a radar device:

[0018] - The receiving channel may include an amplifier equipped with a device automatic gain adjustment to compensate for variations in the intensity of received echoes.

[0019] - The device for generating a radar signal from the transmission channel may include a first signal generator configured to generate the first signal consisting of periodic repetitions of frequency-linearly modulated oscillation trains; and the receiving channel may include: a frequency multiplier configured to receive the first signal as input and multiply its frequency by the factor K; and a mixer configured to receive the radar echoes as input and demodulate them by mixing with a signal from said frequency multiplier in order to obtain a demodulated signal; a low-pass filter configured to attenuate components of said demodulated signal at frequencies that are multiples of said average repetition frequency of the trigger pulses; and an analog-to-digital converter having an acquisition rate suitable for converting a signal at said average repetition frequency.

[0020] - The receiving channel may include at least one analog converter - digital having an analog bandwidth of width at least equal to the spectral width of said received echoes but an acquisition rate adapted to convert a signal at said average repetition frequency of the trigger pulses, said converter being synchronized with said trigger pulses through a delay line.

[0021] - The device for generating a radar signal from the transmission channel may include a said first signal generator configured to generate said first signal consisting of periodic repetitions of frequency-linearly modulated oscillation trains; and the receiving channel may include: a frequency multiplier configured to receive said first signal as input and multiply its frequency by said factor K; and a mixer configured to receive said radar echoes as input and demodulate them by mixing them with a signal from said frequency multiplier to obtain a demodulated signal; and at least one analog-to-digital converter having an analog bandwidth of a width at least equal to the spectral width of said received echoes but an acquisition rate suitable for converting a signal at said average repetition frequency of the trigger pulses, said converter being synchronized with said trigger pulses through a delay line.

[0022] - More specifically, said delay line may exhibit a variable delay.

[0023] - The receiving channel may comprise a plurality of said converters analog-digital synchronized with said trigger pulses through respective delay lines introducing different delays.

[0024] The device for generating a radar signal of the transmitting channel may include a device for applying phase scrambling and the receiving channel may include a phase scrambling device.

[0025] Yet another object of the invention is a method for generating a radar signal comprising the generation of periodically repeated series of periodic oscillation trains, the oscillation trains of the same series having a linearly variable time spacing; the frequency of the oscillations varying linearly from one periodic oscillation train to another in the same series.

[0026] Other features, details and advantages of the invention will become apparent from the description given with reference to the accompanying drawings provided by way of example, which represent, respectively:

[0027] [Fig.1], the functional diagram of a radar generation device according to an embodiment of the invention;

[0028] [Fig.2], the graph of a spectral template that the generated radar signals must respect;

[0029] [Fig.3], the spectrogram of the radar signal generated by the device of the [Fig.1];

[0030] [Fig.4], the functional diagram of a radar device according to an embodiment of the invention;

[0031] [Fig. 5A], the signal at the output of the mixer of the receiving channel of the device [Fig.4], represented in the time domain, in the case of a single target located at a distance of 3.75 m (time of flight: 25 ns);

[0032] [Fig.5B], an enlargement of a portion of [Fig.5A];

[0033] [Fig.6A], the signal at the output of the mixer of the receiving channel of the device [Fig.4], represented in the spectral domain, also in the case of a single target located at a distance of 3.75 m (time of flight: 25 ns);

[0034] [Fig.6B], an enlargement of a portion of [Fig.6A];

[0035] [Fig.7A], [Fig.7B] and [Fig.7C], respectively, the output signals of the channel reception of the mixer of the device of [Fig.4], of the low-pass filter and of a Fourier transform calculation module, in the case of three targets located at 1m, 5m and 12m;

[0036] [Fig.8], the functional diagram of a radar device according to another mode of realization of the invention;

[0037] [Fig.9A], the signal downsampled by the receiving channel of the device [Fig.8], also in the case of three targets located at 1m, 5m and 12m;

[0038] [Fig.9B], an enlargement of a portion of [Fig.9A];

[0039] [Fig. 10], the functional diagram of a radar device according to yet another embodiment of the invention; and

[0040] [Fig. 11 A], the signal downsampled by the receiving channel of the device of the [Fig. 10] in the case of three targets located at distances of 1 m, 5 m and 12 m, respectively;

[0041] [Fig. 1 IB], an enlargement of a portion of [Fig. 1 IA]; and

[0042] [Fig.llC], the signal obtained by Fourier transform of that of [Fig.llA].

[0043] The radar signal generation device of [Fig. 1] comprises a first generator

[112] of a first signal sc(t) in the form of a periodic sequence of linearly frequency modulated oscillation trains (“chirp”, from the English “chirp”, chirping):

[0044] sc = sin ( 2,t ( fpRpf+ V te [ft TMr^( 1 '

[0045] where tpRFo is the oscillation frequency of the first signal at the beginning of each frequency-modulated oscillation train, ^rf the frequency modulation rate Tchirp Let t be the time and Tchirp the repetition period of the oscillation trains. It is understood that a train starts with oscillations at the frequency fpRFo, a frequency that gradually increases up to fpRFo + ®prf; ®prf is therefore the bandwidth of the sprf signal. Alternatively, the frequency can gradually decrease down to fpRFo'BpRF-. In both cases, the oscillation frequency returns to a value of fpRFo after the time Tchiip, or Tchiip + Tguard if a guard interval Tguard is required between two pulse trains.

[0046] The frequency fpRFo is generally on the order of a few MHz, for example between 1 and 100 MHz. In the following, we will take f PRF 0 = 19.4 MHz, BPRF = 1.2 MHz (which gives an average pulse repetition frequency of 20 MHz) and Tchirp = 40 ps.

[0047] The first signal generator 112 may, for example, include a sawtooth waveform generator (voltage ramps) clocked by a clock 111 and driving a voltage-controlled oscillator (not shown).

[0048] The oscillatory signal sc is used by a trigger pulse generator 113 to generate periodically repeated series of trigger pulses synchronously with the oscillations of said first signal. For example, a trigger pulse can be generated corresponding to each oscillation, or at each zero crossing of the first signal (thus two trigger pulses per oscillation), etc. In what follows, we consider a case where a trigger pulse is generated at the beginning of each oscillation of the first signal, which can be achieved using a duty cycle controller. It is easily found that the series of pulses corresponding to each repetition of the oscillation train of the first signal is given by

[0049]

[0050] I । (2) e PA Tchirp] PRFnuil -tchirp z L f

[0051] Where the time positions ts(n) of each pulse are given by

[0052] . / s ~ ir 1 1 (3) = — -1 + J1+-—27— (n-1) ...,nnwx\ i!U \ V /

[0053] Nmax being the number of pulses in each series.

[0054] We note that the spacing A ts(n ) - ts ( n + 1 ) - ts (n) between two impulses varies linearly (at least to a first approximation) in time.

[0055] In [Fig. 1], block 110 designates the generator of periodically repeated series of variable-spaced sPRF trigger pulses. This block comprises the clock 111, the first signal generator 112—which, in the example considered here, consists of a voltage ramp generator and a frequency-controlled oscillator—and the pulse generator 113. Other architectures for implementing this functional block can be considered by those skilled in the art.

[0056] The sPRF trigger pulses activate an oscillator 120 for a brief duration—on the order of magnitude of the pulse duration—leading to the generation of a train of periodic oscillations at an oscillator frequency fc much higher than that of the oscillations of the first signal. Typically, if the oscillation frequency of the first signal is on the order of a few tens of MHz, fc will be on the order of a few GHz. The oscillator 120 therefore generates microwave pulses. "Microwaves" refers to frequencies between 300 MHz and 300 GHz, but the oscillator 120 will preferably operate in the 1 GHz–100 GHz range.

[0057] The waveform generated by the oscillator 120, activated / deactivated by a trigger pulse, may exhibit a spectrum incompatible with current regulations. For this reason, the oscillation trains (which, hereafter, will also be called "microwave pulses") can be shaped by a wave shaper 130. Several microwave pulse envelope shapes are achievable, in order to comply with regulations and spectral masks. In the following, we will consider the case of Gaussian pulses whose envelope is given by [°058] A(n=e-^

[0059] Where F is a dimensionless parameter that determines its position relative to the corresponding trigger pulse (for example, we can take y = 3) and another parameter, having the dimensions of time, determines the duration of the microwave pulse, and therefore its spectral width. If the target bandwidth of the microwave pulse is Bpuke, we obtain

[0060] (5) .aaaaaaaaaaaaaaaaaaaaaaaaaaaaaa

[0061] Where xdB is the reference level (in dB) used for the measurement of the Bpuise band (generally, xdB is chosen to be equal to -3dB or -10 dB).

[0062] The microwave oscillator 120 is a frequency-controlled generator driven by a control signal VtUne such that the frequency fc varies linearly in time over a Tchirp period, that is to say for microwave pulses corresponding to trigger pulses of the same series.

[0063] Advantageously, this control signal is generated by the first generator of signals 112 are used to ensure synchronization between the oscillation frequency variations of oscillator 120 and the generation of trigger pulses. In particular, the Vtune control signal can be a replica of the voltage ramps used in generating the first signal. Alternatively, a separate control signal generator can be used.

[0064] Optionally, a phase scrambler 150 can cooperate with the pulse shaper to apply phase scrambling to the microwave pulses, thereby smoothing their spectrum. A power amplifier 140 can also amplify the signal prior to transmission.

[0065] The signal from the oscillator 120 can be written - without a multiplicative coefficient defining its amplitude, and without considering any possible phase interference:

[0066] Stx ( 0 = E^e-■sin (27Tfc ( n ) ■ ( t - ts(n) ) ) (6)

[0067] Where the dependence of the oscillator frequency fc and the number of pulses, and therefore of time, has been highlighted.

[0068] The oscillator 120 and its control signal Vtune are such that the frequency fc varies linearly with time from one microwave pulse to the next, while remaining constant during a single pulse. More specifically, the evolution of the oscillation frequency fc is advantageously chosen so that the power spectral density of a high-order harmonic k of the minimum repetition frequency fPRFÜ of the trigger pulses remains constant over a spectral band defined by a template. This is illustrated in [Fig. 2] in the case jPRF0 = 2Q MHz and K = 400. In the figure, the dashed line represents a spectral template centered around 8 GHz and the solid line represents the spectrum of the harmonic k = 400 (smoothed by phase scrambling to suppress fast oscillations at 20 MHz, see below).It is understood that spectral efficiency is maximized when the harmonic spectrum remains constant across the entire width of the template. To achieve this, a frequency evolution law for the oscillator 120 is chosen, given by [equation missing].

[0069] r. . r, Bprf / U (7) JC \J PRF0 ' chirp / /

[0070] The spectrum STX(f) of the signal sT x(t) given by (6) and (7) can be written

[0071] = E” ( ei <f+f / n^ - e< ) ■ e-J W)w

[0072] Where "j" is the imaginary unit.

[0073] Fig. 3 is a spectrogram of the sTX(t) signal. The oblique dashed lines highlight the modulated ("chirpized") discrete harmonics whose frequency is given by . / f, bpkf A. “x*PRF0+ Tdùrp )

[0074] The spectrum of equation (8) can be approximated by

[0075] MH ( tT^ï \ (9) -11^ Tch.rp

[0076] Where n is a gate function, which is 1 inside the interval [-1 / 2, F?] and 0 outside. It can be verified that the coefficients ck(t) of the series vary very little over a duration Tchirp; it is therefore possible to replace them with their average values ​​ck over said duration Tchiip. We thus obtain:

[0077] / \ - , , „+sc _ \ ~ WO - \k=ock ' ij 'n \ TMrp )

[0078] K <10)

[0079] Equation (10) highlights the fact that the generated radar signal can be written as a sum of frequency-modulated components around a multiple of the fundamental frequency fpRP(r). The most intense component corresponds to k=K. Using a relatively low-frequency oscillatory signal (the first signal) to generate trigger pulses for a microwave oscillator makes it possible to obtain particularly low phase noise, as demonstrated in (Siligaris 2023). Unlike the case of (Siligaris 2023), however, in the case of the invention the radar signal is pulsed and the frequency of the microwave carrier varies linearly from one pulse to the next.

[0080] The sTX(t) signal is intended to be emitted by a radar antenna and reflected by one or more targets. The echo signals are then detected and demodulated to extract target distance information. If we consider a single target at a distance D of the transmitting and receiving antennas (assumed to be coincident or co-located - case of monostatic radar, but generalization to the bistatic case poses no difficulty), the echo is received with a delay = D / C, where c is the speed of light. The received signal can therefore be written (unless there is a multiplicative factor of amplitude):

[0081] , V t G ^AR' ^chirp ) (11)

[0082] Where “cc” denotes the complex conjugate and the gate function can be omitted since time is defined only over an interval [t^ Tchirp), that is, between the beginning of the reception of the echo and the end of the transmission signal.

[0083] The demodulation of this received signal can be carried out in several ways. Two will be considered in the following (as well as their combination): - Coherent demodulation using a local oscillator signal linearly modulated in frequency; - Non-uniform sampling.

[0084] Figure 4 shows the functional diagram of a radar device according to an embodiment of the invention comprising a transmission channel 1 and a reception channel 2. The transmission channel essentially consists of a radar signal generation device of the type shown in Figure 1. The reception channel 2 performs coherent demodulation of a received echo signal, as explained below.

[0085] The receiving channel 2 includes a low-noise amplifier 210 for pre-amplifying the received signal, followed by an I / Q mixer 220 for converting it to an intermediate frequency. The mixer also receives as input a local oscillator signal Srx, which it multiplies by the pre-amplified echo signal. The local oscillator signal Srx is a linearly frequency-modulated signal that reproduces the K harmonic of the oscillations of the first signal. [00861

[0087] This signal is advantageously obtained by taking, by means of a separator 114, a portion of the first signal inside the module 110 and multiplying its frequency by a factor K, for example by means of a PLL 230.

[0088] By writing the echo signal as the sum of its two quadratures

[0089] sTx(i ■ TAR) - STX^ "tARJ' STX (i~rAR)Q(^

[0090] It is found that the signal at the output of mixer 220 can be written

[0091] «MixtO - (^(I-^arVF^O-TarIq) conj(sRX (t) ) (14)

[0092] Where “conj” denotes the complex conjugation operation. By setting _ bprf on u — T .. *chu p can write, in a more compact way:

[0093] SMIX ( O = O* ' + (15)

[0094] It is observed that, for k=K, a useful intermediate frequency signal is obtained

[0095] SIF(O — ckr' prf^ar + . ej2n{2-K-ctT)ej <p (16)

[0096] Where (p = 2tt ( - kr- fPRF{ / ar + kr - a- ) is a constant phase shift over time. This signal sIF(t) is essentially a sinusoidal signal whose frequency is proportional to the flight time tar, and therefore to the distance from the target.

[0097] For any value of k other than K, there will be other contributions in the spectrum, namely spread spectra dependent on the difference Kk and raised to a higher frequency in absolute value than that of the useful signal sIF(t), dependent on k and with an additional phase shift dependent on the distance and proportional to k. The amplitude of these other contributions will be lower relative to the useful component s1F(t) because these signals will be spectrally spread and weighted by coefficients ck of lesser amplitude than ck = K -

[0098] Figure 5A shows a graph of the I (upper panel) and Q (lower panel) components of the SmixW signal. Figure 5B is a detailed view of this signal obtained by stretching the time axis. Figure 6A is the spectrum of this signal, showing a central peak corresponding to the useful signal at an intermediate frequency—whose position provides access to the target distance—and secondary peaks at higher frequencies (multiples of / PRFq) that are less intense and more spectrally spread out, corresponding to harmonics of a different order than K. Figure 6B is a detailed view of this signal obtained by stretching the time axis to show only the central peak.

[0099] A low-pass filter 250 filters out these unwanted spectral components, and then an analog-to-digital converter 260 samples and digitizes the filtered signal. Thanks to the filtering of the harmonics of fPRFQ, which avoids aliasing effects, the analog-to-digital converter 260 can operate at a relatively low acquisition rate, on the order of fPRFQ (20 MSpS for fPRF^^ MHz). Advantageously, the converter 260 can receive a timing signal from the first signal generator 110, and more specifically from the clock 111.

[0100] The radar device of [Fig. 4] also operates in the presence of multiple targets at different distances. For example, [Fig. 7A] illustrates the signal at the output of mixer 220 in the case of three targets C1, C2, and C3 at distances of 1 m, 5 m, and 12 m, respectively. [Fig. 7B] illustrates the signal at the output of low-pass filter 250, and [Fig. 7C] its spectrum, obtained, for example, at the output of a digital Fast Fourier Transform (FFT) calculation module 270. The three peaks corresponding to the three targets are clearly visible in the spectrum.

[0101] The impulsive nature of the received signal allows for the use of a gain matching system applied to the mixer output, which makes it possible to compensate Analogously, the signal attenuation due to free-space losses is compensated either for the effects of radar cross-section (RCS) disparities present in different range boxes, or for spurious echoes, or any combination of these options. As illustrated in [Fig. 4], the gain matching system comprises variable-gain amplifiers 240 driven by a gain profiler 245. The latter can apply a predefined gain profile or determine it automatically based on the time evolution of the received signal amplitude. Even without gain matching, the temporal separation of the pulses prevents receiver saturation and thus improves radar dynamics. Thanks to the temporal separation of the pulses, the converter resolution 260 can be relatively low, for example, 8 bits.

[0102] Figure 8 shows the functional diagram of a radar device according to another embodiment of the invention comprising a transmit channel 1 and a receive channel 2. The transmit channel essentially consists of a radar signal generation device of the type of Figure 1. The receive channel 2 differs from that of the device in Figure 8 mainly by the absence of the mixer 220 (and therefore also of the frequency multiplier 230), by the presence of a coupler 280 to generate the I and Q components from the received and pre-amplified signal, and by the fact that the analog-to-digital converter is driven by the sPRF trigger pulses from the generator 113 and time-shifted by a duration by a delay line.As will be explained below, a given delay value only allows the detection of an echo signal corresponding to a time of flight Tak “ Tg«ie. To detect targets at unknown distances, a variable delay line can be used for sequential acquisition, or a plurality of delay lines introducing different delays rgate^ ... Tgate.%... rgate,L to drive respective converters 260i ... 260L, each equipped with its own transform calculation module 270i ... 270L. This second solution is shown in the figure. These converters, operating in parallel, have a low acquisition rate (in the example considered, 20 MSps, where 1 MSps = 10⁶ samples per second) and a modest resolution (e.g., 8 bits). They are therefore small and have low power consumption, which allows their number to be increased.It is also possible to combine approaches, using several variable delay lines in parallel.

[0103] An analog-to-digital converter, time-shifted by a trigger pulse with a delay, acquires the echo signal—filtered by the 250' low-pass filter—at times

[0104] . / } (17)

[0105] It should be noted that the low-pass filter 250' has a cutoff frequency much higher than that of the filter 250 of the embodiment of [Fig.4], because it operates on a radio frequency signal.

[0106] The analog-to-digital converter has a wide analog bandwidth up to the microwave carrier (for example at 8GHz) but its sampling rate is much lower (on the order of fpRFO, for example 20 MHz), which largely defines its energy consumption and feasibility.

[0107] Equation (17) defines a subsampling of the microwave echo signal, which allows the component corresponding to a time of flight to be extracted, as illustrated in [Fig. 9A] and [Fig. 9B]. In [Fig. 9B], in particular, the groups of three echo pulses corresponding to the three targets are clearly distinguishable.

[0108] The embodiment of [Fig. 10] combines frequency conversion by means of the mixer 220 and the frequency multiplier, and direct detection by sampling by means of converters 260i ... 260L timed by trigger pulses offset by means of delay lines 290i .. .290L. This makes it possible to avoid the use of a low-pass filter.

[0109] Figures 11A and 1IB illustrate the sampling of the intermediate frequency signal according to this embodiment of the invention. Figure 1IC shows the Fourier transform of the signals sampled with three offsets corresponding to the time-of-flights for targets at 1, 5, and 12 meters.

[0110] The invention has been described with reference to particular embodiments, but variations are possible. For example, the generation of two I and Q components of the signals is not essential (but advantageous, as it allows for a reduction in the acquisition rate). Similarly, the frequency ranges indicated are given only as a non-limiting example.

[0111] The invention is particularly suitable for the implementation of "on-a-chip" radars in which the radar device (or at least the transmission channel and / or the acquisition channel) is monolithically integrated, but it is not limited to this case. References

[0112] (Antide 2020) E. Antide, M. Zarudniev, O. Michel et M. Pelissier, "Comparative Study of Radar Architectures for Human Vital Signs Measurement," 2020 IEEE Radar Conférence (RadarConf20), Florence, Italy, 2020, pp. 1-6, doi: 10.1109 / RadarConf2043947.2020.9266569.

[0113] (Liu 209) Y.-H. Liu et al., "9.3 A680 qW Burst-Chirp UWB Radar Transceiver for Vital Signs and Occupancy Sensing up to 15m Distance," 2019 IEEE International Solid- State Circuits Conférence - (ISSCC), San Francisco, CA, USA, 2019, pp. 166-168, doi: 10.1109 / ISSCC.2019.8662536.

[0114] (Andersen 2017) N. Andersen et al., "A 118-mW Pulse-Based Radar SoC in 55-nm CMOS for Non-Contact Human Vital Signs Détection," in IEEE Journal of Solid-State Circuits, vol. 52, no. 12, pp. 3421-3433, Dec. 2017, doi: 10.1109 / JSSC.2017.2764051.

[0115] (Siligaris 2023) Siligaris, A., Bossuet, A., Barrau, L., Antide, E., Gonzalez-Jimenez, J. L., Dehos, C., & Zarudniev, M. (2023, September). Fast Chirping 58-64 GHz FMCW Radar Transceiver using D-PROT Multiplier in CMOS 45nm RFSOI for Vital Signs Détection. In ESSCIRC 2023—IEEE 49th European Solid State Circuits Conférence (ESSCIRC) (pp. 505–508). IEEE.

Claims

Demands

1. 1. A radar signal generation device comprising: - a generator (110) of periodically repeated series of trigger pulses (sPRF), the pulses of the same series having a linearly variable time spacing; and - an oscillator (120) configured to receive said trigger pulses as input and to generate a train of periodic oscillations corresponding to each said trigger pulse; characterized in that - said oscillator (120) has a variable frequency depending on a control signal (Vtune); and in that - the device also comprises a generator (112) of said control signal, adapted to linearly vary in time the frequency of the oscillator (120) from one train of periodic oscillations to another for trains of periodic oscillations triggered by trigger pulses belonging to the same series.

2. 2. Device according to claim 1 wherein the oscillator (120) and the generator (112) of said control signal are configured so that the average frequency of the oscillations of said periodic oscillation trains is a multiple, by a factor K>1, of an average repetition frequency of the trigger pulses.

3. 3. Device according to claim 2 in which said trigger pulse series generator (110) comprises a first signal generator (112) and a trigger pulse generator (113), the first signal generator (112) being configured to generate a first signal consisting of periodic repetitions of frequency linearly modulated oscillation trains, the trigger signal generator being configured to receive said first signal as input and to generate said trigger pulses synchronously with the oscillations of said first signal.

4. 4. Device according to claim 3 wherein the oscillator (120) and the generator of said control signal (112) are configured such that the frequency of the periodic oscillation train generated corresponding to an nth trigger pulse of said series is given by: tc(n) = K^fPRF0±^—ts(n) j where fpRFo is the frequency of the oscillations of the first signal at the beginning of each frequency-modulated oscillation train, bprf is the frequency modulation rate Tchirp of said oscillations and ts(n) is a generation time of said nth trigger pulse, with ts(n=1) = 0.

5. 5. Device according to any one of claims 3 or 4 wherein said first signal generator (112) is configured so that the frequency modulation of the oscillations of said first signal is driven by said control signal.

6. 6. Device according to any one of claims 2 to 5 wherein the K factor is greater than or equal to 10 and preferably between 100 and 1000.

7. 7. Device according to any one of the preceding claims also comprising a pulse shaper (130) for modifying the time profile of said periodic oscillation trains.

8. 8. Device according to any one of the preceding claims wherein said oscillator (120) is a microwave oscillator.

9. 9. Device according to any one of the preceding claims also comprising a device (150) for applying phase scrambling to said periodic oscillation trains.

10. 10. Radar device comprising: - a transmit channel (1) comprising a device for generating a radar signal according to any one of the preceding claims; and - a receive channel (2) configured to receive echoes of said radar signal and to demodulate them synchronously with their generation so as to extract time-of-flight information.

11. 11. Device according to claim 10 in which the receiving channel (2) includes an amplifier (240) equipped with an automatic gain adjustment device (245) to compensate for variations in the intensity of the received echoes.

12. 12. Device according to any one of claims 10 and 11 wherein: - the radar signal generation device of the transmission channel (1) is a device according to any one of claims 3 to 6, or any one of claims 7 to 9 when it depends on claim 3, comprising said first signal generator (112) configured to generate said first signal consisting of periodic repetitions of oscillation trains linearly modulated in frequency; and - the receiving channel includes: - a frequency multiplier (230) configured to receive said first signal as input and multiply its frequency by said factor K; and - a mixer (220) configured to receive said radar echoes as input and demodulate them by mixing with a signal from said frequency multiplier (230) in order to obtain a demodulated signal; - a low-pass filter (250) configured to attenuate components of said demodulated signal at frequencies that are multiples of said average repetition frequency of the trigger pulses; and - an analog-to-digital converter (260, 260i, 260L) having an acquisition rate suitable for converting a signal at said average repetition frequency.

13. 13. Device according to any one of claims 10 and 11 wherein: - the radar signal generation device of the transmit channel (1) is a device according to claim 2; and - the receive channel (2) comprises at least one analog-to-digital converter (260i,260l) having an analog bandwidth of width at least equal to the spectral width of said received echoes but an acquisition rate adapted to convert a signal at said average repetition frequency of the trigger pulses, said converter being synchronized with said trigger pulses through a delay line (290i, 290l).

14. 14. A device according to any one of claims 10 and 11, wherein: - the radar signal generation device of the transmitting channel (1) is a device according to any one of claims 3 to 6, or any one of claims 7 to 9 when it depends on claim 3, comprising said first signal generator (112) configured to generate said first signal consisting of periodic repetitions of frequency-linearly modulated oscillation trains; and - the receiving channel (2) comprises: - a frequency multiplier (230) configured to receive said first signal as input and multiply its frequency by said factor K; and - a mixer (220) configured to receive said radar echoes as input and demodulate them by mixing with a signal from said frequency multiplier (230) in order to obtain a demodulated signal; and - at least one analog-to-digital converter (260i, 260L) having an analog bandwidth of width at least equal to the spectral width of said received echoes but an acquisition rate adapted to convert a signal at said average repetition frequency of the trigger pulses, said converter being synchronized with said trigger pulses through a delay line (290b 290L).

15. 15. Device according to any one of claims 13 or 14 wherein said delay line has a variable delay.

16. 16. Device according to any one of claims 13 or 14 wherein the receiving channel comprises a plurality of said analog-to-digital converters (260i, 260L) synchronized with said trigger pulses through respective delay lines introducing different delays.

17. 17. Device according to any one of claims 10 to 16 wherein the radar signal generation device of the transmit channel (1) includes a device for applying phase scrambling (150) and the receive channel includes a phase scrambling device (150).

18. 18. A method for generating a radar signal comprising the generation of periodically repeated series of periodic oscillation trains, the oscillation trains of the same series having a linearly variable time spacing; the frequency of the oscillations varying linearly from one periodic oscillation train to another in the same series.

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