Pulsed radar using a hybrid pulsed and frequency modulation approach

Abstract

Pulsed radar by hybrid pulsed and frequency modulation approach. Radar device comprising: - a transmit channel (1) configured to generate a radar signal (IE); and - a receive channel (2) configured to receive echoes (SE) and demodulate them synchronously so as to extract time-of-flight information; characterized in that the transmit channel (1) and the receive channel (2) comprise: - a shared local oscillator (1204), having a variable frequency according to a control signal; - a generator (1202) of said control signal, adapted so that the frequency of said local oscillator varies in time either linearly or linearly in steps, with possible temporal scrambling;and- means for shaping (101, 201) pulses to generate a series of transmit pulses (IE) and a corresponding series of receive demodulation pulses (IDR), each receive demodulation pulse having a duration greater than that of the corresponding transmit pulse and defining a respective time range for receiving echoes. Figure for the abstract: Fig. 1;

Description

Title of the invention: Pulsed radar using a hybrid pulsed and frequency modulation approach

[0001] The invention relates to the field of radar, and more particularly to radar-on-a-chip, which can be used, in particular, for measuring a patient's vital signs or as a presence detector, see for example (Antide 2020). These radars must exhibit both high spatial resolution (on the order of a few centimeters) and low power consumption (a few tens of mW).

[0002] A commonly used technique for these applications is that of frequency-modulated continuous-wave, duty-cycled (FMCW-DC), see for example (Liu 2019) and (Siligaris 2023). Indeed, this technique exploits the principle of compressing a wave emitted with a wide bandwidth to a narrow bandwidth in the intermediate frequency, which makes it possible to use analog-to-digital converters (ADCs) with a relatively low acquisition rate, of a few MSps or tens of MSps (IMSps = 10⁶ samples per second - acronym for "Mega Samples per second").However, when dealing with targets exhibiting highly diverse radar cross sections (RCS), it will be necessary to use ADCs with high dynamic range, for example 9 to 12 bits or more, and therefore relatively high power consumption.

[0003] An alternative is to use the ultra-wideband radio pulse (IR-UWB) technique, in which a short pulse is emitted and the time of flight of its echo is measured to determine the distance to the target. See, for example, (Andersen 2017). One advantage of this technique is that, since the echo signals are separated in time, it is possible to apply automatic gain control (AGC) to enable the acquisition of a highly contrasting environment in the presence of very diverse RCS objects while limiting the dynamic range of the ADCs. On the other hand, obtaining good spatial resolution relies on the use of a high acquisition rate, with several GSpS (IGSps = 10⁹ samples per second).

[0004] The invention aims to overcome, in whole or in part, the aforementioned drawbacks of the prior art. More particularly, it aims to enable the use of ADCs with lower dynamic range than FMCW-DC radars and a lower acquisition rate than IR-UWB radars, without sacrificing either the spatial resolution or the RCS dynamic range of detectable targets.

[0005] According to the invention, this objective is achieved through a hybrid approach combining the use of a pulsed radar signal and the principle of frequency modulation. More particularly, the invention uses a radar signal consisting of a series of transmission pulses in which the carrier frequency varies, for example, linearly, from one pulse to the next. Demodulation pulses, generated simultaneously with the transmission pulses by a shared local oscillator, are used to demodulate the echoes of the transmission pulses. Since the radar signal is pulsed, the echo signals are separated in time, as in the case of the IR-UWB technique, allowing gain adjustment that permits the use of relatively low dynamic range ADCs even in the presence of a contrasting environment.Furthermore, the ADC acquisition rate depends on the pulse repetition frequency and not on the spatial resolution; it can therefore be lower than in the conventional IR-UWB technique. Advantageously, the technique of the invention is compatible with the use of frequency, pulse repetition period, and / or phase scrambling, which is desirable particularly for security reasons (immunity to interference and electromagnetic attacks).

[0006] Also, an object of the invention is a radar device comprising: - a transmission channel configured to generate a radar signal; and - a reception channel configured to receive echoes of said radar signal and to demodulate them synchronously with their generation so as to extract time-of-flight information; characterized in that the transmission channel and the reception channel comprise: - a shared local oscillator, exhibiting a variable frequency depending on a control signal; - a generator of said control signal, adapted so that the frequency of said local oscillator varies over time either linearly, linearly in steps, or by taking a plurality of discrete values ​​obtainable by scrambling a linear variation in steps; and - means for shaping a signal generated by said local oscillator at said variable frequency, adapted to generate a series of transmission pulses and a corresponding series of demodulation pulses in reception, each demodulation pulse in reception having a duration greater than that of the corresponding transmission pulse and defining a respective time range for receiving echoes.

[0007] According to particular embodiments of such a device:

[0008] - The formatting means may include: - a first pulse shaper, belonging to said transmission channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and to provide at its output said transmission pulses; and - a second pulse shaper, belonging to said reception channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and to provide at its output said demodulation pulses in reception.

[0009] - The radar device may also include a control device for said local oscillator configured to restart said local oscillator during each generation of a transmit pulse and the corresponding receive demodulation pulse.

[0010] - The formatting means may include: - said control device for said local oscillator, which is then configured to shape the rising edges of the signal generated by the latter at said variable frequency; and - a falling edge shaping device, arranged on said transmission channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and to provide at its output said transmission pulses; said falling edge shaping device also being configured to generate a reset signal of said local oscillator.

[0011] - The radar device may also include a period jammer of pulse repetition, configured to control said local oscillator control device so as to vary pseudo-randomly the interval between two successive restarts of said local oscillator.

[0012] - The radar device may also include a phase jammer for apply the same phase scrambling to the transmit pulses and the receive demodulation pulses.

[0013] - The receiving path may include a mixer configured to receive in input of said echoes of said radar signal and one of said demodulation pulses at the receiver and provide at the output a mixed signal, a low-pass filter to filter said mixed signal and an analog-to-digital converter to convert the filtered mixed signal to digital format. More particularly, said low-pass filter may include a windowed integrator, configured to integrate said mixed signal over successive integration time windows, each said integration time window being contained within the echo reception time range defined by said demodulation pulse at the receiver, and to be reset at the end of said integration window. Said windowed integrator may also have a time-varying integration gain in a predetermined or adaptive manner. By Furthermore, said control signal generator can be adapted so that the frequency of said local oscillator varies in time by taking a plurality of discrete values ​​obtainable by scrambling a linear variation in steps; and the receiving channel can also include a means for frequency scrambling of the filtered mix signal converted to digital format.

[0014] - The receiving channel may include an amplifier equipped with a device automatic gain adjustment to compensate for variations in the intensity of received echoes.

[0015] - Said local oscillator may be a microwave oscillator.

[0016] Other features, details and advantages of the invention will become apparent from the description given with reference to the accompanying drawings provided by way of example, which represent, respectively:

[0017] [Fig. 1], the functional diagram of a device according to a first embodiment of the invention;

[0018] [Fig.2], transmission pulses and reception demodulation pulses corresponding;

[0019] [Fig.3], a graph illustrating the temporal sequence of emission pulses and demodulation in reception, as well as of the signal generated by the local oscillator;

[0020] [Fig.4], the functional diagram of a device according to a second embodiment of the invention;

[0021] [Fig.5A], the spectrum of an emission signal;

[0022] [Fig.5B] and [Fig.5C], the spectrum of the mixed signal in reception before and after low-pass filtering, respectively, in the presence of a single target;

[0023] [Fig.6A], [Fig.6B], [Fig.6C] and [Fig.6D], the mixing signal in the domain of the time in the presence of a single target before and after low-pass filtering ([Fig.6A]: in-phase component before filtering; [Fig.6B]: filtered in-phase component; [Fig.6C]: quadrature component before filtering; [Fig.6D]: filtered quadrature component];

[0024] [Fig.7A], [Fig.7B], [Fig.7C], [Fig.7D], the spectrum of the mixed signal in reception in the presence of two targets, respectively: unfiltered and without gain profiling ([Fig.7A]); after low-pass filtering but without gain profiling ([Fig.7B]); unfiltered but with gain profiling ([Fig.7C]); after low-pass filtering and with gain profiling ([Fig.7C]);

[0025] [Fig.8], three different types of local oscillator control signals;

[0026] [Fig.9], the functional diagram of the receiving channel of a device according to a third embodiment of the invention; and

[0027] [Fig. 10], the use of a windowed integrator to simultaneously perform low-pass filtering, gain profiling and suppression of unwanted signals.

[0028] The radar device of [Fig. 1] comprises a transmit channel 1, for generating a radar signal consisting of so-called transmit pulses IE, and a receive channel 2, for receiving echoes of these pulses reflected by targets and processing them to extract time-of-flight information. These two channels have shared components 12.

[0029] The components shared between the transmit and receive channels include a clock 1200 that generates a timing signal for the device, determining the repetition period (PRP) of the transmit pulses. Optionally, a time scrambler 1201 introduces pseudo-random variations in this repetition period; the interval between the nth transmit pulse and the next is then denoted by PRPn (see [Fig. 2]). In this case, the smallest possible PRP determines the maximum target distance that can be measured.

[0030] The timing signal, possibly time-scrambled, drives a triggering device 1203 for a frequency-controlled local oscillator 1204. At each timing signal, the device 1203 restarts the local oscillator 1204, then stops it before the arrival of the next timing signal. The initial phase of the oscillator 1204 is the same at each restart. Its oscillation frequency, however, varies linearly from one restart to the next. This is made possible by a control signal generator 1202 for the oscillator, driven by the timing signal (possibly time-scrambled). In contrast, the frequency of the local oscillator remains constant between its triggering and stopping. Generally, the frequency of the local oscillator is preferably in the microwave range.This refers to frequencies between 300 MHz and 300 GHz or, more restrictively, the range 1 GHz - 100 GHz.

[0031] The SO oscillator signal generated by the local oscillator 1204—consisting of an in-phase component I and a quadrature component Q—is fed as input to a first pulse shaper 101 belonging solely to the transmit channel 1, configured to generate, at each triggering of the local oscillator, a transmit pulse IE typically having a duration on the order of a few nanoseconds (ns). The upper part of [Fig. 2] illustrates a transmit pulse IEn comprising a rising edge FME and a falling edge FDE, followed—after a PRP interval—by another transmit pulse IEn+i. The latter has an envelope identical to that of IEn, but a carrier frequency of a different type, as explained above.

[0032] The I and Q components of the IE emission pulses are combined by a combiner 102, then supplied as input to a power amplifier 103 to be subsequently transmitted by an antenna not shown.

[0033] The SO oscillator signal is also supplied as input to a second pulse shaper 201 belonging solely to the receive channel 2, configured to generate, at each triggering of the local oscillator, a so-called "receive demodulation" (IDR) pulse, preferably of a significantly longer duration (for example, by at least a factor of 10) than that of the transmit (IE) pulses. The duration of the IDR pulse sets an upper limit on the pulse repetition frequency and determines the acquisition "depth," that is, the maximum target distance that can be detected.

[0034] The lower part of [Fig. 2] illustrates a receive demodulation pulse IDRn comprising a rising edge FMR and a falling edge FDR, followed – after a PRP interval – by another receive demodulation pulse IDRn+i. The latter has an envelope identical to that of IDRn, but a carrier frequency of a different type, as explained above. In contrast, the carriers of the receive demodulation pulses IDRn and IDRn+i are identical to those of the corresponding transmit pulses IEn and IEn+i.

[0035] A phase scrambler 1205 may optionally be provided to apply identical polarity reversals, in a pseudo-random sequence, to the IE and IDR pulses. If the reversals are applied independently to the two quadratures of these pulses, the result is QPSK modulation by a pseudo-random signal. Alternatively, the scrambler 1205 may apply phase jumps that can take more than two values.

[0036] The IDR pulses are fed into a mixer 203 which also receives, on another input, an echo signal SE, captured by an antenna (not shown) and amplified by a low-noise amplifier 202. Initially, the echo signal SEn, which is mixed with the receive demodulation pulse IDRn, is considered to originate from the reflection of the transmit pulse IEn by a target at a distance d and therefore corresponds to a replica of said transmit pulse delayed by r = 2dlc, where c is the speed of light. It is understood that, under these conditions, the receive demodulation pulse defines a reception window for the echo signals.

[0037] Figure 3 illustrates the temporal sequence of the envelopes of the transmit pulses IE and receive demodulation pulses IDR, as well as the signal generated by the local oscillator SO. At an initial time t0, the local oscillator 1204 receives the control signal that defines its oscillation frequency, then, at a time t1 approximately 1 ns later, it receives a trigger signal that starts the oscillation. It can be seen that the amplitude of the SO signal gradually increases before stabilizing at a time t2 and maintaining a constant value until the oscillator stops at a time t3, several tens of nanoseconds later. This shutdown occurs approximately 10 ns before the arrival of the next trigger signal. The rising edges of the transmit (IE) and receive (IDR) pulses begin once the SO signal amplitude has stabilized. While the transmit (IE) pulse is very brief (e.g., 6 ns), the receive demodulation pulse continues until shortly (e.g., 1 ns) before the oscillator shuts down.

[0038] The SM mixing signal from the mixer 203 is first filtered by an optional high-pass filter, the purpose of which is to remove DC or very low-frequency components resulting from direct coupling between the transmitter and receiver, as well as other parasitic effects. Next, the signal is preferably amplified by a variable-gain amplifier 205, which adjusts the gain to compensate for differences in the intensity of the various received echoes. The variation can follow a predefined profile—for example, increasing over time, because later echoes correspond to a more distant target and therefore greater attenuation—or it can be adaptive.

[0039] The SM mix signal then passes through a low-pass filter 206, with a bandwidth on the order of a few MHz (depending on the frequency slope used in transmission and the desired maximum echo distance). The filtered SMF mix signal at the output of the filter 206 is then sampled and converted to digital format by an analog-to-digital converter (ADC) 207. Given the small bandwidth of the filter 206, the converter 207 can have a relatively low acquisition frequency, for example, 20 MSpS (1 MSpS = 10⁶ samples / second). The use of a variable-gain amplifier makes it possible to limit the converter's resolution—for example, to 12 bits—while maintaining acceptable dynamic range.

[0040] The signal converted to digital format is then processed by a processor 208 to extract time-of-flight information from the echo signals. As will be explained later, this processing can consist of simply calculating a Fourier transform.

[0041] Figure 4 shows the functional diagram of a device according to a second embodiment of the invention. This device differs from that of Figure 1 in that the local oscillator 1204 is triggered by a device 1206 which simultaneously shapes the rising edge of the IE and IDR pulses. The falling edge of the IE pulses is shaped by a device 105 receiving the oscillator signal as input, while the falling edge of the IDR pulses is simply defined by the oscillator stopping. In other words, pulse generation occurs partially in baseband, instead of entirely in the microwave domain as in the first embodiment. Spectrum control is, however, more difficult.

[0042] Document FR 3 099 910 discloses a circuit that can be used to implement pulse shapers 101, 201 and 105. This circuit is compatible with binary phase scrambling.

[0043] Document FR 3 015 153 describes a triggered microwave oscillator with controlled oscillation frequency, which can be used to implement the local oscillator 1204.

[0044] The operation of the device in [Fig.1] and [Fig.4] will now be illustrated using an analytical model and numerical simulations.

[0045] First, a reference baseband transmission pulse p0(t) is defined whose frequency characteristics are compatible with regulatory templates and the minimum bandwidth required, and which has the shortest possible time support. For example, a Gaussian pulse with the expression: [°°46]

[0047] Where the duration parameter Tp is related to the BXiffi band of the pulse by

[0048] (2)

[0049] xdB being the reference level for the measurement of the band. We typically take xdB=-10 dB and BXdIi=500 MHz which gives Tp~ 1.367ns and an effective pulse duration at 99% of its amplitude equal to 4.292^1^-5.867^. A significantly shorter pulse is possible; the order of magnitude to be retained in practice is -2—.

[0050] The reference emission pulse is then time-shifted so that its effective support is on [0, PTp}, with for example P = 4.292.

[0051] We now consider the case where the local oscillator 1204 is restarted at each pulse from the same initial state, with a phase conventionally taken to be 0 at each pulse start and a frequency updated at each pulse following a linear ramp rising with slope _ Bdwp where Bchiro and — t ,. 1 is the chirp frequency excursion and T^rp the chirp duration, the frequency remaining constant between two restarts. We also consider the PRP interval between two oscillator restarts to be constant and equal to T; therefore, the number of pulses is... The demodulation pulse in receive mode is taken from length equal to T and phase interference is not taken into account. The analytical expression of the emitted radar signal, in the form of a series of emission pulses, is therefore given by

[0052] s ( t ) = ( t-nT ) (3)

[0053] With œ (n) - wrn + 27iBchirp = wm + n27raT = œm + narT with ar = 2îra

[0054] The analytical expression of the sequence of demodulation pulses in IDR reception is written 100551 LO^ty^^d-nT)ÿ-jiÀh^t-nD (4)

[0056] With Ry(t) a unit rectangular time gate of duration T centered on I.

[0057] Considering the case of a reflection of the radar signal by a single target at a distance d, the mixing signal SM at the mixer output is obtained by calculating the product of the radar signal delayed by r = 2J / c and the signal LO^t. We obtain

[0058] SM ( t ) -s(tr) LORx (t) - (t-nT-t) ( t-nT ) (5)

[0059] It is assumed that there is no interference between consecutive PRPs, that is to say that the transmit pulse TEn is received during the duration of the receive demodulation pulse TDRn, which assumes t + fiTP <T. On a alors :

[0060] SM(t) = (t-nT-t)Rt(t-nT)ej<^nV-nT-^ej^»it-nr) _(t-nT-t)ej"Ta'T (6)

[0061] The left-hand term is a fixed-phase term that can be neglected. The sum term is an impulse signal whose phase of consecutive impulses rotates by arT, which in the frequency domain is equivalent to having a first harmonic at p _ _ _ ar. The other frequency components have a period of 4 and Therefore, for A + p for all integers k, the object's distance information is present in this first harmonic, which is extracted by the low-pass filter 206, typically having a bandwidth of less than 4. The sampling rate of the analog-to-digital converter 207 is determined by this bandwidth. In the presence of several echoes at different distances, we obtain as many harmonics, resolved or unresolved.

[0062] Fig. 5A illustrates the power spectrum of a radar signal with a bandwidth of 500 MHz, a pulse repetition frequency of 10 MHz, a chirp defined by a_q 005^- • In dotted line, the spectral template which must be respected, and which is indeed respected.

[0063] Figure 5B illustrates the spectrum of the SM mixing signal in the presence of a target at a distance such that the delay T is 20 ns. The presence of discrete harmonics is noted, the lowest frequency of which is extracted by the low-pass filter 206. Figure 5C shows the spectrum of the filtered SMF signal, in which the higher-order harmonics are attenuated by 20 dB or more, and thus effectively suppressed. [Fig.6A] and [Fig.6C] show, respectively, the I and Q components of the SM signal in the time domain, and [Fig.B] and [Fig.6C] the I and Q components of the filtered SMF signal.

[0064] Figures 7A and 7B show the spectra of the SM and SMF signals, respectively, in the case of two reflections at 20 ns and 60 ns, with a power difference of 19 dB and in the absence of gain profiling. The splitting of the harmonics can be seen, but the component corresponding to the largest delay is difficult to detect. As illustrated in Figures 7C and 7D, the use of amplifier 205 with automatic gain adjustment makes it possible to compensate for variations in the intensity of the received echoes.

[0065] Certain assumptions of the analytical model can now be relaxed.

[0066] First, it is assumed that the condition that there is no interference between consecutive PRPs, i.e. T + [3Tp < T], is not met. In this case, there is aliasing of the spectrum of the mixed signal, because an echo is demodulated by a signal of a frequency different from that of its carrier.

[0067] First, we note that 0 < T < T is equivalent to CtT < FT < 0 and since _ a- nt we have B <w t-. n. Il y a donc un degré de liberté offert par la bande parcourue n s rT < u The chirp is used to determine the frequency difference between the intermediate frequency range (i.e., at the mixer output) and the PRF. In the previous example (PRF = 10 MHz, a = 0.025 GHz, Bchirp = 50 MHz), N = 200, and the minimum IF is -2.5 MHz, or PRF / 4. Harmonic filtering is possible because the lowest harmonic is at 7.5 MHz. If Bchirp is -1 GHz, then the minimum IF is -5 MHz and the lowest harmonic is +5 MHz, which would require complex filtering.

[0068] Let us return to the case (k+l)T>T> kT with k> 1 and assume that there is no phase interference. Then, apart from the edge effect caused by the "lost" pulses, the spectrum of the SM signal is "folded" at FT_kT. We verify that pp + K. Put another way, FT is not a continuous function of T; it is not a "classical" spectral folding.

[0069] We always assume (k + 1) T > T > kT with k > 1 and that there is phase scrambling of the BPSK type. In this case, the unscrambling sequence is misaligned since in the basic architecture it is synchronous with the scrambling sequence. Therefore, the pulses will not be summed coherently.

[0070] At this point, it is appropriate to enrich the model of the SM signal at the mixer output by taking into account the BPSK scrambling and unscrambling operations with the binary code b(n) e{-1,1}:

[0071] SM (t) = E^b (n),p0 (t - nT - t) (n).RT (t -nT) (7)

[0072] If 0 < t < T, the operation is transparent: [0°73] SM (t) = s (t - t), LOrx (t) = E^b (n)b (n), pQ (t - nT - t) i

[0074] If (k+l)T>T>kTaveck> 1:

[0075] SM(t) = £^b(n)b(n+k) &W1,

[0076] Since b(n) is ideally a white random process (sequence), then E[b(n)b(n+k)] = 0, Vk*0 and E[m(t)] = 0, so by ergodicity m(t) has zero mean. In practice, b(n) is not perfectly white and m(t) resembles interference noise spreading the power over the entire band as a function of the covariance of b(n).

[0077] In an embodiment that handles cases where (k+1)T>T> kT with k > 1, it is necessary to parallelize the desoldering operation for all desired values ​​of k. In this case, it would preferably be appropriate to generate a demodulation pulse at the receiver without phase scrambling, which requires a minor modification of the architecture of [Fig. 1] and [Fig. 4], and to perform the phase desoldering after the receiver mixer by parallelizing the baseband chain and delaying the desoldering sequence b(n) by k values. This constitutes an additional option.

[0078] Finally, there are overlapping PRP portions where the mixer output is corrupted: (kT - pTp < t < kT) the received pulse being mixed with a phase-discontinuous demodulation signal. This also applies, by extension, to PRP portions in which the demodulation signal is intentionally turned off. A change in the PRP can allow these "dead" zones to be addressed.

[0079] We now consider the case of temporal interference (pseudo-random variations in the spacing between pulses). Numerical simulations make it possible to verify that this has no impact on the frequency of the fundamental harmonic as long as the amplitude of the spacing variation is not excessive.

[0080] The following assumption concerns the restart of the local oscillator and its initial phase. In the proposed analytical model, the phase is reset to 0 (or any constant value) at each pulse, regardless of the generated frequency. The analytical model clearly shows that if this assumption is removed, then the expression for the mixing signal becomes:

[0081] SM(t) ^E^KCt-nT-^RTCt-nT)^^^ (8) 11—VU

[0082] which gives the same result, the phase term canceling out at the mixer outlet.

[0083] Finally, we are interested in the case where the oscillator frequency evolves continuously and linearly, including during the generation of the IE and IDR pulses, which therefore exhibit a "chirp". We can then write 100841 [°°851 UMt) =OT(t-nT)e*»^

[0086] With œ ) = + t = œm + art

[0087] After mixing, we obtain (still with the other initial assumptions, including phase resetting):

[0088] SM(t) = _nT-t) (11) [00S9] This signal is therefore a sinusoid of frequency FT = -ar "sampled" by a finite comb of pulses spaced by T. The result is indeed a line spectrum of frequencies p (A) weighted by the pulse spectrum. The difference The case considered previously only arises from the evolution of the local oscillator frequency during the pulse duration, which can be calculated: if the pulse lasts pTp, the frequency changes by "pTp, i.e., 0.147 MHz with 0TP = 5.867 ns and a = 0.025 GHz / ps, and the difference is bounded by 0.3°. We see that this difference is marginal considering the bandwidths under consideration.

[0090] Up to this point, we have considered the case where the oscillator control signal generated by device 1202 is a linear ramp, either increasing or decreasing. In fact, the frequency excursion must be limited, so the control signal will more likely be sawtooth or triangular (alternating between an increasing and decreasing ramp). Since, preferably, the oscillator frequency varies discretely from one trigger to the next, the control signal can also vary in steps. In [Fig. S], reference numeral 1202' designates a device for generating such a stepwise linear control signal. The values ​​taken by the oscillator control voltage are designated by the integers from 0 to 5; a stepwise linear ramp therefore corresponds to the sequence {0, 1, 2, 3, 4, 5}.According to an alternative embodiment of the invention, these discrete values ​​can be subject to jamming, which results in frequency jamming of the radar signal; in [Fig. S], reference numeral 1202” designates a device for generating such a control signal, corresponding to the sequence {4, 1, 2, 5, 3, 0}. Alternatively, certain values ​​can be omitted, for example the frequency {4, 1, 5, 3, 0} can be used.

[0091] The primary advantage of this variant relates to problems of coexistence with other radars of the same type, or to security (sequence unknown to an attacker and modifiable at will). Another advantage is the possibility of transmitting on several antennas. (MIMO operation) by using orthogonal frequency hopping sequences between antennas. By doing so, transmission can occur simultaneously on several frequency sub-bands in parallel, and each receiving channel only retrieves the correlation results with the sequence from its corresponding transmitter.

[0092] The difficulty caused by frequency interference is the need to compensate for it in the receiver; otherwise, the mixed signal will not be obtained because there will be random phase jumps between the low-frequency components of the mixed signal corresponding to successive pulses. This compensation cannot be achieved in the analog domain. To achieve it in the digital domain without having to significantly increase the acquisition rate of the analog-to-digital converter 207, it is possible to implement low-pass filtering by means of a windowed integrator 210 driven by the clock signal, as illustrated in [Fig. 9]. Since the integrator is reset to zero each time the local oscillator is reset, each digitized sample contains the useful information, low-pass filtered by the integrator over a time PRPn.A debugging module 211 applies the permutation to the samples which puts them back in the "correct" order (the one in which the frequency of the oscillator signal evolves linearly).

[0093] Even independently of the frequency jamming case, the use of a windowed integrator can prove advantageous because such a device can replace, in whole or in part, the variable-gain amplifier 205 or even the high-pass filter 204. Indeed, as illustrated in [Fig. 10], the integrator has an integration gain G that can vary in a predefined or adaptive manner to compensate for variations in the intensity of echoes received from different targets. The example in [Fig. 10] corresponds to the case where the echo signal consists of reflections from targets with substantially equivalent radar cross-sections, but located at different distances.In this case, later reflections are generally weaker because they undergo greater attenuation, which can be compensated for by an increasing integration gain G within the integration window FL. Furthermore, unwanted SEI signals intended to be suppressed by the 204 high-pass filter arrive at the receiving channel very early, as they are primarily due to direct coupling with the transmitter. These signals can therefore be suppressed, at least partially, by shifting the start of the integration window relative to the beginning of the PRP period by a few nanoseconds.

[0094] The invention has been described with reference to particular embodiments, but variations are possible. For example, the generation of two I and Q components of the signals is not essential (but advantageous, as it allows for a reduction in the acquisition rate). Similarly, the frequency and time ranges indicated are given only as non-limiting examples. Furthermore, the use of two quadrature components are not essential; if only component I is used, combiner 102 is not necessary.

[0095] The duration(IE)“duration(IDR) condition is not essential. In the extreme, it is even possible to take the same duration for both pulses.

[0096] In certain embodiments, particularly operating at THz (Terahertz) frequencies, the low-noise amplifier 202 at the input of the receiving channel may be omitted.

[0097] In the presence of time interference in the PRP, the analog-to-digital converter 207 and / or the optional windowed integrator 210 can be driven by the scrambled clock signal. However, this is not essential because the frequency of the filtered mixing signal depends only on the delay between the echo signal and the demodulation pulse at the receiver, and is therefore not affected by any time interference.

[0098] The processor 208 and, where applicable, the frequency debugging means 211 can be implemented using dedicated digital integrated circuits (FPGA, ASIC) or, more advantageously, a suitably programmed microprocessor. In the latter case, the frequency debugging means 211 can be implemented by software.

[0099] The invention is particularly suitable for the implementation of "on-a-chip" radars in which the radar device (or at least the transmission channel and / or the acquisition channel) is monolithically integrated, but it is not limited to this case. References

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[0101] (Liu 209) Y.-H. Liu et al., "9.3 A680 qW Burst-Chirp UWB Radar Transceiver for Vital Signs and Occupancy Sensing up to 15m Distance," 2019 IEEE International Solid-State Circuits Conference - (ISSCC), San Francisco, CA, USA, 2019, pp. 166-168, doi: 10.1109 / ISSCC.2019.8662536.

[0102] (Andersen 2017) N. Andersen et al., “A 118-mW Pulse-Based Radar SoC in 55-nm CMOS for Non-Contact Human Vital Signs Detection,” in IEEE Journal of Solid-State Circuits, vol. 52, no. 12, pp. 3421-3433, Dec. 2017, doi: 10.1109 / JSSC.2017.2764051.

[0103] (Siligaris 2023) Siligaris, A., Bossuet, A., Barrau, L., Antide, E., Gonzalez-Jimenez, J.L., Dehos, C., & Zarudniev, M. (2023, September). Fast Chirping 58-64 GHz FMCW Radar Transceiver using D-PROT Multiplier in CMOS 45nm RFSOI for Vital Signs Detection. In ESSC1RC 2023-1EEE 49th European Solid State Circuits Conférence (ESSCIRC) (pp. 505-508). IEEE.

Claims

Demands

1. Radar device comprising: - a transmission channel (1) configured to generate a radar signal (IE); and - a receiving channel (2) configured to receive echoes (SE) of said radar signal and to demodulate them synchronously with their generation so as to extract time-of-flight information; wherein the transmitting channel (1) and the receiving channel (2) comprise: - a shared local oscillator (1204), exhibiting a variable frequency depending on a control signal; - a generator (1202) of said control signal, adapted so that the frequency of said local oscillator varies linearly in time, or varies linearly in steps from one pulse to the next, or takes on a plurality of discrete values ​​obtainable by scrambling a linear variation from one pulse to the next in steps; and - means for shaping (101, 201, 1206, 105) a signal generated by said local oscillator (1204) at said variable frequency, adapted to generate a series of transmit pulses (IE) forming said radar signal and a corresponding series of receive demodulation pulses (IDR), each receive demodulation pulse having a duration greater than or equal to that of the corresponding transmit pulse and defining a respective time range for receiving echoes; said receiving channel (2) further comprising a mixer (203) configured to receive as input said echoes (SE) of said radar signal and one of said receive demodulation pulses (IDR) and to provide as output a mix signal (SM); characterized in that: - said receiving channel (2) also includes a low-pass filter (206, 210) configured to filter said mix signal by extracting its first harmonic component and an analog-to-digital converter (207) to convert the filtered mix signal (SMF) to digital format.

2. A radar device according to claim 1, wherein the shaping means comprise: - a first pulse shaper (101), belonging to said transmission channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and to provide at its output said transmission pulses (IE); and - a second pulse shaper (201), belonging to said reception channel, configured to receive as input said signal generated by said local oscillator at said variable frequency and to provide at its output said reception demodulation pulses (IDR).

3. Radar device according to any one of the preceding claims, comprising a control device (1203, 1206) of said local oscillator configured to restart said local oscillator (1204) at each generation of a transmit pulse and the corresponding receive demodulation pulse.

4. Radar device according to claim 3 when it depends directly on claim 1 in which the shaping means comprise: - said driver device (1206) of said local oscillator (1204), which is configured to shape rising edges of the signal generated by the latter at said variable frequency; and - a falling edge shaping device (105), arranged on said transmission channel, configured to receive at input said signal generated by said local oscillator at said variable frequency and to provide at its output said transmission pulses; said falling edge shaping device (105) also being configured to generate a reset signal of said local oscillator.

5. Radar device according to any one of claims 3 or 4 also comprising a pulse repetition period jammer (1201), configured to control said local oscillator (1204) driver device so as to vary pseudorandomly the interval between two successive restarts of said local oscillator.

6. Radar device according to any one of the preceding claims also comprising a phase scrambler (1025) for applying the same phase scrambling to the transmit pulses (IE) and the receive demodulation pulses (IDR).

7. Radar device according to any one of the preceding claims wherein said low-pass filter comprises a windowed integrator (210), configured to integrate said mix signal over successive integration time windows (IF), each said integration time window being contained within the echo reception time range defined by a said receive demodulation pulse (IDR), and to be reset at the end of said integration window.

8. Radar device according to claim 7 in which said windowed integrator (210) has a time-varying integration gain in a predetermined or adaptive manner.

9. Radar device according to any one of claims 7 or 8 wherein said control signal generator (1202”) is adapted so that the frequency of said local oscillator varies in time by taking a plurality of discrete values ​​obtainable by scrambling a stepwise linear variation; and wherein the receiving channel (2) also includes a frequency scrambling means (211) of the filtered mix signal converted to digital format.

10. Radar device according to any one of the preceding claims in which the receiving channel includes an amplifier (205) equipped with an automatic gain adjustment device to compensate for variations in the intensity of the received echoes.

11. Radar device according to any one of the preceding claims wherein said local oscillator (1204) is a microwave oscillator.

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