Bias circuit of a digital-to-analog converter

Abstract

The invention relates to an electronic circuit comprising: an output node adapted to provide an output current through a load circuit; a plurality of elementary source branches connected in parallel; a bias circuit comprising: a current mirror formed by a first bias branch traversed by a reference current and a second bias branch; the second bias branch comprises: a follower transistor, a first bias cascode transistor and a second bias current copying transistor; a first amplifier circuit configured to copy the electrical potential of the output node onto the drain of the first bias cascode transistor; a second amplifier circuit configured to generate a voltage on the gate of the first bias cascode transistor to regulate the voltage of its source to a predefined setpoint voltage. Figure for abstract: Fig. 2

Description

Title of the invention: Bias circuit of a digital-to-analog converter

[0001] The invention relates to the field of circuits for converting a digital signal into an analog signal, more particularly the design of an analog-to-digital current converter circuit with a high linearity constraint.

[0002] A digital-to-analog converter is an electronic device that converts digital signals into analog signals. It is used when digital data needs to be converted into an analog form, for example for use in audio systems or to control analog devices. The digital-to-analog converter works by converting digital data, usually expressed in binary, into a corresponding analog voltage and / or current.

[0003] Digital-to-analog converters are used in many applications such as audio systems, industrial control systems, wireless communications, and measuring instruments. They are essential for enabling interaction between digital and analog systems by converting digital data into analog signals that can be understood by analog devices.

[0004] More particularly, a current digital-to-analog converter circuit is an electronic device that transforms digital signals into proportional analog currents. Unlike a voltage digital-to-analog converter that generates an analog voltage at the output, a current digital-to-analog converter produces an analog current at the output. Ideally, the output current is directly proportional to the digital input value. This current can then be converted into a voltage across a load resistor.

[0005] The conversion process in a current digital-to-analog converter circuit involves the use of a network of a plurality of elementary current source branches. Each digital input value selects one (or more) source branches constituting a specific path for the current by activating or deactivating corresponding switches. This makes it possible to generate different output currents proportional to the digital input values.

[0006] The design of a current digital-to-analog converter is mainly constrained by the pairing of the transistors, the dependence of the current on the supply voltage or ground, as well as the output impedance of the elementary current source branches that make up the converter. With regard to the pairing and dependence of the power supplies, this can be adjusted by using appropriate dimensions for transistors in elementary current sources.

[0007] However, for an N-bit current digital-to-analog converter, it is necessary to have 2N-1 elementary current source branches. The output impedance of the current digital-to-analog converter depends on the number of activated current sources. During a code ramp going from 0 to 2N-1, the variation of the output impedance of the converter causes a systematic integrated non-linearity error.

[0008] In order to understand the linearity problem in current digital-to-analog converters, [Fig.l] shows the following curves: - The first curve Cl shows the theoretical variation of the output current Is of the converter according to the number of activated elementary sources. The current Is varies between zero and an ideal maximum current value; - The second curve C2 shows the real non-linear variation of the output current Is of the converter according to the number of elementary sources activated. The current Is varies between zero and a real maximum current value; - The third curve C3 presents a theoretical linear model of the converter output current Is according to the number of activated elementary sources. The current Is varies between zero and the said actual maximum current value.

[0009] For a digital-to-analog converter, the actual maximum current value Imax ree imeasured is less than the theoretically calculated ideal maximum current value Imax_ ideai. The non-linearity is maximum for an intermediate number of activated elementary sources, between the minimum number and the maximum number of activated elementary sources. The INL (Acronym for Integrated Non Linearity) is a measurement that evaluates the linearity of a digital-to-analog converter. It is a measure of the largest deviation observed between the second curve C2 and the third curve C3 corresponding to perfectly linear operation between zero and the actual maximum current value Imax reei. More precisely, the INL quantifies the deviations between the actual analog current values ​​produced by the converter and the ideal values ​​of the third curve C3 expected over the entire conversion range.INL is usually expressed in LSB (Least Significant Bit). An INL of 0 LSB means that the DAC produces perfectly linear voltage values ​​without any errors.

[0010] In this sense, several problems have been clearly identified with the current digital-to-analog converters known in the state of the art; and more particularly the problems relating to errors due to the non-linearity of the behavior of the analog-to-digital converter for a large number of activated elementary source branches.

[0011] Furthermore, when designing a digital-to-analog converter circuit, the output impedance must be greater than a critical threshold which depends on the specified INL, the load resistance and the number of bits of the digital-to-analog converter. Typically, for a specified INL of the order of / 2 LSB (Least Significant Bit), the target value of the output impedance becomes difficult to achieve for digital-to-analog converters beyond 8-10 bits.

[0012] More generally, the invention can be applied to converters constrained in linearity, by allowing the reduction of the gap between the theoretical behavior and the real behavior.

[0013] We will begin by introducing the solutions known to those skilled in the art for improving the linearity of an analog-to-digital converter having a plurality of elementary current source branches.

[0014] The first solution according to the state of the art consists of stacking a plurality of cascode transistors in each branch. The cascode transistors are sized so as to increase the output impedance of the converter circuit and thus reduce the amplitude of the non-linearity. The disadvantage of this solution consists not only in a considerable increase in the surface area occupied by the circuit but also in a reduction in the variation range of the output voltage across the load resistance at the output of the converter.

[0015] The second solution according to the state of the art consists of adding a digital correction module to correct the conversion errors due to the non-linearity of the behavior of the digital-to-analog converter in current. The digital correction circuit can use techniques such as gamma correction, linearity correction, interpolation correction to adjust the values ​​of the analog signals produced by the converter. The disadvantage of this solution is that it requires the introduction of additional complexity to the system, which results in a considerable increase in the surface area occupied by the circuit and an increase in its energy consumption.More specifically, the state-of-the-art solution is based on the construction of a Look Up Table which consists of matching the digital code received as input to a modified code which makes it possible to ensure that the measured curve comes as close as possible to the ideal curve. This requires the implementation of a large memory to store this table. This results in a considerable increase in the surface area occupied by the circuit.

[0016] To overcome the limitations of existing solutions, the invention proposes an analog-digital converter circuit in which the bias circuit comprises at least one amplifier circuit allowing the increase of the output impedance seen by the elementary current source branches. Several embodiments of the invention are presented with implementation variants of the proposed solution.

[0017] The subject of the invention is an electronic circuit configured to convert an N-bit digital input signal into an output current, with N a non-zero natural integer, said circuit comprising: - a power supply node adapted to receive a supply voltage, and an electrical ground; - an output node adapted to supply said output current through a target load circuit mounted between on the one hand said output node and on the other hand a first reference node chosen from the electrical ground or the power supply node; - a plurality of elementary source branches, mounted in parallel between the output node and a second reference node distinct from said first reference node and chosen from electrical ground or the power supply node; each elementary source branch comprising: • a bit-driven control transistor having a drain connected to the output node; • an elementary cascode transistor and an elementary current copying transistor mounted in series with the control transistor; - a polarization circuit comprising: • a current mirror formed by a first polarization branch traversed by a reference current and a second polarization branch; the first polarization branch comprises a first generator of said reference current and a first polarization current copying transistor receiving the reference current; the second bias branch comprises in this order: a follower transistor, a first bias cascode transistor and a second bias current copying transistor connected in series; the gate of the second bias current copying transistor and the gate of each elementary current copying transistor being connected to the gate of the first bias current copying transistor; the gate of each elementary cascode transistor being connected to the gate of the first bias cascode transistor; • a first amplifier circuit configured to copy the electrical potential of the output node onto the drain of the first cascode bias transistor; • a second amplifier circuit configured to generate a voltage on the gate of the first bias cascode transistor to regulate the voltage of the source of said first bias cascode transistor to a predefined setpoint voltage.

[0018] According to a particular aspect of the invention, the first amplifier circuit is an operational amplifier having a non-inverting input connected to the output node, an inverting input connected to the drain of the first bias cascode transistor and an output connected to the gate of the follower transistor.

[0019] According to a particular aspect of the invention, the second amplifier circuit is an operational amplifier having a non-inverting input configured to receive a first bias voltage, an inverting input connected to the source of the first bias cascode transistor and an output connected to the gate of the first bias cascode transistor and intended to generate said regulated voltage.

[0020] According to a particular aspect of the invention, the second amplifier circuit comprises a second current generator and a first amplification transistor powered by said second dedicated current generator and having: - a gate connected to the source of the first bias cascode transistor; - and a drain connected to the gate of the first bias cascode transistor.

[0021] According to a particular aspect of the invention, the first polarization branch comprises a second polarization cascode transistor mounted between the first reference current generator and the first polarization current copy transistor.

[0022] According to a particular aspect of the invention, the biasing circuit further comprises a third amplifier circuit having a non-inverting input configured to receive a second bias voltage, an inverting input connected to the source of the second biasing cascode transistor and an output connected to the gate of the second biasing cascode transistor.

[0023] According to a particular aspect of the invention, the second bias voltage corresponds to the first bias voltage.

[0024] According to a particular aspect of the invention, the biasing circuit further comprises a second amplification transistor powered by a third dedicated current generator and having a gate connected to the source of the second cascode transistor. biasing and having a drain connected to the gate of the second biasing cascode transistor.

[0025] The invention also relates to a photonic device with configurable phase comprising: - a waveguide for guiding an input light beam generated by a laser source; - a transmitting or receiving antenna connected to the waveguide; - a resistive element intended to heat the waveguide by Joule effect so as to control the propagation index of the waveguide to configure the phase of the out-of-phase light beam; - an electronic circuit according to the invention configured to inject a control current through the resistive element from a digital control signal to control heating of at least one section of the waveguide; the resistive element corresponding to said target load circuit.

[0026] The invention also relates to a phase-controlled optical network configured to direct a light beam in a predetermined direction comprising: a plurality of configurable phase photonic devices according to the invention powered by a common laser source; each configurable phase photonic device being controlled by a dedicated digital control signal generated by control means.

[0027] Other features and advantages of the present invention will become more apparent upon reading the following description in relation to the following appended drawings.

[0028] [Fig-1] [Fig.l] illustrates curves of the behavior of a converter digital-analog current according to the state of the art. This figure is already described in a previous section.

[0029] [Fig.2] [Fig.2] illustrates an electronic diagram of the digital-to-analog converter according to a first embodiment of the invention.

[0030] [Fig.3] [Fig.3] illustrates an electronic diagram of the digital-to-analog converter according to a second embodiment of the invention.

[0031] [Fig.4] [Fig.4] illustrates an electronic diagram of the digital-to-analog converter according to a third embodiment of the invention.

[0032] [Fig.5a] [Fig.5a] illustrates a first photonic device implementing the converter circuit according to the invention.

[0033] [Fig.5b] [Fig.5b] illustrates a second photonic device implementing the converter circuit according to the invention.

[0034] [Fig.2] illustrates an electronic diagram of the digital-to-analog converter circuit DI according to a first embodiment of the invention. The circuit DI is configured to convert an input digital signal BW = [Bit0 Biti Bit2Bit3... BitN J of N bits of rank i=0 to Nl into an output current Lut-, with N a non-zero natural integer. The circuit DI comprises an output node 11 adapted to provide an analog output signal Vout, a supply node 12 adapted to receive a supply voltage VDD, a plurality of elementary source branches BCj of rank j=0 to 2n- 1 and a biasing circuit 20.

[0035] A target load circuit is connected to the digital-to-analog converter circuit D1. The target load circuit is intended to receive the output current Lut- The target load circuit may be, by way of indicative and non-limiting example, a resistive element or an inductive element or a capacitive element or an electronic circuit to be powered by the output current Lut- In the remainder of the description, the target load circuit is produced by a resistive element RI but the invention remains compatible with any load circuit receiving the output current Lut according to the application context of the invention. A target load circuit formed by at least one inductive element makes it possible, for example, to produce a power converter. A target load circuit formed by at least one capacitive element makes it possible, for example, to produce a sampling circuit.A target load circuit to be powered by the output current Lut is for example an amplifier or a logic circuit.

[0036] In the example illustrated, the resistive element RI is mounted between the supply node 12 and the output node 1. The resistive element RI is intended to be traversed by the output current Lut, the intensity of which depends on the digital input signal BW. The resistive element RI thus makes it possible to convert the output current Lut, a voltage drop between the supply node 12 and the output node 1, and thus into an analog output signal Vout.

[0037] Alternatively, the resistive element RI uses the output current Lut to produce a quantity of heat by Joule effect. This is useful for applications requiring local and adjustable heating around the resistive element RL.

[0038] The digital-to-analog converter circuit D1 comprises 2N-1 elementary source branches BQ connected in parallel between the output node 1 and the electrical ground GND. Each elementary source branch BQ comprises a control transistor Mlj, an elementary cascode transistor M2j and an elementary current copy transistor M3j.

[0039] For each elementary source branch BQ of rank j, the drain of the control transistor Mlj is connected to the output node 11 common to all the source branches; the drain of the elementary cascode transistor M2j is connected to the source of the control transistor Mlj; the drain of the elementary current copy transistor M3j is connected to the source of the elementary cascode transistor M2j; the source of the elementary current copy transistor M3j is connected to the electrical ground GND; the gate of the control transistor Mlj is controlled by a bit of the digital input signal BW and the respective gates of the elementary cascode transistor M2j and the elementary current copy transistor M3j are biased by the bias circuit 20. Thus the output impedance rds of the two transistors M2j, M3j of the elementary source branch BQ is partly determined by the bias circuit 20. The combination of the transistors M2j, M3j of the elementary source branch BQ behaves like a current source having an output impedance rds. The control transistor Mlj plays the role of a current switch in the elementary source branch BQ. When the control transistor Mlj is in the on state, a current path is established from the supply node 12 to electrical ground GND through the load impedance RI and then the elementary source branch BQ associated with said control transistor.Thus, each elementary source branch BQ has the capacity to generate an elementary current Isrc through the load impedance RL. In the case where the sizing of the transistors of the source branches is identical, the elementary currents are identical. In this case, ideally, when the bits of the digital input signal BW activate M source branches, the output current Lut flowing through the load impedance RI is equal to M times the elementary current Isrc. The transistors of each elementary source branch BQ are sized so as to operate in saturation mode for a predetermined maximum output current value Iout, a predetermined supply voltage VDD and a predetermined output impedance value.

[0040] To take into account the weight of the bits in the digital input signal BW = [Bit0 Biti Bit2Bit3 BitN J, the distribution of the bits with respect to the control transistors Mlj is carried out in the following manner: the bit of rank i=0 Bit0 having the lowest weight (as an illustrative and non-limiting example) controls the gate of the control transistor M10 of the elementary source branch BC0; the bit of rank i=1 Biti having the following weight controls the gates of the control transistors Mil and M12 of the 2|H following source branches BQ and BC2; the bit of rank i=2 Bit2 having the following weight controls the gates of the control transistors M13 to M16 of the 21"2 following source branches BC3 to BC6 and so on up to the bit BitN of rank N1 which controls the gates of the control transistors of the last 2N 1 source branches BQ. The sequential order of the distribution has been described to facilitate the understanding of the control distribution but it is not obligatory.Generally speaking, the bit of rank i=0 to Nl controls the gates of the control transistors of 21 source branches BQ chosen from the 2N-1 source branches of the converter circuit DI. Thus, when the bit of rank i Bit, (and therefore of weight i) is in the logic state “1”, the 21 branches. associated sources BC are conducting. This induces the injection of a current associated with said bit Bif; the injected current is proportional to the weight of said bit Bit, and is equal to 21 * Isrc.

[0041] It is also possible to add a digital circuit (not shown) configured to convert the digital input signal BW to convert the most significant bits into thermometric codes to control the gates of the control transistors Mlj which reduces the disadvantages relating to the differential non-linearity (DNL Differential Non Linearity) of the converter.

[0042] The biasing circuit 20 comprises a current mirror formed by a first biasing branch BS1 traversed by a reference current Iref and a second biasing branch BS2 coupled to the first biasing branch BS1. The first biasing branch BS1 comprises a first generator SCI of said reference current Iref, a first biasing current copying transistor M2.

[0043] The second polarization branch BS2 comprises in this order: a follower transistor M3, a first polarization cascode transistor M4 and a second polarization current copy transistor M5 connected in series from the supply node 12 to the electrical ground GND.

[0044] The illustrated specific implementation of the current mirror formed by the polarization branches BS1, BS2 is set out for informational purposes and is not limiting. The invention remains compatible with all current mirror architectures in general.

[0045] Advantageously, and optionally, the first bias branch BS1 comprises a second bias cascode transistor ML. The second bias cascode transistor M1 makes it possible to improve the current copying from the first bias branch BS1 to the second bias branch BS2. In an implementation without a second bias cascode transistor M1, the first bias current copying transistor M2 is mounted as a diode directly connected to the first generator SCI.

[0046] More particularly, the first current generator SCI is connected between the power supply node 12 and the drain of the second bias cascode transistor ML. The drain of the first bias current copy transistor M2 is connected to the source of the second bias cascode transistor ML. The source of the first bias current copy transistor M2 is connected to the electrical ground GND. The gate of the bias current copy transistor M2 is connected to the output of the first current generator SCI. In the first embodiment, the second bias cascode transistor M1 is biased by a reference voltage (also called setpoint voltage) Vcasc>ref coming from an external source not shown.

[0047] The gate of the first bias current copying transistor M2 is connected to the drain of the second bias cascode transistor M1 and to the gate of the second bias current copying transistor M5. The cascode arrangement of the second bias cascode transistor M1 makes it possible to impose a drain voltage on the second bias current copying transistor M2. The drain of the follower transistor M3 is connected to the power supply node 12. The drain of the first bias cascode transistor M4 is connected to the source of the follower transistor M3. The drain of the second bias current copying transistor M5 is connected to the source of the first bias cascode transistor M4. The source of the second bias current copying transistor M5 is connected to the electrical ground GND.By connecting the gates of the first copy transistor M2 and the second copy transistor M5 and having a very close potential on the drains of M2 and M5, we ensure that we copy the reference current Iref (or a multiple of Iref) from the first polarization branch BS1 into the second polarization branch BS2, in particular because we thus limit the Early effect.

[0048] For each elementary source branch BQ, the gate of the elementary cascode transistor M2j is connected to the gate of the first bias cascode transistor M4 of the second bias branch BS2. This makes it possible to couple each elementary source branch BQ to the second bias branch BS2. In addition, for each elementary source branch BQ, the gate of the elementary current copy transistor M3j is connected to the gate of the first bias current copy transistor M2 of the first bias branch BS1. This makes it possible to couple each elementary source branch BQ to the first bias branch BSL. The double coupling of each elementary source branch BQ makes it possible to copy the reference current (or a multiple of the current) to any one of the source branches BQ when the latter is in the on state.

[0049] The bias circuit 20 also comprises a first amplifier circuit RC1 configured to copy the output voltage Vout onto the drain of the first bias cascode transistor M4. This makes it possible to obtain a behavior of the second bias branch BS2 similar to that of the source branches BQ which see the variations of the output voltage Vout during the operation of the digital-to-analog converter circuit DI. Improving the similarity of the voltage variations seen by the second bias branch BS2 and the source branches BQ makes it possible to improve the linearity of the digital-to-analog converter DI. For example, the amplifier circuit RC1 is an operational amplifier having a non-inverting input connected to the output node 11, an inverting input connected to the drain of the first cascode bias transistor M4 and an output connected to the gate of the follower transistor M3.

[0050] The bias circuit 20 further comprises a second amplifier circuit RC2, configured to generate a regulated voltage Vcasc>src on the gate of the first bias cascode transistor M4 from at least the source voltage of said first bias cascode transistor M4. The combination of the second amplifier circuit RC2 with the bias cascode transistor M4 forms a regulated cascode. In addition, the second amplifier circuit RC2 is mounted so as to regulate the gate voltage of the first bias cascode transistor M4 and thus maintain the drain of the second bias current copy transistor M5 at a first predetermined bias voltage Vcasc.

[0051] According to the first embodiment, the second amplifier circuit RC2 is produced by an operational amplifier having a non-inverting input configured to receive a first bias voltage Vcasc, an inverting input connected to the source of the first bias cascode transistor M4 and an output connected to the gate of the first bias cascode transistor M4 and intended to generate the regulated voltage Vcasc>src. This assembly of the second amplifier circuit RC2 makes it possible to increase the output impedance of the converter circuit DI and thus improve the linearity of the output current Iout as a function of the number of activated source branches BQ. Indeed, the use of the second amplifier RC2, of gain A, mounted in “regulated cascode” makes it possible to multiply the output impedance rout by a factor of A+1 compared to a cascode assembly according to the state of the art, without the second amplifier RC2.Advantageously, the output power of the second amplifier circuit RC2 is more than four times that of the first amplifier circuit RC1. This allows the second amplifier circuit RC2 to drive the gates of the 2n- 1 load transistors M2j in addition to the gate of the first bias cascode transistor M4.

[0052] We have described a combination of technical elements formed by: on the one hand, the first amplifier RC1 which copies the output voltage Vout into the second bias branch BS2; and on the other hand the second amplifier circuit RC2 which regulates the gate voltage of the first bias cascode transistor M4. This combination makes it possible to obtain improved linearity of the output current Lut as a function of the number of activated source branches BQ (and therefore as a function of the bits in logic state “1”). This improves the linearity of the output current while avoiding enlarging or adding several cascode stages in the source branches BQ. The solution according to the invention then makes it possible to gain in surface area occupied by the converter circuit compared to state-of-the-art solutions without degrading linearity.

[0053] Advantageously, and optionally, each elementary source branch BCj is not limited to a single elementary cascode transistor M2j and thus comprises a plurality of elementary cascode transistors M2j, M2j', M2j”... Advantageously, and optionally, the second bias branch BS2 is not limited to a single first bias cascode transistor M4 and thus comprises a plurality of bias cascode transistors M4, M4', M4”... The bias of at least one bias cascode transistor among said plurality of bias cascode transistors is regulated by at least one amplifier circuit RC2 mounted as described previously.

[0054] [Fig. 3] illustrates an electronic diagram of the digital-to-analog converter DI according to a second embodiment of the invention. The structural and functional characteristics described for the first embodiment remain valid for the second embodiment. In the illustrated embodiment, the bias circuit 20 further comprises a third amplifier circuit RC3 having a non-inverting input configured to receive a second bias voltage Vcasc, an inverting input connected to the source of the second bias cascode transistor M1 and an output connected to the gate of the second bias cascode transistor ML. The third amplifier circuit RC3 then makes it possible to control the gate of the second bias cascode transistor M1 of the first bias branch BS1 and to maintain the drain of the first bias current copy transistor M2 at the second predetermined bias voltage Vcasc.The combination of the third amplifier circuit RC3 with the second bias cascode transistor M1 forms a regulated cascode circuit. This improves the symmetry and thus the current copying between the first bias branch BS1 and the second bias branch BS2. In the illustrated case, and optionally, the second bias voltage also corresponds to the first bias voltage Vcasc of the second amplifier circuit RC2.

[0055] [Fig.4] illustrates an electronic diagram of the digital-to-analog converter DI according to a third embodiment of the invention. The structural and functional characteristics described for the first and second embodiments remain valid for the third embodiment. The difference with the other embodiments consists of an alternative implementation of the second amplifier circuit and / or the third amplifier circuit while keeping the same technical advantages relating to these two elements.

[0056] In this embodiment, the second amplifier circuit comprises a second current generator IRC2 and a first amplification transistor MRC2 powered by said second dedicated current generator IRC2. The gate of the first amplification transistor MRC2 is connected to the drain of the second bias current copy transistor M5 (and therefore to the source of said first bias cascode transistor M4). The drain of the first amplification transistor MRC2 is connected to the gate of the first bias cascode transistor M4. In this embodiment, the first bias voltage Vcasc corresponds to the gate-source voltage VGs,mrc2 of the first amplification transistor MRC2. The first amplification transistor MRC2 is mounted as a common source amplifier.The choice of the intensity of the current IRC2 and the dimensioning in width and length of the first amplification transistor MRC2 make it possible to generate a regulated voltage Vcasc>src on the gate of the first polarization cascode transistor M4 from the source voltage of said first polarization cascode transistor M4.

[0057] Similarly, in this embodiment, the third amplifier circuit comprises a third current generator IRC3 and a second amplification transistor MRC3 powered by said third dedicated current generator IRC3. The gate of the second amplification transistor MRC2 is connected to the drain of the first bias current copy transistor M2 (and therefore to the source of the second bias cascode transistor M1). The drain of the second amplification transistor MRC3 is connected to the gate of the second bias cascode transistor ML. In this embodiment, the second bias voltage Vcasc corresponds to the gate-source voltage VGs,mrc 3 of the second amplification transistor MRC3. The second amplification transistor MRC3 is mounted as a common source amplifier.The choice of the intensity of the current IRC3 and the dimensioning in width and length of the second amplification transistor MRC3 make it possible to generate a regulated voltage Vcasc>ref on the gate of the second polarization cascode transistor Ml from the voltage of the source of said second polarization cascode transistor ML.

[0058] We emphasize that the implementation choices for producing the second and third amplifier circuits are independent. It is possible to opt for:

[0059] on one side a differential operational amplifier to produce the second amplifier circuit and on the other side a common-source amplification transistor assembly to produce the third amplifier circuit; or vice versa.

[0060] Generally speaking, for all embodiments of the invention, the combination of the first amplifier circuit RC1 and the second amplifier circuit RC2 (or Mrc 2) presents a synergy making it possible to regulate at the same time the biasing of the drain and the source of the first biasing cascode transistor. M4 and thus regulate the polarization of the gate of said first cascode polarization transistor M4. The regulation of the polarization of said gate makes it possible to reduce the gap between the actual behavior and the theoretical behavior of the converter circuit of the digital-to-analog converter D1.

[0061] The invention has been explained with NMOS type transistors as a non-limiting illustrative example. Nothing prevents the same solution from being implemented with PMOS type transistors. Alternatively, the person skilled in the art adapts the architecture proposed and described in Figures 2 to 4 with an implementation based on PMOS transistors and / or a complementary mixed NMOS / PMOS implementation. In the case of an exclusively PMOS implementation, the target load circuit is mounted between the output node 11 and the electrical ground GND; the elementary source branches BCj are mounted in parallel between the output node 11 and the power supply node 12.

[0062] [Fig.5a] illustrates a first DP photonic device implementing the converter circuit according to the invention.

[0063] The invention is applicable in the context of a first phase-configurable photonic device DP. The photonic device DP comprises a waveguide WG, a transmitting antenna GC, a resistive element RI and an electronic circuit D1 digital-to-analog converter according to any one of the embodiments of the invention.

[0064] The WG waveguide is configured to guide an input light beam REMin generated by a SR laser source. The WG waveguide is a microstructure integrated on a substrate made of a semiconductor material. The laser source may be external or integrated on the same semiconductor substrate.

[0065] The transmitting antenna GC is connected to the output of the waveguide WG and configured to transmit a phase-shifted light beam REMout. The transmitting antenna GC is made by a diffraction grating coupler (translation of the term in English Grating Coupler) integrated on the semiconductor substrate.

[0066] The resistive element RI corresponds to the target load circuit mounted at the output node 11 of the electronic circuit D1 digital-to-analog converter. The resistive element RI is intended to heat the waveguide WG by the Joule effect so as to control the propagation index of the waveguide WG to configure the phase of the phase-shifted light beam REMout.

[0067] The local modification of the propagation index makes it possible to modify the phase of the fraction of the signal emitted by the laser source SR which passes through the waveguide WG and which continues to the transmission antenna GC. The resistance of the resistive element RI is between 10Q and 100KQ. The digital-analog converter circuit D1 is configured to control the current flowing through the resistive element RI from a digital control signal BWC. The digital control signal BWC is generated by a control unit CONT. The digital control signal BWC is coded according to the desired heating which depends on the targeted phase shift.

[0068] [Fig.5b] illustrates a second OPA photonic device implementing the converter circuit according to the invention. The second OPA photonic device is an optical phased array. The invention is of particular interest in the context of a photonic device of the optical phased array type. An optical phased array is a photonic device that allows a light beam to be directed in a predetermined direction without using mechanical parts.

[0069] The second photonic device OPA comprises a plurality of phase-configurable DPm photonic devices of rank m=0 to Ml. The phase-configurable DPm photonic devices form a photonic network. Each phase-configurable DPm photonic device receives a dedicated digital control signal BWCm from the control unit CONT. All the waveguides WG of the DPm photonic devices are powered by a common laser source SR. The common laser source SR generates an input light beam REMin which is split into M fractions. Each fraction is guided by the waveguide WG of an associated DPm photonic device. Ideally, the phase-configurable DPm photonic devices are configured so as to create a constant phase shift (modulo 2.pi) for each phase-configurable DPm photonic device of rank m relative to the next phase-configurable DPm photonic device of rank m+1.The output light beam is obtained by the superposition of the different phase-shifted light beams REMOjREMi up toREMM.

[0070] As a non-limiting example, the photonic device D2 comprises an array of 256 DPm photonic devices with configurable phase and thus 256 digital-to-analog converter circuits DI in a 5x5mm2 CMOS circuit. In this context, the solution according to the invention has a particular advantage linked to a reduced surface area of ​​the 256 digital-to-analog converter circuits DI combined with improved linearity. The improved linearity makes it possible to inject a more precise phase shift between the different antennas of the OPA.

[0071] The implementation of the converter circuit according to the invention has been described for a photonic transmission device for informational and non-limiting purposes. The invention remains compatible and also has the same advantages for a photonic device in which the GC antenna is configured for reception. In this case, the GC antennas receive a flow from the scene and it is possible to favor a reception direction by modulating the phases of the antennas.

Claims

1. Claims Electronic circuit (Dl) configured to convert an N-bit digital input signal (BW) into an output current (Iout), with N a non-zero natural integer, said circuit (Dl) comprising: - a power supply node (12) adapted to receive a supply voltage (VDD), and an electrical ground (GND); - an output node (11) adapted to supply said output current ie (IOut) through a target load circuit (RI) mounted between on the one hand said output node (11) and on the other hand a first reference node chosen from the electrical ground (GN D) or the power supply node (12); - a plurality of elementary source branches (BC0,BCi,B CK), mounted in parallel between the output node (11) and a second reference node distinct from said first reference node and chosen from the electrical ground (GND) or the power supply node (12); each elementary source branch (B Ci) comprising: • a control transistor (M10, M11) controlled by a bit and having a drain connected to the output node (11); • an elementary cascode transistor (M20, M21) and an elementary current copy transistor (M30, M31) connected in series with the control transistor (M10, M11); - a polarization circuit (20) comprising: • a current mirror formed by a first polarization branch (BS1) traversed by a reference current (Iref) and a second polarization branch (BS2); the first polarization branch (BS1) comprises a first generator (SCI) of said reference current (Iref) and a first polarization current copy transistor (M2) receiving the reference current (Iref); the second polarization branch (BS2) comprises in this order: a follower transistor (M3), a first cascode polarization transistor (M4) and a second current copying transistor bias transistor (M5) connected in series; the gate of the second bias current copy transistor (M5) and the gate of each elementary current copy transistor (M30, M31) being connected to the gate of the first bias current copy transistor (M2); the gate of each elementary cascode transistor (M20, M21) being connected to the gate of the first bias cascode transistor (M4); • a first amplifier circuit (RC1) configured to copy the electrical potential of the output node (Vout) onto the drain of the first bias cascode transistor (M4); • a second amplifier circuit (RC2, MRC2) configured to generate a voltage (Vcasc>src) on the gate of the first bias cascode transistor (M4) to regulate the source voltage of said first bias cascode transistor (M4) to a predefined setpoint voltage (Vcasc).

2. Electronic circuit (Dl) according to claim 1 wherein the first amplifier circuit (RC1) is an operational amplifier having a non-inverting input connected to the output node (11), an inverting input connected to the drain of the first bias cascode transistor (M4) and an output connected to the gate of the follower transistor (M3).

3. Electronic circuit (Dl) according to any one of claims 1 or 2 wherein the second amplifier circuit (RC2) is an operational amplifier having a non-inverting input configured to receive a first bias voltage (Vcasc), an inverting input connected to the source of the first bias cascode transistor (M4) and an output connected to the gate of the first bias cascode transistor (M4) and intended to generate said regulated voltage (Vcasc>src).

4. Electronic circuit (Dl) according to any one of claims 1 or 2 wherein the second amplifier circuit (RC2) comprises a second current generator (IRC2) and a first amplification transistor (MRC2) powered by said second dedicated current generator (IRC2) and having: - a gate connected to the source of the first polarization casco transistor (M4); - and a drain connected to the gate of the first transistor case polarization mode (M4).

5. Electronic circuit (Dl) according to any one of claims 1 to 4 in which the first polarization branch (BS1) comprises a second polarization cascode transistor (Ml) mounted between the first generator (SCI) of the reference current (Iref) and the first polarization current copy transistor (M2).

6. Electronic circuit (Dl) according to claim 5 wherein the bias circuit (20) further comprises a third amplifier circuit (RC3) having a non-inverting input configured to receive a second bias voltage (Vcasc), an inverting input connected to the source of the second bias cascode transistor (Ml) and an output connected to the gate of the second bias cascode transistor (Ml).

7. Electronic circuit (Dl) according to claim 6 in combination with claim 3 wherein the second bias voltage (Vcasc) corresponds to the first bias voltage (Vcasc).

8. Electronic circuit (Dl) according to claim 5 wherein the bias circuit (20) further comprises a second amplification transistor (MRC3) powered by a third dedicated current generator (IRC3) and having a gate connected to the source of the second bias cascode transistor (Ml) and having a drain connected to the gate of the second bias cascode transistor (Ml).

9. A phase-configurable photonic device (DP) comprising: - a waveguide (WG) for guiding an input light beam (REMin) generated by a laser source (SR); - a transmitting or receiving antenna (GC) connected to the waveguide (WG); - a resistive element (RI) intended to heat the waveguide (WG) by Joule effect so as to control the propagation index of the waveguide (WG) to configure the phase of the out-of-phase light beam (REM); - an electronic circuit (Dl) according to any one of claims 1 to 8 configured to inject a control current (Iout) through the resistive element (RI) from a digital control signal (BWC) to control the heating

10. formation of at least one section of the waveguide (WG); the resistive element (RI) corresponding to said target load circuit. Optical phased array (OPA) configured to direct a light beam (REMtot) in a predetermined direction comprising: a plurality of phase-configurable photonic devices (DPi, DP2, DP3) according to claim 9 powered by a common laser source (SR); each phase-configurable photonic device (DP0, DPi, DP2, DP3) being controlled by a dedicated digital control signal (BWC0, BWCi, BWC2, BWC3) generated by control means (CONT).

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