Differential signal transmission apparatus and method for manufacturing differential signal transmission apparatus

Abstract

To provide a differential signal transmission apparatus with suppressed impedance mismatch and low energy loss.SOLUTION: In a differential signal transmission apparatus in which a first differential transmission line consisting of positive and negative signal lines arranged parallel to each other on a dielectric substrate with a ground plane and a second differential transmission line arranged on the dielectric substrate and having a portion where the spacing between the positive and negative signal lines varies, are connected at the junction point, the values of self-inductance and self-capacitance per unit length of the positive and negative signal lines of the second differential transmission line vary according to the spacing between the positive and negative signal lines.SELECTED DRAWING: Figure 18

Description

[Technical field]

[0001] The present disclosure relates to a differential signal transmission device and a method for manufacturing the same. [Background technology]

[0002] As a signal line for high-speed communication, a differential transmission line in which a positive signal line and a negative signal line are wired in a pair is often used from the viewpoint of stabilizing signal quality and suppressing unnecessary RF radiation. In order to ensure sufficient performance in this differential transmission line, it is necessary to maintain the symmetry of the positive signal line and the negative signal line and at the same time to keep the characteristic impedance of the signal line constant. However, there are some parts, such as wiring parts to connectors, that cannot be physically connected continuously in the same shape. In such a case, it is necessary to make the characteristic impedance the same along the differential transmission line. For example, in Patent Document 1, in a region where the distance between the positive and negative signal lines in a differential transmission line on a printed circuit board is widened, a ground is inserted between the positive and negative signal lines to increase the ground capacitance of the signal line to compensate for the decrease in mutual inductance between the two signal lines, thereby keeping the differential impedance approximately constant.

[0003] Furthermore, Patent Document 2 discloses a method for matching the impedances of both differential transmission lines in each mode when two differential transmission lines having different characteristic impedances in the differential mode and the common mode are connected. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] JP 2004-274005 A [Patent Document 2] JP 2012-44248 A Summary of the Invention [Problem to be solved by the invention]

[0005] In the technology disclosed in Patent Document 1, the mutual inductance becomes nearly zero after the region where the gap between the positive and negative signal lines widens, but the self-inductance of both signal lines is constant in the region before and after the gap because the thickness of the signal lines does not change. An increase in the ground capacitance of both positive and negative signal lines has the effect of reducing both the differential impedance and the common impedance, but a decrease in mutual inductance contributes to an increase in the differential impedance and a decrease in the common impedance, so even if it is possible to somehow match the differential impedance before and after the region where the gap between the positive and negative signal lines widens, there is a problem in that the mismatch in the common impedance becomes large.

[0006] Furthermore, in the technique disclosed in Patent Document 2, the differential transmission line itself is not optimized, and impedance matching is performed using a matching circuit with resistors, which causes a problem of energy loss.

[0007] The present disclosure is devised to solve the above-mentioned problems, and has an object to provide a differential signal transmission device in which impedance mismatch is suppressed and energy loss is reduced. [Means for solving the problem]

[0008] The differential signal transmission device disclosed herein is a differential signal transmission device in which a first differential transmission line composed of a positive signal line and a negative signal line arranged parallel to each other on a dielectric substrate having a ground plane, and a second differential transmission line arranged on the dielectric substrate and having a portion where the distance between the positive signal line and the negative signal line varies, are connected at a connection point, in which the values ​​of self-inductance and self-capacitance per unit length of the positive signal line and the negative signal line of the second differential transmission line vary according to the distance between the positive signal line and the negative signal line.

[0009] Furthermore, in a differential signal transmission device in which a first differential transmission line is formed on one side of a dielectric substrate and has a positive signal line and a negative signal line arranged in parallel, and a second differential transmission line has a positive signal line and a negative signal line which have a portion whose spacing varies and whose respective self-inductance and self-capacitance do not vary with position, and which has a ground plane on at least the other side or an inner layer of the dielectric substrate, in the second differential transmission line, the spacing between the positive signal line and the negative signal line increases with increasing distance from the connection point, and in the second differential transmission line, an intermediate plane made of a non-magnetic metal is arranged between the positive signal line and the negative signal line so that the spacing between the positive signal line and the negative signal line decreases with increasing distance from the connection point, and a conductor removal region where no conductor is formed is provided in a region of the ground plane facing the region in which the intermediate plane is formed.

[0010] Also, in a differential signal transmission device in which a first differential transmission line is formed on one surface of a dielectric substrate, and has a positive signal line and a negative signal line arranged in parallel, and a second differential transmission line has a positive signal line and a negative signal line which have a portion whose spacing varies and whose respective self-inductance and self-capacitance do not vary with position, and which has a ground plane on at least the other surface or an inner layer of the dielectric substrate, in the second differential transmission line, the spacing between the positive signal line and the negative signal line increases with increasing distance from the connection point, and in the second differential transmission line, an intermediate plane of insulating magnetic material is arranged between the positive signal line and the negative signal line so that the spacing between the positive signal line and the negative signal line decreases with increasing distance from the connection point.

[0011] A manufacturing method for a differential signal transmission device disclosed herein includes a first differential transmission line having a positive side signal line and a negative side signal line, and a second differential transmission line having a positive side signal line and a negative side signal line in a physically different arrangement from the first differential transmission line, which are connected at a connection point. The manufacturing method for a differential signal transmission device disclosed herein includes a step of determining circuit parameters of the second differential transmission line, for a differential mode characteristic impedance and a common mode characteristic impedance of the first differential transmission line, by adjusting three circuit parameters other than one of self-inductance, self-capacitance, mutual inductance, and mutual capacitance of the positive side signal line and the negative side signal line of the second differential transmission line, so that the differential mode characteristic impedance of the second differential transmission line becomes the same as the differential mode characteristic impedance of the first differential transmission line, and the common mode characteristic impedance of the second differential transmission line becomes the same as the common mode characteristic impedance of the first differential transmission line. Effect of the Invention

[0012] According to the differential signal transmission device of the present disclosure, it is possible to suppress mismatch in differential mode characteristic impedance and mismatch in common mode characteristic impedance, and a differential signal transmission device with low energy loss can be provided. [Brief description of the drawings]

[0013] [Figure 1] 1A and 1B are perspective views including cross sections showing modeled configurations of transmission lines, where FIG. 1A shows the configuration of a single-ended transmission line, and FIG. 1B shows the configuration of a differential transmission line. [Diagram 2] 1 is a diagram showing an example of the relationship between mutual inductance Lm0, self-capacitance Cs, and mutual capacitance Cm0 per unit length of a differential transmission line. [Diagram 3] 1 is a diagram showing an example of the relationship between mutual inductance Lm0 and self-inductance L0 per unit length of a differential transmission line. [Figure 4]This is a model diagram that expresses the traveling wave and reflected wave between the input port and output port of a differential transmission line using mixed-mode S-parameters. [Diagram 5] This is a model diagram that represents the traveling wave and the reflected wave when two differential transmission lines are continuously connected. [Figure 6] This is a model diagram that expresses the traveling wave and reflected wave between the input port and the output port of a differential transmission line using mixed-mode T parameters. [Figure 7] 11 is a diagram showing a calculation example of the frequency characteristic of the differential mode reflection coefficient SDD11 when differential transmission lines having the same differential mode and the same common mode characteristic impedance are continuously connected in a case where there is no mode conversion component. FIG. [Figure 8] 10 is a diagram showing a calculation example of the frequency characteristic of the differential mode transmission coefficient SDD21 when differential transmission lines having characteristic impedances of the same differential mode and the same common mode are continuously connected in a case where there is no mode conversion component. FIG. [Figure 9] FIG. 13 is a diagram showing an example of calculation of the frequency characteristics of the differential mode reflection coefficient SDD11 when differential transmission lines are continuously connected in which the differential mode characteristic impedances match when there is no mode conversion component, but the common mode characteristic impedances are different. [Figure 10] FIG. 13 is a diagram showing an example of calculation of the frequency characteristics of the differential mode transmission coefficient SDD21 when differential transmission lines are continuously connected in which the differential mode characteristic impedances match when there is no mode conversion component, but the common mode characteristic impedances are different. [Figure 11] 1 is a diagram showing frequency characteristics of each mode conversion component assumed for evaluating the influence when mode conversion occurs in a differential transmission line. [Figure 12] 12 is a diagram showing a calculation example of the frequency characteristic of the differential mode reflection coefficient SDD11 when differential transmission lines having the same differential mode and the same common mode characteristic impedance are continuously connected in a case where the mode conversion component assumed in FIG. 11 is present. FIG. [Figure 13]12 is a diagram showing a calculation example of the frequency characteristic of the differential mode transmission coefficient SDD21 when differential transmission lines having the same differential mode and the same common mode characteristic impedance are continuously connected in a case where the mode conversion component assumed in FIG. 11 is present. FIG. [Figure 14] FIG. 12 is a diagram showing an example of calculation of the frequency characteristics of the differential mode reflection coefficient SDD11 when differential transmission lines are continuously connected in which the differential mode characteristic impedances match when there is a mode conversion component assumed in FIG. 11, but the common mode characteristic impedances are different. [Figure 15] FIG. 12 is a diagram showing an example of calculation of the frequency characteristics of the differential mode transmission coefficient SDD21 when differential transmission lines are continuously connected in which the differential mode characteristic impedances match when there is a mode conversion component assumed in FIG. 11, but the common mode characteristic impedances are different. [Figure 16] 1 is a model diagram for explaining mode conversion in a first differential transmission line. FIG. [Figure 17] 1 is a schematic diagram showing a configuration of a differential signal transmission device in which two differential transmission lines are connected. [Figure 18] 1 is a plan view showing a configuration of a differential signal transmission device according to a first embodiment. [Figure 19] 4 is a plan view showing another configuration of the differential signal transmission device in accordance with the first embodiment. FIG. [Figure 20] 10 is a plan view showing still another configuration of the differential signal transmission device in accordance with the first embodiment. FIG. [Figure 21] FIG. 11 is a plan view showing a configuration of a differential signal transmission device according to a second embodiment. [Figure 22] FIG. 11 is a plan view showing another configuration of the differential signal transmission device according to the second embodiment. [Figure 23] FIG. 11 is a plan view showing a configuration of a differential signal transmission device according to a third embodiment. [Figure 24] FIG. 11 is a plan view showing another configuration of the differential signal transmission device according to the third embodiment. [Diagram 25] FIG. 13 is a diagram showing a table of circuit parameters of each differential transmission line assumed for consideration of a differential signal transmission device according to embodiment 4. [Figure 26] FIG. 13 is a diagram showing an example of the relationship between the mutual capacitance between the positive and negative signal lines at an arbitrary point x of the second differential transmission line, calculated on the assumption that the differential mode characteristic impedances of the first differential transmission line and the second differential transmission line match, or the common mode characteristic impedances match, and the ratio of the mutual inductance in the first differential transmission line to the mutual inductance between the positive and negative signal lines at the arbitrary point x of the second differential transmission line, in a differential signal transmission device according to embodiment 4. [Figure 27] FIG. 27 is a diagram showing an example of a relationship between the common-mode characteristic impedance of each of the second differential transmission lines calculated from both mutual capacitances in FIG. 26, the differential-mode characteristic impedance of the second differential transmission line calculated using the mutual capacitance calculated from the differential-mode characteristic impedance agreement in FIG. 26, and the ratio of the mutual inductance in the first differential transmission line to the mutual inductance between the positive and negative signal lines at an arbitrary point x of the second differential transmission line in a differential signal transmission device according to embodiment 4. [Figure 28] FIG. 28 is a diagram showing an example of the relationship between the common mode return loss and the differential mode return loss calculated from the characteristic impedance of both common modes and the characteristic impedance of the differential mode in FIG. 27, and the ratio of the mutual inductance in the first differential transmission line to the mutual inductance between the positive and negative signal lines at an arbitrary point x of the second differential transmission line in a differential signal transmission device according to embodiment 4. [Figure 29] FIG. 11 is a plan view showing a configuration of a differential signal transmission device according to a fourth embodiment. [Diagram 30] 13 is a perspective view including a cross section showing a configuration of a second differential transmission line of a differential signal transmission device according to a fourth embodiment. FIG. [Diagram 31] FIG. 11 is a plan view showing another configuration of the differential signal transmission device according to the fourth embodiment. [Diagram 32] 13 is a perspective view including a cross section showing a second differential transmission line of another configuration of the differential signal transmission device according to the fourth embodiment. FIG. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0014] The present disclosure discloses a technology for suppressing mismatch by analyzing differential transmission lines in a signal transmission device using differential transmission lines, in which the configuration of the differential transmission line changes during transmission, for example, by having a portion where the distance between the positive signal line and the negative signal line changes, while focusing not only on the differential impedance, i.e., the characteristic impedance of the differential mode, but also on the common impedance, i.e., the characteristic impedance of the common mode.

[0015] <Definition of each characteristic impedance> First, we will model and analyze the differential transmission line. In a single-ended transmission line consisting of a signal line 1, an insulator (dielectric substrate) 3, and a conductive ground plane 4, with the structure shown in Figure 1A, if we assume that the self-inductance per unit length is L0 [H / m], the capacitance of the signal line to the ground plane 4 is C0 [F / m], and the transmission line is lossless, the characteristic impedance Z0 [Ω] is given by the following formula.

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[0016] Next, consider a differential transmission line with the structure shown in Figure 1B, consisting of positive signal line 1a, negative signal line 1b, insulator (dielectric substrate) 3, and conductive ground plane 4. Here too, we assume that the transmission line is lossless. In addition, positive signal line 1a and negative signal line 1b are symmetrical with respect to the center line between positive signal line 1a and negative signal line 1b. Let the self-inductance per unit length of the positive signal line 1a and the negative signal line 1b be L0 [H / m], the same as in Figure 1A; the capacitance (self-capacitance) of both the positive and negative signal lines to the ground plane 4 be Cs [F / m]; the mutual inductance between the positive signal line 1a and the negative signal line 1b be Lm0 [H / m]; the mutual capacitance between the midpoint potential of the positive and negative signal lines and each of the positive and negative signal lines (twice the capacitance between the positive and negative signal lines) be Cm0 [F / m]; the characteristic impedance in the differential mode is Zdif [Ω]; the characteristic impedance in the common mode is Zcom [Ω]; and the characteristic impedance equivalent to single-ended (SE) is Zs [Ω]. Then, Zs [Ω], the characteristic impedance per side of the differential mode transmission line ZD [Ω], and the characteristic impedance per side of the common mode transmission line ZC [Ω] are given by the following equations.

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[0017] Here, assuming Lm0 << L0 and Cm0 << Cs,

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[0018] On the other hand, from equations (1-3) and (1-4),

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[0019] Here, if Cs, Lm0, and Cm0 are expressed using Zs, ZC, ZD, and L0, taking into account equation (1-2),

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[0020] On the other hand, when assuming the transmission of a TEM wave, the propagation velocity v0 must be the same for differential mode transmission, common mode transmission, and single-ended transmission, so

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[0021] Also,

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[0022]

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[0023]

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[0024]

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[0025] The Lm0 dependence of L0, Cs, and Cm0 calculated using equations (7-1), (7-2), and (7-3) is shown in the graphs of Figures 2 and 3. When this relationship is satisfied, ZC, Z0, and ZD can each be maintained at a constant value, but in order to achieve this, for example, when Lm0 decreases (increases), the other three parameters must also decrease (increase). When a parameter other than Lm0 decreases (increases), the remaining three parameters must also decrease (increase).

[0026] <Continuous connection of differential transmission lines> Generally, the traveling waves and reflected waves in a high-frequency circuit consisting of a differential transmission line between ports 1 and 2 are expressed by the mixed-mode S-parameters shown in Figure 4 and the following equation (10), where adx is the differential traveling wave, bdx is the differential reflected wave, acx is the common traveling wave, bcx is the common reflected wave, and x is the port number 1 or 2.

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[0027] When the first differential transmission line 1 and the second differential transmission line 2 shown in Figure 5 are cascaded, let us consider what the transmission characteristics expressed by the mixed-mode S-parameters after cascading will be. With the mixed-mode S-parameters in equation (10) as they are, the input and output (port 1 or port 2) are distributed on the left and right sides, so the characteristics after cascading cannot be obtained by matrix multiplication. To obtain the transmission characteristics by matrix multiplication, it is necessary to define the "mixed-mode T-parameters" in which the forward wave components and reflected wave components of ports 1 and 2 are separated on both sides, as shown in Figure 6 and equation (11), convert to this equation, and obtain each parameter of the matrix of the mixed-mode T-parameters. Here, the suffix r represents the reflected wave and t represents the forward wave. Also, at port 2, which is the connection point between the first differential transmission line 1 and the second differential transmission line 2, the other incident wave becomes its own reflected wave, and the other reflected wave becomes its own incident wave. In other words, bd2 = ad2, ad2 = bd2, bc2 = ac2, ac2 = bc2 are established.

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[0028] The mixed mode T parameters are calculated from equation (10). By rearranging equation (10) by putting port 1 on the left side and port 2 on the right side, we get

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[0029] Where:

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[0030] Using these, equation (10) becomes:

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[0031] By using the mixed-mode T-parameter defined by equation (15), the input / output relationship of the differential transmission line is clarified. If the mixed-mode T-parameter of the first differential transmission line 1 between port 1 and port 2 is T1, and the mixed-mode T-parameter of the second differential transmission line 2 between port 2 and port 3 is T2, then:

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[0032] Next, we consider the conversion from the mixed-mode T-parameters to the mixed-mode S-parameters (Stotal). We consider the conversion from the mixed-mode T-parameters (Ttotal) after the cascade connection of the first differential transmission lines 1 and 2 expressed by equation (12) back to the mixed-mode S-parameters.

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[0033]

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[0034]

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[0035] Using these, equation (15) becomes:

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[0036] In summary, the procedure for obtaining the mixed-mode S-parameters (Stotal) in a two-stage cascaded differential transmission line is as follows: 1: Convert each line's mixed-mode S-parameters to mixed-mode T-parameters 2: Matrix multiplication is performed on the two mixed mode T parameters obtained. (Calculate T = T1 x T2 to calculate the overall mixed mode T parameter) 3: The product of the mixed-mode T-parameters is converted back to mixed-mode S-parameters to obtain the overall mixed-mode S-parameters (conversion of mixed-mode T-parameters to mixed-mode S-parameters is performed). By using the converted Stotal, the transmission characteristics of a two-stage connected differential transmission line can be evaluated.

[0037] <When ignoring mode conversion components> As a special case, when there are no differential-to-common and common-to-differential mode conversion components, equation (10) can be written as follows:

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[0038] Its mixed-mode T parameters can be calculated as follows:

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[0039] Where:

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[0040] Here, assuming that the mixed-mode T parameter of the first differential transmission line 1 is T1a and the mixed-mode T parameter of the second differential transmission line 2 is T2a,

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[0041] Next, consider converting the mixed-mode T-parameters Ttotala after the first differential transmission lines 1 and 2 are continuously connected, expressed by equation (16a), back to the mixed-mode S-parameters Stotala.

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[0042]

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[0043] <Calculation example when mode conversion components can be ignored> Calculation examples of the differential mode reflection coefficient SDD11 and the differential mode transmission coefficient SDD21 when the mode conversion component can be ignored are shown in Fig. 7 to Fig. 10. Fig. 7 and Fig. 8 show SDD11 and SDD21, respectively, when the same transmissions with Zdif = 100.0 [Ω] and Zcom = 27.7 [Ω] are cascade-connected (condition A). Fig. 9 and Fig. 10 show SDD11 and SDD21, respectively, when the first differential transmission line 1 has Zdif1 = 100.0 [Ω] and Zcom1 = 27.7 [Ω], and the second differential transmission line 2 has Zdif2 = 100.0 [Ω] and Zcom2 = 55.4 [Ω], and when the differential transmission lines with the same Zdif but different Zcom are continuously connected (condition B). In each figure, "_1" indicates the first differential transmission line 1, "_2" indicates the second differential transmission line 2, and "_1&2" indicates the attribute of the cascade connection line of the first differential transmission line 1 and the second differential transmission line 2. Comparing the characteristics of Figures 7 and 9, no degradation of the differential mode reflection coefficient SDD11 due to the mismatch in Zcom is observed.

[0044] <Calculation example when mode conversion component cannot be ignored> Next, assuming that the transmission coefficients of the first differential transmission line 1 (only port 1 → port 2 or port 2 → port 1, not including the reflection components of port 1 → port 1 or port 2 → port 2) are as shown in FIG. 11, calculation examples of the differential mode reflection coefficient SDD11 and the differential mode transmission coefficient SDD21 under the above-mentioned (condition A) are shown in FIG. 12 and FIG. 13, and calculation examples of the differential mode reflection coefficient SDD11 and the differential mode transmission coefficient SDD21 under (condition B) are shown in FIG. 14 and FIG. 15. Generally, the higher the frequency, the more prominent the mode conversion, and it is gradually becoming a component that cannot be ignored in the transmission lines for high-speed communication these days. Comparing the graphs of FIG. 12 and FIG. 14, it can be seen that the differential mode reflection coefficient SDD11 increases, i.e., worsens, when Zcom does not match (condition B) at frequencies above 1 GHz compared to when they match (condition A). This can be explained as follows using FIG. 16.

[0045] In Fig. 16, the right-pointing arrows represent traveling waves, and the left-pointing arrows represent reflected waves. What we want to send to the load is a differential traveling wave, and the more differential reflected waves there are, the worse the transmission efficiency becomes, i.e., the larger the SDD11 becomes. The reflected waves that make up the SDD11 are as follows: (a) Differential signal reflected at port 1 of the first differential transmission line 1 (b-1) A differential signal that travels through the first differential transmission line 1 and is reflected (SDD11_2) at port 2 of the second differential transmission line 2 (b-2) A signal that is mode converted from differential to common (SCD21_1) in the first differential transmission line 1, and then converted from common to differential mode (SDC11_2) at port 2 of the second differential transmission line 2 and reflected. (c) After being mode converted from differential to common (SCD21_2) within the first differential transmission line 1, the common mode signal is reflected (SCC11_2) at port 2 of the second differential transmission line 2, and then the common mode signal is converted from common to differential mode (SDC12_1) within the first differential transmission line 1. These are shown in Figure 16. Here, because the common-mode characteristic impedance Zcom differs between the first differential transmission line 1 and the second differential transmission line 2, the reflection of the common-mode signal at port 2 increases, and the effect of (c) above becomes greater. This is thought to result in a deterioration (increase) of SDD11 in the high-frequency range where SCD21_1 and SDC12_1 become large. Even when Zcom is the same (condition A), there is the effect of (b-2) (the effect of SCD21_1), so SDD11 deteriorates (increases) slightly in the high-frequency range, but this is significantly smaller than (condition B).

[0046] As described above, when a mode conversion component exists and there is a mismatch in Zcom between the first differential transmission line 1 and the second differential transmission line 2, SDD11 deteriorates (increases), adversely affecting the transmission of differential mode signals.

[0047] <Embodiment> For signal lines in high-speed communications, differential transmission lines, in which positive and negative signal lines are paired, are often used to stabilize signal quality and suppress unwanted RF radiation. To ensure sufficient performance, however, it is necessary to maintain symmetry between the positive and negative signal lines while also keeping the characteristic impedance of the signal lines constant.

[0048] In actual design, there are constraints of physical layout, for example, in the wiring example of PCB (Print Circuit Board) shown in Fig. 17, even if the positive side signal line 1a and the negative side signal line 1b are laid out close to each other and electrical coupling is ensured between points A and B (first differential transmission line 1), in the area to the right of point B (second differential transmission line 2), the connection point 11a of the positive side signal line 2a and the connection point 11b of the negative side signal line 2b of the connector 10 are separated due to the layout of the connector 10 to be connected, and it may be necessary to wire the positive side signal line 2a and the negative side signal line 2b gradually separated. In the first differential transmission line 1 and the second differential transmission line 2, the positive side signal line and the negative side signal line are also capacitively coupled to the ground layer of the PCB (not shown).

[0049] Consider the case where the first differential transmission line 1 and the second differential transmission line 2 are connected at x = 0 (point B) as shown in Figure 17. In the second differential transmission line 2, the mutual inductance and mutual capacitance gradually decrease as the distance between the positive signal line 2a and the negative signal line 2b increases. On the other hand, if the self-inductance determined by the width, thickness, material, etc. of the positive signal line and the negative signal line pattern, and the self-capacitance determined by the distance between the signal line pattern and the ground layer of the PCB and the dielectric constant of the insulator between them are constant for both the first differential transmission line 1 and the second differential transmission line 2, as is clear from the above-mentioned formulas (1-3) and (1-4), compared with the value of the first differential transmission line 1, ZD increases and ZC decreases in the second differential transmission line 2, so that matching between the first differential transmission line 1 and the second differential transmission line 2 becomes difficult to achieve for both differential signals and common-mode signals. The differential mode characteristic impedance Zdif = 2ZD, and the common mode characteristic impedance Zcom = ZC / 2, so the difference becomes even greater.

[0050] Since the purpose of the differential transmission line is to transmit a differential signal, it goes without saying that the differential mode characteristic impedance ZD of the first differential transmission line 1 and the second differential transmission line 2 must be matched. However, if mode conversion from differential mode to common mode occurs in the first differential transmission line 1, and the common mode characteristic impedance ZC of both differential transmission lines does not match, the common mode signal converted from the differential mode is reflected at the second differential transmission line 2 (point B), and the reflected wave is converted from common mode to differential mode in the first differential transmission line 1, which increases the reflection of the differential signal at the input (point A) of the first differential transmission line 1. In other words, as described above in the case where the mode conversion component cannot be ignored, if mode conversion from differential mode to common mode and from common mode to differential mode occurs in the first differential transmission line 1, the mismatch in the common mode characteristic impedance of the positive signal line and the negative signal line increases (deteriorates) the reflection coefficient SDD11 of the differential mode at point A. Normally, it is impossible to make the layout of the positive signal line and the negative signal line completely symmetrical, so mode conversion always occurs.

[0051] As a result of the above analysis, it was found that the challenge is to match not only the differential mode characteristic impedance but also the common mode characteristic impedance as closely as possible when connecting differential transmission lines. Below, we will disclose various configurations of differential transmission lines that will minimize the reflection coefficient of the differential mode by feeding back the results of the analysis to the actual configuration of the differential transmission line.

[0052] Embodiment 1 The first embodiment is an embodiment using a microstrip line (MSL). FIG. 18 shows a configuration example using a microstrip line having a ground plane 4 facing the signal line. FIG. 18a shows a signal line surface layer in which a signal line with a constant thickness is formed, and FIG. 18b shows a ground layer in which the ground plane 4 is formed on the inner layer or back side. If it is necessary to separately wire the positive signal line 2a and the negative signal line 2b to the right of point B, that is, in the second differential transmission line 2, the greater the separation distance, the smaller the mutual inductance Lm and mutual capacitance Cm will be. At this time, as discussed in the above <Definition of each characteristic impedance>, in order to match the characteristic impedance of the differential mode with the characteristic impedance of the common mode, it is necessary to also reduce the other two parameters, the self-inductance L0 of the signal line and the electrostatic capacitance Cs of the signal line to ground. First, by tapering the signal line so that it becomes thicker (wider) as the distance between the positive signal line 2a and the negative signal line 2b increases, L0 can be gradually reduced. When the signal line becomes thicker (the area per unit length increases), the Cs of the signal line increases, but it is necessary to reduce Cs by more than the increase. To achieve this, Cs can be reduced by removing the ground plane (conductor) 4 directly below the thickened signal line (forming conductor removal areas 40a, 40b). Even if the ground plane (conductor) directly below is removed, the signal line will be coupled with the surrounding ground plane, so Cs will not become 0 and can be calculated by numerical analysis, etc. Specifically, L0(x) at displacement x = x is calculated from the pattern width that can be realized by the PCB pattern design, and the value and the differential mode characteristic impedance ZD [Ω] and common mode characteristic impedance ZC [Ω] of the first differential transmission line 1 can be substituted into equations (6-1a) to (6-3a) to obtain other parameters Cs(x), Lm0(x), and Cm0(x) at x = x. Of course, calculations can also be performed using equations (7-1) to (7-3). Circuit parameters such as the distance between the positive signal line 2a and the negative signal line 2b and the shapes of the conductor removed regions 40a and 40b may be set so as to obtain the obtained Cs(x), Lm0(x), and Cm0(x).

[0053]

number

[0054] The second differential transmission line 2 of the differential signal transmission device designed according to this design is ultimately configured as in formula (5-3), that is, in the second differential transmission line 2, the ratio Lm0 / L0 of the mutual inductance Lm0 per unit length between the positive signal line 2a and the negative signal line 2b to the self-inductance L0 per unit length of each of the positive signal line 2a and the negative signal line 2b, and the ratio Cm0 / Cs of the mutual capacitance Cm0 per unit length between the positive signal line 2a and the negative signal line 2b to the self-capacitance Cs per unit length of each of the positive signal line 2a and the negative signal line 2b are constant regardless of the distance between the positive signal line 2a and the negative signal line 2b. This configuration is the same not only in this first embodiment, but also in the differential signal transmission devices described in the following second and third embodiments.

[0055] FIG. 19 shows another example of a configuration using microstrip lines. FIG. 19a shows the signal line surface layer of a dielectric substrate on which a signal line of a constant thickness is formed, and FIG. 19b shows a ground layer on which a ground plane 4 is formed on the inner layer or back side. In FIG. 19, the positive signal line 2a and the negative signal line 2b of the second differential transmission line 2 are tapered to make them thinner, as opposed to FIG. 18, to increase the self-inductance L0. In this case, the other three parameters need to be increased, so an intermediate plane 5 made of a magnetic metal (iron, nickel, etc.) is provided between the two signal lines, and the intermediate plane 5 is brought asymptotically close to the positive signal line 2a and the negative signal line 2b (the value of d(x) is reduced as it moves away from point B), thereby gradually increasing the mutual capacitance Cm and mutual inductance Lm between the signal lines. If a non-magnetic metal (copper, aluminum, etc.) is used for the intermediate plane, Lm cannot be increased, and the impedance correction effect is reduced. If the intermediate plane 5 is connected to the ground, it will not contribute to an increase in Cm, but will contribute to an increase in Cs. Therefore, the ground layer (conductor) in the same area opposite the intermediate plane 5 is removed (forming the conductor removed area 40) so that the intermediate plane 5 only contributes to an increase in Cm. Furthermore, in the area of ​​the second differential transmission line 2 to the right of point B, side ground planes 6a and 6b are provided on the sides of the positive signal line 2a and the negative signal line 2b, respectively, and are connected to the respective signal lines to increase Cs. The distance between the side ground plane 6a and the positive signal line 2a, and the distance between the side ground plane 6b and the negative signal line 2b are made smaller as they move away from the connection point b. The side ground planes 6a and 6b on the opposite side to the intermediate plane 5 are firmly connected to the ground plane 4 by vias 60, and are electrically low impedance zero potential surfaces. The calculation of each circuit parameter is performed by the same method as described in FIG. 18, and by adjusting the distance between the intermediate plane 5 and the side ground planes 6a and 6b.

[0056] FIG. 20 shows another example of the configuration using microstrip lines. In the configuration of FIG. 20, a new inner layer shown in FIG. 20b is provided between the signal line surface layer shown in FIG. 20a and the ground layer shown in FIG. 21c. In the configuration shown in FIG. 19, if the increase in Cs is insufficient for the increase in L0, a new inner layer shown in FIG. 20b is provided, and no ground plane is provided between points A and B (X<0), and a ground plane 4c is provided only in the area to the right of point B (X>0), which is the area of ​​the second differential transmission line, thereby making Cs larger. The reason for not providing a plane between points A and B is to avoid an effect on the characteristic impedance of the first differential transmission line 1. The ground plane 4c is firmly connected to the ground plane 4 in FIG. 20c and the side ground planes 6a and 6b of the signal line surface layer by vias 60, and is electrically connected to zero potential with low impedance, and the conductor immediately below the intermediate plane 5 is removed (forming a conductor-removed area 40c).

[0057] Embodiment 2 The second embodiment is an embodiment using a stripline (SL). FIG. 21 shows an example of a configuration using a stripline. The layer in which a signal line with a constant thickness is arranged is the inner layer (signal line inner layer 2) shown in FIG. 21b, and the ground layer 1 on which the ground plane 4a shown in FIG. 21a is formed and the ground layer 2 on which the ground plane 4b shown in FIG. 21c is formed are arranged above and below the signal line inner layer 2. The widths of the positive side signal line 2a and the negative side signal line 2b in the second differential transmission line are made wider as they move away from the connection point B, and conductor-removed regions 40a, 40b in which the conductors are removed are formed in the regions of the ground plane 4a and the ground plane 4b facing each signal line. The ground planes 4a and 4b sandwiching the signal line inner layer 2 are electrically connected firmly by, for example, vias 60, and are at the same potential.

[0058] Another example of the configuration using striplines is shown in Fig. 22. The signal lines have a constant thickness and are the inner layer (signal line inner layer 2) shown in Fig. 22b. The signal line inner layer 2 has a ground layer 1 with a ground plane 4a formed thereon shown in Fig. 22a and a ground layer 2 with a ground plane 4b formed thereon shown in Fig. 22c. The widths of the positive signal line 2a and the negative signal line 2b in the second differential transmission line are narrowed with increasing distance from the connection point B, and an intermediate plane 5 made of magnetic metal is placed between the positive signal line 2a and the negative signal line 2b. Furthermore, a side ground plane 6a and a side ground plane 6b are placed on the opposite sides of the intermediate plane 5 of the positive signal line 2a and the negative signal line 2b, respectively. 19, the distances between the positive signal line 2a and the negative signal line 2b and the intermediate plane 5, the distance between the positive signal line 2a and the side ground plane 6a, and the distance between the negative signal line 2b and the side ground plane 6b decrease with increasing distance from connection point B. The ground planes 4a and 4b sandwiching the signal line inner layer 2, and the side ground planes 6a and 6b in the signal line inner layer 2 are firmly connected electrically by vias 60 or the like. In addition, the conductors in the areas of the ground planes 4a and 4b facing the intermediate plane 5 are removed (forming conductor-removed areas 40).

[0059] When the self-capacitance Cs of the signal line is insufficient in the widened gap region between the positive signal line 2a and the negative signal line 2b, the value of Cs can be increased by inserting a layer between the ground layer 1 and the signal line inner layer 2, and between the signal line inner layer 2 and the ground layer 2, as shown in Fig. 20b, in which there is no ground plane (conductor) between points A and B (x < 0), and a ground plane 4c in which a conductor-removed region 40c in which the conductor in the opposing region of the intermediate plane 5 is removed is formed only in the portion to the right of point B (x > 0), which is the region of the second differential transmission line. The reason for not providing a plane between points A and B is to avoid an effect on the characteristic impedance of the first differential transmission line 1.

[0060] Embodiment 3 The third embodiment is an embodiment using a coplanar line (CPW). FIG. 23 shows an example of a configuration using a coplanar line. FIG. 23a shows a surface layer (signal line surface layer) on which a signal line with a constant thickness is formed, and FIG. 23b shows a ground layer on which a ground plane 4 is formed on an inner layer or back surface. In the case of a coplanar line, the electrical coupling between the signal line on the signal line surface layer and the ground plane is weak, and side ground planes 61a, 62a, 61b, and 62b are formed on the signal line surface layer, and the positive signal line 2a and the negative signal line 2b are strongly coupled to the side ground planes adjacent thereto. The side ground planes 61a, 62a, 61b, and 62b adjacent thereto in the same layer as the positive signal lines 1a and 2a and the negative signal lines 1b and 2b are connected to the ground plane 4 via vias, respectively, and are set to zero potential with low electrical impedance. In the region of the second differential transmission line 2 to the right of the cascade connection point B, the self-inductance L0 is reduced by increasing the width of the signal line as it moves away from point B, but in this region, the distance between the signal line and the adjacent ground plane in the same layer (the distance between the positive signal line 2a and the side ground plane 62a, and the distance between the negative signal line 2b and the side ground plane 62b) is gradually increased to reduce the self-capacitance Cs. The mutual inductance Lm and the mutual capacitance Cm are gradually reduced in the region to the right of point B. As described in the first embodiment, L0(x) at displacement x = x is calculated, and the value, the differential mode characteristic impedance ZD [Ω], and the common mode characteristic impedance ZC [Ω] of the first differential transmission line 1 are substituted into equations (6-1a) to (6-3a) to obtain other parameters Cs(x), Lm0(x), and Cm0(x) at x = x. Of course, the calculations may be performed using equations (7-1) to (7-3). The distance between the positive signal line 2a and the side ground plane 62a, and the distance between the negative signal line 2b and the side ground plane 62b are determined so that Cs(x), Lm0(x), and Cm0(x) take the calculated values.

[0061] FIG. 24 shows another example of the configuration using coplanar lines. FIG. 24a shows a surface layer (signal line surface layer) on which a signal line with a constant thickness is formed, and FIG. 24b shows a ground layer on which a ground plane 4 is formed on the inner layer or back side. As with the configuration of the microstrip line shown in FIG. 19, the self-inductance L0 is increased by tapering the positive signal line 2a and the negative signal line 2b of the second differential transmission line 2. The idea of ​​providing an intermediate plane 5 made of a magnetic metal between the positive signal line 2a and the negative signal line 2b of the second differential transmission line 2, and decreasing the distance between the positive signal line 2a and the negative signal line 2b and the intermediate plane 5 and the distance between the side ground plane 62a and the side ground plane 62b as they move away from the continuous connection point B, and the calculation of each circuit parameter are the same as those explained in the case of FIG. 19.

[0062] Embodiment 4 The fourth embodiment is an embodiment of a configuration in which the self-inductance L0 and the self-capacitance Cs of the second differential transmission line are constant and the same values ​​as the self-inductance L0 and the self-capacitance Cs of the first differential transmission line. As described above, in order to match the characteristic impedances ZD and ZC of the first differential transmission line and the second differential transmission line, in the region in which these impedances may vary, it is ideal to obtain the circuit parameters L0(x), Cs(x), Lm0(x), and Cm0(x) at x=x using equations (6-1a) to (6-3a) and change them according to the displacement x, but this is considered to be physically difficult in many cases. In this case, in the second differential transmission line 2, L0 and Cs are constant, that is, the line width, thickness, and distance to ground of the positive side signal line and the negative side signal line are constant, and the two parameters Lm0(x) and Cm0(x) are complementarily changed, so that the characteristic impedance of the differential mode and the characteristic impedance of the common mode in the cascade connection line can be matched to some extent. The reason is that, for example, when Lm0 decreases in equations (1-3) and (1-4), the coefficient sign is opposite, so by increasing Cm0, it is possible to move in the direction that alleviates the fluctuations in ZD and ZC.

[0063] In the positive signal lines 1a, 2a and negative signal lines 2a, 2b of the microstrip lines shown in Figure 29 described later, the circuit parameters of the first differential transmission line 1 and the second differential transmission line 2 are as shown in the table shown in Figure 25, and the first differential transmission line 1 and the second differential transmission line 2 are connected at x = 0 (point B in Figure 29).

[0064] When considering matching the differential mode characteristic impedance and the common mode characteristic impedance between the first differential transmission line 1 and the second differential transmission line 2, from equations (1-3) and (1-4),

number

[0065] From equation (8-1), solving for Cm(x) and Lm(x), let these be CmZd(x) and LmZd(x), and

number

[0066] From equation (8-2), solving for Cm(x) and Lm(x), let these be CmZc(x) and LmZc(x), and

number

[0067] Here, in the second differential transmission line 2, an example of CmZd(x) and CmZc(x) calculated as a function of Lm0 / L0 from equations (8-3a) and (8-4a) is shown in the graph in Figure 26, which shows that the two are in a negative proportional relationship. Lm(x) / L0 assumes that the mutual inductance gradually decreases as the positive signal line 2a and the negative signal line 2b move away from each other. Also, #1: The differential mode characteristic impedance ZD2 (#1) of the single line of the second differential transmission line 2 calculated from CmZd(x) and L0, Cs, and Lm(x) #2: Common mode characteristic impedance ZC2(#2) of the single line of the second differential transmission line 2 calculated from CmZc(x) and L0, Cs, and Lm(x) #3: Common mode characteristic impedance ZC2 (#3) of the single line of the second differential transmission line 2 calculated from CmZd(x) and L0, Cs, and Lm(x) #4: The differential mode characteristic impedance ZD1 (#4) of the single line of the first differential transmission line 1 #5: Common mode characteristic impedance ZC1 (#5) of the single line of the first differential transmission line 1 is also shown in the graph of Figure 27. From this, since L0 and Cs are fixed values, it can be seen that as Lm0 / L0 decreases, both ZD2 and ZC2 become more deviated from ZD1, but even if ZC2, which should be calculated using CmZc(x), is calculated (#3) using CmZd(x), the degree of deviation from ZD1 is larger than that of ZC2 calculated (#2) using CmZc(x), but it can be seen that there is not much difference. In addition, the differential mode and common mode return loss of the connection point (point B) of the first differential transmission line 1 and the second differential transmission line 2 obtained from #1, #2, and #3 are shown in the graph of Figure 28. Because there is not much difference between the common mode return loss (#2) at ZC2 calculated from CmZc(x) and the common mode return loss (#3) at ZC2 calculated using CmZd(x), it can be said that even if the common mode is not taken into account and CmZd(x) is used for the purpose of matching the differential mode characteristic impedance, the impact on the common mode return loss is small. Using this idea, we will consider specific methods of realizing this below.

[0068] <Use non-magnetic metal as intermediate plane> Examples of configurations based on microstrip lines are shown in Figures 29 and 30. Figure 29a shows the surface layer on which the signal lines are formed (signal line surface layer), Figure 29b shows the ground layer on which the ground plane 4 is formed on the inner layer or back side, and Figure 30 shows an enlarged perspective view including a cross section perpendicular to the board surface at the position of x=x of the second differential transmission line 2. In order to compensate for the effect on ZD and ZC caused by the decrease in mutual inductance Lm(x) in the region of the second differential transmission line 2 by increasing the mutual capacitance Cm(x), a non-magnetic metal intermediate plane 5 is laid out between the positive signal line 2a and the negative signal line 2b. Without the non-magnetic metal intermediate plane 5, Lm(x) would decrease and Cm(x) would also decrease as the x coordinate moves to the right. When a non-magnetic metal intermediate plane 5 is provided, the distance d(x) between the intermediate plane 5 and the positive signal line 2a and the negative signal line 2b is decreased as x increases so that the increase in Cm(x) can compensate for the decrease in Lm(x) due to electrostatic coupling between the intermediate plane 5 and each of the positive signal line 2a and the negative signal line 2b, and Cm(x) is gradually increased based on formula (8-3a). In addition, in order to keep the self-capacitance Cs at a constant value, a conductor-removed region 40 is formed in the ground plane 4 by removing the conductor, so that only Cm(x) increases. With this method, it is impossible to completely match ZD(x) and ZC(x) with those of the first differential transmission line 1, but a certain improvement effect can be expected.

[0069] <Uses magnetic insulator as intermediate plane> Similarly, examples of configurations based on microstrip lines are shown in Fig. 31 and Fig. 32. Fig. 31a shows the surface layer (signal line surface layer) on which the signal lines are formed, Fig. 31b shows the ground layer formed on the inner layer or back side, and Fig. 32 shows an enlarged perspective view including a cross section perpendicular to the board surface at the position of x=x of the second differential transmission line 2. In order to compensate for the effect on ZD and ZC caused by the decrease in mutual capacitance Cm(x) in the region of the second differential transmission line 2 by increasing the mutual inductance Lm(x), an intermediate plane 51 made of a magnetic insulator is laid out between the positive signal line 2a and the negative signal line 2b. Without the intermediate plane 51 made of a magnetic insulator, Cm(x) decreases and Lm(x) also decreases as the x coordinate moves to the right. A magnetic insulating intermediate plane 51 is provided, and the distance d(x) between the intermediate plane 51 and the positive signal line 2a and the negative signal line 2b is decreased as x increases so that the increase in Lm(x) can complement the decrease in Cm(x) due to the magnetic coupling between the intermediate plane 51 and the positive signal line 2a and the negative signal line 2b, respectively, and Lm(x) is gradually increased based on the formula (8-3b). However, since there is also electrostatic coupling due to the existence of the intermediate plane 51, the decrease in Cm due to the increase in the distance between the positive signal line 2a and the negative signal line 2b is mitigated to some extent. Also, in order to keep the self-inductance L0 at a constant value, if the conductor of the ground layer is magnetic, it is necessary to remove it so that only Lm(x) increases, but in a normal PCB, the conductor of the ground layer is copper, i.e., a non-magnetic material, so there is no need to remove it. With this method, it is impossible to completely match ZD(x) and ZC(x) with those of the first differential transmission line 1, but a certain improvement effect can be expected.

[0070] The configurations shown in Figs. 29 to 32 show examples in which they are applied to a microstrip line, but they can also be applied to a stripline or a coplanar line.

[0071] To summarize the above embodiments 1 to 4, a technology is disclosed in which, in a second differential transmission line having a portion where the distance between the positive signal line 2a and the negative signal line 2b varies, by adjusting at least three parameters other than the circuit parameters of the self-inductance, self-capacitance, mutual inductance, and mutual capacitance of at least the positive signal line 2a and the negative signal line 2b, the differential mode characteristic impedance of the second differential transmission line becomes the same or approximately the same as the differential mode characteristic impedance of the first differential transmission line, and the common mode characteristic impedance of the second differential transmission line becomes the same or approximately the same as the common mode characteristic impedance of the first differential transmission line.

[0072] In the first to third embodiments, in a second differential transmission line having a portion where the distance between a positive signal line 2a and a negative signal line 2b changes, the self-inductance and self-capacitance of at least the positive signal line 2a and the negative signal line 2b are changed according to the distance, thereby compensating for the effects of the mutual inductance and mutual capacitance that change according to the change in distance, or, in addition to the change in distance, an intermediate plane made of magnetic metal between both signal lines, thereby making the differential mode characteristic impedance of the second differential transmission line the same as the differential mode characteristic impedance of the first differential transmission line and making the common mode characteristic impedance of the second differential transmission line the same as the common mode characteristic impedance of the first differential transmission line.

[0073] In addition, in embodiment 4, in a second differential transmission line having a portion where the distance between the positive signal line 2a and the negative signal line 2b changes, the self-inductance and self-capacitance of the positive signal line 2a and the negative signal line 2b do not change depending on the position, but the mutual inductance or mutual capacitance between the positive signal line 2a and the negative signal line 2b is compensated for by mutual capacitance or mutual inductance, thereby disclosing a technology in which the differential mode characteristic impedance of the second differential transmission line is made approximately the same as the differential mode characteristic impedance of the first differential transmission line and the common mode characteristic impedance of the second differential transmission line is made approximately the same as the common mode characteristic impedance of the first differential transmission line.

[0074] By using the differential transmission line configuration and calculation method described above and setting each circuit parameter (self-inductance, self-capacitance, mutual inductance, and mutual capacitance of the positive and negative signal lines), the following effects can be expected. When two differential transmission lines are connected in series, Since it is possible to match or nearly match not only the differential mode characteristic impedance but also the common mode characteristic impedance, when mode conversion occurs, it is possible to suppress reflection of the differential mode signal at the input terminal due to common mode reflection, i.e., transmission loss. Using the mixed-mode T-parameters, the mixed-mode S-parameters of the cascaded line can be calculated from each mixed-mode S-parameter.

[0075] Although various exemplary embodiments and examples are described in this disclosure, various features, aspects, and functions described in one or more embodiments are not limited to application to a specific embodiment, but can be applied to the embodiments alone or in various combinations. Therefore, countless modifications not illustrated are expected within the scope of the technology disclosed in this specification. For example, the following are included: when at least one component is modified, added, or omitted, and further, when at least one component is extracted and combined with a component of another embodiment. [Explanation of symbols]

[0076] 1 first differential transmission line, 1a (first differential transmission line) positive signal line, 1b (first differential transmission line) negative signal line, 2 second differential transmission line, 2a (second differential transmission line) positive signal line, 2b (second differential transmission line) negative signal line, 3 dielectric substrate, 4, 4a, 4b ground plane, 40, 40a, 40b, 40c conductor removal area, 5, 51 middle plane, 6a, 6b, 61a, 61b, 62a, 62b side ground plane

Claims

1. A differential signal transmission device in which a first differential transmission line configured with a positive side signal line and a negative side signal line arranged parallel to each other on a dielectric substrate having a ground plane, and a second differential transmission line arranged on the dielectric substrate and having a portion where the distance between the positive side signal line and the negative side signal line changes are connected at a connection point, A differential signal transmission device in which the values ​​of self-inductance and self-capacitance per unit length of the positive signal line and the negative signal line of the second differential transmission line change according to the distance between the positive signal line and the negative signal line.

2. 2. The differential signal transmission device according to claim 1, wherein in the second differential transmission line, a distance between the positive signal line and the negative signal line increases with increasing distance from the connection point, and widths of the positive signal line and the negative signal line increase with increasing distance from the connection point.

3. 3. The differential signal transmission device according to claim 2, wherein the first differential transmission line and the second differential transmission line are configured by microstrip lines having the positive side signal line and the negative side signal line, respectively, formed on one surface of the dielectric substrate, and a first ground plane of a conductor formed on the other surface or an inner layer of the dielectric substrate, and a conductor removed region where no conductor is formed is provided in a region of the first ground plane facing a region in which the positive side signal line and the negative side signal line of the second differential transmission line are formed.

4. 3. The differential signal transmission device according to claim 2, wherein the first differential transmission line and the second differential transmission line are configured as striplines in which the positive signal line and the negative signal line are formed in a signal line inner layer sandwiched between a first dielectric substrate and a second dielectric substrate, a first ground plane made of a conductor is provided on a surface or inner layer of the first dielectric substrate opposite to the signal line inner layer, and a second ground plane made of a conductor is provided on a surface or inner layer of the second dielectric substrate opposite to the signal line inner layer, and a conductor removed region where no conductor is provided is provided in regions of the first ground plane and the second ground plane facing a region in which the positive signal line and the negative signal line of the second differential transmission line are formed.

5. 3. The differential signal transmission device according to claim 2, wherein the first differential transmission line and the second differential transmission line are coplanar lines having the positive signal line and the negative signal line formed on one surface of the dielectric substrate, a conductor side ground plane formed on the one surface of the dielectric substrate and arranged along each of the positive signal line and the negative signal line, and a conductor first ground plane formed on the other surface or on an inner layer of the dielectric substrate, and a distance between the positive signal line and the negative signal line in the second differential transmission line and the side ground plane increases with increasing distance from the connection point.

6. 2. The differential signal transmission device according to claim 1, wherein in the second differential transmission line, a distance between the positive signal line and the negative signal line increases with increasing distance from the connection point, and widths of the positive signal line and the negative signal line decrease with increasing distance from the connection point.

7. in the second differential transmission line, an intermediate plane made of a magnetic metal is disposed between the positive signal line and the negative signal line such that a distance between the positive signal line and the negative signal line decreases with increasing distance from the connection point; 7. The differential signal transmission device according to claim 6, wherein a side ground plane is arranged along the opposite side of the positive signal line and the negative signal line from the intermediate plane such that the spacing between the positive signal line and the negative signal line decreases with increasing distance from the connection point.

8. 8. The differential signal transmission device according to claim 7, wherein the first differential transmission line and the second differential transmission line are configured by microstrip lines having the positive side signal line and the negative side signal line, respectively, formed on one surface of the dielectric substrate, and a first ground plane of a conductor formed on the other surface or an inner layer of the dielectric substrate, and a conductor removal region in which no conductor is formed is provided in a region of the first ground plane facing a region in which the intermediate plane is disposed.

9. 9. The differential signal transmission device according to claim 8, wherein an inner layer ground plane of a conductor is formed in a region between one surface of the dielectric substrate and a layer on which the first ground plane is formed and corresponding to a region of the second differential transmission line, and a conductor removal region in which no conductor is formed is provided in a region of the inner layer ground plane facing the intermediate plane.

10. 8. The differential signal transmission device according to claim 7, wherein the first differential transmission line and the second differential transmission line are configured as striplines, in which the positive side signal line and the negative side signal line are formed in a signal line inner layer sandwiched between a first dielectric substrate and a second dielectric substrate, and have a first ground plane made of a conductor on a surface or inner layer of the first dielectric substrate opposite to the signal line inner layer, and a second ground plane made of a conductor on a surface or inner layer of the second dielectric substrate opposite to the signal line inner layer, and have a conductor removed region where no conductor is formed in regions of the first ground plane and the second ground plane facing a region where the intermediate plane is disposed.

11. 11. The differential signal transmission device according to claim 10, wherein a conductor inner layer ground plane is formed in an area between the signal line inner layer and the first ground plane, and between the signal line inner layer and the second ground plane, the area corresponding to the area of ​​the second differential transmission line, and the area of ​​each of the inner layer ground planes facing the intermediate plane has a conductor removal area where no conductor is formed.

12. 7. The differential signal transmission device according to claim 6, wherein the first differential transmission line and the second differential transmission line are configured as coplanar lines having the positive signal line and the negative signal line formed on one side of the dielectric substrate, a conductor side ground plane formed on the one side of the dielectric substrate and arranged along each of the positive signal line and the negative signal line, and a conductor first ground plane formed on the other side or in an inner layer of the dielectric substrate, wherein a distance between each of the side ground planes and the positive signal line and the negative signal line in the second differential transmission line decreases with increasing distance from the connection point, and an intermediate plane made of a magnetic metal is arranged between the positive signal line and the negative signal line in the second differential transmission line so that the distance between the positive signal line and the negative signal line decreases with increasing distance from the connection point.

13. 13. The differential signal transmission device according to claim 1, wherein a ratio of the self-inductance per unit length of the positive signal line and the negative signal line to the mutual inductance per unit length between the positive signal line and the negative signal line of the second differential transmission line, and a ratio of the self-capacitance per unit length of the positive signal line and the negative signal line to the mutual capacitance per unit length between the positive signal line and the negative signal line are constant regardless of the spacing between the positive signal line and the negative signal line.

14. A differential signal transmission device in which a first differential transmission line is formed on one surface of a dielectric substrate, the first differential transmission line having a positive signal line and a negative signal line arranged in parallel, and a second differential transmission line having a positive signal line and a negative signal line having a portion whose interval varies and whose respective self-inductance and self-capacitance do not vary with position, are connected at a connection point, and the device has a ground plane on at least the other surface or an inner layer of the dielectric substrate, In the second differential transmission line, a distance between the positive signal line and the negative signal line increases with increasing distance from the connection point, in the second differential transmission line, an intermediate plane made of a non-magnetic metal is disposed between the positive signal line and the negative signal line such that a distance between the positive signal line and the negative signal line decreases with increasing distance from the connection point; A differential signal transmission device having a conductor removed area where no conductors are formed in an area of ​​the ground plane opposite to an area where the intermediate plane is formed.

15. A differential signal transmission device in which a first differential transmission line is formed on one surface of a dielectric substrate, the first differential transmission line having a positive signal line and a negative signal line arranged in parallel, and a second differential transmission line having a positive signal line and a negative signal line having a portion whose interval varies and whose respective self-inductance and self-capacitance do not vary with position, are connected at a connection point, and the device has a ground plane on at least the other surface or an inner layer of the dielectric substrate, In the second differential transmission line, a distance between the positive signal line and the negative signal line increases with increasing distance from the connection point, A differential signal transmission device in which an intermediate plane of insulating magnetic material is arranged between the positive signal line and the negative signal line in the second differential transmission line such that the distance between the positive signal line and the negative signal line decreases with increasing distance from the connection point.

16. 1. A method for manufacturing a differential signal transmission device in which a first differential transmission line having a positive side signal line and a negative side signal line and a second differential transmission line having a positive side signal line and a negative side signal line that are arranged physically differently from the first differential transmission line are connected at a connection point, A method for manufacturing a differential signal transmission device, comprising a step of determining circuit parameters of the second differential transmission line, by adjusting three circuit parameters other than one of the self-inductance, self-capacitance, mutual inductance, and mutual capacitance of the positive signal line and the negative signal line of the second differential transmission line, with respect to the differential mode characteristic impedance and common mode characteristic impedance of the first differential transmission line, so that the differential mode characteristic impedance of the second differential transmission line becomes the same as the differential mode characteristic impedance of the first differential transmission line, and the common mode characteristic impedance of the second differential transmission line becomes the same as the common mode characteristic impedance of the first differential transmission line.

17. A method for manufacturing a differential signal transmission device according to any one of claims 1 to 12, comprising the steps of: a relationship between a self-inductance L0 [H / m], a self-capacitance Cs [F / m], a mutual inductance Lm [H / m], and a mutual capacitance Cm [F / m] per unit length at each position relative to the connection point of the positive signal line and the negative signal line in the second differential transmission line, and a differential mode characteristic impedance ZD [Ω] and a common mode characteristic impedance ZC [Ω] of the first differential transmission line; [0010] A method for manufacturing a differential signal transmission device, comprising the step of determining circuit parameters of the second differential transmission line based on the

18. 16. A method for manufacturing a differential signal transmission device according to claim 14 or 15, wherein, when a self-inductance L0 [H / m] and a self-capacitance Cs [F / m] per unit length in the first differential transmission line and the second differential transmission line are constant, and a mutual inductance per unit length in the first differential transmission line is Lm0 [H / m] and a mutual capacitance per unit length in the first differential transmission line is Cm0 [F / m], the mutual inductance Lm [H / m] and the mutual capacitance Cm [F / m] per unit length at each position relative to the connection point of the positive side signal line and the negative side signal line in the second differential transmission line are [0025] A method for manufacturing a differential signal transmission device, comprising steps determined based on the above.

19. A method for manufacturing a differential signal transmission device according to any one of claims 1 to 12, 14, and 15, comprising the steps of: The input point of the first differential transmission line is defined as port 1, the connection point as port 2, and the output point of the second differential transmission line as port 3. b is a reflected wave, a is a traveling wave, d and D are differential modes, c and C are common modes, the first digit of the subscript of the reflected wave b and the traveling wave a is the transmission mode, the second digit is the corresponding port number, the first digit of the mixed mode S parameter factor is the output transmission mode, the second digit is the input transmission mode, the third digit is the output port number, and the fourth digit is the input port number. Using a mixed mode S parameter S1 from port 1 to port 2 and a mixed mode S parameter S2 from port 2 to port 3, the transmission from port 1 to port 2 and from port 2 to port 3 are [0030] Expressed as, A mixed-mode T parameter T1 between the port 1 and the port 2 and a mixed-mode T parameter T2 between the port 2 and the port 3 are [0045] [0050] [006] [0070] By defining the above, the mixed mode T parameter Ttotal between the port 1 and the port 3 is the product of T1 and T2, and the following subscripts are defined as r for reflected wave, t for forward wave, the first digit of the mixed mode T parameter factor below is the output transmission mode, the second digit is the input transmission mode, the third digit is the distinction between reflected wave and forward wave in the output wave, and the fourth digit is the distinction between reflected wave and forward wave in the input wave, [0080] From this equation, the mixed mode S-parameter from port 1 to port 3, Stotal, is expressed as follows: [0097] [0089] and evaluating the transmission characteristics from the port 1 to the port 3 using the Stotal obtained by the conversion. A method for manufacturing a differential signal transmission device having the above structure.

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