Ac motor control device, ac motor driving device and electrically driven apparatus using the same

Abstract

To provide an AC motor control device, an AC motor driving device and an electrically driven apparatus using them, which can accurately reproduce a motor current fundamental wave component.SOLUTION: An AC motor control device comprises: a PWM signal generation unit (8) that generates a control signal which is composed of a PWM pulse signal and is for controlling an electric power conversion device by controlling an electric power conversion device (2) outputting AC power to an AC motor (1) and by comparing between an AC voltage command and a carrier wave; a current detection unit (20) that detects a DC bus current at predetermined detection timing; and a current detection timing calculation unit (17a) that calculates detection timing. A motor current at a time point in which a carrier wave arrives at a mountain or a valley is reproduced at timing calculated by the current detection timing calculation unit, and the electric power conversion device is controlled on the basis of the reproduced motor current.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to an AC motor control device that controls an AC motor, an AC motor drive device that is equipped with this AC motor control device and drives an AC motor, and electric equipment using this AC motor drive device. [Background technology]

[0002] In various fields, such as general industry, home appliances, automobiles, and railways, AC motors are becoming faster and more compact in order to produce higher output.

[0003] For high-speed rotation control of an AC motor, synchronous PWM control (see, for example, Patent Document 1) is used, which synchronizes the phase relationship between the carrier wave and the voltage command and changes the PWM carrier frequency and the number of pulses for each electrical angle frequency. Synchronous PWM control suppresses the generation of harmonics.

[0004] As AC motors become smaller, position sensorless technology that estimates the magnetic pole position of an AC motor without using a sensor is being applied in order to further reduce the size of devices that use AC motors.As a position sensorless technology, a conventional technology that estimates the magnetic pole position based on a calculated value of the axis error (Δθc) between the real axis (dq axis) and the control axis (dc-qc axis) of the AC motor is known (for example, see Patent Document 2).

[0005] Furthermore, to detect the motor current, a technique is used that reproduces the three-phase motor current based on the shunt current flowing through the DC bus of the inverter without using a phase current sensor (see, for example, Patent Document 3 (FIGS. 3 and 4)). [Prior art documents] [Patent documents]

[0006] [Patent Document 1] JP 2005-237194 A [Patent Document 2] JP 2015-33310 A [Patent Document 3] International Publication No. 2015 / 025622 Summary of the Invention [Problem to be solved by the invention]

[0007] In synchronous PWM control, the current ripple of the motor current increases in the high-speed range of the AC motor. In particular, when the inductance of the AC motor decreases as the AC motor becomes smaller, the current ripple becomes more pronounced. This causes a large error between the motor current reproduced based on the shunt current and the fundamental wave component.

[0008] Such current detection errors reduce the accuracy of current control of the AC motor. In addition, in the position sensorless control based on the above-mentioned axis error, the axis error is calculated using the dq-axis current values ​​(see "Equation (3)" in Patent Document 2), so the estimation accuracy of the magnetic pole position is reduced.

[0009] Therefore, the present invention provides an AC motor control device and an AC motor drive device that can reproduce the fundamental wave component of the motor current with high accuracy, and an electrically-driven device using the same. [Means for solving the problem]

[0010] In order to solve the above problems, an AC motor control device according to the present invention controls a power conversion device that outputs AC power to an AC motor, and includes a PWM signal generating unit that generates a control signal for controlling the power conversion device, which is composed of a PWM pulse signal, by comparing an AC voltage command with a carrier wave, a current detecting unit that detects a DC bus current flowing on the DC side of the power conversion device at a predetermined detection timing, and a current detection timing calculating unit that calculates the detection timing. Furthermore, the AC motor control device according to the present invention reproduces the motor current at the time when the carrier wave becomes a peak or a valley based on the DC bus current detected by the current detecting unit at the detection timing calculated by the current detection timing calculating unit, and controls the power conversion device based on the reproduced motor current.

[0011] In order to solve the above problems, an AC motor drive device according to the present invention includes a power conversion device that outputs AC power to an AC motor, and a control device that controls the power conversion device, and the control device is the AC motor control device according to the present invention.

[0012] In order to solve the above problems, an electric device according to the present invention comprises a mechanical load driven by an AC motor, a power conversion device that outputs AC power to the AC motor, and a control device that controls the power conversion device, wherein the control device is the AC motor control device according to the present invention. Effect of the Invention

[0013] According to the present invention, the fundamental wave component of the motor current can be reproduced with high accuracy.

[0014] Problems, configurations and effects other than those described above will become apparent from the following description of the embodiments. [Brief description of the drawings]

[0015] [Figure 1] 1 is a functional block diagram showing a configuration of an AC motor driving device according to a first embodiment. [Diagram 2] FIG. 2 is a coordinate plane diagram showing the relationship between coordinate axes and axis error (Δθ) in a rotating coordinate system. [Diagram 3] 4 is a graph showing an example of the relationship between a rotation frequency command value fr* and a carrier frequency fs in synchronous PWM control. [Figure 4] FIG. 2 is a waveform diagram showing an example of a shunt current Is and motor currents Iu, Iv, and Iw. [Diagram 5] 2 is a circuit diagram of a main circuit of the power conversion device 2 (FIG. 1), showing an example of a flow state of a shunt current Is. [Figure 6] FIG. 4 is a waveform diagram showing an example of a motor current reproduced from a shunt current Is. [Figure 7] FIG. 11 is a waveform diagram showing another example of a motor current reproduced from the shunt current Is. [Figure 8]5 is a waveform diagram similar to FIG. 4, showing an example of the shunt current Is and the motor currents Iu, Iv, and Iw. [Figure 9] 4 is a waveform diagram showing an example of a shunt current Is and a motor current, illustrating a calculation means for detecting the timing of detecting the shunt current Is in the first embodiment. FIG. [Figure 10] FIG. 11 is a functional block diagram showing a configuration of an AC motor driving device according to a second embodiment. [Figure 11] 13 is a waveform diagram showing an example of the shunt current Is and the motor current, illustrating a calculation means for detecting the timing of the shunt current Is in the second embodiment. FIG. [Figure 12] FIG. 11 is a functional block diagram showing a configuration of an AC motor driving device according to a third embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0016] Hereinafter, the embodiments of the present invention will be described with reference to the following Examples 1 to 3 with reference to the drawings.

[0017] In the various figures, the same reference numbers indicate the same components or components with similar functions. EXAMPLES

[0018] FIG. 1 is a functional block diagram showing a configuration of an AC motor driving device according to a first embodiment of the present invention.

[0019] The AC motor driving device according to the first embodiment is composed of a power conversion device 2 having a three-phase inverter circuit and applying a three-phase AC voltage to an AC motor 1, a DC power supply 18 providing DC power to the power conversion device 2, a shunt resistor 19 on the DC side of the power conversion device 2 for detecting a DC bus current flowing between the DC power supply 18 and the power conversion device 2, and a control device 3 for controlling the speed of the AC motor 1 by controlling the switching of the power conversion device 2 based on the detected value of the DC bus current.

[0020] In the first embodiment, a permanent magnet synchronous motor is applied as the AC motor 1. The AC motor 1 drives a mechanical load 21 such as a blower or a compressor.

[0021] The control device 3 generates control signals Sup, Sun, Svp, Svn, Swp, and Swn for controlling switching of the power conversion device 2 by vector control based on the detection value of the DC bus current detected by the shunt resistor 19. The control signals Sup, Sun, Svp, Svn, Swp, and Swn respectively control on / off switching of semiconductor switching elements (IGBTs in FIG. 5) in the U-phase upper arm, U-phase lower arm, V-phase upper arm, V-phase lower arm, W-phase upper arm, and W-phase lower arm in the three-phase inverter circuit (see FIG. 5). As a result, the power conversion device 2 converts the DC power input from the DC power source 18 into three-phase AC power and outputs a three-phase AC voltage to the AC motor 1.

[0022] In vector control, coordinate conversion is performed between a rotating coordinate system and a fixed coordinate system according to the rotational position of the AC motor 1. In the first embodiment, the rotational position of the AC motor 1 is estimated by position sensorless control as described below without using a rotational position sensor (e.g., an encoder) or a magnetic pole position sensor (e.g., a Hall element).

[0023] In the first embodiment, the motor current used in the vector control and the position sensorless control is detected based on the DC bus current by a so-called one-shunt resistor method without using a phase current sensor (for example, a CT).

[0024] As described later, in the first embodiment, the timing at which the DC bus current flows, which is equal to the motor current at the timing of the peak or valley of the carrier wave, is calculated, and the DC bus current is detected at the calculated timing. As is well known, the motor current at the timing of the peak or valley of the carrier wave is less affected by the current ripple. Therefore, the detected value of the DC bus current is also less affected by the current ripple. Therefore, the fundamental wave component of the motor current of each phase can be detected from the DC bus current. This fundamental wave component is used in vector control and position sensorless control. This improves the accuracy and stability of the control of the AC motor 1.

[0025] In the first embodiment, the control device 3 includes a computer system such as a microcomputer, and the computer system executes a predetermined program to perform vector control, position sensorless control, and motor current detection based on the DC bus current.

[0026] The operation of the control device 3 regarding the vector control will be described below with reference to FIG.

[0027] The three-phase (U, V, and W) motor currents Iu, Iv, and Iw (output of current processing unit 9) reproduced from the DC bus current are converted by coordinate conversion unit 10 into d-axis current Idc and q-axis current Iqc in a rotating coordinate system whose coordinate axes are the dc-qc axes set within the control device 3.

[0028] The current command generating unit 5 generates the q-axis current command Iq* and the d-axis current command Id* from the q-axis current Iqc without using a speed controller (ASR) or a PI controller. In the first embodiment, the current command generating unit 5 generates Iq* from Iqc via a first-order lag element, and changes Id* to a predetermined value according to the magnitude of Iqc.

[0029] The rotational speed command generating unit 4 generates a rotational speed command ωr* for the AC motor 1. For example, the rotational speed command generating unit 4 outputs a preset rotational speed value as ωr*, or outputs a rotational speed value selected from a plurality of preset rotational speed values ​​as ωr*. The rotational speed command generating unit 4 may generate ωr* based on a preset rotational speed pattern.

[0030] The vector control unit 6 generates a d-axis voltage command Vdc* and a q-axis voltage command Vqc* using a predetermined voltage equation based on the rotation speed command ωr*, the q-axis current command Iq*, and the d-axis current command Id*. Note that the voltage equation is expressed using the motor constants (d-axis inductance, q-axis inductance, winding resistance, and induced voltage constant) of the AC motor 1, but since this is well known, detailed description will be omitted.

[0031] Coordinate conversion unit 7 converts the d-axis voltage command Vdc* and the q-axis voltage command Vqc* in a rotating coordinate system whose coordinate axes are the dc and qc axes into three-phase (U, V and W phases) AC voltage commands Vu*, Vv*, Vw* on fixed axes.

[0032] Here, the position sensorless control in the first embodiment will be described.

[0033] In the first embodiment, the control device 3 executes position sensorless control by the axis error estimator 11 and the PLL controller 12 shown in FIG.

[0034] FIG. 2 is a coordinate plane diagram showing the relationship between the coordinate axes and the axis error (Δθ) in a rotating coordinate system.

[0035] As shown in FIG. 2, the axis error Δθ is defined as the phase shift between the dc-qc axes, which are the coordinate axes for control, and the dq axes, which are the actual coordinate axes set for the rotor of the AC motor 1.

[0036] In the first embodiment, the direction of the main magnetic flux of the permanent magnet included in the rotor of the AC motor 1 is set to the d-axis.

[0037] In addition, "ω1" in FIG. 2 represents the rotation speed of the rotor of the AC motor 1 (an estimated value in the first embodiment).

[0038] The axis error estimator 11 (FIG. 1) estimates the axis error Δθ by calculating the right-hand side of equation (1) based on the rotational speed ω1 (estimated value), the d-axis voltage command Vdc*, the q-axis voltage command Vqc*, the d-axis current Idc, and the q-axis current Iqc.

[0039]

number

[0040] In equation (1), R and Lq are the winding resistance and the q-axis inductance of AC motor 1, respectively.

[0041] It is well known that in a permanent magnet motor such as the AC motor 1, the axis error Δθ is expressed by equation (1). Therefore, further detailed explanation of equation (1) will be omitted.

[0042] The PLL control unit 12 (FIG. 1) uses PLL (Phase Locked Loop) control to estimate a rotation speed ω1 that makes the axis error Δθ zero, based on the axis error Δθ estimated by the axis error estimating unit 11.

[0043] The integrator 13 (FIG. 1) calculates the rotational position θdc of the rotor of the AC motor 1 by integrating the rotational speed ω1 calculated by the PLL controller 12. The rotational position θdc is used for coordinate conversion in the coordinate converters 7 and 10.

[0044] As described above, the control device 3 converts the d-axis voltage command Vdc* and the q-axis voltage command Vqc* generated by vector control into three-phase AC voltage commands Vu*, Vv*, and Vw* by the coordinate conversion unit 7. Furthermore, as described next, the control device 3 generates control signals Sup, Sun, Svp, Svn, Swp, and Swn in accordance with Vu*, Vv*, and Vw*.

[0045] The PWM signal generating unit 8 (FIG. 1) generates control signals Sup, Sun, Svp, Svn, Swp, and Swn consisting of PWM pulse signals based on three-phase (U, V, and W phase) AC voltage commands Vu*, Vv*, and Vw*. In the first embodiment, the PWM signal generating unit 8 generates Sup, Sun, Svp, Svn, Swp, and Swn consisting of PWM pulse signals by comparing a carrier wave Sc (e.g., a triangular wave) generated by the carrier generating unit 15 with Vu*, Vv*, and Vw*.

[0046] In the first embodiment, in the high speed range of the AC motor 1, the PWM signal generating unit 8 generates the control signals Sup, Sun, Svp, Svn, Swp, and Swn by synchronous PWM control.

[0047] In synchronous PWM control, the period of the carrier wave is an integer multiple of the period of the sine wave voltage command, and the phases of the carrier wave and the voltage command are synchronized. In synchronous PWM control, when the output frequency of the power conversion device 2 is changed, the carrier frequency is also changed according to the output frequency. In this case, the number of pulses of the PWM pulse signal per half period of the voltage command is constant. Furthermore, in synchronous PWM control, the number of pulses may be switched according to the range of the output frequency.

[0048] FIG. 3 is a graph showing an example of the relationship between the rotation frequency command value fr* (=ωr* / 2π) and the carrier frequency fs in synchronous PWM control that switches the number of pulses.

[0049] In this example, in the low-speed range, a PWM pulse signal is generated by asynchronous PWM control. In the medium-high speed range, the control switches to synchronous PWM control, and the number of pulses is reduced as the frequency range becomes higher. fs is set to 3n times fr* (n: natural number). This suppresses the generation of harmonics.

[0050] As shown in FIG. 1, in the first embodiment, the synchronous PWM control unit 14 calculates a carrier period Ts in synchronous PWM control based on the d-axis voltage command Vdc*, the q-axis voltage command Vqc*, the rotational speed command ωr*, and the rotational position θdc, and instructs the carrier generating unit 15 to generate a carrier wave Sc so that the carrier period is Ts and the phases of the carrier wave and the voltage command are synchronized.

[0051] The carrier generating unit 15 generates an interrupt signal Trg at the timing when the carrier wave Sc reaches a maximum value (peak) and a minimum value (valley). The control device 3 executes control processes related to vector control, position sensorless control, synchronous PWM control, and detection of the motor current according to Trg.

[0052] In the first embodiment, the number of pulses in the synchronous PWM control is set to three.

[0053] Next, detection of the motor current by the one-shunt resistor method in the first embodiment will be described with reference to FIGS.

[0054] As shown in FIG. 1, in order to detect the motor current of the AC motor 1, the control device 3 includes a current detection unit 20, a current processing unit 9, a current gradient information calculation unit 16, and a current detection timing calculation unit 17a.

[0055] The current detection unit 20 samples the voltage drop due to the shunt resistor 19, calculates the DC bus current in the power conversion device 2 based on the sampled voltage drop, and outputs the calculated value as the shunt current Is. The timing (Tad) at which the current detection unit 20 samples the voltage drop due to the shunt resistor 19 is set by the current detection timing calculation unit 17a.

[0056] The current processing unit 9 reproduces the three-phase motor currents Iu, Iv, Iw from the shunt current Is output by the current detection unit 20 using the three-phase AC voltage commands Vu*, Vv*, Vw* and the control signals Sup, Sun, Svp, Svn, Swp, Swn.

[0057] The current processing unit 9 outputs three-phase motor currents Iu, Iv, Iw close to the fundamental wave components by the current gradient information calculation unit 16 and the current detection timing calculation unit 17a. In the first embodiment, such three-phase motor currents Iu, Iv, Iw are converted into d-axis current Idc and q-axis current Iqc by the coordinate conversion unit 10 (FIG. 1), and these Idc and Iqc are used for vector control and position sensorless control. This improves the accuracy and stability of the vector control and position sensorless control.

[0058] FIG. 4 is a waveform diagram showing an example of the shunt current Is and the motor currents Iu, Iv, and Iw. In addition, in FIG. 4, the waveforms of the three-phase AC voltage commands Vu*, Vv*, Vw* and carrier wave Sc, as well as the waveforms of the control signals Sup, Svp, Swp are also shown.

[0059] The detection timing shown in Fig. 4 shows an example of the timing at which the current detection unit 20 (Fig. 1) detects the shunt current Is flowing through the shunt resistor 19 (the detection timing in the first embodiment will be described later (Fig. 9)). This detection timing corresponds to, for example, the timing at which an A / D conversion function of the microcomputer constituting the control device 3 is activated. In addition, this detection timing is the timing before and after the timing at which the control signal of the phase that is the middle phase among the three-phase AC voltage commands Vu*, Vv*, Vw* changes.

[0060] The shunt current Is flowing through the shunt resistor 19 at the timing before and after the timing when the control signal for the phase serving as the intermediate phase changes will be described with reference to FIG.

[0061] FIG. 5 is a circuit diagram of a main circuit of the power conversion device 2 (FIG. 1), showing an example of a flow state of the shunt current Is.

[0062] The intermediate phase is the V phase. That is, this is the case where "Vu*>Vv*>Vw*" in FIG.

[0063] The semiconductor switching elements (IGBTs in the first embodiment) constituting the three-phase inverter circuit, which is the main circuit, are turned on and off by control signals Sup, Sun, Svp, Svn, Swp, and Swn. In each phase, the semiconductor switching elements of the upper arm and the semiconductor switching elements of the lower arm are turned on and off in a complementary manner.

[0064] In mode 1, the semiconductor switching elements in the U-phase upper arm, the V-phase upper arm, and the W-phase upper arm are in the ON state, and the other semiconductor switching elements are in the OFF state. In Fig. 4, this corresponds to the case where Sup, Svp, and Swp are all "1", i.e., ON control signals. In such mode 1, no shunt current Is flows through the shunt resistor 19 (Is = 0).

[0065] In mode 2, the semiconductor switching elements in the U-phase upper arm, the V-phase upper arm, and the W-phase lower arm are in the ON state, and the other semiconductor switching elements are in the OFF state. In Fig. 4, this corresponds to the case where Sup and Svp are "1", i.e., ON control signals, and Swp is "0", i.e., an OFF control signal. In such mode 2, a shunt current Is equal to the W-phase motor current flows through shunt resistor 19 (Is = -Iw: the direction toward AC motor 1 is defined as the positive direction).

[0066] In mode 3, the semiconductor switching elements in the U-phase upper arm, the V-phase lower arm, and the W-phase lower arm are in the ON state, and the other semiconductor switching elements are in the OFF state. In Fig. 4, this corresponds to the case where Sup is "1", i.e., an ON control signal, and Svp and Swp are "0", i.e., OFF control signals. In such mode 3, a shunt current Is equal to the U-phase motor current flows through shunt resistor 19 (Is = Iu).

[0067] In mode 4, the semiconductor switching elements in the U-phase lower arm, the V-phase lower arm, and the W-phase lower arm are in the ON state, and the other semiconductor switching elements are in the OFF state. In Fig. 4, this corresponds to the case where Sup, Svp, and Swp are all "0", that is, OFF control signals. In such mode 1, no shunt current Is flows through shunt resistor 19 (Is = 0).

[0068] The Is in the above-mentioned modes 2 and 3 corresponds to the Is before and after the timing when the control signal Svp changes from 1 (ON) to 0 (OFF) during the period when the V phase is an intermediate phase in FIG. 4. Therefore, if Is is detected at such timing, -Iw and Iu can be detected. At this time, Iv can be calculated from the three-phase current relational equation "Iu+Iv+Iw=0". In other words, Iu, Iv, and Iw can be reproduced from Is.

[0069] As described above, when detecting the shunt current Is at the timing before and after the timing when the control signal of the middle phase among the three-phase AC voltage commands Vu*, Vv*, Vw* changes, in order to ensure the simultaneity of detection of the two-phase motor currents detected based on Is, the detection timing of the shunt current Is is set near the peak or bottom of Is as shown in Fig. 4. Therefore, as shown in Fig. 4, the detected motor currents (Iu, Iv, Iw) are set near the peak or bottom of the current ripple in the motor current. This is a value near the bottom.

[0070] Therefore, the motor currents Iu, Iv, and Iw reproduced from the shunt current Is contain detection errors for the fundamental wave components of the motor currents. In particular, in a low-inductance motor, which is advantageous for high speed operation, the current ripple is large and the detection error becomes large.

[0071] FIG. 6 is a waveform diagram showing an example of a motor current reproduced from the shunt current Is.

[0072] In Fig. 6, Iu(n) is the U-phase motor current reproduced from the shunt current Is (hereinafter referred to as the "reproduced current"). Note that Fig. 6 also shows the actual U-phase motor current (hereinafter referred to as the "actual current") Iu and the fundamental component of Iu (hereinafter referred to as the "fundamental component") Iuo. Note that Fig. 6 shows one cycle of waveforms for each of Iu(n), Iu, and Iuo.

[0073] In Fig. 6, the reproduced current Iu(n) has a stepped waveform. This is because the control device 3 executes control processing in discrete time using a microcomputer. The plotted points in the waveform of Iu(n) indicate the time points at which the motor current is reproduced from the shunt current Is.

[0074] As each plot point indicates, the reproduced current Iu(n) represents the actual current Iu including the current ripple, but has a detection error with respect to the fundamental wave component Iuo. Therefore, if the d-axis current Idc and q-axis current Iqc converted from the reproduced current Iu(n) are used, the accuracy and stability of vector control and sensorless control may decrease.

[0075] FIG. 7 is a waveform diagram showing another example of the motor current reproduced from the shunt current Is.

[0076] In FIG. 7, Iu(n)' is the reproduced current on the assumption that the current value near the center of the current ripple of Iu can be sampled from Is.

[0077] As can be seen by comparing Figure 7 and Figure 6, Iu(n)' and the fundamental component I u0 The difference Δ' between Iu(n) and the fundamental component I u0 is smaller than the difference Δ between

[0078] FIG. 8 is a waveform diagram similar to FIG. 4, showing an example of the shunt current Is and the motor currents Iu, Iv, and Iw.

[0079] FIG. 8 shows detection timings when sampling current values ​​near the center of the ripple in the motor current.

[0080] As shown in Fig. 8, the detection timings of the motor current are the timings of the peaks (maximum values) and valleys (minimum values) of the carrier wave Sc. The current values ​​of the motor current detected at these detection timings are less affected by the ripples in the motor current and are close to the fundamental wave component.

[0081] At the timing of the peaks, the control signals Sup, Svp, and Swp are all 0, i.e., OFF. In this case, no shunt current Is flows (Is=0), as in mode 4 shown in Figure 5. At the timing of the valleys, the control signals Sup, Svp, and Swp are all 1, i.e., ON. In this case, no shunt current Is flows (Is=0), as in mode 1 shown in Figure 5.

[0082] In this way, the shunt current Is does not flow at the timings of the peaks (maximum values) and valleys (minimum values) of the carrier wave Sc. In Fig. 8, the shunt current Is flows intermittently, and Is = 0 in the time width set by the voltage command and the carrier wave, centered on the timings of the peaks and valleys. Therefore, even if the shunt current Is is sampled at the timings of the peaks and valleys, it is difficult to detect the fundamental wave component of the motor current.

[0083] Therefore, in the first embodiment, during the period when the shunt current Is flows (Is≠0), the timing at which the current value of the shunt current Is becomes the same as the current value of the motor current at the timing of the peak and valley of the carrier wave Sc is calculated. Based on the shunt current Is detected at the calculated timing, the fundamental wave component of the motor current is detected.

[0084] FIG. 9 is a waveform diagram showing an example of the shunt current Is and the motor current, which indicates a calculation means for detecting the timing of the shunt current Is in the first embodiment.

[0085] In FIG. 9, as an example, the detection timing T A’ This shows a means for calculating

[0086] The control device 3 in the first embodiment (FIG. 1) uses the current detection timing calculation unit 17a and the current gradient information calculation unit 16 to calculate the detection timing of the shunt current Is.

[0087] 9, the maximum phase, middle phase, and minimum phase of the three-phase voltage command are the U phase, V phase, and W phase, respectively, before the timing of the crest of the carrier wave Sc, and are the U phase, W phase, and V phase, respectively, after the timing of the crest. At the timing of the crest, the middle phase switches from the V phase to the W phase, and the minimum phase switches from the W phase to the V phase.

[0088] For this reason, a shunt current Is equal to the U-phase motor current Iu flows from when the carrier wave Sc exceeds the V-phase AC voltage command Vv*, which is the middle phase, until it becomes equal to the U-phase AC voltage command Vu*, which is the maximum phase. At this time, Iu including the ripple increases and reaches a peak when Sc becomes equal to Vu*.

[0089] While Sc exceeds Vu*, Is = 0. At this time, Iu including the ripple decreases and becomes a current value equal to the fundamental wave component at the timing of the peak of Sc.

[0090] The period T2 from the timing T(u) when Iu reaches its peak, ie, when Sc is equal to Vu*, to the timing of the peak of Sc, ie, the timing of point A in Iu, is expressed by equation (2).

[0091]

number

[0092] In equation (2), Ts is the period of the carrier wave Sc.

[0093] In Fig. 9, Iu has the same current value at point A' as at point A before it reaches its peak value. In this case, if the period between point A' and the peak in Iu is T1, the amount of change ΔI in Iu during period T2 and period T1 is equal. Therefore, ΔI is expressed by equations (3) and (4).

[0094]

number

[0095]

number

[0096] K2 and K1 are the slopes, or time change rates, of Iu in periods T2 and T1, respectively. K2 and K1 are calculated based on the three-phase voltage equation of the AC motor 1 (FIG. 1).

[0097] Timing T at point A' A’ In this case, Is is equal to Iu. Therefore, the timing T A’ If Is is sampled at point A, the current value of Iu at point A, that is, the current value equal to the fundamental component of Iu, can be obtained.

[0098] Timing T A’ can be calculated using equation (5).

[0099]

number

[0100] Based on Figure 9, T A’ = T(u) - T1. By calculating T1 using equations (2) to (4), we obtain equation (5).

[0101] T(u) is calculated based on the U-phase AC voltage command Vu* and the carrier wave Sc. K1 and K2 are calculated based on the three-phase voltage equation of the AC motor 1 (FIG. 1) as described above. Ts is set by the synchronous PWM control unit 14. Therefore, the timing T A’ can be calculated.

[0102] The control device 3 in the first embodiment (FIG. 1) uses the current detection timing calculation unit 17a and the current gradient information calculation unit 16 to calculate the detection timing of the shunt current Is.

[0103] As shown in FIG. 1, the current detection timing calculation unit 17a receives the carrier period Ts, the three-phase AC voltage commands Vu*, Vv*, Vw*, the rotational position θdc, and the gradients of the three-phase motor currents (K1, K2 in FIG. 9) generated by the current gradient information calculation unit 16, and calculates and outputs the current detection timing Tad (FIG. 1) using equation (5) in accordance with these parameters (Tad=T A’ The current detection timing calculation unit 17a calculates Tad at the timing when the interrupt signal Trg is input.

[0104] The operation of the current detection timing calculation unit 17a will be described below for the case of FIG.

[0105] When the interrupt signal Trg indicates the timing of a peak or a valley, the current detection timing calculation unit 17a determines the maximum phase and minimum phase of the voltage command in the input three-phase AC voltage commands Vu*, Vv*, and Vw*. In the case of FIG. 9, when the current detection timing calculation unit 17a receives Trg indicating the timing of a peak, it determines that the U-phase voltage command Vu* is the maximum phase of the voltage command, and determines the detection timing T of Is for the U-phase motor current Iu, which is the maximum phase. A’ Calculate the following.

[0106] The current detection timing calculation unit 17a calculates the magnitude of Vu*, which is a sine wave, based on θdc, and calculates the timing T(u) of the point A' at which the magnitude of Sc and the magnitude of Vu* match before the crest timing. For example, if the ratio of the peak-to-peak (peak-to-valley) voltage value of Sc to the magnitude of Vu* is equal to the ratio of Ts / 2 to T(u), then T(u) can be calculated. It is assumed that the peak-to-peak (peak-to-valley) voltage value of Sc and the voltage amplitude value (output voltage) of Vu* are set in advance in the current detection timing calculation unit 17a.

[0107] A gradient (time rate of change) K2 of Iu at the timing of the peak of Sc (point A in FIG. 9) and a gradient (time rate of change) K1 of Iu at the timing before point A when Is is flowing and Iu has the same current value as at point A (point A' in FIG. 9) are calculated in advance using a three-phase voltage equation of AC motor 1. Current gradient information calculation unit 16 has table data that associates the previously calculated current gradient (K1, K2 in FIG. 9), the motor current (Iu in FIG. 9) that is the target of timing calculation in current detection timing calculation unit 17a, and whether it is the maximum phase or minimum phase of the voltage command (or whether it is a peak or a valley of Sc (whether Trg indicates a peak or a valley)).

[0108] When the current detection timing calculation unit 17a refers to the table data held by the current gradient information calculation unit 16, the current gradient information calculation unit 16 determines that the current detection timing calculation unit 17a is T A’ The current detection timing calculation unit 17a uses the current slope input from the current slope information calculation unit 16 and T(u) calculated as described above to calculate T A’ Calculate.

[0109] The current detection timing calculation unit 17a calculates the T A’ is set in the current detection unit 20. The current detection unit 20 detects the set timing T A’ As a result, the current detection unit 20 detects Is which is less affected by the current ripple component, that is, a current value equivalent to the fundamental component of Iu.

[0110] The above-mentioned three-phase voltage equation is a known three-phase voltage equation including a winding resistance, an inductance, and an induced voltage constant as motor constants. The above-mentioned current gradient can be calculated by calculating a time differential term of the current in the three-phase voltage equation according to the three-phase voltage command value.

[0111] 9, the Is detection timing for detecting the fundamental wave component is described for the U-phase current when the voltage command maximum phase is the U-phase, but the Is detection timing is calculated similarly for the V-phase current when the voltage command maximum phase is the V-phase, and for the W-phase current when the voltage command maximum phase is the W-phase. Also, the Is detection timing is calculated similarly for the U-phase current, V-phase current, and W-phase current when the voltage command minimum phase is the U-phase, V-phase, and W-phase, respectively.

[0112] As described above, according to the first embodiment, the motor current at the time when the carrier wave Sc becomes a peak (maximum) or a valley (minimum) is reproduced based on the shunt current Is, i.e., the DC bus current, detected by the current detection unit 20 at the detection timing calculated by the current detection timing calculation unit 17a. This makes it possible to reproduce the fundamental wave component of the motor current with high accuracy. This improves the accuracy of the current control of the AC motor 1 and improves the estimation accuracy of the magnetic pole position of the AC motor 1. Furthermore, it is possible to stably drive a low inductance motor with a large current ripple at high speed.

[0113] Moreover, the current detection timing calculation unit 17a calculates, as the detection timing, the timing at which the current value of the motor current becomes the same as the time when the carrier wave Sc becomes a peak (maximum) or a valley (minimum) during the period when the DC bus current flows, thereby enabling the fundamental wave component of the motor current to be reproduced with high accuracy from the detection value of the DC bus current. EXAMPLES

[0114] Hereinafter, the second embodiment of the present invention will be described with reference to FIGS. 10 and 11, focusing mainly on the differences from the first embodiment.

[0115] Second Embodiment FIG. 10 is a functional block diagram showing a configuration of an AC motor driving device according to a second embodiment of the present invention.

[0116] In the second embodiment, as described later, the current detection timing calculation unit 17b calculates the timings (times B and C in FIG. 11) at which the shunt current Is is detected before and after the timings of the peaks and valleys of the carrier wave Sc. The average value calculation unit 22 calculates average values ​​(Iu', Iv', Iw') of the three-phase motor currents (Iu, Iv, Iu) reproduced by the current processing unit 9, based on the Is detected at the timings at the two points in time calculated by the current detection timing calculation unit 17b. As described later, the average values ​​of the three-phase motor currents at the two points in time calculated by the average value calculation unit 22 are equal to the fundamental wave component of the motor current.

[0117] FIG. 11 is a waveform diagram showing an example of the shunt current Is and the motor current, illustrating a calculation means for the detection timing of the shunt current Is in the second embodiment.

[0118] In FIG. 11, as an example, the detection timing T B ,T C In FIG. 11, T B ,T C are the timings of points B and C, respectively.

[0119] The period when Is does not flow, including the peak of Sc, i.e., the peak value of Iu, I up and bottom value I ub The period between time points B and I is T3. up The period between the time B , I ub The period between time and time C is ΔT C If the carrier period is Ts, the detection timing T B ,T C are expressed by equations (6) and (7), respectively.

[0120]

number

[0121]

number

[0122] In addition, the shunt current at time points B and C is Is(B) and Is(C), and the slope (time rate of change) of Iu during the period when Is flows, including time point B, is K. B , the slope (time rate of change) of Iu during the period when Is flows, including time C, is K C Then, the peak value Iup and bottom value Iub of Iu are expressed by equations (8) and (9), respectively.

[0123]

number

[0124]

number

[0125] Furthermore, the current value of Iu at time A, i.e., the fundamental component I u0 I up and I ub is the average value of and is expressed by equation (10).

[0126]

number

[0127] Here, ΔT C is set so as to avoid the delay in the rise of Is caused by the turn-on characteristics of the semiconductor switching elements constituting the main circuit of the power conversion device 2, and the influence of ringing that occurs when Is rises. Furthermore, in the second embodiment (FIG. 11), ΔT B and ΔTC There is a relationship as shown in equation (11).

[0128]

number

[0129] According to equation (11), ΔT set as above C Depending on the B is set.

[0130] Equation (12) is obtained from equations (8) to (11).

[0131]

number

[0132] As shown in equation (12), the fundamental component I u0 is the detection timing T B and T C It can be calculated by the average value of the current Is detected at

[0133] The control device 3 in the second embodiment (FIG. 10) uses the current detection timing calculation unit 17b and the current gradient information calculation unit 16 to calculate the detection timing of the shunt current Is.

[0134] The current detection timing calculation unit 17b calculates the carrier period Ts, the three-phase AC voltage commands Vu*, Vv*, and Vw*, the rotational position θdc, and the gradient of the three-phase motor current generated by the current gradient information calculation unit 16 (K in FIG. 11). B ,K C ) is input, and the current detection timing Tad (FIG. 10) is calculated and output using equations (6), (7), and (11) according to these parameters (in FIG. 11, Tad=T B ,T C The current detection timing calculation unit 17b calculates Tad at the timing when the interrupt signal Trg is input.

[0135] The operation of the current detection timing calculation unit 17b will be described below for the case of FIG.

[0136] When the interrupt signal Trg indicates the timing of a peak or a valley, the current detection timing calculation unit 17b determines the maximum phase and minimum phase of the voltage command in the input three-phase AC voltage commands Vu*, Vv*, and Vw*. In the case of FIG. 11, when the current detection timing calculation unit 17b receives Trg indicating the timing of a peak, it determines that the U-phase voltage command Vu* is the maximum phase of the voltage command, and determines the detection timing T of Is for the U-phase motor current Iu, which is the maximum phase. B ,T C Calculate the following.

[0137] T3 / 2 in the formulas (6) and (7) corresponds to T2 in the first embodiment (FIG. 9). Therefore, the current detection timing calculation unit 17b calculates T3 / 2 by the calculation means of T2 in the first embodiment. In addition, the current detection timing calculation unit 17b calculates ΔT C This data is, for example, table data indicating the correspondence between the motor current (Iu, Iv, Iw (Iu in FIG. 11)) for which the detection timing of Is is to be calculated, whether it is the maximum phase or the minimum phase, and ΔTc. The current detection timing calculation unit 17b refers to this table data and calculates ΔT B ΔT used to set C Read out.

[0138] The slope (time rate of change) of Iu at points B and C in Fig. 11, respectively, B and K. C is calculated in advance using the three-phase voltage equation of the AC motor 1. The current gradient information calculation unit 16 calculates the current gradient (K B ,K C ) and table data correlating the motor current (Iu in FIG. 11) for which detection timing calculation is performed in current detection timing calculation unit 17b with whether it is the maximum phase or minimum phase of the voltage command (or whether it is a peak or a valley of Sc (whether Trg indicates a peak or a valley)).

[0139] When the current detection timing calculation unit 17b refers to the table data held by the current gradient information calculation unit 16, the current gradient information calculation unit 16 determines that the current detection timing calculation unit 17b has a time ΔT B ,T B ,T C The current slope (K B ,K C The current detection timing calculation unit 17b outputs the current gradient (K B ,K C ) and the aforementioned ΔT C Using equation (11), ΔT B Furthermore, the current detection timing calculation unit 17b calculates the current gradient (K B ,K C ), ΔT C , current slope (K B ,K C ) and ΔT C ΔT calculated from B Using the above T3 / 2, the detection timing T of Is is calculated by equations (6) and (7). B ,T C Calculate the following.

[0140] The current detection timing calculation unit 17b calculates T(K B ,K C ) is set in the current detection unit 20. The current detection unit 20 detects the set timing T B ,T C Detect Is with.

[0141] The average value calculation unit 22 (FIG. 10) detects the current at timing T B ,T C The average value of the above-mentioned Is(B) and Is(C) is calculated by equation (12) and output. u0 corresponds to Iu' output by the average value calculation unit 22 (FIG. 10).

[0142] This makes it possible to detect the fundamental wave component of the three-phase motor current, which is less affected by the current ripple component, based on the detected value of Is.

[0143] The above-mentioned three-phase voltage equation is a known three-phase voltage equation including a winding resistance, an inductance, and an induced voltage constant as motor constants. The above-mentioned current gradient can be calculated by calculating a time differential term of the current in the three-phase voltage equation according to the three-phase voltage command value.

[0144] 11 has described the Is detection timing for detecting the fundamental wave component for the U-phase current when the voltage command maximum phase is the U-phase, but the Is detection timing is similarly calculated for the V-phase current when the voltage command maximum phase is the V-phase, and for the W-phase current when the voltage command maximum phase is the W-phase. Also, when the voltage command minimum phase is the U-phase, V-phase, or W-phase, the Is detection timing is similarly calculated for the U-phase current, V-phase current, and W-phase current, respectively.

[0145] As described above, according to the second embodiment, similarly to the first embodiment, the motor current at the time when the carrier wave Sc becomes a peak (maximum) or a valley (minimum) is reproduced based on the shunt current Is, i.e., the DC bus current, detected by the current detection unit 20 at the detection timing calculated by the current detection timing calculation unit 17b. Furthermore, according to the second embodiment, the current detection timing calculation unit 17b calculates, as the detection timing, a plurality of timings at which the average value of the detected DC bus current becomes the same as the motor current at the time when the carrier wave Sc becomes a peak (maximum) or a valley (minimum) during the period when the DC bus flows. This makes it possible to reproduce the fundamental wave component of the motor current with high accuracy. EXAMPLES

[0146] Third Embodiment FIG. 12 is a functional block diagram showing a configuration of an AC motor driving device according to a third embodiment of the present invention.

[0147] The following describes the differences from the first embodiment.

[0148] As shown in FIG. 12, AC motor 1 drives blower 22 in a vacuum cleaner as mechanical load 21 (FIG. 1) in the first embodiment.

[0149] Moreover, the AC motor driving device according to the third embodiment includes an operation switch 23, as shown in Fig. 12. Operation switch 23 is used to select one of "strong", "weak" and "off" as an operation mode of the vacuum cleaner.

[0150] The rotation speed command generating unit 4 generates a rotation speed command ωr* in accordance with an operation mode selected by the operation switch 23. For example, the rotation speed commands ωr1*, ωr2* (<ωr*1) and zero are generated for the operation modes "strong", "weak" and "off", respectively.

[0151] According to the third embodiment, a small AC motor can be stably and precisely controlled to operate at high speed. This allows the small AC motor to be applied to a vacuum cleaner, thereby making the vacuum cleaner more compact.

[0152] Although the control device 3 (FIG. 1) of the first embodiment is applied to the control device 3 (FIG. 12) of the third embodiment, the control device 3 (FIG. 10) of the second embodiment may be applied.

[0153] The AC motor drive devices according to the first to third embodiments can be applied not only to vacuum cleaners but also to electric devices in which a mechanical load is driven by an AC motor, such as electric compressors, electric automobiles, and electric railcars.

[0154] The present invention is not limited to the above-described embodiments, but includes various modifications. For example, the above-described embodiments have been described in detail to clearly explain the present invention, and the present invention is not necessarily limited to those including all of the configurations described. In addition, it is possible to add, delete, or replace part of the configuration of each embodiment with other configurations.

[0155] For example, the number of pulses in the synchronous PWM control is not limited to three, and may be another number of pulses. [Explanation of symbols]

[0156] 1...AC motor 2. Power conversion device 3. Control device 4...Rotational speed command generator 5...Current command generation section 6...Vector control unit 7... Coordinate conversion section 8...PWM signal generation section 9...Current processing section 10... Coordinate conversion section 11...Axis error estimator 12...PLL control section 13...Integrator part 14...Synchronous PWM control section 15…Carrier generation section 16...Current gradient information calculation section 17a, 17b...Current detection timing calculation unit 18…DC power supply 19...Shunt resistor 20...Current detection section 21. Load 22...Average value calculation section 23...Operation switch

Claims

1. In an AC motor control device that controls a power conversion device that outputs AC power to an AC motor, a PWM signal generating unit that generates a control signal for controlling the power conversion device, the control signal being a PWM pulse signal, by comparing an AC voltage command with a carrier wave; a current detection unit that detects a DC bus current flowing on a DC side of the power conversion device at a predetermined detection timing; a current detection timing calculation unit that calculates the detection timing; Equipped with an AC motor control device that reproduces a motor current at a point in time when the carrier wave becomes a peak or a valley based on the DC bus current detected by the current detection unit at the detection timing calculated by the current detection timing calculation unit, and controls the power conversion device based on the reproduced motor current.

2. 2. The AC motor control device according to claim 1, The current detection timing calculation unit calculates, as the detection timing, the timing at which the current value of the motor current becomes the same as the time point during a period in which the DC bus current flows.

3. 3. The AC motor control device according to claim 2, 2. An AC motor control device, comprising: a first period during which the DC bus current flows, the first period including the time point, and the second period immediately preceding a period during which the DC bus current does not flow.

4. 4. The AC motor control device according to claim 3, a current gradient information calculation unit that calculates a first gradient of the motor current when the current value of the motor current becomes the same as that at the time point, and a second gradient of the motor current at the time point, The AC motor control device, wherein the current detection timing calculation unit calculates the detection timing by using the first gradient and the second gradient.

5. 2. The AC motor control device according to claim 1, The current detection timing calculation unit The AC motor control device calculates, as the detection timing, a plurality of timings at which an average value of the detected DC bus current during a period during which the DC bus current flows becomes the same as the motor current at the time point.

6. 6. The AC motor control device according to claim 5, 13. An AC motor control device, wherein the period during which the DC bus current flows is a period immediately before and after a period during which the DC bus current does not flow, including the time point.

7. 7. The AC motor control device according to claim 6, a current gradient information calculation unit that calculates a first gradient of the motor current in the immediately preceding period and a second gradient of the motor current in the immediately following period, The AC motor control device, wherein the current detection timing calculation unit calculates the detection timing by using the first gradient and the second gradient.

8. 2. The AC motor control device according to claim 1, An AC motor control device, characterized in that the AC voltage command is generated by vector control based on the reproduced motor current.

9. 9. The AC motor control device according to claim 8, 2. An AC motor control device, comprising: a rotational position used for coordinate transformation in the vector control, the rotational position being estimated by sensorless control.

10. 2. The AC motor control device according to claim 1, The AC motor control device, wherein the PWM signal generating unit generates the control signal by synchronous PWM control.

11. a power conversion device that outputs AC power to an AC motor; A control device that controls the power conversion device; In an AC motor drive device comprising:

2. An AC motor drive device, wherein the control device is an AC motor control device as defined in claim 1.

12. An AC motor; a mechanical load driven by the AC motor; and a power conversion device that outputs AC power to an AC motor; A control device that controls the power conversion device; In an electrically powered device comprising:

2. An electrically-driven device, wherein the control device is an AC motor control device as defined in claim 1.

13. The electrically powered device according to claim 12, The electrically-driven appliance, wherein the mechanical load is a blower in a vacuum cleaner.

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