Switching amplifier

Abstract

To suppress the surge voltage of the first surge voltage caused by square waves and the second surge voltage caused by harmonic frequency components, to broaden the frequency characteristics of the output of the switching amplifier and to make the output frequency variable.SOLUTION: A switching amplifier in push-pull configuration with narrow bandwidth filter on the output side comprises a partial resonance circuit to suppress the first surge voltage caused by square waves and a LCR series resonator to suppress the second surge voltage caused by harmonic frequency components, and the waveform shape of the drain-source voltage Vds of the switching element is a trapezoidal waveform.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a switching amplifier using a switching mode system such as class D, class E, class F, inverse class F, or class EF, and in particular to a switching amplifier suitable for a variable frequency high frequency power supply that can vary the output frequency over a wide band. [Background technology]

[0002] A switching amplifier generates a square wave through switching operation and amplifies the output by modulating the pulse width of the generated square wave according to the input signal level. One circuit method for switching amplifiers is to operate two switching elements in opposite phases and increase the output current by alternately outputting and drawing in current, and a large capacity output can be obtained by applying a push-pull circuit.

[0003] In a push-pull switching amplifier of a class D amplifier, the drain-source voltage waveform generated by the switching operation is a square wave. The drain-source voltage waveform is distorted (surge) from a square wave due to leakage inductance. It is known that in order to make the drain-source voltage waveform a square wave, the degree of coupling of the output transformer is increased to reduce the leakage inductance (Non-Patent Document 1). [Prior art documents] [Non-patent literature]

[0004] [Non-Patent Document 1] Microsemi PPG Application Note1812 / 13.56 MHz, Class D Push-Pull, 2KW RF Generator with Microsemi DRF1300 Power MOSFET Hybrid / By Gui Choi 2011 September Summary of the Invention [Problem to be solved by the invention]

[0005] In a switching amplifier, a problem occurs in that a surge voltage (hereinafter referred to as a first surge voltage) occurs due to leakage inductance of an output transformer or the like during switching operation to generate a square wave.

[0006] Furthermore, the output waveform of the output transformer of a switching amplifier contains harmonic frequency components in addition to the fundamental frequency components. As a harmonic countermeasure to remove these harmonic frequency components from the output, it is possible to provide a narrow-band filter such as an LPF on the secondary side of the output transformer. The narrow-band filter passes the fundamental frequency components while blocking the harmonic frequency components. These harmonic frequency components are reflected by the narrow-band filter and returned to the switching element. The returned harmonic frequency components act as a surge voltage (hereinafter referred to as the second surge voltage) on the switching element. In a VHF switching amplifier using a switching element such as an LDMOS with a wide frequency band up to several hundred MHz, the frequency of the harmonic frequency components reflected by the narrow-band filter and returned is also within the frequency band of the switching element, so they are affected by the surge voltage, and the problem of the second surge voltage becomes more pronounced.

[0007] 14A and 14B are diagrams for explaining the second surge voltage due to the harmonic frequency component. In a circuit example of a push-pull switching amplifier of a class-D amplifier, Fig. 14A shows an example of a circuit without a narrow-band filter, and Fig. 14B shows an example of a circuit with a narrow-band filter.

[0008] 14A includes a switching unit 111 having two switching elements 111A and 111B, an output transformer 112 connected between the drains of the switching elements 111A and 111B, and a DC power supply 113 connected to a center tap of a primary winding 112a of the output transformer 112. The switching amplifier 101a configures a push-pull circuit 114 by alternately switching the switching elements 111A and 111B using gate signals Pg of opposite phases from signal sources 110A and 110B. A current is transmitted to a secondary winding 112b of the output transformer 112 by the push-pull circuit 114. The transmitted current is supplied to a load 117 connected to an output terminal 116.

[0009] 14A, the drain-source voltages VdsA and VdsB of switching elements 111A and 111B are switched to produce a square wave If0 and a current Inf0 of harmonic frequency components, which are transmitted to and consumed by load 117. In this configuration, the output waveform also becomes a square wave, and is therefore not suitable for use as a high-frequency power supply for semiconductor manufacturing equipment that must output a sine wave waveform.

[0010] On the other hand, switching amplifier 101b in Fig. 14B has a sine wave output waveform by providing narrow band filter 115 on the secondary side of output transformer 112. Although the configuration with narrow band filter 115 can obtain a sine wave output voltage, it is not possible to maintain the waveforms of drain-source voltages VdsA and VdsB as square waves. The inventors of the present application have found that the reason why the square wave waveforms are not maintained is that current Inf0 of harmonic frequency components contained in drain-source voltages VdsA and VdsB is reflected by narrow band filter 115 and returns to switching elements 111A and 111B.

[0011] In VHF band amplifiers, switching elements with a wide frequency band of the device itself, such as LDMOS, are used to cover a wide frequency band. Therefore, in switching amplifiers with wide output frequency characteristics, it was not possible to proactively provide a narrow-band filter on the secondary side of the output transformer in order to prevent harmonic frequency components from being reflected back. This caused the problem that it was not possible to achieve both a wide band and a sine wave output waveform.

[0012] Therefore, in VHF band amplifiers, two surge voltages are generated during switching operation: a first surge voltage due to leakage inductance of the output transformer, etc., and a second surge voltage due to harmonic frequency components reflected by the narrow-band filter. These surge voltages are factors that prevent the output frequency of high-frequency power supplies that use switching amplifiers from being made broadband.

[0013] In order to widen the frequency characteristics of the output of a switching amplifier and obtain a stable output even when the output frequency is variable, it is required that the waveform shape of the drain-source voltage Vds of the switching element be a trapezoidal waveform in which the first surge voltage caused by a square wave and the second surge voltage caused by harmonic frequency components are suppressed.

[0014] The present invention aims to solve the above-mentioned problems of the conventional technology, suppress the first surge voltage caused by a square wave and the second surge voltage caused by harmonic frequency components, widen the frequency characteristic of the output of a switching amplifier, and make the output frequency variable. [Means for solving the problem]

[0015] The switching amplifier according to the present invention is a push-pull switching amplifier equipped with a narrow-band filter on the output side, and includes a partial resonant circuit for suppressing a first surge voltage caused by a square wave, and an LCR series resonant element for suppressing a second surge voltage caused by harmonic frequency components, and the waveform shape of the drain-source voltage Vds of the switching element is made trapezoidal, thereby widening the frequency characteristic of the output of the switching amplifier and making the output frequency variable.

[0016] (Drain-source voltage Vds with trapezoidal waveform) The trapezoidal waveform has an inclined waveform in which the voltage rises and the section that requires a certain time width during the rise is formed, and the crest value is maintained constant in the section from when the voltage rises until when it falls.

[0017] The ramp waveform suppresses surge voltages and harmonic frequency components in the drain-source voltage Vds of the switching element that occur during the voltage rise and fall. A waveform that maintains a constant peak value keeps the drain-source voltage Vds constant even when the output frequency is varied and swept over a wide band, so the output gain can be kept constant.

[0018] (Switching amplifier configuration) The switching amplifier according to the present invention comprises: (a) A transformer-coupled push-pull circuit consisting of a pair of switching elements that alternately switch on and off with input signals of opposite phases, a primary winding that inputs the output signal of the switching elements to both ends, a secondary winding that extracts the output signal, and a transformer with an external power supply connected to the center tap of the primary winding (b) A narrow-band filter connected to the output of the transformer secondary winding. (c) A pair of partial resonant circuits formed by an inductance component on the drain side of each switching element and a capacitance component between the drain and source of each switching element (d) An LCR series resonator consisting of a series circuit of inductance, capacitance, and resistance connected between the drains of each switching element. The device comprises the following components: The switching element provided in the switching amplifier according to the present invention can be a MOSFET, and in the case of a power MOSFET, an LDMOS or VDMOS having a DMOS structure can be used.

[0019] (Push-pull circuit) In a push-pull circuit, a pair of switching elements alternate between on and off switching operations using input signals of opposite phases. This alternating switching operation alternates between each switching element being conductive to an external power source connected to the center tap of the transformer's primary winding, causing an inductance current to flow in the primary winding in alternately opposite directions. This inductance current causes a load current to flow in the secondary winding of the transformer due to transformer coupling. The inductance current flowing in the primary winding alternately flows in the opposite direction. This inductance current on the primary winding side corresponds to the secondary winding side alternately outputting and drawing in current, so the transmitted load current flows in the secondary winding.

[0020] The load current induced in the secondary winding contains fundamental and harmonic frequency components according to the switching frequency of the switching element. Because the even harmonic frequency components cancel each other out due to the characteristics of the push-pull circuit, the load current contains only odd harmonic frequency components.

[0021] (narrow band filter) The narrow-band filter passes the fundamental frequency component contained in the load current of the secondary winding and blocks the harmonic frequency components. The harmonic frequency components are reflected by the narrow-band filter and returned to the switching element. The harmonic frequency components returned to the switching element become a surge voltage (second surge voltage) for the switching element.

[0022] (LCR series resonator) The LCR series resonator is composed of a series circuit of inductance, capacitance, and resistance, and is connected between the drains of a pair of switching elements. The resonant frequency of the series circuit is set to match the frequency of the harmonic frequency components of the load current.

[0023] (Parallel connection of LCR series resonators) The LCR series resonator can widen the frequency band of harmonic frequency components of the load current by configuring a plurality of series circuits with different resonant frequencies connected in parallel.

[0024] The LCR series resonator, which is formed by connecting two series circuits in parallel, comprises a first series circuit having a resonant frequency lower than the harmonic frequency n×f0 of the high frequency output, relative to the fundamental frequency f0 of the high frequency output output from the secondary side of the transformer, and a second series circuit having a resonant frequency higher than the harmonic frequency n×f0 of the high frequency output.

[0025] When the LCR series resonator is composed of a plurality of series circuits connected in parallel, the number of series circuits connected in parallel is not limited to two and may be any number.

[0026] (Suppression of the second surge voltage by an LCR series resonator) The harmonic frequency components reflected by the narrowband filter flow to ground via the LCR series resonator and the switching element in the ON state. At this time, the harmonic frequency components are consumed and attenuated by the resistance of the series circuit. This attenuation of the harmonic frequency components suppresses the second surge voltage caused by the harmonic frequency components applied to the switching element. By suppressing the second surge voltage, fluctuations in the peak value of the drain-source voltage Vds of the switching element are suppressed and become almost constant.

[0027] (partial resonant circuit) The partial resonant circuit is formed of an inductance component on the drain side of the switching element and a capacitance component between the drain and source of the switching element. The partial resonant circuit is formed in each switching element of the pair of switching elements.

[0028] The inductance component (Ls) of each partial resonant circuit is either a series inductance (Ls1) connected in series between the switching element and one end of the transformer, a leakage inductance (Lleak) of the output transformer, or a wiring inductance (Lline) between the switching element and one end of the transformer, or any combination thereof.

[0029] The capacitance component (Ctotal) of each partial resonant circuit is the output parasitic capacitance (Coss) of the switching element, or a combined capacitance of the output parasitic capacitance (Coss) of the switching element and an additional capacitance (Cp) connected in parallel to the switching element.

[0030] When the switching element changes from the on state to the off state, the charge of the capacitance component (Ctotal) that was discharged in the on state is charged by the first resonance due to the partial resonance of the primary current of the transformer. During this charging, the drain-source voltage Vds of the switching element increases with a time constant τr determined by the inductance component (Ls) and the capacitance component (Ctotal).

[0031] On the other hand, when the switching element changes from the OFF state to the ON state, the charge of the capacitance component (Ctotal) that was charged in the OFF state is discharged by the second resonance due to the partial resonance of the primary current of the transformer. During this discharge, the drain-source voltage Vds of the switching element decreases with a time constant τf determined by the inductance component (Ls) and the capacitance component (Ctotal).

[0032] During the dead time period, the capacitance component (Ctotal) of the first resonance is the capacitance component (Ctotal) of two switching elements connected in series. On the other hand, the capacitance component (Ctotal) of the second resonance is the capacitance component (Ctotal) of only one switching element, so the time constant τr of the first resonance is smaller than the time constant τf of the second resonance, and the voltage rises faster. The first and second resonances cause the drain-source voltage Vds to have a trapezoidal waveform.

[0033] (Suppression of the first surge voltage by partial resonant circuit and LCR series resonant element) Due to partial resonance of the primary current of the transformer, the drain-source voltage Vds of the switching element increases or decreases with the time constants τr and τf during the dead time period, so that the surge voltage generated in the drain-source voltage Vds is suppressed.

[0034] During the period excluding the dead time, the charging current of the first resonance and the discharging current of the second resonance due to partial resonance flow between the two drains via the LCR series resonator. The charging and discharging current that flows through the LCR series resonator is consumed and attenuated by the resistance Rr of the LCR series resonator. During the period excluding the dead time, the charging and discharging current due to partial resonance increases and decreases with the time constants τr and τf, and is consumed and attenuated by the LCR series resonator, thereby suppressing the occurrence of the first surge voltage in the drain-source voltage Vds.

[0035] Therefore, the first surge voltage caused by the square wave is suppressed by the partial resonant circuit and the LCR series resonant body, and the waveform of the drain-source voltage Vds becomes a trapezoidal wave.

[0036] (Resonant frequency and bandwidth of partial resonant circuit) The resonant frequency of the pair of partial resonant circuits reduces the impedance in the frequency range where the impedance is high by the partial resonant circuits, in the range from the fundamental frequency f0 to the third harmonic frequency 3×f0 of the high frequency output output from the secondary side of the transformer, and sets a predetermined time constant for the rise and fall of the drain-source voltage Vds of the LDMOS.

[0037] (Resonant frequency and bandwidth of LCR series resonator) When the fundamental frequency of the high-frequency output from the secondary side of the transformer is f0, the resonant frequency of the LCR series resonator is the harmonic frequency n×f0 (n is an integer), and the bandwidth of the resonant frequency is set to the frequency range (n×f0-min to n×f0-max) according to the variable frequency range (f0-min to f0-max) of the fundamental frequency f0.

[0038] (Action of the switching amplifier of the present invention) The switching amplifier according to the present invention suppresses surge voltages, broadens the frequency characteristic of the output of the switching amplifier, and makes the output frequency variable, thereby achieving the following secondary effects.

[0039] Eliminates the need for leakage countermeasures for output transformers. Prior art documents disclose that in order to obtain a square-wave drain-source voltage waveform, the degree of coupling of the output transformer is increased to reduce leakage inductance, and a structure is shown in which a magnetic core is prepared as the material for the output transformer and a twisted-wire winding is wound around the core to reduce leakage inductance. This is a common method of increasing the degree of coupling to reduce leakage inductance. Another method is to select a material with a high relative permeability μ for the core material. However, there is a problem in that it is difficult to design a core material that can withstand continuous high output voltage because losses due to the core material increase in proportion to the relative permeability μ.

[0040] According to the present invention, it is possible to eliminate the need for a core material with high relative magnetic permeability or a twisted wire structure as a leakage countermeasure for an output transformer, and it is possible to realize an amplifier capable of continuously outputting high output power.

[0041] Generally, even when using a coreless, air-core type output transformer which has a large leakage inductance, or a parallel-winding type planar transformer which has poor coupling due to its winding structure, it is possible to insert an LPF (low-pass filter) on the secondary side of the transformer, thereby realizing a class-D push-pull amplifier with low output harmonics.

[0042] The peak value of the drain-source voltage Vds can be made almost the same over the entire wide band, making it possible to maintain constant power gain.

[0043] According to the present invention, the waveform of the drain-source voltage Vds always has a similar peak value, so that output fluctuations can be suppressed. The fluctuations in power gain due to frequency variation that are prominent in class AB and class C amplifiers are suppressed, and flat output characteristics can be obtained.

[0044] Even if the leakage inductance is large, a stable trapezoidal Vds waveform can be obtained, enabling a wide bandwidth.

[0045] A low-pass filter (LPF) can be inserted on the secondary side of the output transformer to reduce output harmonics.

[0046] The surge voltage due to partial resonance can be suppressed, and the switching drain-source voltage Vds can be prevented from exceeding the element withstand voltage of the switching element. Effect of the Invention

[0047] As described above, according to the present invention, it is possible to suppress the first surge voltage caused by a square wave and the second surge voltage caused by harmonic frequency components, to broaden the frequency characteristics of the output of the switching amplifier, and to make the output frequency variable. [Brief description of the drawings]

[0048] [Figure 1] FIG. 1 is a diagram for explaining a first configuration of a switching amplifier according to the present invention. [Diagram 2] FIG. 2 is a diagram for explaining a second configuration of the switching amplifier of the present invention. [Figure 3A] FIG. 11 is a diagram for comparing the pass characteristics of an LCR series resonator and an LC series resonator. [Figure 3B] FIG. 2 is a diagram showing an equivalent circuit of an LCR series resonator and an LC series resonator. [Figure 4] 3A and 3B are diagrams for explaining signals at various parts of the switching amplifier of the present invention. [Figure 5A] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 5B] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 6A] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 6B] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 7A] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 7B] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 8A] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 8B] 5A to 5C are diagrams for explaining an example of operation in each section of the switching amplifier of the present invention. [Figure 9] 10 is a diagram for explaining a trapezoidal waveform of a drain-source voltage of a switching element. FIG. [Figure 10] FIG. 13 is a diagram showing an example of a waveform when the fundamental frequency f0 is 45 MHz. [Figure 11]FIG. 13 is a diagram showing an example of a waveform when the fundamental frequency f0 is 35 MHz. [Figure 12A] FIG. 1 is a diagram for explaining the broadening of the output band by an LCR series resonator. [Figure 12B] FIG. 1 is a diagram for explaining the broadening of the output band by an LCR series resonator. [Figure 13A] 1 is a Smith chart for explaining the relationship between a partial resonance frequency band and a high impedance frequency band. [Figure 13B] 10 is a frequency characteristic diagram for explaining the relationship between a partial resonance frequency band and a high impedance frequency band. FIG. [Figure 14A] FIG. 11 is a diagram for explaining a second surge voltage caused by a harmonic frequency component. [Figure 14B] FIG. 11 is a diagram for explaining a second surge voltage caused by a harmonic frequency component. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0049] (1) Schematic configuration of a switching amplifier The schematic configuration of a switching amplifier of the present invention will be described below with reference to Figures 1 and 2. Figure 1 shows a first configuration example having one LCR series resonator, and Figure 2 shows a second configuration example having two LCR series resonators connected in parallel.

[0050] (1-1: First configuration example) The first configuration example of the switching amplifier 1 includes a switching section 11, an output transformer 12, an external power supply consisting of a DC power supply 13, and a switching amplifier consisting of a narrow-band filter 15, as well as an LCR series resonator 20 and a partial resonant circuit 21 (21A, 21B).

[0051] The switching unit 11 has two switching elements. MOSFETs are used as the switching elements, and power MOSFETs are used for large currents. DMOSs such as LDMOSs and VDMOSs are used as power MOSFETs. Here, an LDMOS is used as an example of a switching element, and the two switching elements provided in the switching unit 11 are represented as LDMOS (11A) and LDMOS (11B).

[0052] The gate terminal of the LDMOS (11A) receives a gate signal Pg from the signal source 10A and switches the on / off state between the drain terminal and the grounded source terminal. Similarly, the gate terminal of the LDMOS (11B) receives a gate signal Pg from the signal source 10B and switches the on / off state between the drain terminal and the grounded source terminal. By making the gate signals Pg of the signal sources 10A and 10B have an opposite phase relationship, the outputs of the LDMOS (11A) and LDMOS (11B) are opposite phases.

[0053] An output transformer 12 is connected between the drain terminal of the LDMOS (11A) and the drain terminal of the LDMOS (11B), a DC power supply 13 is connected to the center tap of the output transformer 12, and the outputs of the LDMOS (11A) and LDMOS (11B) are in opposite phase to form a push-pull circuit 14. The push-pull circuit 14 alternates between outputting and drawing current in the secondary winding 12b of the output transformer 12, transmitting the load current which is the output current.

[0054] A narrow-band filter 15 such as an LPF is provided between the secondary winding 12b of the output transformer 12 and the output terminal 16. The narrow-band filter 15 blocks the harmonic frequency n×f0 contained in the load current of the output current and passes the fundamental frequency f0 to the output terminal 16.

[0055] In the push-pull circuit 14, even-order harmonic frequency components cancel each other out, so the harmonic frequency n×f0 flowing through the secondary winding 12b becomes an odd-order harmonic frequency. Furthermore, the higher the order of the harmonic frequency, the smaller the peak value of the harmonic frequency. Therefore, in designing the narrow-band filter 15, by providing it with frequency characteristics that block the third harmonic frequency 3×f0 as the harmonic frequency n×f0, the harmonic frequency components can be sufficiently blocked.

[0056] The switching amplifier according to the present invention characteristically comprises an LCR series resonator 20 as a configuration for suppressing a second surge voltage generated due to harmonic frequency components contained in the load current of the output current, and characteristically comprises an LCR series resonator 20 and a partial resonant circuit 21 (21A, 21B) as a configuration for suppressing a first surge voltage generated due to high frequency components contained in the drain-source voltage Vds, which is the output voltage of the LDMOS switching element.

[0057] (LCR series resonator) The LCR series resonator 20 is composed of a series circuit of an inductance Lr, a capacitance Cr, and a resistance Rr, and is connected between the drain terminals of the LDMOS (11A) and LDMOS (11B), and is connected in parallel to the primary winding 12a of the output transformer 12.

[0058] The resonant frequency of the series circuit of the LCR series resonator 20 is set to match the frequency of the harmonic frequency components of the load current.

[0059] In a push-pull circuit, the harmonic frequency components contained in the load current are odd-order harmonic frequency components, and the peak values ​​of the fifth-order and higher harmonic frequency components are smaller than the peak value of the third-order harmonic frequency, so the resonant frequency of the series circuit is set to a frequency approximately three times the fundamental frequency f0 and its bandwidth [f0-min, f0-max], or approximately two to four times the frequency, taking into account the deviation of the resonant constant. Note that f0-min is the lower limit frequency of the load current, and f0-max is the upper limit frequency of the load current.

[0060] The resistor Rr, which is the resistance of the series circuit, is set to, for example, 0.5Ω[ohm] to 15Ω[ohm] so that sufficient attenuation is obtained at the center frequency of the third harmonic frequency and the attenuation effect is obtained over a wide band within the bandwidth. The setting range of the resistor Rr is such that the resistance value as seen from the primary side of the output transformer is approximately equal to the load impedance of the fundamental wave.

[0061] When the turns ratio of the output transformer is m:n, the primary resistance is R1, and the secondary resistance is R2, R1 = (m / n) 2 ×R2. In this relationship, for example, when m=1, n=2, and R2=50[ohm], the primary resistance R1 is 12.5[ohm], so 15[ohm] is set as the upper limit. Also, as the fundamental frequency increases, the load impedance suitable for the LDMOS characteristics decreases, so 0.5[ohm] is set as the lower limit.

[0062] The LCR series resonator 20 attenuates the harmonic frequency components reflected by the narrowband filter and returned to the LDMOS. This keeps the peak value of the drain-source voltage of the LDMOS constant, so that the output gain characteristics can be kept constant even when the load current frequency varies over a wide band, making it possible to achieve a stable output over a wide band.

[0063] (partial resonant circuit) The partial resonant circuit 21 includes a partial resonant circuit 21A on the LDMOS (11A) side, and a partial resonant circuit 21B on the LDMOS (11B) side.

[0064] The partial resonant circuit 21A is formed of a series circuit of an inductance component Ls-A on the drain terminal side of the LDMOS (11A) and a capacitance component CA of the LDMOS (11A).

[0065] The inductance component Ls-A is any one of a series inductance Ls1-A connected in series between the LDMOS (11A) and one end of the output transformer 12, a leakage inductance Lleak of the output transformer 12, and a wiring inductance Lline-A between the LDMOS (11A) and one end of the output transformer 12, or a combination of any one of them. The series inductance Ls1-A is connected as necessary according to the resonant frequency of the partial resonant circuit 21A.

[0066] The capacitance component CA of the LDMOS (11A) can be the parasitic capacitance Coss-A of the LDMOS (11A), or the combined capacitance Ctotal-A of the additional capacitance Cp-A and the parasitic capacitance Coss-A connected in parallel between the drain and source terminals of the LDMOS (11A).

[0067] The partial resonant circuit 21B, like the partial resonant circuit 21A, is configured by a series circuit of a series inductance component Ls-B on the drain end side of the LDMOS (11B) and a capacitance component CB of the LDMOS (11B).

[0068] The inductance component Ls-B is any one of the following: a series inductance Ls1-B connected in series between the LDMOS (11B) and the other end of the output transformer 12, a leakage inductance Lleak of the output transformer 12, and a wiring inductance Lline-B between the LDMOS (11B) and the other end of the output transformer 12, or a combination of any one of them. The series inductance Ls1-B is connected as necessary according to the resonant frequency of the partial resonant circuit 21B.

[0069] The capacitance component CB of the LDMOS (11B) can be the parasitic capacitance Coss-B of the LDMOS (11B), or a combined capacitance Ctotal-B of the additional capacitance Cp-B and the parasitic capacitance Coss-B connected in parallel between the drain and source terminals of the LDMOS (11B). The additional capacitances Cp-A and Cp-B may be either lumped constants or distributed constants.

[0070] The resonant frequencies of the partial resonant circuits 21A and 21B reduce the impedance in the frequency range in which the impedance is high due to the partial resonant circuits, within the range from the fundamental frequency f0 to the third harmonic frequency 3×f0 of the high frequency output output from the secondary side of the output transformer 12, and set a predetermined time constant for the rise and fall of the drain-source voltage Vds of the LDMOS.

[0071] The drain-source voltage Vds is sloped to have a trapezoidal waveform with a predetermined time width at the rising and falling edges, thereby suppressing high frequency components that occur at the rising and falling edges. At this time, the partial resonant circuits 21A and 21B partially resonate in the vicinity of a frequency region where the impedance between the drains becomes high, thereby reducing the impedance.

[0072] (Series resonators and partial resonant circuits) The series resonator and the partial resonant circuit have different resonant frequencies that are suppressed by resonance. The resonant frequency of the series resonator is a frequency for suppressing surge voltage by reducing the harmonic frequency components contained in the high frequency output. On the other hand, the resonant frequency of the partial resonant circuit is a frequency for suppressing surge voltage by reducing impedance at a frequency that results in high impedance by making the drain-source voltage Vds a trapezoidal waveform with sloping rise and fall characteristics.

[0073] When the series resonator resonates, the impedance is only the resistance Rr, and a parallel damping circuit is formed by this resistance Rr, the inductance component Ls of the partial resonant circuit, and the capacitance component C. The damping rate ζ of this parallel damping circuit is ζ = (1 / (2 × Rr)) × (Ls / C), and the smaller the resistance Rr is, the larger the damping rate ζ becomes, thereby enhancing the damping effect.

[0074] The resonant frequency of the series resonator is set to the harmonic frequency bandwidth [n×f0min~n×f0max] according to the variable frequency range [f0min~f0max] when the fundamental frequency f0 is varied. Therefore, when the frequency of the high frequency output is changed, multiple resonant frequencies are set according to the changed frequency.

[0075] On the other hand, the resonant frequency of the partial resonant circuit is set so that the drain-source voltage Vds has a trapezoidal waveform with a sloped portion in a frequency range near the frequency at which the impedance becomes high. The inductance component Ls and the capacitance component C are set according to the set resonant frequency. The frequency range in which the resonant frequency is set does not vary significantly even if the frequency of the high-frequency output is changed. Therefore, even if the frequency of the high-frequency output is changed, there is no need to change the resonant frequency according to the changed frequency, and it is possible to set the resonant frequency at only one point.

[0076] (1-2: Second configuration example) The second configuration example of the switching amplifier 1, like the first configuration example, includes a switching section 11, an output transformer 12, a DC power supply 13, a narrow-band filter 15, an LCR series resonator 20, and a partial resonant circuit 21 (21A, 21B).

[0077] In the switching amplifier 2 of the second configuration example, two LCR series resonators 20A and 20B are connected in parallel between the drain terminal of the LDMOS (11A) and the drain terminal of the LDMOS (11B) and to the primary winding 12a of the output transformer 12.

[0078] The first LCR series resonator 20A and the second LCR series resonator 20B have a series circuit similar to that of the LCR series resonator 20 of the first configuration example, and the first LCR series resonator 20A is composed of a series circuit of inductance Lr1, capacitance Cr1, and resistance Rr1, and the second LCR series resonator 20B is composed of a series circuit of inductance Lr2, capacitance Cr2, and resistance Rr2.

[0079] The first LCR series resonator 20A and the second LCR series resonator 20B are set to have different resonant frequencies of the series circuits, thereby widening the frequency band in which harmonic frequency components of the load current are attenuated.

[0080] The first LCR series resonator 20A has a resonant frequency lower than the harmonic frequency n×f0 of the high frequency output relative to the fundamental frequency f0 of the high frequency output from the secondary side of the transformer, and the second LCR series resonator 20B has a resonant frequency higher than the harmonic frequency n×f0 of the high frequency output. By shifting the resonant frequencies of the LCR series resonators connected in parallel, the band of harmonic frequencies to be attenuated is expanded.

[0081] Here, two LCR series resonators are connected in parallel, but the number of LCR series resonators connected in parallel is not limited to two and may be any number, and the number of LCR series resonators connected in parallel and the frequency value of each resonant frequency are set according to the bandwidth of the harmonic frequency to be output. The suppression of harmonic frequencies by the LCR series resonator 20 will be described later.

[0082] (1-3: Comparison between LCR series resonator and LC series resonator) The surge voltage suppression effect of an LCR series resonator will be explained in comparison with an LC series resonator. Figure 3A shows the attenuation characteristics of a configuration in which two LC series resonators are connected in parallel. The attenuation characteristics of Figure 3A show the characteristics of a configuration in which two LCR series resonators are connected in parallel, as represented by the equivalent circuit shown in Figure 3B.

[0083] In Fig. 3A, the solid line indicates the pass characteristic of the LCR series resonator, and the dashed line indicates the pass characteristic of the LC series resonator, both of which are set to resonate at the same frequency. In Fig. 3A, the pass characteristic is shown by the attenuation rate versus frequency, with the attenuation rate increasing downward on the vertical axis.

[0084] Since the drain-source voltage Vds contains odd-order harmonics, the resonant frequencies of the LCR series resonator and the LC series resonator are designed to resonate at frequencies two to four times the fundamental frequency f0, including a frequency about three times the bandwidth (f0-min to f0-max). In Fig. 3A, the attenuation frequency region is the frequency range in which the impedance of the series resonator appears to be 50Ω [ohm] or less, and the attenuation rate at which the amplitude becomes 1 / 2 is -6dB or less.

[0085] By using two series resonators with different frequency ranges, the frequency range of the resonant frequency can be broadened. In order to broaden the frequency range of the resonant frequency, an LC series resonator is inappropriate, whereas an LCR series resonator is suitable.

[0086] According to the pass characteristic of the LC series resonator without resistor Rr, shown by the dashed line in Fig. 3A, there is no attenuation effect at frequencies including the third harmonic ((2-4) x f0). If there is a frequency range in which no attenuation effect is achieved, frequency components in that frequency range appear in the drain-source voltage Vds, which can cause a surge voltage to occur. If the surge voltage exceeds the withstand voltage of an element such as an LDMOS, it can cause the element to be discarded.

[0087] On the other hand, according to the pass characteristic of the LCR series resonator equipped with resistor Rr shown by the solid line in FIG. 3A, there is a sufficient attenuation rate even at frequencies including the third harmonic ((2 to 4) × f0), and it is possible to provide an attenuation effect over a wide frequency range.

[0088] (2) Example of switching amplifier operation An example of the operation of the switching amplifier according to the present invention will be described with reference to Figures 4 to 8. Figure 4 shows signals at each part of the switching amplifier, and Figures 5 to 8 show an example of the operation in each section of the switching amplifier.

[0089] Fig. 4A shows the gate-source voltage VgsA of the LDMOS (11A), and Fig. 4B shows the gate-source voltage VgsB of the LDMOS (11B). Fig. 4C shows the drain-source voltage VdsA of the LDMOS (11A), and Fig. 4D shows the drain-source voltage VdsB of the LDMOS (11B). Fig. 4E shows the current IdsA flowing from the LDMOS (11A) to the primary winding 12a of the output transformer 12, and the current IdsB flowing from the LDMOS (11B) to the primary winding 12a of the output transformer 12. The current IdsA is indicated by a solid line, and the current IdsB is indicated by a dashed line.

[0090] The LDMOS (11A) and LDMOS (11B) switch between on and off with the gate-source voltages VgsA and VgsB at the threshold voltage, and the gate-source voltages VgsA and VgsB switch in opposite phases to each other, thereby performing push-pull operation. In order to avoid a period in which the two LDMOSs (11A) and (11B) constituting the push-pull circuit are simultaneously in the on state, a dead time period is provided in which both gate-source voltages become 0V when each gate-source voltage switches, and the two LDMOSs (11A) and (11B) are both in the off state.

[0091] In the following, an example of the operation of the LDMOS (11A) and the LDMOS (11B) will be described by dividing it into eight sections, sections T1 to T8. Sections T1 and T5 are dead time sections in which both LDMOS (11A) and (11B) are in the off state. Sections T2, T3, and T4 are sections in which LDMOS (11A) is in the off state and LDMOS (11B) is in the on state. Meanwhile, sections T6, T7, and T8 are sections in which LDMOS (11A) is in the on state and LDMOS (11B) is in the off state.

[0092] Sections T1 and T2 are transition sections in which the drain-source voltage VdsA of the LDMOS (11A) rises as the LDMOS (11A) is turned off, and sections T5 and T6 are transition sections in which the drain-source voltage VdsB of the LDMOS (11B) rises as the LDMOS (11B) is turned off.

[0093] Sections T4 and T5 are transition sections in which the drain-source voltage VdsA of the LDMOS (11A) drops due to resonance of the LDMOS (11A) and turning off of the LDMOS (11B), and sections T8 and T1 are transition sections in which the drain-source voltage VdsB of the LDMOS (11B) drops due to resonance of the LDMOS (11B) and turning off of the LDMOS (11A).

[0094] (Section T1) Section T1 is a dead time section in which both the LDMOS 11A and the LDMOS 11B are in the off state. Note that section T1 is a state in which the LDMOS 11A is turned off from the on state.

[0095] The operating state of section T1 will be described with reference to Fig. 5A. When the LDMOS (11A) is turned off, a current IdsA flows toward the LDMOS (11A) side in the partial resonant circuit 21A on the LDMOS (11A) side due to the energy of the primary winding 12a. The capacitance component CA of the partial resonant circuit 21A is charged by the current IdsA. As the capacitance component CA is charged, the drain-source voltage VdsA rises with a time constant τr determined by the LC constant of the capacitance component CA and the inductance component Ls-A.

[0096] The drain-source voltage VdsA rises with a time constant τr, so that the rising edge of the drain-source voltage VdsA can be delayed. By delaying the rising edge of the drain-source voltage VdsA, the high-frequency components generated by the steep rise of the drain-source voltage VdsA are reduced, and the surge voltage at the rising edge is suppressed.

[0097] At this time, the capacitance component CA of the partial resonant circuit 21A is connected in series with the capacitance component CB of the partial resonant circuit 21B. This series connection reduces the LC constant, and the rise time tr of the drain-source voltage VdsA becomes slightly shorter than the fall time tf.

[0098] On the other hand, in the partial resonant circuit 21B on the LDMOS (11B) side, a current IdsB flows from the LDMOS (11B) side toward the DC power supply 13 side, due to the current IdsA of the partial resonant circuit 21A. The charge stored in the capacitance component CB of the partial resonant circuit 21B is discharged by the flow of the current IdsB.

[0099] As a result of the capacitance component CB being discharged, the drain-source voltage VdsB drops toward 0 V with a time constant τf determined by the LC constant of the capacitance component CA and the series inductance Ls1-B.

[0100] (Section T2) In the section T2, the LDMOS (11A) is in the off state and the LDMOS (11B) is in the on state.

[0101] The operating state of section T2 will be described with reference to Fig. 5B. The LDMOS (11A) is in the off state, and due to the resonance of the partial resonant circuit 21A on the LDMOS (11A) side, the charging voltage of the drain-source voltage VdsA increases with a time constant τr, and the current IdsA decreases.

[0102] As in section T1, in section T2, the drain-source voltage VdsA rises with a time constant τr, delaying the rise of the drain-source voltage VdsA. This reduces the high-frequency components that are generated when the drain-source voltage VdsA rises sharply, and suppresses the surge voltage at the time of rise. Sections T1 and T2 cause the rising waveform of the drain-source voltage VdsA to become a trapezoidal waveform.

[0103] Since the LDMOS (11B) is in the on state, the drain-source voltage VdsB is maintained at 0 V. The current IdsB of the partial resonant circuit 21B switches from the current direction in the section T1 to flow from the DC power supply 13 toward the ground side.

[0104] (Section T3) In the section T3, the LDMOS (11A) is in the off state and the LDMOS (11B) is in the on state.

[0105] The operating state of section T3 will be described with reference to Fig. 6A. Since LDMOS (11B) is in the on state, current IdsB starts to flow toward LDMOS (11B), but since LDMOS (11A) is in the off state, the charge of capacitance component CA of LDMOS (11A) is maintained in a charged state, and the peak value of drain-source voltage VdsA is maintained almost constant.

[0106] Thereafter, on the LDMOS (11A) side, the current IdsA is attracted by the current IdsB and flows from the LDMOS (11A) toward the DC power supply 13, and the drain-source voltage VdsA oscillates due to resonance by the partial resonant circuit 21A as well as resonance including the inductance component of the primary winding 12a of the output transformer 12. In the middle period of the section T3 accompanied by resonant oscillation, the peak value of the drain-source voltage VdsA is almost constant, and the surge voltage generated by a sudden change in the load current is suppressed by the partial resonance.

[0107] In the latter part of the section T3, the charge of the capacitance component CA of the LDMOS (11A) changes to discharge, and the peak value of the drain-source voltage VdsA starts to decrease. A load current Iload accompanied by harmonic frequency components flows through the secondary winding 12b of the output transformer 12 as a result of the current IdsB flowing through the primary winding 12a. The load current Iload passes through the narrow-band filter 15 and flows from the output terminal 16 to the load 17. Meanwhile, the harmonic frequency components are reflected by the narrow-band filter 15 and flow through the LCR series resonator 20 via the output transformer 12, and then flow through the LDMOS (11B) in the on-state. At this time, the harmonic frequency components are reduced by the resistance Rr of the LCR series resonator 20, and the surge voltage generated by the harmonic frequency components is suppressed.

[0108] (Section T4) In section T4, like section T3, the LDMOS (11A) is in the OFF state and the LDMOS (11B) is in the ON state.

[0109] The operating state of section T4 will be described with reference to Fig. 6B. In section T4, following the latter half of section T3, the drain-source voltage VdsA shows a downward trend due to resonance including the inductance component of the primary winding 12a of the output transformer 12, and the surge voltage is suppressed by resonance of the partial resonant circuit 21A.

[0110] (Section T5) Section T5 is a dead time section in which both the LDMOS (11A) and the LDMOS (11B) are in the off state, and the LDMOS (11B) is turned off from the on state.

[0111] The operating state of section T5 will be described with reference to Fig. 7A. When the LDMOS (11B) is turned off, a current IdsB flows toward the LDMOS (11B) side in the partial resonant circuit 21B on the LDMOS (11B) side due to the energy of the primary winding 12a.

[0112] In the partial resonant circuit 21A on the LDMOS (11A) side, a current IdsA flows from the LDMOS (11A) side toward the DC power supply 13 side due to the current IdsB of the partial resonant circuit 21B. The flow of the current IdsA discharges the charge stored in the capacitance component CA of the partial resonant circuit 21A. As the capacitance component CA is discharged, the drain-source voltage VdsA drops toward 0V with a time constant τf.

[0113] On the other hand, the capacitance component CB of the partial resonant circuit 21B is charged by the current IdsB. As the capacitance component CB is charged, the drain-source voltage VdsB rises with a time constant τr determined by the LC constant of the capacitance component CB and the inductance component Ls-B.

[0114] The drain-source voltage VdsB rises with a time constant τr, so that the rising edge of the drain-source voltage VdsB can be delayed. By delaying the rising edge of the drain-source voltage VdsB, the high-frequency components generated by the steep rise of the drain-source voltage VdsB are reduced, and the surge voltage at the rising edge is suppressed.

[0115] At this time, the capacitance component CB of the partial resonant circuit 21B is connected in series with the capacitance component CA of the partial resonant circuit 21A. This series connection reduces the LC constant, and the rise time tr of the drain-source voltage VdsB becomes slightly shorter than the fall time tf.

[0116] (Section T6) In a section T6, the LDMOS (11A) is in an on state and the LDMOS (11B) is in an off state.

[0117] The operating state of section T6 will be described with reference to Fig. 7B. The drain-source voltage VdsA is maintained at 0 V because the LDMOS (11A) is in the on state. The current IdsA of the partial resonant circuit 21A switches from the current direction of section T5 and flows from the DC power supply 13 to the ground side.

[0118] The LDMOS (11B) is in an off state, and due to resonance of the partial resonant circuit 21B on the LDMOS (11B) side, the charging voltage of the drain-source voltage VdsB increases with a time constant τr, and the current IdsB decreases.

[0119] As in section T5, in section T6, the drain-source voltage VdsB rises with a time constant τr, delaying the rise of the drain-source voltage VdsB. This reduces the high-frequency components that are generated by the steep rise of the drain-source voltage VdsB, and suppresses the surge voltage at the time of the rise. Sections T5 and T6 make the rising waveform of the drain-source voltage VdsB a trapezoidal waveform.

[0120] (Section T7) In a section T7, the LDMOS (11A) is in an on state and the LDMOS (11B) is in an off state.

[0121] The operation state of section T7 will be described with reference to Fig. 8A. Since LDMOS (11A) is in the on state, current IdsA starts to flow toward LDMOS (11A), but since LDMOS (11B) is in the off state, the charge of capacitance component CB of LDMOS (11B) is maintained in a charged state, and the peak value of drain-source voltage VdsB is maintained almost constant.

[0122] Thereafter, on the LDMOS (11B) side, the current IdsB is attracted by the current IdsA and flows from the LDMOS (11AB) to the DC power supply 13, and the drain-source voltage VdsB oscillates due to resonance by the partial resonant circuit 21B as well as resonance including the inductance component of the primary winding 12a of the output transformer 12. In the middle period of the section T7 accompanied by resonant oscillation, the peak value of the drain-source voltage VdsB is almost constant, and the surge voltage generated by a sudden change in the load current is suppressed by the partial resonance.

[0123] In the latter part of section T7, the charge of the capacitance component CB of the LDMOS (11B) changes to discharge, and the peak value of the drain-source voltage VdsB tends to decrease. A load current Iload accompanied by harmonic frequency components flows in the secondary winding 12b of the output transformer 12 due to the current IdsA flowing in the primary winding 12a. The load current Iload passes through the narrow-band filter 15 and flows from the output terminal 16 to the load 17. Meanwhile, the harmonic frequency components are reflected by the narrow-band filter 15 and flow to the LCR series resonator 20 via the output transformer 12, and then flow to the LDMOS (11A) in the on-state. At this time, the harmonic frequency components are reduced by the resistance Rr of the LCR series resonator 20, and the surge voltage generated by the harmonic frequency components is suppressed.

[0124] (Section T8) In section T8, like section T7, the LDMOS (11A) is in the ON state and the LDMOS (11B) is in the OFF state.

[0125] The operating state in section T8 will be described with reference to Fig. 8B. In section T8, following the latter half of section T7, the drain-source voltage VdsB shows a downward trend due to resonance including the inductance component of the primary winding 12a of the output transformer 12, and the surge voltage is suppressed by resonance of the partial resonant circuit 21B.

[0126] (3) Trapezoidal waveform The trapezoidal waveform of the drain-source voltage of the switching element according to the present invention will be described. The trapezoidal waveform shown in Fig. 9 is the drain-source voltage VdsA of the LDMOS (11A) between sections T1 and T5, and shows two resonances by the partial resonant circuit 21 and the LCR series resonator 20, and the surge voltage suppression by the two resonances.

[0127] (Suppression of the first surge voltage) In the sections T1 to T5 and T6 to T8, the partial resonant circuit 21 reduces high-frequency components contained in the load current, and suppresses the first surge voltage generated by the high-frequency components.

[0128] The load current flows as a current IdsA in the partial resonant circuit 21A, and flows as a current IdsB in the partial resonant circuit 21B.

[0129] In sections T1 and T2, the current IdsA charges the capacitance component CA of the partial resonant circuit 21A in both sections T1 and T2, and the current IdsB discharges the capacitance component CB of the partial resonant circuit 21B in section T1.

[0130] In section T3, the current IdsA changes the capacitance component CA of the partial resonant circuit 21A from a charged state to a discharged state.

[0131] In sections T4 and T5, the current IdsA discharges the capacitance component CA of the partial resonant circuit 21A in both sections T1 and T2, and the current IdsB charges the capacitance component CB of the partial resonant circuit 21B in section T5.

[0132] In the partial resonant circuit 21, the drain-source voltage Vds rises and falls in a trapezoidal shape with a rise time tr and a fall time tf due to charging and discharging in the sections T1, T2 and T4, T5, reducing the high-frequency components contained in the current Ids and suppressing the first surge voltage generated by the high-frequency components. In the partial resonant circuit 21, the high-frequency components contained in the current Ids are reduced by the resistance of the load impedance in association with the resonant operation in the section T3, suppressing the first surge voltage generated by the high-frequency components.

[0133] (Suppression of the second surge voltage) In section T3, the partial resonant circuit 21 reduces the harmonic frequency components contained in the load current, and suppresses the second surge voltage generated by the harmonic frequency components.

[0134] In section T3, a load current flows through the second side of the output transformer 12. The load current includes a frequency component of the fundamental frequency f0 and a frequency component of the harmonic frequency n×f0. By setting the resonant frequency of the LCR series resonator 20 to correspond to the harmonic frequency n×f0, the LCR series resonator 20 captures the frequency component of the harmonic frequency n×f0 reflected by the narrowband filter and dissipates and attenuates it through the resistor Rr. In this way, the LCR series resonator 20 suppresses the second surge voltage generated by the harmonic frequency component.

[0135] (4) Wideband The broadening of the output band of a switching amplifier according to the present invention will be described below. The broadening of the output band in partial resonance and in an LCR series resonator will be described below.

[0136] (Wideband partial resonance) The broadening of the output band by partial resonance will be described with reference to Figures 10 and 11. Figure 10 shows an example of a waveform when the fundamental frequency f0 is 45 MHz, and Figure 11 shows an example of a waveform when the fundamental frequency f0 is 35 MHz.

[0137] Also, Figures 10A and 11A show currents IdsA and IdsB, Figures 10B and 11B show drain-source voltages VdsA and VdsB, Figures 10C and 11C show gate-source voltages VgsA and VgsB, and Figures 10D and 11D show voltages VR1 and VR2 across resistor Rr of the LCR DC resonator.

[0138] The resonant frequency of the partial resonant circuit is designed to be in the range of 1 to 3 times the fundamental frequency f0, and preferably about 1.5 to 2.5 times. When the resonant frequency of the partial resonant circuit is set in the above frequency range, even when the fundamental frequency f0 is varied, it is desired that the total loss, taking into account the switching loss and the increase in loss opportunity due to the increase in frequency, is suitable for the output of the switching amplifier.

[0139] The waveform examples in FIG. 10 and FIG. 11 show examples in which the fundamental wave frequency f0 is set to 40 MHz and the frequency is varied between 35 MHz and 45 MHz.

[0140] When the frequency is 35 MHz, there is a slight voltage rise in the waveform of the residual drain-source voltage Vds just before the LDMOS switches from the off state to the on state due to partial resonance. However, although this voltage rise increases the switching loss in one cycle, the effect on the total loss is small because the frequency is low.

[0141] On the other hand, when the frequency is 45MHz, the LDMOS gate-source voltage Vgs is applied when the drain-source voltage Vds is near its lowest point, so switching is performed by ZVS, and the switching loss per cycle is reduced. Although the frequency increases, the effect on the overall loss is small because the switching loss is reduced. When the partial resonance frequency exceeds three times the fundamental frequency f0, the residual drain-source voltage Vds increases further, resulting in a decrease in overall efficiency.

[0142] (Wideband LCR series resonator) The broadening of the output bandwidth by the LCR series resonator will be explained with reference to Fig. 12. Fig. 12 shows a resonance curve expressed by admittance or current.

[0143] FIG. 12A shows a resonance curve when the LCR series resonator is configured as one series circuit, and FIG. 12B shows a resonance curve when the LCR series resonator is configured as two series circuits connected in parallel.

[0144] The curve on the left in FIG. 12A represents the passband of the variable frequency band, with the lower limit frequency f0-min and upper limit frequency f0-max for the fundamental frequency f0 being set by frequencies close to the maximum value 1 of the resonance curve.

[0145] For the passband of the variable frequency band, the curve on the right side of Fig. 12A is the resonance curve of the series circuit of the harmonic frequency band when the LCR series resonator is configured as one series circuit. The resonance curve of the series circuit of the harmonic frequency band is set so that the lower limit frequency n×f0-min and the upper limit frequency n×f0-max for the harmonic frequency n×f0 (n is an integer) are set in a frequency range where the series resonator of the series circuit of the harmonic frequency band is 50Ω [ohm] or less (standard 1) and the current flowing through the series resonator is 1 or more times.

[0146] The lower limit frequency n×f0-min and the upper limit frequency n×f0-max set the frequency range of the harmonic frequency band. This frequency range is set, for example, to ±1.5% or more of the harmonic frequency n×f0, totaling 3% or more. By setting this harmonic frequency band, even if the frequency changes within the variable frequency band, the admittance or current will be more than 1 times the standard of 50Ω [ohm], and high loss (attenuation) can be expected, making it possible to widen the range.

[0147] The curve on the left side of Figure 12B, like the curve on the left side of Figure 12A, represents the passband of the variable frequency band, and the lower limit frequency f0-min and upper limit frequency f0-max for the fundamental frequency f0 are set by frequencies that are close to the maximum value of the resonance curve, 1.

[0148] In contrast to the resonance curves of the variable frequency band, the two curves on the right side of Fig. 12B are resonance curves of the series circuit of the harmonic frequency band when the LCR series resonator is configured with two series circuits. The resonance curve of the series circuit of the harmonic frequency band is configured by shifting the center frequencies f1 and f2 using the two resonance curves of the series circuit of the harmonic frequency band on the right side of Fig. 12A.

[0149] The resonance curve of the series circuit in one harmonic frequency band is set to a center frequency f1 shifted to the lower frequency side, and the resonance curve of the series circuit in the other harmonic frequency band is set to a center frequency f2 shifted to the higher frequency side, and the two resonance curves are set at a frequency at which the current flowing through the series resonator in the harmonic frequency band, which is a combination of the lower limit frequency f0-min and the upper limit frequency f0-max for the harmonic frequency n × f0 (n is an integer), is equal to or greater than 1.

[0150] The lower limit frequency n×f0-min and upper limit frequency n×f0-max set as the frequency range of the harmonic frequency band are set to, for example, ±3% or more of the harmonic frequency n×f0, and a total of 6% or more. By using two series circuits, the frequency range of the harmonic frequency band can be expanded. By setting this harmonic frequency band, even if the frequency changes within the variable frequency band, the admittance or current is more than 1 times the standard of 50Ω [ohm], and high loss (attenuation) can be expected, making it possible to widen the range.

[0151] (Partial resonance frequency band and high impedance frequency band) The relationship between the partial resonance frequency band and the high impedance frequency band will be described with reference to Fig. 13. Fig. 13A shows a Smith chart, and Fig. 13B shows a frequency characteristic diagram of the attenuation rate.

[0152] The Smith chart in Fig. 13A shows a constant SWR circle S that indicates the reflection state with the load. When the frequency is swept, the impedance is represented by a locus that moves clockwise from the low frequency side to the high frequency side on the constant SWR circle S. In the constant SWR circle S shown in Fig. 13A, in the frequency range from the fundamental frequency f0 to the harmonic frequency n x f0, there is a high impedance band fb where the impedance is high between the fundamental frequency f0 and the harmonic frequency n x f0.

[0153] Therefore, a switching amplifier with a variable frequency output range from the fundamental frequency f0 to the harmonic frequency n×f0 has two frequencies and frequency ranges where the impedance is high: the high impedance band fb and the harmonic frequency n×f0. High impedance can cause surge voltages.

[0154] In order to suppress the surge voltage in the high impedance band fb, the impedance in the frequency range near the high impedance band fb is reduced. In the present invention, the partial resonant frequency band FB2 is set by a partial resonant circuit to reduce the impedance and suppress the surge voltage in the high impedance band fb.

[0155] In order to suppress surge voltages in a frequency range centered on the harmonic frequency n×f0, the impedance in this frequency range is reduced. In the present invention, the LCR series resonator band FB3 is set by the LCR series resonator to reduce the impedance and suppress surge voltages in the frequency range centered on the harmonic frequency n×f0.

[0156] Fig. 13B shows the attenuation rate vs. frequency characteristic of the Smith chart of Fig. 13A. The partial resonant frequency band FB2 corresponds to the high impedance band fb, and the LCR series resonator band FB3 corresponds to the frequency range centered on the harmonic frequency n×f0. By setting the attenuation rate high in the partial resonant frequency band FB2 and the LCR series resonator band FB3, the surge voltage is suppressed. [Industrial Applicability]

[0157] A switching amplifier according to the present invention can be applied to high-frequency power sources (RF generators) used in semiconductor manufacturing equipment, liquid crystal panel manufacturing equipment, and the like. [Explanation of symbols]

[0158] 1,2 Switching Amplifier 10A signal source 10B signal source 11 Switching section 12 Output transformer 12a primary winding 12b Secondary Winding 13 DC power supply 14 Push-pull circuit 15 Narrowband Filters 16 Output terminal 17 Load 20, 20a, 20b LCR series resonator 21,21A,21B partial resonant circuit 101a, 101b Switching amplifier 110A,110B signal source 111 Switching section 111A, 111B Switching element 112 Output transformer 112a Primary Winding 112b Secondary Winding 113 DC power supply 114 Push-pull circuit 115 Narrow Band Filter 116 Output terminal 117 Load C, CA, CB Capacitance components Coss-A, Coss-B parasitic capacitance Cp-A, Cp-B additional capacity Cr,Cr1,Cr2 Capacitance Ctotal-A,Ctotal-B combined capacity FB2 partial resonance frequency band FB3 LCR series resonator band Ids, IdsA, IdsB current Iload Load current Lline-A, Lline-B wiring inductance Lr, Lr1, Lr2 inductance Lleak Leakage inductance Ls, Ls-A, Ls-B Inductance components Ls1-A, Ls1-B Series inductance Pg Gate signal R1 Primary resistance Rr,Rr1,Rr2 Resistance S Constant SWR Circle T1, T2, T3, T4, T5, T6, T7, T8 Section Vds, VdsA, VdsB Drain-source voltage Vgs, VgsA, VgsB Gate-source voltage f0 Fundamental frequency f0-max Upper limit frequency f0-min Lower limit frequency f1,f2 center frequency fb High impedance band τf, τr time constants

Claims

1. a transformer-coupled push-pull circuit including a pair of switching elements which alternately perform on / off switching operations in response to input signals of opposite phases, a primary winding to both ends of which is input with an output signal of the switching element, a secondary winding to which output signals of opposite phases are taken out, and an output transformer having an external power supply connected to a center tap of the primary winding; a narrow-band filter connected to an output end of the secondary winding of the output transformer; a pair of partial resonant circuits each including an inductance component on the drain side of each of the switching elements and a capacitance component between the drain and source of each of the switching elements; an LCR series resonator configured by a series circuit of an inductance, a capacitance, and a resistance, connected between drain terminals of the switching elements; A switching amplifier comprising:

2. an inductance component included in each partial resonant circuit is any one of a series inductance connected in series between each switching element and one end of the output transformer, a leakage inductance of the output transformer, and a wiring inductance between each switching element and one end of the output transformer, or any combination thereof; The capacitance component of each partial resonant circuit is an output parasitic capacitance of the switching element, or a combined capacitance of the output parasitic capacitance of the switching element and an additional capacitance connected in parallel to the switching element.

2. The switching amplifier of claim 1.

3. The resonant frequency of each partial resonant circuit is In the range of the fundamental frequency f0 to the third harmonic frequency (3×f0) of the high frequency output output from the secondary side of the output transformer, The partial resonant circuit is set to reduce the impedance in a frequency range where the impedance is high, and The frequency at which the time constants of the rise and fall of the drain-source voltage of the switching element are set as the set values.

2. The switching amplifier of claim 1.

4. With respect to the fundamental frequency f0 of the high frequency output output from the secondary side of the output transformer, The resonant frequency of the LCR series resonator is a harmonic frequency n×f0 (n is an integer), The bandwidth of the resonant frequency is set to a frequency range of (n×f0-min to n×f0-max) according to the variable frequency range (f0-min to f0-max) of the fundamental frequency f0; 2. The switching amplifier of claim 1.

5. A plurality of the LCR series resonators having different resonant frequencies are connected in parallel.

2. The switching amplifier of claim 1.

6. With respect to the fundamental frequency f0 of the high frequency output output from the secondary side of the output transformer, a first LCR series resonator having a resonant frequency on the lower frequency side than a harmonic frequency n×f0 of the high frequency output; a second LCR series resonator having a resonant frequency on the higher frequency side than the harmonic frequency n×f0 of the high frequency output; 6. The switching amplifier of claim 5.

7. The switching element is a DMOS.

7. A switching amplifier according to claim 1.

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