Laser phase tracking method and system for internal reflection subtraction in phase encoding lidar

Abstract

To provide a system and a method related to photodetection and distance measurement in an optical field, for Doppler compensation and internal reflection subtraction when an autonomous vehicle is operated.SOLUTION: A system and a method for photodetection in autonomous vehicle include the steps of: modulating an optical signal from a laser to generate a modulated optical signal; and transmitting the modulated optical signal to an object. The system and method include the steps of: receiving a returned optical signal in response to the step of transmitting the modulated optical signal; mixing the returned optical signal with a reference optical signal related to the optical signal from the laser to generate the mixed optical signal; and detecting the mixed optical signal to generate an electric signal. On the basis of the electric signal and the modulated optical signal, internal reflection parameter of the returned optical signal from one or more optical components is determined. The parameter can be used for operating a vehicle.SELECTED DRAWING: Figure 2

Description

[Technical field]

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS This application claims priority to and the benefit of U.S. Provisional Patent Application No. 62 / 760,437, filed November 13, 2018. This application also claims priority to and the benefit of U.S. Provisional Patent Application No. 62 / 783,749, filed December 21, 2018. The entire disclosures of U.S. Provisional Patent Application No. 62 / 760,437 and U.S. Provisional Patent Application No. 62 / 783,749 are incorporated herein by reference in their entireties. [Background technology]

[0002] Optical detection of distance using lasers, often referred to by the mnemonic LIDAR and sometimes also referred to as laser RADAR (Radio-Wave Detection and Ranging), for light detection and ranging, is used for a variety of applications ranging from altitude measurement to imaging to collision avoidance. LIDAR provides finer scale range distance resolution with smaller beam sizes than conventional microwave ranging systems such as RADAR. Optical detection of distance can be achieved with several other techniques including direct distance measurement based on the round trip travel time of a light pulse about an object, chirp detection based on the frequency difference between a transmitted chirped light signal and the signal scattered back from the object, and phase-encoded detection based on a sequence of single frequency phase changes distinguishable from natural signals. Summary of the Invention [Problem to be solved by the invention]

[0003] Aspects of the present disclosure relate generally to Light Detection and Ranging (LIDAR) in the field of optics, and more particularly to systems and methods for Doppler Compensation and Internal Reflection Subtraction during operation of an Autonomous Vehicle.

[0004] One disclosed embodiment relates to a system for internal reflection subtraction. In some embodiments, the system includes a laser source providing an optical signal. In some embodiments, the system includes a modulator modulating the optical signal from the laser to generate a modulated optical signal. In some embodiments, the system includes an optical coupler generating a Reference Optical Signal related to the optical signal from the laser, transmitting the modulated optical signal toward an object, and receiving a returned optical signal responsive to transmitting the modulated optical signal toward the object. In some embodiments, the system includes an optical mixer mixing the returned optical signal with a reference optical signal to generate a mixed optical signal, the reference optical signal being related to the optical signal from the laser. In some embodiments, the system includes an optical detector detecting the mixed optical signal to generate an electrical signal. In some embodiments, the system includes a processor determining a parameter of internal reflection of the returned optical signal from one or more optical components based on the electrical signal and the modulated optical signal, and operating a vehicle based on the parameter of internal reflection.

[0005] In another aspect, the present disclosure relates to a method for internal reflection subtraction. The method includes modulating an optical signal from a laser to generate a modulated optical signal. In some embodiments, the method includes transmitting the modulated optical signal toward an object. In some embodiments, the method includes receiving a returned optical signal in response to transmitting the modulated optical signal. In some embodiments, the method includes mixing the returned optical signal with a reference optical signal related to the optical signal from the laser to generate a mixed optical signal. In some embodiments, the method includes detecting the mixed optical signal to generate an electrical signal. In some embodiments, the method includes determining a parameter of internal reflection of the returned optical signal from one or more optical components based on the electrical signal and the modulated optical signal. In some embodiments, the method includes operating a vehicle based on the parameter of internal reflection.

[0006] In another aspect, the present disclosure relates to a system for Doppler compensation. In some embodiments, the system includes a laser source providing an optical signal at a carrier frequency. In some embodiments, the system includes a phase modulator. In some embodiments, the system includes an optical mixer. In some embodiments, the system includes a first optical detector. In some embodiments, the system includes a processor that controls the phase modulator to generate an optical output signal including an Optical Broadband Phase-Encoded Optical Signal. In some embodiments, the processor receives the returned optical signal and receives a real part of the mixed optical signal from the first optical detector as a result of mixing the returned optical signal with a reference optical signal. In some embodiments, the output optical signal or the reference optical signal includes a pilot tone having a pilot frequency offset by a known signed non-zero frequency from the carrier frequency. In some embodiments, the processor determines a peak frequency of the returned optical signal based at least in part on a Fourier transform of a digital signal including the real part of the mixed optical signal. In some embodiments, the processor determines a signed Doppler Frequency Shift of the returned optical signal based on the peak frequency and the pilot frequency, and in some embodiments, the processor operates an apparatus based on the signed Doppler Frequency Shift.

[0007] In another aspect, the present disclosure relates to a method for Doppler compensation. In some embodiments, the method includes generating an optical output signal by phase modulation of an optical signal at a carrier frequency from a laser using a wideband RF phase encoding signal. In some embodiments, the output optical signal includes an optical wideband phase encoding optical signal. In some embodiments, the method includes transmitting the output optical signal. In some embodiments, the method includes receiving a returned optical signal in response to transmitting the output optical signal. In some embodiments, the method includes outputting a mixed optical signal by mixing the returned optical signal with a reference optical signal, where the output optical signal or the reference optical signal includes a pilot tone having a pilot frequency offset from the carrier frequency by a non-zero frequency having a known code. In some embodiments, the method includes detecting a real part of the mixed optical signal with a first optical detector. In some embodiments, the method includes determining, on a processor, a peak frequency of the returned optical signal based at least in part on a Fourier transform of a digital signal including the real part of the mixed optical signal. In some embodiments, the method includes determining, on a processor, a Doppler frequency shift having a sign of the returned optical signal based on the peak frequency and the pilot frequency, In some embodiments, the method includes operating an apparatus based on the Doppler frequency shift having a sign.

[0008] Still other aspects, features and advantages will become readily apparent from the following detailed description, by simply illustrating a number of specific embodiments and implementations, including the best modes contemplated for carrying out the present invention. Also, other embodiments may have other different features and advantages, and many of these details may be modified in various obvious aspects, all without departing from the spirit and scope of the present invention. For example, any components discussed herein for Doppler compensation and / or internal reflection subtraction may be combined as a single embodiment performing the operation and / or function of the respective components. Any operations for any method of Doppler compensation and / or internal reflection subtraction as discussed herein may be combined as a single embodiment. Thus, the drawings and descriptions should be regarded as illustrative, rather than restrictive, in nature. [Brief description of the drawings]

[0009] BRIEF DESCRIPTION OF THE DRAWINGS Embodiments are illustrated in the accompanying drawings, in which: FIG.

[0010] [Figure 1a] 1 is a schematic graph illustrating an exemplary transmitted optical phase encoding signal for distance measurement according to one embodiment; [Figure 1b] 1 is a schematic graph illustrating the example transmitted optical signal of FIG. 1a together with a returned optical signal as a series of binary digits for distance measurement of FIG. 1a according to one embodiment; [Figure 1c] 1 is a schematic graph illustrating an example cross-correlation of a reference signal with two returned signals according to one embodiment; [Diagram 2] FIG. 1 is a block diagram illustrating example components of a high-resolution LIDAR system according to one embodiment; [Diagram 3]FIG. 1 is a block diagram illustrating exemplary components of a Doppler Compensated Phase-Encoded LIDAR System using In-Phase / Quadrature (I / Q) processing according to one embodiment; [Figure 4a] 1 is a schematic graph illustrating an example spectrum of an output optical signal according to one embodiment; [Figure 4b] 1 is a schematic graph illustrating an example spectrum of a Doppler shifted optical return signal when IQ processing is performed according to one embodiment; [Figure 4c] 1 is a schematic graph illustrating an example spectrum of a Doppler shifted optical return signal when no IQ processing is performed according to one embodiment; [Figure 5a] FIG. 1 is a block diagram illustrating an example Doppler compensated LIDAR system using non-carrier pilot tones in accordance with one embodiment; [Figure 5b] FIG. 1 is a block diagram illustrating an example Doppler compensated LIDAR system using non-carrier pilot tones in accordance with one embodiment; [Figure 6a] 1 is a schematic graph illustrating an example spectrum of an SSB output optical signal according to one embodiment; [Figure 6b] 1 is a schematic graph illustrating an example spectrum of a Doppler shifted SSB optical return signal when IQ processing is performed according to one embodiment; [Figure 6c] 1 is a schematic graph illustrating an example spectrum of a Doppler shifted SSB optical return signal when no IQ processing is performed according to one embodiment; [Figure 7] 1 is a flow chart illustrating an example operation for using a Doppler compensated SSB phase encoding LIDAR system with codes according to one embodiment; [Figure 8] 1 is a graph illustrating an example transmitted optical signal having non-carrier pilot tones according to one embodiment; [Figure 9a] 1 is a flowchart illustrating an exemplary method for operating a phase-encoding LIDAR system to determine and compensate for internal reflections, according to one embodiment; [Figure 9b] 1 is a flowchart illustrating an exemplary method for operating a phase-encoding LIDAR system to determine and compensate for internal reflections, according to one embodiment; [Figure 10a] 1 is a graph illustrating an example range signal before and after correction to remove returns from internal reflections according to one embodiment; [Figure 10b] 1 is a graph illustrating an example range signal before and after correction to remove returns from internal reflections according to one embodiment; [Figure 10c] 1 is a graph illustrating an example range signal before and after correction to remove returns from internal reflections according to one embodiment; [Figure 10d] 1 is a graph illustrating an example range signal before and after correction to remove returns from internal reflections according to one embodiment; [Figure 10e] 1 is a graph illustrating an example range signal before and after correction to remove returns from internal reflections according to one embodiment; [Figure 11] 1 is a flowchart illustrating an exemplary method for phase tracking internal reflections before subtraction according to one embodiment; [Figure 12] 1 is a flow chart illustrating an exemplary method for phase tracking internal reflections before subtraction according to another embodiment; [Figure 13] 1 is a graph illustrating an exemplary internal reflection subtraction according to one embodiment; [Figure 14] FIG. 1 is a block diagram illustrating a computer system upon which an embodiment of the present invention may be implemented; [Figure 15] 1 illustrates an example chipset in which one embodiment of the present invention may be implemented. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0011] To achieve acceptable range accuracy and detection sensitivity, Direct Long Range LIDAR systems use short pulse lasers with low Pulse Repetition Rate and very high pulse peak power. High pulse power can lead to rapid performance degradation of optical components. Chirped and Phase-Encoded LIDAR systems use long optical pulses with relatively low peak optical power. In this configuration, range accuracy increases with the chirp bandwidth or length and the bandwidth of the phase code, rather than the pulse duration, so excellent range accuracy can still be obtained.

[0012] LIDAR detection has also been used using phase-encoded microwave signals modulated onto an optical carrier, where the bandwidth (B) is proportional to the inverse of the duration (τ) of the pulse carrying each phase (B=1 / τ), and any phase-encoded signal is composed of a large number of these pulses. This technique relies on correlating a series of phases (or phase changes) of a particular frequency in the return signal with a series of phases (or phase changes) in the transmitted signal. The time delay associated with the peak in the correlation is related to the distance due to the speed of light in the medium. The distance resolution is proportional to the pulse width (τ). Advantages of this technique include the smaller number of components required and the use of mass-produced hardware parts developed for phase-encoded microwave and optical communications.

[0013] Various coherent ranging techniques have been demonstrated to provide single shot range and Doppler information for moving targets. In both coded and chirp waveform designs, the Doppler effect of the moving target must be understood and in any fashion compensated for to allow accurate range measurements. In the coded case, Doppler compensation is essential prior to cross correlation due to the "Thumb-Tack" ambiguity function. Without this operation, range measurements would be difficult to observe or impossible. Techniques have been demonstrated to overcome these difficulties. In the inventors' prior work, techniques for Doppler shift detection for phase-encoding LIDAR and range correction based on the detected Doppler shift have been disclosed and published in WIPO (World Intellectual Property Organization) Patent Cooperation Treaty (PCT) International Publication WO2018 / 144853 by Crouch et al., entitled "Method and System for Doppler Detection and Doppler Correction of Optical Phase-Encoded Range Detection" (hereinafter "Crouch I"), and PCT International Application No. PCT / US2018 / 041388 by Crouch et al., entitled "Method and System for Time Separated Quadrature Detection of Doppler Effects in Optical Range Measurements" (hereinafter "Crouch II"), the entire contents of each of which are incorporated herein by reference as if fully set forth herein.

[0014] However, phase-encoding LIDAR systems that implement the aforementioned approach to signed Doppler detection often experience difficulty in consistently resolving Doppler detection ambiguities. For this reason, there is a long-felt need to resolve detection ambiguities in a manner that improves the performance of phase-encoding LIDAR systems to compensate for the Doppler effect in optical ranging.

[0015] Additionally, reflections from internal optics associated with a phase-encoding LIDAR system (or any LIDAR system) (also referred to herein as "internal reflections" or "internal optical reflections") often further degrade the measurement accuracy of a phase-encoding LIDAR system. For example, strong back reflections from circulator optics can increase the noise floor (through unwanted side lobe structures) above the shot noise, limiting the dynamic range of the measurement and obscuring small target returns. Conventional phase-encoding LIDAR systems estimate and digitally remove the stationary signal (internal back reflections) using a technique called Circulator Subtraction, which has proven to be a very effective method of observing weak non-stationary target signals.

[0016] However, circular subtraction requires a frequency stable (>π / 10 phase drift) laser for several (>25) measurements, where one measurement corresponds to one transmitted optical signal (pulse or chirp or phase sequence) and one received optical signal (pulse or chirp or phase sequence) received in response thereto (if any). Because such requirements are too burdensome and / or costly for many LIDAR applications (e.g., autonomous vehicles), there is a long-felt need to provide alternative techniques and / or equipment configurations to mitigate (e.g., remove, subtract) internal reflections to improve the performance of phase-encoding LIDAR systems that perform optical ranging.

[0017] Thus, the present disclosure relates to a system and method for compensating for the negative effects of any internal optical reflections and Doppler shifts of the returned signal from one or more internal optical components. That is, the present disclosure provides a system (e.g., a phase encoding LIDAR system) and method that adds a Non-Carrier Pilot Tone to the output optical signal or the Reference Optical Signal so that the Doppler shift can correct the cross-correlation calculation, thereby improving the system's ability to determine distance. The system may further improve its ability to determine distance by subtracting (removing) the internal reflections of the returned signal from the optical components.

[0018] In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of the present specification. However, it will be apparent to one of ordinary skill in the art that the present invention may be practiced without such specific details. In other instances, well-known structures and devices are shown in block diagram form in order to avoid unnecessarily obscuring the present invention. 1. Overview of Phase-Encoded Detection

[0019] FIG. 1a is a schematic graph 110 showing an exemplary transmitted optical phase encoding signal for distance measurement according to one embodiment. The horizontal axis 112 represents time in arbitrary units from a starting time of 0 (Zero). The left vertical axis 114a represents the power during the transmitted signal in arbitrary units, and the right vertical axis 114b represents the phase of the transmitted signal in arbitrary units. To most simply illustrate the technology of phase encoding LIDAR, binary phase encoding is described. Trace 115 shows the power for the left vertical axis 114a, which is constant during the transmitted signal and drops to 0 beyond the transmitted signal. The dotted trace 116 shows the phase of the signal for a continuous wave signal.

[0020] As can be seen, the trace is In Phase with the Carrier for part of the transmitted signal (phase=0), then fluctuates by a symbol Δφ during a short time interval (phase=Δφ), repeatedly switching back and forth between the two phase values ​​for the transmitted signal, as indicated by ellipsis 117. The shortest interval of constant phase is a parameter of the encoding called the pulse duration (τ), which is usually the duration of a number of periods of the lowest frequency in the band. The inverse (1 / τ) is the Baud Rate, with each Baud representing a symbol. The number of these constant phase pulses in time of the transmitted signal (N) is the number of symbols (N), which indicates the length of the encoding. In the case of Binary encoding, there are two phase values, and the shortest interval of phase can be considered as 0 for one value and 1 for the other, so a symbol is one Bit, and the Baud Rate is also called the Bit Rate. In the case of multiple phase encoding, there are many phase values. For example, four phase values ​​such as Δφ*{0, 1, 2 and 3} are the same as {0, π / 2, π and 3π / 2} respectively for Δφ=π / 2 (90 degrees), so the four phase values ​​can represent 0, 1, 2 and 3 respectively. In this example, each symbol represents the same amount of information in two bits, and the bit rate is twice the baud rate.

[0021] Phase-Shift Keying (PSK) refers to a digital modulation scheme that conveys data by changing (modulating) the phase of a reference signal (carrier wave), as shown in Figure 1a. The modulation is stamped by varying the sine and cosine inputs at precise times. In Radio Frequency (RF), PSK is widely used for wireless local area networks (LANs), radio frequency identification (RFID) and Bluetooth communications. Alternatively, instead of operating relative to a constant Reference Wave, the transmission can operate relative to itself. The phase change of a single transmitted waveform can be considered a symbol. In this system, the demodulator determines the phase change of the received signal (relative to the Reference Wave), not the phase itself. Because the scheme relies on the difference between successive phases, this is called Differential Phase-Shift Keying (DPSK). DPSK is much simpler to implement than plain PSK because the demodulator does not need to have a copy of a reference signal to determine the exact phase of the received signal (hence it is a non-coherent, or asynchronous, method).

[0022] For optical distance measurement applications, the carrier frequency is the optical frequency f c and RFf 0is modulated onto an Optical Carrier. The number of symbols (N) and duration (τ) are selected to achieve the required range accuracy and resolution. The pattern of symbols is selected to be distinguishable from noise and other sources of coded signal. Thus, a strong correlation between the transmitted signal and the returned signal is a strong indication of the reflected or backscattered signal. The transmitted signal consists of one or more symbol blocks, each long enough to provide a strong correlation with the reflected or backscattered return even in the presence of noise. In the following discussion, the transmitted signal may consist of M blocks of N symbols per block, where M and N are assumed to be non-negative integers.

[0023] FIG. 1b is a schematic graph 120 showing an exemplary transmitted signal as a series of binary digits along with a returned optical signal for distance measurement according to one embodiment. The horizontal axis 122 represents time in arbitrary units after a starting time at 0. The vertical axis 124a represents the frequency (f c +f 0 ) in arbitrary units. The vertical axis 124b represents the amplitude of the optical transmit signal at frequency (f c +f 0 ) in arbitrary units and is offset from axis 124a to separate the traces. Trace 125 shows a transmitted signal of M*N binary symbols with phase changes as shown in FIG. 1a to generate successive codes, starting with 00011010 and indicated by the ellipses. Trace 126 shows an ideal (noise-free) return signal scattered from a non-moving object (so the return is not Doppler shifted). Although the amplitude is reduced, the code 00011010 is recognizable. Trace 127 shows an ideal (noise-free) return signal scattered from a moving object and therefore Doppler shifted. The return is at the appropriate optical frequency (f c +f 0) and is not well detected within the expected frequency band and therefore is greatly reduced in amplitude.

[0024] The observed frequency of the return (f') is the exact frequency of the return due to the Doppler effect (f = f c +f 0 ) is different.

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[0025] FIG. 1c is a schematic graph 130 showing an example cross-correlation of a transmitted signal with two returned signals according to one embodiment. In Phase Coded Ranging, the arrival of the phase-coded return is detected in the return signal by cross-correlating the transmitted signal or other reference signal with the returned signal, and is essentially implemented by cross-correlating the code for the RF signal with an electrical signal from an Optical Detector using Heterodyne detection, and down-mixing it back into the RF band. The horizontal axis 132 represents the lag time, in arbitrary units, applied to the coded signal before performing the cross-correlation calculation with the returned signal. The vertical axis 134 shows the amplitude of the cross-correlation calculation. The cross-correlation for any one lag is calculated by convolving the two traces, i.e., multiplying the corresponding values ​​in the two traces, adding for all points in the traces, and then repeating for each time lag. Alternatively, cross-correlation can be achieved by multiplying the Fourier transforms of each of the two traces, followed by an Inverse Fourier Transform. Efficient hardware and software implementations for the Fast Fourier Transform (FFT) can be widely used for both the forward and inverse Fourier transforms.

[0026] It should be noted that the cross-correlation calculations can be done using analog or digital electrical signals after the return amplitude and phase are detected from the photodetector. To move the signal at the photodetector to an RF frequency distance where it can be easily digitized, the optical return signal is optically mixed with a reference signal before impinging on the detector. A phase-encoded copy of the transmitted optical signal can be used as the reference signal, but it is also possible, and often preferred, to use a Continuous Wave Carrier Frequency Optical Signal output by the laser as the reference signal and capture both the amplitude and phase of the electrical signal output by the detector.

[0027] Trace 136 shows the cross-correlation with an ideal (noise-free) return signal reflected from a non-moving object (hence the return is not Doppler shifted). The peak occurs at a time Δt after the start of the transmitted signal. This indicates that the return signal contains a transmitted phase code version starting at time Δt. The range R for the reflecting (or backscattering) object is calculated from the Two Way Travel Time Delay based on the speed of light (C) in the medium, as given by Equation 3:

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[0028] The dotted trace 137 shows the cross-correlation with an ideal (noise-free) return signal scattered from a moving object (hence the return is Doppler shifted). The return signal does not contain phase encoding in the appropriate frequency bin, the correlation holds for all time lags, and the peak is not easily detected, and in the presence of noise is often not detectable. Therefore, Δt is not easily determined and the range R is not easily generated. 2. Overview of light detection hardware

[0029] To explain how to implement the phase encoding detection approach, several general and specific hardware approaches are described. Figure 2 is a block diagram showing exemplary components of a LIDAR system Doppler compensated according to one embodiment. The LIDAR system 200 includes a laser source 212, a splitter 216, a modulator 282a, a reference path 220, a scanning optics 218, a processing system 250, an acquisition system 240, and a detector array 230. The laser source 212 emits a phase or frequency modulated carrier wave 201 from the modulator 282a before or after the splitter 216 to generate a phase encoded or chirped optical signal 203 having a duration (D). The splitter 216 separates the modulated (or, as shown, unmodulated) optical signal. A target beam 205, also referred to herein as a Transmitted Signal or Output Optical Signal, is generated, which may have most of the energy of the beam 201. Also generated is a modulated or unmodulated reference beam 207a that may have a small amount of energy but enough to mix well with the return beam 291 scattered from an object (not shown). In the illustrated embodiment, the reference beam 207a is individually modulated with a modulator 282b. However, in some embodiments, the modulator 282b is omitted. The reference beam 207a passes through a reference path 220 and is directed to one or more detectors as a reference beam 207b. In some embodiments, the reference path 220 introduces a known delay sufficient for the reference beam 207b to reach the detector array 230 along with light scattered from objects external to the LIDAR. In some embodiments, the reference beam 207b is referred to as a Local Oscillator (LO) signal, referring to an older approach in which the reference beam 207b was generated locally from a separate oscillator.

[0030] In various embodiments, ranging from less flexible to more flexible approaches, the criteria are: 1) placing a mirror in the scene to reflect a portion of the transmit beam off the detector array so that the path lengths are closely matched; 2) using Fiber Delay to closely match the path lengths and broadcast a reference beam using optics near the detector array as proposed in FIG. 2 with or without path length adjustment to compensate for observed or predicted phase or frequency differences for a particular distance; 3) using a time delay of a Frequency Shifting device (Acousto-Optic Modulator) or waveform modulation of a local oscillator to generate a separate modulation to compensate for path length mismatch, or some combination of the scattered or reflected field. In some embodiments, the object is close enough and the transmit duration is long enough so that the return overlaps well with the reference signal without delay.

[0031] The transmit signal is often sent to illuminate the region of interest via some Scanning Optic 218. The detector array is a one-dimensional (1D) or two-dimensional (2D) array of paired or unpaired detectors arranged in a plane approximately perpendicular to a paired or unpaired single detector or a returned beam 291 from the object. The reference beam 207b and the return beam 291 are combined with zero or more optical mixers to generate an optical characteristic signal that is appropriately detected. The frequency, phase or amplitude or some combination of the interference pattern is recorded multiple times during the signal duration (D) for each detector by the acquisition system 240. The number of temporal samples processed per signal duration affects the Down-Range Extent. The number is often a practical consideration selected based on the number of symbols per signal, the Signal Repetition Rate and the Available Camera Frame Rate. The frame rate is the sampling bandwidth, also called the "Digitizer Frequency". The only fundamental limit to the Range Extent is the coherence length of the laser and the length of the chirp or unique phase code before it repeats (for unambiguous distance measurements). This is possible because the returned Heterodyne signal or digital record of the Bit can be compared and cross-correlated with any portion of the transmitted Bit from the previous transmission history.

[0032] The acquired data is made available to a processing system 250, such as a computer system as described below with reference to FIG. 14 or a chipset as described below with reference to FIG. 10. The processing system 250 includes a Doppler compensation module 270a and a phase tracked internal reflection subtraction module 270b (shown in FIG. 2 as internal reflection subtraction module 270b). The Doppler compensation module 270a determines the sign and magnitude of the Doppler shift and determines a corrected distance based thereon along with any other corrections described herein. The internal reflection subtraction module 270b determines a phase tracked internal reflection correction that is used by the processing system 250 along with any other corrections known to provide a corrected distance measurement (e.g., a signal generated from the Doppler compensation module 270a). In some embodiments, the data processing also provides an estimate of the Doppler shift, which is the amount of shift in the frequency of the return signal due to object motion. In some embodiments, the processing system 250 also provides a scanning signal to drive the scanning optics 218 and includes a modulation signal module 272 for transmitting one or more electrical signals that drive the modulators 282a, 282b, as shown in FIG. 2.

[0033] Any known device or system can be used to implement the laser source 212, the modulators 282a, 282b, the beam splitter 216, the reference path 220, the optical mixer 284, the detector array 230, the scanning optics 218, or the acquisition system 240. Optical coupling for focal point or flood on the target through the pupil plane is not shown. As used herein, an optical coupler is any component that affects the propagation of light in space coordinates to direct light from one component to another, such as vacuum, air, glass, crystal, mirrors, lenses, optical circulators, beam splitters, phase plates, polarizers, optical fibers, optical mixers, alone or in some combination, among others. 3. Phase-encoding Optical Detection

[0034] In some embodiments, an Electro-Optic Modulator provides the modulation. The system is configured to generate a phase code of length (M*N) and symbol duration (τ) appropriate for the required down-range resolution, for a total duration of D=M*N*τ, as described in more detail below for various embodiments. For example, the total number of pulses (M*N) in a 3D imaging application is in the range of about 500 to about 4,000. Since the processing is performed in the digital domain, it is advantageous to select M*N as a power of 2, for example in the interval 512 to 4096. When no averaging is performed, M is 1. If there is a random noise contribution, it is advantageous to have M of about 10. As a result, N is in the range of 512 to 4096 for M=1 and in the range of about 50 to about 400 for M=10. For baud rates of 500 Mbps to 1 Gbps, the duration of this code is between about 50 nanoseconds and 8 microseconds. It should be noted that the Range Window can be made to extend up to several kilometers under such conditions, and the Doppler resolution can also be quite high (depending on the duration of the transmitted signal). Although the processes, equipment, and data structures are depicted in Figures 2, 3, and 5 as essential blocks in particular arrangements for illustrative purposes, in other embodiments, one or more of the processes or data structures, or portions thereof, may be the same or may be included on a different host, arranged in a different manner in one or more databases, or may be omitted, or one or more different processes or data structures may be the same or may be included on a different host. For example, splitter 216 and reference path 220 may include zero or more optical couplers. 4. Doppler-compensated LIDAR with internal reflection subtraction

[0035] 3 is a block diagram showing exemplary components of a phase encoding LIDAR system using the previous approach. An object 390 is shown to illustrate the operation of the LIDAR system 300, but the object 390 is not part of the LIDAR system 300. The system includes a laser 310 and a phase modulator 320, a beam splitter 312, an optical mixer 360 such as a circulator and combiner, a scanning optics 322 including transmit / receive optics such as a circulator 323, and a photodetector 330 (also referred to as "Optical Detector 330"). The output of the beam splitter 312 is used in the local oscillator reference signal 314 (also referred to as "LO 314") and is input to the optical mixer 360. A return optical signal 324 is output by the circulator 323 in the scanning optics 322 and sent to the optical mixer 360. The result is a mixed optical signal that is directed to the photodetector 330, which outputs an electrical signal 332 that is input to the processing system 350. The processing system 350 includes a digital code module 372 for generating a digital code used to drive the phase modulator 320. The processing system 350 also includes a Doppler compensation module 370a used to induce a Doppler shift having a code based on the digital code and / or the electrical signal 332. The processing system 350 also includes a phase tracking internal reflection subtraction module 370b for determining cross correlation and range based on the digital code and / or the electrical signal 332.

[0036] In previous approaches, the method of determining the sign of the Doppler shift involved separately processing the real and imaginary components of the mathematical representation of the signal. In an embodiment described in more detail below, the use of Single Sideband Phase Modulation eliminates the need for separate processing. However, such systems can also be used.

[0037] A sinusoid with phase modulation in electrical engineering (corresponding to the angular modulation between the real and imaginary parts of the mathematical function (exp(iωt))) can be decomposed into or synthesized from two amplitude-modulated sinusoids offset in phase by 1 / 4 cycle (π / 2 radians). All three functions have the same frequency. The amplitude-modulated sinusoids are known as the in-phase component (I) with a phase of 0 and the quadrature-phase component (Q) with a phase of π / 2.

[0038] The laser emits light at a carrier frequency f c The modulated laser light signal (L) is mathematically represented by Equation 4:

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[0039] Mathematically, this expression is: real part = I OR cos(ωt) and the imaginary part = I OI sin(ωt), where I OR is the real part of the intensity (in-phase), and I OI is the imaginary part. The phase of the oscillation is given by the angle between the real and imaginary parts. Hence, L=I OR cos(ωt)+iI OI sin(ωt), and I O is the root of the sum of the squares of the real and imaginary parts, that is, I O 2 =I OR 2 +I OI 2Splitter 312 directs a small portion of the signal strength for use as a reference signal (called a local oscillator) LO given by Equation 5.

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[0040] In this embodiment, a modulation signal module in the processing system transmits an electrical signal representing the digital code of the symbol to be imposed as a phase change on an optical carrier, represented as B(t), where B(t) switches between 0 and π / 2 as a function of t. A phase modulator 320 acting as modulator 282a takes a digital line from a Field-Programmable Gate Array (FPGA), amplifies it, and imparts the phase change to the optical carrier by driving an EO phase modulator. Then the transmitted optical signal T is given by Equation 6a:

number

[0041] In various embodiments using current technology, this pilot peak useful for Doppler detection is not at the carrier frequency, i.e., f 0 ≠f c However, instead, a non-carrier pilot tone f P (e.g., f 0 =f P , and ωp=2πf P The pilot peak is a single tone with a known code offset from the carrier frequency. And the known pilot frequency f P The pilot tones are called non-carrier (NC) pilot tones having the following structure: Then, Equation 6a is replaced by Equation 6b. where P is a non-carrier pilot signal and ωc is the frequency at the center of the code, which in some embodiments is still at the carrier frequency, but in other embodiments using a single sideband phase modulator, is at another frequency. In some embodiments, as described in more detail below, an NC pilot tone is added to the reference optical signal instead of the transmitted output optical signal.

[0042] The returned signal is directed to an optical mixer where it is mixed with a reference optical signal LO given by Equations 5a through 5c. The returned signal R from the kth object intercepted by the transmitted beam is expressed as a signal having a carrier pilot tone f 0 =f C is given by Equation 7a.

number

[0043] For a non-carrier pilot tone and a single sideband code, the return is expressed as Equation 7c and the electric field is expressed as Equation 7d.

[0044] At the Optical Mixer 360, the Coincident Signals (e.g., LO 314 and Return Optical Signal 324) generate a Mixed Optical Signal 362 with a beat frequency related to the difference in frequency, phase and amplitude of the two optical signals being mixed, and an output that depends on the function of the Optical Mixer 360. As used herein, Down-Mixing means Optical Heterodyne Detection, which is an embodiment of heterodyne detection using a nonlinear optical process. In Optical Heterodyne Detection, referred to herein as "Down-Mixing", an optical signal of interest at some optical frequency is nonlinearly mixed with a reference "Local Oscillator" LO set at a close-by frequency. The desired result is a Difference Frequency that carries the information (amplitude, phase and frequency modulation) of the original optical frequency signals, but oscillates at a more easily processed lower frequency, conveniently referred to herein as the beat frequency in the RF band. In some embodiments, the beat frequency is within the RF band that can be output from the photodetector as an electrical signal 332, such as an electrical analog signal that can be easily digitized by an RF Analog to Digital Converter (ADC). The electrical signal 332 is input to the processing system 350 and, together with the digital code from the digital code module 372, is used by the Doppler compensation module 370a to determine the sign of the Doppler shift and the resulting velocity and direction of the object, in some embodiments. In some embodiments, the electrical signal 332 is input to the processing system 350 and, together with the digital code from the digital code module 372, is used by the phase tracking internal reflection subtraction module 370b so that the processing system 350 determines the cross correlation and distance.

[0045] In some embodiments, the raw signal is processed to find the Doppler peak, whose frequency ω Dis used to correct the correlation calculation and determine accurate distance. However, in such an embodiment, the lack of I / O detection would result in a real-valued time domain signal at the Doppler shift frequency for both positive and negative Doppler shifts. Analysis of this signal would not reveal the sign of the Doppler shift. In another embodiment, optical mixers and processing determine the in-phase and quadrature-phase components and ω D After using such a separation to predict ahead, we use ω to correct the cross-correlation calculation to infer Δt. D It has proven useful when configured to use ω D The value of Δt is used to indicate the velocity of the object. The value of Δt is then used to determine and provide the distance to the object using Equation 3 above. Separation of the I and Q signals by an optical mixer allows the sign of the Doppler shift to be unambiguously determined, as described in International Patent Application PCT / US2018 / 016632, published August 9, 2018 as WO2018 / 144853, the entire contents of which are incorporated by reference herein as if fully set forth herein. 4.1 Doppler-compensated LIDAR - Spectrum of Doppler-shifted return signals

[0046] Fig. 4a is a schematic graph 140a illustrating an exemplary spectrum of a transmitted signal, and Fig. 4b is a schematic graph 440a illustrating an exemplary spectrum of a Doppler-shifted complex return signal. The horizontal axis 442 represents the optical carrier f c The vertical axis 444 represents the amplitude of a particular narrow frequency bin, also called the spectral density, in arbitrary units relative to 0. A typical phase modulator modulates an RF phase modulated signal with a carrier frequency f c This is called Dual Sideband Modulation. Trace 445 shows that, as in a normal phase encoding implementation, c RFf corresponding to 0= 0 (an electric field with frequency 0 is also called Direct Current (DC)). The modulation is c Even if the carrier frequency is lowered, there is still a residual spike in the spectral density at that frequency, which dominates the power spectrum and is used to sense the Doppler shift, as explained below. The residual carrier spike is referred to in this disclosure as the carrier "pilot" because it guides the Doppler shift detection.

[0047] In FIG. 4b, trace 446 shows an ideal (noise-free) complex return signal backscattered from an object moving towards the LIDAR system and thus Doppler-shifted (called blue-shifted) at higher frequencies. The return is at the appropriate RF frequency f 0 = 0 (DC), but instead has a peak at the shifted frequency f s to Δf D When a time-domain signal is sampled using IQ techniques, the sign of the Doppler shift (positive or negative relative to DC) is evident in the resulting power spectrum.

[0048] In another embodiment described in Crouch I, it has been found to be advantageous if the optical mixer and processing is configured to determine in-phase (I) and quadrature (Q) components. IQ detection by one or more components (e.g., optical mixer 360 in FIG. 3) can be utilized to resolve the sign of the Doppler shift.

[0049] In an exemplary hardware embodiment described in Crouch I, a 90 Degree Optical Hybrid Optical Mixer allows I / Q detection of an optically down-mixed signal into two channels that are digitized. This system allows very flexible "software defined" measurement architectures to arise. Another hardware embodiment described in Crouch I includes the use of a 3x3 Multimode Interference (MMI) structure. Such devices are more compact than Free-Space 90-Degree Hybrids. These generate a 120 degree phase shift at each output port. Each port is then independently detected and digitized before software-based reconstruction of the complex signal. This approach requires at least two detectors and two Analog to Digital Converters (ADCs) to support the measurement. This presents a nontrivial challenge for scaling in a highly integrated design where the optical hybrid, detectors, electrical routing, and digitizers all take up space.

[0050] In another embodiment described in Crouch II, advantages are obtained if the in-phase (I) and quadrature (Q) components are sensed at different times. A reference signal can be made to be in phase with a signal transmitted during one time interval to measure an electrical signal related to the real part of the returned optical signal at the photodetector 330, and the reference signal can be made to be in quadrature with a signal transmitted during another non-overlapping time interval to measure an electrical signal related to the complex part of the returned optical signal at the photodetector 330. A complex signal can then be digitally generated in the Doppler compensation module 370a from the two measured electrical signals. The digitally constructed complex signal can then be used to determine the appropriately signed Doppler shift.

[0051] In this embodiment, the I / Q separation has the sign ω D ω D is used to correct the cross-correlation calculation to derive Δt. The estimate should be within the frequency resolution of the system, set by the duration of the processing interval. The Doppler-shifted and time-delayed (after downmixing) code term for the electric field due to the kth return is given by Equation 7e. Doppler Estimation ω Dk’ The correction with is done by vector multiplication in the time domain (or the CIRCSHIFT method in the frequency domain). The vector multiplication is shown in Equation 7f. It can be seen that in Equation 7f, a residual phase term remains, but the oscillating Doppler component of the argument is removed (compensated).

[0052] Also, ω D A value having the sign of Δt is used to provide the velocity and direction of the object at least in the vector linking the object to the LIDAR system 300. The value of Δt is then used to determine and provide the distance to the object using Equation 3 above. I and Q signal separation by an optical mixer allows the sign of the Doppler shift to be unambiguously determined. In some such Doppler compensation embodiments where the I and Q components are determined separately, Δf is calculated by taking the spectrum of the transmitted and returned signals, searching for peaks in each, and subtracting the frequency of the corresponding peak. D Instead of finding , it is more efficient to take the cross spectrum of the in-phase and quadrature components of the returned signal that has been downmixed in the RF band.

[0053] In practice, the complex return, which represents all of the in-phase and quadrature-phase (I / Q) components of the return, is often not measured, and the spectrum of the real-valued return is expressed as +Δf D and -Δf D, and therefore the direction of Doppler shift and the direction of target movement are not discernible from a single return. FIG. 4c is a schematic diagram showing an exemplary return signal spectrum when no IQ processing is performed according to one embodiment. If only the real part of the mixed optical signal is sampled (Non-IQ Detection), the resulting power spectrum will look like the sum of the two traces. The carrier pilots are visible on both sides of the spectrum. The Doppler shifted carrier pilot tones are evident and can be corrected for before the correlation operation for distance measurement, but the peaks appear on both sides of the spectrum and are symmetric about DC (RFf 0 =RFf c =0), the code may not be recoverable.

[0054] In all these cases, the method uses an electro-optic phase modulator to induce a dual sideband waveform. With these modulators, the modulation depth is selected to maintain constant energy at the carrier frequency. A residual carrier acts as a "pilot" tone in the transmitted waveform. When this CW "pilot" is Doppler shifted, the result is a beat note that can be used to estimate the Doppler shift from the target. Doppler estimation based on this pilot tone allows efficient Doppler correction of the coded signal prior to cross-correlation with a reference code (a two-step, Doppler-followed range estimation process). Electro-optic modulation architectures beyond the basic phase modulator are common in fiber optic communications.

[0055] The general architecture (see Shimotsu or Laoyssa references) allows the realization of SSBSC (Single-Sideband Suppressed Carrier) phase modulation. In the disclosed new embodiment, the NC pilot tone is moved far away from the carrier frequency (RFf o ≠0), f cIn some embodiments, this is done using single sideband modulation to generate a non-carrier pilot and / or phase encoding signal, and a non-carrier pilot tone is introduced. That is, in some embodiments, the transmitted (optical output) signal is generated by single sideband (SSB) modulation of an optical signal at a carrier frequency from a laser using a broadband RF phase encoding signal, and the output optical signal includes an Optical Broadband Phase-Encoded Optical Signal at a single side of the carrier frequency and a pilot tone at a frequency within the optical broadband.

[0056] With proper waveform design, these SSBSC phase modulators can be used to generate pilot tones at frequencies other than the carrier frequency, which allows for non-baseband centered operation of the Doppler shift estimation described above. Non-carrier pilot tones are referred to as engineered pilot tones or simply NC pilot tones in the following. Furthermore, since the NC pilot tones are present only on one side of the carrier (single sideband), the Doppler shift will appear in the signal spectrum without sign ambiguity even without IQ separation. Thus, the sign of the Doppler shift (approaching target vs. receding target) will be accurately resolved. Similar results occur when the NC pilot tones are introduced into the reference optical signal instead of the transmitted optical signal. The Doppler correction based on the Doppler shift with the restored sign is the same as described in the previous method.

[0057] In various embodiments, a variety of different components are used to introduce a non-carrier pilot tone into one of the output optical signals transmitted via scanning optics 322 for interaction with object 390 or into the reference optical signal for mixing with the return signal 324 at optical mixer 360. 4.2 Doppler-compensated LIDAR using NC pilot tones and internal reflection subtraction

[0058] FIG. 5a is a block diagram illustrating an exemplary Doppler compensated LIDAR system using non-carrier pilot tones according to one embodiment. Although an object 390 is shown for illustrative purposes, the object 390 is not part of the LIDAR system 500a. The scanning optics 322, the circulator 323, the optical mixer 360, the photodetector 330, the digital code module 372, and the object 390 are as described above for FIG. 3. However, here, the laser 310 and the beam splitter 312 are replaced with an optical source 510a having one or more lasers and other optical couplers to generate an optical base signal OB 511a and a reference signal REF 514a for phase modulation. The optical source 510a includes a non-carrier pilot tone generator 588a (labeled "NC pilot tone generator 588a" in FIG. 5a) configured to generate a pilot tone that is different from the carrier frequency of the laser used to generate the optical base signal 511a. This configuration may be accomplished as a separate laser with a different drive function or its own drive function for the previous laser 310. The dual sideband phase modulator 380 is replaced with one or more phase modulators 580a, including zero or more single sideband (SSB) or single sideband with carrier suppression (SSBSC) phase modulators, or some combination.

[0059] The processing system 350 is replaced with a processing system 550a including a Doppler compensation module 570a with a single sideband (SSB) phase processing module 574a. The processing system 550a also includes a phase tracking internal reflection subtraction module 572a for determining cross correlation and distance based on the electrical signal 532a. The NC pilot tone generator 588a is controlled by the SSB phase processing module 574a, which provides a driving function. The SSB phase processing module 574a also determines a Doppler shift having a sign based on the electrical signal 532a. The electrical signal 532a is output by the detector 330 when struck by the mixed optical signal 562a output by the optical mixer 360 when the REF 514a and the return signal 524a are simultaneously generated in the optical mixer 360. The return signal 524a is received in response to transmitting an optical output signal from the phase modulator 580a through the scanning optics 322.

[0060] In some of these embodiments, the optical source 510a uses a first laser driven to generate only the carrier frequency optical signal. The optical source 510a uses another second laser driven by the SSB phase processing module 574a to generate a non-carrier pilot tone as the NC pilot tone generator 588a. In some of these embodiments, the optical source 510a includes a splitter that splits and directs the carrier frequency optical signal to the phase modulator 580a as a component of the optical base OB 511a and to the optical mixer 360 as a component of the optical reference signal REF 514a. In some of these embodiments, the NC pilot tone optical signal from the NC pilot tone generator 588a is added to the optical base signal OB 511a or the optical reference signal REF 514b using a combiner optical coupler, but not to both. In some embodiments, the REF 514a includes only the NC pilot tone without the carrier frequency optical signal, and the optical base signal OB 511a includes only the carrier frequency optical signal.

[0061] In some embodiments, the optical source includes a single laser that generates a carrier frequency optical signal and a splitter that directs most of the energy to phase modulator 580a as OB 511a. The optical source 510a also includes an acousto-optic modulator (AOM) to shift the optical frequency of the low energy beam emitted by the splitter to a NC pilot tone that is then output to optical mixer 360 as REF 514a. In this embodiment, the AOM is controlled by SSB phase processing module 574a by driving the source to generate an acoustic frequency sufficient to shift the optical frequency to the desired pilot frequency at the AOM.

[0062] The phase code is phase modulated onto the optical-based signal OB 511a using phase modulator 580a. SSB phase processing module 574a determines the signed Doppler shift based on the electrical signal 532a, the known pilot frequency of the NC pilot tone, and whether the pilot tone is in REF 514a or in electrical signal 532a as a result of being added to the optical-based signal OB 511a. The existing code in the Doppler detection module uses the signed Doppler shift from SSB phase processing module 574a to correct the range and provide the range and appropriately signed relative velocity.

[0063] FIG. 5b is a block diagram illustrating an exemplary Doppler compensated LIDAR system using non-carrier pilot tones according to another embodiment. In this embodiment, the phase modulator is configured to generate NC pilot tones and thus includes an NC pilot tone generator 588b. The NC pilot tone generator 588a of the laser source 510a is omitted from the optical source 510b. Although the object 390 is shown for illustrative purposes, the object 390 is not part of the LIDAR system 500b. The scanning optics 322, the circulator 323, the optical mixer 360, the photodetector 330, the digital code module 372 and the object 390 are as described above with respect to FIG. 3. Here, the optical source 510a is replaced with an optical source 510b having one or more lasers and other optical couplers to generate an optical base signal OB 511b and a reference signal REF 514b for phase modulation.

[0064] Phase modulator 580a is replaced with phase modulator 580b, which includes NC pilot tone generator 588b configured to generate a pilot tone at a different frequency than the carrier. NC pilot tone generator 588b includes one or more single sideband (SSB) or single sideband with carrier suppression (SSBSC) phase modulators or other optical components, or some combination. In some embodiments, a single SSBSC phase modulator provides both the phase code and the pilot tone.

[0065] In some embodiments, the phase modulator 580b includes a splitter, a combiner, a dual sideband phase modulator that applies a phase code signal, and an SSB phase modulator as an NC pilot tone generator 588b. The splitter splits the light-based signal OB511a into two beams of approximately equal intensity, sending one beam to the dual sideband phase modulator and the other to the SSB phase modulator as an NC pilot tone generator 588b. The outputs from the two modulators are combined in an optical combiner to generate an output optical signal that is sent via the scanning optics 322.

[0066] In some embodiments, the phase modulator includes a single DSB or SSB or SSBSC phase modulator for applying a phase code and an acousto-optic modulator (AOM) as the NC pilot tone generator 588b for shifting the optical frequency of the light-based signal having the carrier frequency to the NC pilot tone. In this embodiment, the AOM is controlled by the SSB phase processing module 574b by driving an acoustic source to generate an acoustic frequency sufficient to move the optical frequency to the desired pilot frequency in the AOM.

[0067] The processing system 350 is replaced with a processing system 550b including a Doppler compensation module 570b with a single sideband phase processing module 574b. The processing system 550b also includes a phase tracking internal reflection subtraction module 572b for determining cross correlation and distance based on the electrical signal 532a. The NC pilot tone generator 588b is controlled by the SSB phase processing module 574b, which provides a phase code and phase encoding LIDAR signal to generate the pilot tone. The SSB phase processing module also determines a Doppler shift having a code based on the electrical signal 532b. The electrical signal 532b is output by the detector 330 when struck by the mixed optical signal 562b output by the optical mixer 360 when the REF 514b and the return signal 524b are simultaneously generated at the mixer. The return signal 524b is received in response to transmitting the optical output signal from the phase modulator 580b through the scanning optics 322. 4.3 Doppler-compensated LIDAR: Spectrum of Doppler-shifted SSB return signals

[0068] 6a is a schematic graph 640a illustrating an exemplary spectrum 645 of an SSB output optical signal according to one embodiment. The transmitted spectrum 645 in the carrier suppressed single sideband case is entirely at the carrier frequency f c (RF=0, DC) on one side, RFf 0 ≠0. The spectrum also has a code peak (fP =f 0 ) with non-carrier pilot tones f P The non-carrier pilot tones exhibit spikes in the spectral density at , but need not be at peaks in other embodiments. The non-carrier pilot tones not only guide the detection of the Doppler shift, but also provide an unambiguous determination of the Doppler shift code, as described in more detail below.

[0069] 6b is a schematic graph 640b illustrating an exemplary spectrum of a Doppler shifted SSB optical return signal when IQ processing is performed according to one embodiment. It shows the Doppler shifted spectrum retro-reflected from an object as trace 646 in the carrier suppressed single sideband case. The Doppler shift and sign are calculated using the shifted non-carrier pilot tones 646a (f PS The dominant spectral peak measured at the pilot frequency f P (see Equation 9 below). These new calculations of signed Doppler shifts are used in the SSB Phase Processing modules 574a, 574b described above. These signed Doppler shifts are used to correct the object's relative velocity and distance calculations, as in the previous approach scheme.

[0070] FIG. 6c is a schematic graph 640c showing an example spectrum of a Doppler shifted SSB return signal spectrum when no IQ processing is performed according to one embodiment. The resulting spectrum 647 is the actual response 646 and its reflection around the carrier frequency. It should be noted that the carrier frequency corresponds to the DC component of the RF signal at 0 frequency when converting to an RF signal both when driving the modulation and after downmixing in an optical mixer. This is evident in both halves of the return spectrum because the non-carrier pilot tone frequency 647a is off the axis of symmetry. In the case of non-IQ sampling, the resulting power spectrum 647 across both sides is the power spectrum of the up- and down-shifted pilot tones f above and below the carrier frequency, respectively.PSU , f PSL The two peaks have two pilot tone peaks 647a, 647b, which are represented by the Doppler shift Δf D This allows for the restoration of

number

[0071] FIG. 7 is a flow chart illustrating an example method of using a single-sideband phase encoding LIDAR system to determine and compensate for Doppler shift effects with sign on range according to one embodiment. Although the operations are shown in FIG. 7 as integrated steps in a particular sequence for illustrative purposes, in other embodiments, one or more of the operations or portions thereof may be performed in a different sequence, superimposed in time series or parallel, omitted, one or more additional operations may be added, or the method may be modified in some combinatorial manner. In some embodiments, some or all of the operations of method 700 may be performed by one or more LIDAR systems, such as LIDAR system 200 in FIG. 2, LIDAR system 300 in FIG. 3, LIDAR system 500a in FIG. 5a, and / or LIDAR system 500b in FIG. 5b. For example, in some embodiments, operations 703 and 710 through 733 may be performed by processing system 350 in FIG. 3 and / or processing system 550 in FIG. 5a and FIG. 5b.

[0072] In operation 701, a transceiver, such as a LIDAR system, is configured to transmit a phase-encoded optical signal based at least in part on an input of a phase code sequence. Also, a portion (e.g., 1% to 10%) of the unmodulated input optical signal or the phase-encoded transmit signal from the laser is directed to a reference optical path. The phase-encoded optical signal or the reference signal includes a NC pilot tone, but not all of the two signals do. For example, the LIDAR is configured as shown in FIG. 5a or FIG. 5b, or some combination. Also, the transceiver is configured to receive a backscattered optical signal from an external object illuminated by the transmitted signal. In some embodiments, operation 701 includes configuring other optical components in hardware to further provide one or more of the following operations. It should be noted that the transmitted signal does not need to be a beam. A diverging signal will reliably capture many different ranges and Doppler values ​​within a single range profile. However, it does not provide cross range resolution within the illuminated spot. However, it is advantageous to use narrow beams that provide the inherent sparsity provided in conjunction with point-by-point scanning to provide cross-range resolution useful for identifying objects.

[0073] In operation 703, a code composed of a sequence of M*N symbols representing M blocks of N symbols each selected from an alphabet of N symbols is generated for use in distance measurements, with no duplication of sequences of N symbols among the M blocks. Also, in some embodiments, a Fourier transform of an RF signal using such phase encoding is determined during operation 703 for use in determining a Doppler shift, as described in Crouch I.

[0074] In operation 705, a first portion of the laser output, as expressed in Equation 4, is modulated using a code received from the digital code module 372 using a single sideband phase modulator, such as an SSBSC phase modulator, to generate a transmitted single sideband phase encoding signal having a non-carrier pilot tone, as expressed in Equation 6, and is directed to a spot within the scene where an object or portion of an object may or may not be present. Also in operation 705, a second portion of the laser output is directed along a reference path as a reference signal, also referred to as a local oscillator signal, as expressed in Equation 5a or 5c.

[0075] In operation 707, any travel time delay Δt and Doppler shift ω as shown in Equation 7a or 7b are D The backscattered return signal R having the signal LO is mixed with a reference signal LO as shown in Equation 5a or Equation 5c to output one or more mixed optical signals 362. In some embodiments with I / Q processing, the mixed signal provides information about the in-phase and quadrature-phase components.

[0076] In operation 708, the mixed optical signal is directed to and detected by one or more photodetectors to convert the optical signal into one or more corresponding electrical signals.

[0077] In some embodiments, averaging and internal reflection subtraction is performed on several other return signals S(t) to remove spurious copies of the transmitted optical signal generated by internal optics along the return signal path, such as the circulator 323. Such spurious copies reduce correlation with the real return from external objects, thereby masking the real return as being nearly imperceptible. If averaging is performed on P different illumination spots, and on the returns such that a single object is not included in all such illumination spots, the average is dominated by spurious copies of the code generated by the internal optics. Then, the spurious copies of the code can be removed from the returned signal, leaving only the real return in the corrected complex electrical signal S(t). P is a number large enough to ensure that the same object is not illuminated in all spots. Values ​​as small as P=100 are computationally advantageous for Graphic Processing Unit (GPU) implementations, while values ​​as large as P=1,000 are preferred and suitable for Field Programmable Gate Array (FPGA) implementations. In an exemplary embodiment, P is about 100. In other embodiments, depending on the application, P can range from about 10 to about 5,000.

[0078] In some embodiments of operation 708, it is determined whether P returns are received. If not, control returns to operation 705 to illuminate another spot. If so, an average signal is calculated (in some embodiments, phase weighted) and the average is subtracted from the return signal to remove the effects of internal reflections.

[0079] In operation 710, the Doppler shift with its sign is determined and used to correct the return signal. If separate I / Q sensing and processing is not used, the frequency f PSL , f PSUis obtained for the electrical signal based on a mix of the reference signal and the returned signal, and the Doppler shift with sign is determined using Equation 8a, Equation 8b, or Equation 8c. If separate I / Q sensing and processing is used, the frequency f of the single dominant peak is PS is obtained for the returned spectrum. The Doppler shift with sign is then determined using Equation 9.

number

number

[0080] In operation 711, the corrected return signals are used to detect range or relative velocity, or both. In some embodiments with Doppler shifts with multiple signs for a single return, operation 711 includes assuming that a particular return is based on an object or part of an object moving at a particular average velocity over the duration of one transmitted signal, and associating each delay time with one of the Doppler shifts with signs. For a Doppler correction with a given sign, only the range peaks associated with the Doppler correction with that sign will be present in the cross-correlation. Thus, it is not possible in many cases to mis-pair a Doppler with a given range and sign.

[0081] In operation 731 it is determined whether another spot is illuminated. If so, control returns to operation 703. If not, control proceeds to operation 733.

[0082] In operation 733, the device operates based on the corrected distance or speed, or both. In some embodiments, this includes providing an image to a display device showing a Doppler corrected position of any object at the multiple spots illuminated by the transmitted light signal. In some embodiments, this includes communicating data to the device identifying at least one object based on a point cloud of the corrected positions at the multiple spots illuminated by the transmitted light signal. In some embodiments, this includes moving the vehicle to avoid a collision with the object, where a closing speed between the vehicle and the object is determined based on the magnitude of the Doppler effect at the multiple spots illuminated by the transmitted light signal. In some embodiments, this includes identifying a vehicle on a collision course or identifying an object based on a point cloud of the corrected positions at the multiple spots illuminated by the transmitted light signal. 4.5 Binary Phase Code

[0083] In pursuing this approach, an interesting case was noted for binary phase codes. The Nyquist Frequency of the Pulse Rate is f N It should be noted that the half Nyquist frequency (f N / 2) are the frequency-shifted code B(t), the pilot tones 2πf P and φ P We only need four phase states (-π / 2, 0, π / 2, π) to fully represent all of t. Two of these phases (e.g., 0 and π) are used as the binary phase encoding B(t), and the other two (e.g., -π / 2 and π / 2) are phases φ that are different from any of the phases used in the binary code. P Single-sided pilot tone at 2πf P This has advantages over available digital-to-analog waveform sources (e.g., allowing the use of FPGA transceivers over expensive High Bit Depth DACs), limits the RAM requirements needed to buffer the waveform (since the four phases have such a low bit depth), and limits the power utilization of the RF driver (e.g., allowing a power-efficient limiting amplifier to be used instead of a power-hungry linear amplifier).

[0084] When the baud rate (1 / τ) is displayed in BAUD, the Nyquist frequency is BAUD / 2. Half the Nyquist frequency is BAUD / 4, and the corresponding angular frequencies are ω HN = 2πBAUD / 4. Regarding Equation 6b, the amplitude P of the pilot tone and the amplitude S of the phase code component must also be the same. Digitally, the RF waveform D for driving the single sideband phase modulator is achieved through the following operations: Start with the code waveform given by Equation 11a.

number

[0085] 8 is a graph illustrating an exemplary transmitted optical signal with a non-carrier pilot tone according to one embodiment. Trace 845 shows an exemplary frequency shifted spread spectrum binary phase code and pilot tone. This waveform, as described herein, has the interesting attribute of being expressed in only four phase states, thus providing a useful method of limiting electronic and power resources to drive the waveform. 4.6 Methods for Subtracting Internal Reflections

[0086] FIG. 9a is a flow chart illustrating an exemplary method of using a phase encoding LIDAR system to determine and compensate for internal reflection effects on distance according to one embodiment. Although the operations are shown in FIG. 9a and subsequent flow charts FIG. 11 and FIG. 12 as integrated steps in a particular sequence for illustrative purposes, in other embodiments, one or more of the operations or portions thereof may be performed in a different sequence, overlapped in time in series or parallel, omitted, or one or more additional operations may be added, or the method may be modified in some combinatorial manner. In some embodiments, some or all of the operations of method 900a may be performed by one or more LIDAR systems, such as LIDAR system 200 in FIG. 2, LIDAR system 300 in FIG. 3, LIDAR system 500a in FIG. 5a, and / or LIDAR system 500b in FIG. 5b. For example, in some embodiments, operations 903a and 910a through 933a may be performed by processing system 350 in FIG. 3 and / or processing system 550 in FIG. 5a and FIG. 5b.

[0087] In operation 901a, a transceiver, such as a LIDAR system, is configured to transmit a phase-encoded optical signal based on an input of a phase code sequence. A portion (e.g., 1% to 10%) of the unmodulated input optical signal or the phase-encoded transmit signal from the laser is also directed to a reference optical path. The transceiver is also configured to receive a backscattered optical signal from any external object illuminated by the transmitted signal. In some embodiments, operation 901a includes configuring other optical components in hardware to further provide one or more of the following operations. It should be noted that the transmitted signal does not need to be a beam. A diverging signal would reliably capture multiple distinct ranges and Doppler values ​​within a single range profile. However, it does not provide cross-range resolution within the illuminated spot. However, it is advantageous to use a narrow beam, which provides the inherent sparsity provided with point-by-point scanning to provide cross-range resolution useful for identifying objects.

[0088] In operation 903a, a code composed of a sequence of M*N symbols representing M blocks of N symbols each selected from an alphabet of N symbols is generated for use in distance measurement, where there is no duplication of sequences of N symbols among the M blocks, and in some embodiments a Fourier transform of an RF signal having such phase encoding is determined during operation 903a for use in determining a Doppler shift, as described in WO2018 / 144853.

[0089] In operation 905a, a first portion of the laser output, as expressed in Equation 4, is phase encoded using the code received from the digital code module 372 to generate a transmitted phase encoding signal, as expressed in Equation 6a, and is directed to a spot on the scene where an object or portion of an object may or may not be present. Also in operation 905a, a second portion of the laser output is directed along a reference path as a reference signal, also referred to as a local oscillator LO signal, as expressed in Equation 5a or 5b.

[0090] In operation 907a, any travel time delay Δt and Doppler shift ω as shown in Equation 7a are D The backscattered return signal R having the signal LO is mixed with a reference signal LO as shown in Equation 5a or Equation 5b to output one or more mixed optical signals 362. In some embodiments with Doppler shift detection and compensation, the mixed signal provides information about the in-phase and quadrature-phase components.

[0091] In operation 908a, the mixed optical signal is directed to one or more optical detectors to convert the optical signal into one or more corresponding electrical signals, which are detected by the one or more optical detectors. For example, in some embodiments with Doppler correction, as described in WO2018 / 144853, two electrical signals are generated by the detectors. One electrical signal on one channel (Ch1) represents the downmixed in-phase component (I), and another electrical signal on another channel (CH2) represents the downmixed quadrature-phase component (Q). A complex downmixing signal S is calculated based on the two electrical signals. In either case, one or more electrical signals are generated in operation 908a.

[0092] In some embodiments, averaging and internal reflection subtraction is performed on several other return signals S(t) to remove false copies of the phase encoding signal generated by internal optics along the return signal path, such as the scanning optics 322. These false copies reduce correlation with the real returns from external objects, thereby masking the real returns as being nearly imperceptible. When averaging is performed on P different illumination spots, and on the returns such that a single object is not included in all these illumination spots, the average is dominated by false copies of the code generated by the internal optics. Then, the false copies of the code can be removed from the returned signal, leaving only the real returns in the corrected complex electrical signal S(t). P is a number large enough to ensure that the same object is not illuminated at all points. Values ​​as small as P=100 are computationally advantageous for Graphic Processing Unit (GPU) implementations, while values ​​as large as P=1,000 are preferred and suitable for Field Programmable Gate Array (FPGA) implementations. In an exemplary embodiment, P is about 100. In other embodiments, P may range from about 10 to about 5,000, depending on the application. Figure 11 is a block diagram illustrating an example of multiple spot averaging to remove returns from internal optics, according to one embodiment. Operations 909a and 910a perform this correction.

[0093] In operation 909a, it is determined whether P returns have been received. If not, control returns to operation 905a to illuminate another spot. If so, control proceeds to operation 910a. In operation 910a, the average signal S S (t) is calculated by Equation 12a, where each received signal of duration (D) is P It is shown in (t).

number

[0094] In operation 911a, the corrected return signal is used to detect range or Doppler shift or both.

[0095] In step 933a, the device operates based on the corrected distance or speed, or both. In some embodiments, this includes providing an image to a display device showing a Doppler corrected position of any object at the multiple spots illuminated by the transmitted light signal. In some embodiments, this includes communicating data to the device identifying at least one object based on a point cloud of the corrected positions at the multiple spots illuminated by the transmitted light signal. In some embodiments, this includes moving the vehicle to avoid a collision with the object, where a closing speed between the vehicle and the object is determined based on the magnitude of the Doppler effect at the multiple spots illuminated by the transmitted light signal. In some embodiments, this includes identifying a vehicle on a collision course or identifying an object based on a point cloud of the corrected positions at the multiple spots illuminated by the transmitted light signal.

[0096] A range profile often consists of the maximum correlation for each of a number of range bins associated with different lag times. The range bins are differentiated by a range bin index that increases with distance (lag time). The distance (Distance, Range) associated with an index is the index number multiplied by the range bin size. The range bin size is given by the lag time increment multiplied by the speed of light in the medium. In the examples shown below, the range bin size is 1 meter unless otherwise stated.

[0097] As explained in WO2018 / 144853, experiments have shown that retroreflections from optical surfaces in the system contribute a series of very large returns (cross-correlations) in the range profile. These large returns induce large side lobe structures that can obscure smaller returns from the actual target in the range profile (power of cross-correlation with phase code vs. time intervals corresponding to range bins). To solve this problem, the Coherent Processing scheme mentioned above was devised to subtract an estimate of the signal portion associated with the circulator optics. Using the correction, S S (t) is applied by coherently subtracting complex vectors from a subset of the code block, resulting in a very effective reduction in the strength of the circulator distance return and associated side lobes.

[0098] FIG. 9b is a flow chart illustrating an example method of using a phase encoding LIDAR system to determine and compensate for internal reflection effects on distance according to one embodiment. Although the operations are shown in FIG. 9b and the subsequent flow charts FIG. 11 and FIG. 12 as integrated steps in a particular sequence for illustrative purposes, in other embodiments, one or more operations or portions thereof are performed in a different sequence, overlapped in time serially or in parallel, omitted, one or more additional operations are added, or the method is modified in some combinatorial manner. In some embodiments, some or all of the operations of method 900b may be performed by one or more LIDAR systems, such as LIDAR system 200 in FIG. 2, LIDAR system 300 in FIG. 3, LIDAR system 500a in FIG. 5a, and / or LIDAR system 500b in FIG. 5b.

[0099] The method 900b includes an act 902b of modulating an optical signal from a laser to generate a modulated optical signal. The method includes an act 904b of transmitting the modulated optical signal toward an object. The method includes an act 906b of receiving a returned optical signal in response to transmitting the modulated optical signal. The method includes an act 908b of mixing the returned optical signal with a reference optical signal related to the optical signal from the laser to generate a mixed optical signal. The method includes an act 910b of detecting the mixed optical signal to generate an electrical signal. The method includes an act 912b of determining a parameter of an internal reflection of the returned optical signal from one or more optical components based on the electrical signal and the modulated optical signal. The method includes an act 914b of operating a vehicle based on the parameter of the internal reflection.

[0100] 10a-e are graphs showing an example range signal before and after correction to remove returns from internal optics according to one embodiment. In FIG. 10a, the horizontal axis shows range bins in meters from 0 to over 4,000 m. The vertical axis shows decibels (dB, reference y 0 log(y / y 010b shows the cross-correlation power in units of 100 m. The trace shows the correlation peak before subtracting the average of P spots. A large peak is shown at about 100 m and a smaller peak at about 2,000 m. In FIG. 10b, the axes are the same but the trace shows the correlation peak after correction by subtracting the average of P=32 spots. The false peak at about 100 m due to the internal optics (here a circulator used instead of the scanning optics 322) is greatly reduced and the peak at about 2,000 m due to the real external object is somewhat enhanced. This is more clearly shown in FIG. 10c, which expands the area of ​​area A near the return from the internal optics, and in FIG. 10d, which expands the area of ​​area B near the real return from the external object. FIG. 10c shows a solid trace showing a large number of false reflections between 60 and 90 m, which are almost completely removed in the dotted trace showing the corrected distance profile. Clearly, optical retroreflections are often composed of a large number of reflections. For example, the circulator has several retroreflective surfaces that are clearly shown as distinct peaks in the uncorrected Fig. 10c. Fig. 10d shows the solid trace that shows an actual peak at about 1,850 m, which is improved by about 10% in the dotted trace that shows the corrected distance profile.

[0101] For example, in a monostatic detection system with 99.99% anti-reflection coated optics and 1 mW optical power, 100 nW of power would be reflected back to the detector. For a coherent system, this is a significant amount of power, much more than the normal target return. Due to the inherent sidebands of the distance measurements reflected back by the system and code used, this can induce an increase in the noise floor. Figure 10e shows the contribution of the internal reflection cross-correlation along a wide range of distance bins from distance bin index 0 to distance bin index 4,000. In the experimental data presented here, the black trace is an example of a noise floor elevated by the side lobe structure of the strong return. The grey is the noise floor after circulator subtraction, which is essentially shot noise limited. Specifically, before suppression, as given by the dark trace, outside of a peak at an index of about 150, there is a steady contribution of the cross-correlation between -5 dB and +10 dB at all distance bins up to index 4,000. After suppression, as given by the grey trace, the peak at index about 150 is well reduced, and the contribution to the cross-correlation is reduced below -5 dB for all distance bins up to index 4,000. 4.7 Methods for subtracting internal reflections using non-phase-stabilized lasers

[0102] In general, the internal reflection subtraction technique discussed above requires a frequency stable (<π / 10 phase drift) laser over several (>25) measurements. While compatible with many embodiments, in some embodiments such small phase drifts are not usable. For example, in some embodiments, the laser 310 in FIG. 3 is a conventional coherent fiber optic communication laser (e.g., a Micro Integrated Tunable Laser Assembly, "uITLA") stabilized using a feedback loop with a driver that introduces Optical Frequency Dither. Laser dithering often exceeds hundreds of Megahertz at frequencies of several hundred Hertz. This can pose a nontrivial challenge associated with the internal reflection average subtraction technique discussed above, since the phase can drift more than 2π during a typical circulator averaging duration of 20 to 30 measurements used in some embodiments or 10 to 5,000 measurements used in various other embodiments.

[0103] In various embodiments, the internal reflection average subtraction method is modified to account for laser phase changes during internal reflection averaging. The phase sensitivity of coherent LIDAR systems can be exploited to solve this problem. By tracking the optical phase of the retroreflected signal, a phase correction (e.g., by multiplying with a single complex number) can be applied to the signal before averaging so that the signals are in phase.

[0104] FIG. 11 is a flow chart illustrating an exemplary method for phase tracking internal reflections before subtraction according to one embodiment. This method 1100 is computationally intensive, but simple and effective. In this "Long Hand" method, the phase of the retro-reflected signal peak is estimated after cross-correlation with a reference code (phase of the complex-valued range peak in the cross-correlation). The retro-reflected peak occurs at a consistent location in the range profile (e.g., fixed range delay) that can be determined once in a calibration operation. Typically, the peak is within a very short range bin, usually the first 100 or so range bins. Also, this is often the highest and most consistent peak under normal scanner operation without retro-reflection subtraction. The phase of this peak is then removed from the time domain signal before averaging. After averaging, the retro-reflected signal phase of a particular shot is reintroduced before subtracting the signal from the original time domain signal vector. This method is a computationally intensive method for estimating the Evolving Phase.

[0105] Operations 901a through 905a, as described above, configure the transceiver (operation 901a), provide a phase code block (operation 903a), and illuminate the next spot with the transmitted optical signal (operation 905a). In operation 1107, the return signal is received and, in some embodiments, stored in a non-transitory computer readable medium for further processing. In some embodiments, the return is a time series of complex values ​​based on separate measurements of the in-phase (I) and quadrature (Q) components of the return signal.

[0106] In step 1111, the optical phase of the retroreflected signal is determined. Operation 1111 includes the phase encoding signal B(t) and E kp The returned signal E of Equation 7b from the pth spot of the P spots to be averaged shown in (t) kThe result is the range profile complex amplitude RP for spot p as a function of time lag corresponding to range bin index i, as given by Equation 13a: p It is.

number

[0107] In act 1113, the return signal is corrected in the time domain by the phase determined in act 1111, as given by Equation 14a.

number

[0108] In operation 1115, the number of summed signals, n, and the complex sum of the phase corrected signals are accumulated and stored in memory or special registers or some other computer readable medium as a time series of complex numbers. The sum is given by Equation 14b. where n≦P is the number of phase correction signals accumulated so far. In some embodiments, operation 1115 also divides the sum by the number of signals to calculate an average signal after the n signals are combined, as given by Equation 14c. In another embodiment, the division is not performed until the sum of Equation 14b has been accumulated for all P signals.

[0109] In operation 1117, it is determined whether the averaging period has been reached, e.g., n=P. After all P signals have been averaged (n=P) and the average of Equation 14c provides the time domain internal reflection subtracted signal Ss(t) given by Equation 12a. This equivalence is expressed in Equation 14d. In some embodiments, the averaging may be done with a time-based exponential decay to give more weight to more recent shots. If P signals have not accumulated, control returns to operation 905a to illuminate the next spot. Otherwise, control proceeds to operation 1131.

[0110] In operation 1131, a next time domain signal to be corrected for internal reflections is received. In some embodiments, this signal is one of the P signals in the Averaging Ensemble that were previously phase corrected and averaged. In other embodiments, this signal is another signal received before or after the signal in the Averaging Ensemble. The returned signals in the Averaging Ensemble and the returned signals corrected for internal reflections constitute a first plurality of returned signals. The Averaging Ensemble is a subset of the first plurality of signals. When the returned signal corrected for internal reflections is in the Averaging Ensemble, the first plurality and the subset are the same.

[0111] In operation 1133, the phase correction signal Sp(t) stored during the averaging period is converted to a corresponding phase correction φI also stored during the averaging period. p The original received signal E kp (t), which recovers the actual phase lag due to the k illuminated objects. E kp If (t) was stored in act 1107, act 1133 may be omitted.

[0112] In operation 1135, the phase-corrected internal reflection subtracted signal Ss(t) is converted into an internal reflection corrected signal S(t) which is used as the returned signal S(t) in further processing. PC E, which is used to average ensembles of P signals in the time domain to generate kp (t) or other E outside the averaging ensemble. k (t) or (t) k(t) is subtracted from the correction, where S S Since (t) is the phase-corrected average, it can be expressed as Equation 9, which is an improvement over Equation 12b.

number

[0113] In operation 1137, the corrected return signal S(t) is used in further processing, for example to determine range or any Doppler shift, or any quantity derived therefrom, or to contribute to some application involving many such measurements, such as object identification, mapping or navigation, or some combination, among others.

[0114] In act 1139, it is determined whether there are other returned signals in the averaging ensemble to correct for internal reflections. If there are other returned signals, control returns to act 1131. If there are no other returned signals, control proceeds to act 1141. In some embodiments, the returned signal to be corrected for internal reflections is outside the averaging ensemble, act 1139 is omitted, and control proceeds directly from act 1137 to act 1141.

[0115] In operation 1141 it is determined whether more measurements have been made. If more measurements have been made, control returns to operation 905a to illuminate the next spot. If more measurements have not been made, the process ends.

[0116] In systems that do not utilize I / Q detection, the retroreflected signal exhibits amplitude oscillations but not phase oscillations due to laser dithering. A strategy is used to track the Amplitude Evolution (AIp) due to laser dithering, and this amplitude evolution is corrected for each return before averaging. The basic strategy is to track the maximum retroreflected peak value observed for several shots. A "phase" is then calculated for each shot by comparing the retroreflected peak for that shot to the retained maximum value. Subsequent operations follow very similarly to the IQ approach where the amplitude of a given shot is normalized (instead of phased) before averaging. The shots are then averaged. The retroreflected signal estimate (to be subtracted) is this average multiplied by the amplitude of a given shot. The correction is applied as given in Equation 10a.

number

[0117] FIG. 12 is a flow chart illustrating an exemplary method for phase tracking internal reflections prior to subtraction according to another embodiment. The phase can be efficiently measured by calculating the dot product of the transmitted code with the received signal circularly shifted by the expected range index of the retro-reflected signal. The dot product calculates the same amplitude and phase results as the overall cross-correlation, but only at the range bins of interest. This method 1200 saves many calculations. From this phase calculation, the received signal is phase corrected in the time diagram. The corrected time domain signal is then averaged with other phase-corrected time measurements. This averaged time signal is used as the retro-reflected signal to subtract from future time measurements as described above.

[0118] Operations 901a through 905a, as described above, configure the transceiver (operation 901a), prepare the phase code block (operation 903a), and illuminate the next spot with the transmitted optical signal (operation 905a). In operation 1107, the return signal is received and, in some embodiments, stored in a non-transitory computer readable medium for further processing.

[0119] 11, in method 1200, the optical phase of the retroreflected signal is determined. However, unlike method 1100, the phase encoding signal B(t) and the returned signal E kp The full cross-correlation of (t) is not calculated for multiple time lags and associated distance indices. Instead, a dot product is calculated for a single time lag using a single associated time distance index.

[0120] Thus, in act 1211, an expected range (and associated time lag and distance index) is determined for the retro-reflected peak. For example, based on experimental data or a single previous cross-correlation, an internally reflected peak is known to occur at a particular distance and its corresponding distance index, ibr. In this example, act 1211 involves merely retrieving the value of ibr from Storage or any other method of receiving this data. Any method can be used to receive or retrieve this value, including retrieving this value as a constant in program code, retrieving it from a data field on a local or remote computer readable medium, searching in a database or flat file in response to a query message, or unsolicited.

[0121] In operation 1213, the returned signal is shifted back as given by Equation 17a. kp The predicted distance index is circularly shifted to provide (t).

number

[0122] In operation 1215, the dot product between the shifted return and the phase code is calculated to provide the range profile entry at index ibr, as expressed in Equation 17b.

[0123] In operation 1215, the phase of this entry is calculated. p The phase associated with this range index for the pth spot, denoted by , is a complex number given by Equation 13c, as discussed above.

[0124] In operation 1217, the return signal is corrected in the time domain by the phase determined in operation 1215, as given by Equation 14a above, and a sum or average is accumulated for the P returned signals, as described above for Equations 14b and 14c. In some embodiments, the phase corrected signal Sp(t) and the phase corrections are stored in a computer readable medium for later retrieval. In some of these embodiments, the received signal E kp (t) instead of the phase correction signal Sp(t) and the phase correction φI p is stored. Also, in operation 1217, the complex sum of the phase corrected signal and the summed n signals is accumulated and stored in a memory or special register or some other computer readable medium. The sum is given by Equation 14a or Equation 14b, discussed above. In some embodiments, operation 1217 also divides the sum by the number of signals to calculate the average signal after the n signals are combined, as given by Equation 14c. In other embodiments, the division is not performed until the sum of Equation 14b has been accumulated for all P signals.

[0125] In act 1119, it is determined whether the averaging period has been reached, e.g., n=P. After all P signals have been averaged (n=P), the average of Equation 14c provides the time domain internal reflection subtracted signal Ss(t) given by Equation 12a. This equivalent is expressed in Equation 14d. If P signals have not been accumulated, control returns to act 905a to illuminate the next spot. Otherwise, control proceeds to act 1131. Acts 1131 through 1141 are as described above for method 1100 in FIG.

[0126] 13 is a graph illustrating an exemplary internal reflection subtraction according to one embodiment. The horizontal axis represents distance index i (referred to as "measurement index" in the drawings). The vertical axis represents Suppression in dB, which is a measure of the reduction in cross-correlation after subtraction of Ss(t) calculated by method 700. The suppression at each index i is given in Equation 18.

number

[0127] In some implementations, the code is broken down into sections (i.e., 256 sample chunks) before performing one or more of the operations mentioned above (e.g., phase estimation and tracking using the dot product trick). If the SNR allows, the phase expansion (and thus the laser frequency) can be digitally tracked at a faster rate of time. This can slightly improve the phase tracking circulator subtraction. 5. Computer Hardware Overview

[0128] FIG. 14 is a block diagram illustrating a computer system 1400 on which an embodiment of the present invention can be implemented. The computer system 1400 includes a communication mechanism, such as a bus 1410, for communicating information between other internal and external components of the computer system 1400. The information is represented as a measurable current physical signal, typically a voltage, but in other embodiments includes phenomena such as magnetic interactions, electromagnetic interactions, pressure interactions, chemical interactions, molecular atomic interactions, and quantum interactions. For example, north and south pole magnetic fields, or zero and non-zero voltages indicate two states (0, 1) of a binary number (bit). Other phenomena may indicate higher base numbers. The superposition of multiple simultaneous quantum states before measurement indicates a quantum bit (qubit). One or more numeric sequences constitute digital data, which is used to indicate a code for a number or letter. In some embodiments, information, referred to as analog data, is represented on a near continuum of measurable values ​​within a particular range. The computer system 1400, or portions thereof, constitute a means for performing one or more steps of one or more methods described herein.

[0129] A sequence of binary digits constitutes digital data used to represent codes for numbers or letters. The bus 1410 may include many parallel conductors of information so that information may be rapidly transmitted between devices coupled to the bus 1410. One or more processors 1402 are coupled to the bus 1410 for processing information. The processor 1402 performs a series of operations on the information. The series of operations includes obtaining information from the bus 1410 and placing information on the bus 1410. The series of operations also typically includes comparing two or more units of information, shifting the position of the units of information, combining two or more units of information, such as adding or multiplying. The sequence of operations performed by the processor 1402 constitutes a computer instruction word.

[0130] The computer system 1400 also includes a memory 1404 coupled to the bus 1410. The memory 1404, such as a Random Access Memory (RAM) or other dynamic storage device, stores information including computer instructions. Dynamic memory allows information stored therein to be changed by the computer system 1400. RAM allows units of information stored at locations called memory addresses to be stored and retrieved independently of information at adjacent addresses. The memory 1404 is also used by the processor 1402 to store temporary values ​​during execution of computer instructions. The computer system 1400 also includes a Read Only Memory (ROM) 1406 or other static storage device coupled to the bus 1410 for storing static information including instructions that are not changed by the computer system 1400. A non-volatile (permanent) storage device 1408, such as a magnetic or optical disk, may also be coupled to the bus 1410 for storing information including instructions that persist even when the computer system 1400 is turned off or loses power.

[0131] Information, including commands, is provided to the bus 1410 for use by the processor from external input devices 1412, such as a keyboard including alphabetic keys operated by a human user, or sensors. The sensors detect conditions in their surroundings and convert such detections into signals compatible with those used to represent information in the computer system 1400. Other external devices coupled to the bus 1410 that are primarily used for human interaction include a display device 1414, such as a cathode ray tube (CRT) or liquid crystal display (LCD), for providing images, and a pointing device 1416, such as a mouse or trackball or cursor direction keys, for controlling the position of a small cursor image provided on the display 1414 and for issuing commands related to graphical elements provided on the display 1414.

[0132] In the illustrated embodiment, special purpose hardware such as Application Specific Integrated Circuits (ASICs) 1420 are coupled to bus 1410. The special purpose hardware is configured to perform operations sufficiently quickly for a special purpose that would not be performed quickly by processor 1402. Examples of ASICs include graphics accelerator cards for generating images for display 1414, Cryptographic Boards for encrypting and decrypting messages sent over a network, voice recognition, and interfacing with specialized external devices such as robotic arms and medical scanning equipment that repeatedly perform some complex sequence of operations that are more efficiently implemented in hardware.

[0133] The computer system 1400 also includes one or more examples of a communication interface 1470 coupled to the bus 1410. The communication interface 1470 provides a two-way communication coupling for various external devices that operate with their own processors, such as printers, scanners, and external disks. Typically, the coupling is performed using a network link 1478 that is connected to a local network 1480 to which various external devices having their own processors are connected. For example, the communication interface 1470 can be a parallel port or a serial port on a personal computer or a Universal Serial Bus (USB) port. In some embodiments, the communication interface 1470 is a telephone modem that provides a telecommunications connection to an Integrated Services Digital Network (ISDN) card or a Digital Subscriber Line (DSL) card or a corresponding type of telephone line. In some embodiments, the communication interface 1470 is a cable modem that converts signals on the bus 1410 into signals for communication connection via coaxial cable or optical signals for communication connection via fiber optic cable. As another example, the communication interface 1470 can be a LAN card that provides a data communication connection to a compatible Local Area Network (LAN) such as Ethernet. Wireless links can also be implemented. Carrier waves, such as acoustic and electromagnetic waves, including radio waves, optical waves, and infrared waves, pass through space without wires or cables. Signals include artificial changes in the amplitude, frequency, phase, polarization, or other physical properties of the carrier wave. For wireless links, the communication interface 1470 transmits and receives electrical, acoustic, or electromagnetic signals, including infrared and optical signals, that carry information streams such as digital data.

[0134] The term computer-readable medium is used herein to refer to any medium that participates in providing information, including instructions for execution, to the processor 1402. Such media may take a variety of forms, including but not limited to non-volatile media, volatile media, and transmission media. Non-volatile media include, for example, optical or magnetic disks, such as the storage device 1408. Volatile media include, for example, dynamic memory 1404. Transmission media include, for example, coaxial cables, copper wire, fiber optic cables, and waves that pass through space without wires or cables, such as radio waves, acoustic waves, and electromagnetic waves, including light waves and infrared waves. The term computer-readable storage medium is used herein to refer to any medium that participates in providing information to the processor 1402, excluding transmission media.

[0135] Common forms of computer readable media include, for example, a floppy disk, a flexible disk, a hard disk, a magnetic tape or any other magnetic medium, a compact disk ROM (CD-ROM), a digital video disk (DVD) or other optical medium, a punch card, a paper tape or any other physical medium having a pattern of holes, a RAM, a programmable ROM (PROM), an erasable PROM (EPROM), a FLASH®-EPROM or other memory chip or cartridge, a carrier wave, or any other medium from which a computer can read. The term computer readable non-transitory storage medium is used herein to refer to any medium that participates in providing information to the processor 1402, excluding carrier waves and other signals.

[0136] The logic encoded in one or more tangible media may include processor instructions on a computer-readable storage medium and / or special purpose hardware such as ASIC 1420.

[0137] The network link 1478 typically provides information communication through one or more networks to other devices that use or process the information. For example, the network link 1478 may provide a connection through a local network 1480 to a host computer 1482 or to equipment 1484 operated by an Internet Service Provider (ISP). The ISP equipment 1484 provides data communication services through a public, worldwide, packet-switched communication network of networks now commonly referred to as the Internet 1490. A computer connected to the Internet, referred to as a server 1492, provides services in response to information received over the Internet. For example, the server 1492 provides information representing video data for presentation on the display 1414.

[0138] The present invention relates to the use of computer system 1400 for implementing the techniques described herein. According to one embodiment of the present invention, the techniques are performed by computer system 1400 in response to processor 1402 executing one or more sequences of one or more instructions contained in memory 1404. These instructions, also referred to as software and program code, may be read into memory 1404 from another computer-readable medium, such as storage device 1408. Execution of the sequences of instructions contained in memory 1404 causes processor 1402 to perform the method steps described herein. In other embodiments, hardware, such as ASIC 1420, may be used in place of or in combination with software to implement the present invention. Thus, embodiments of the present invention are not limited to any specific combination of hardware and software.

[0139] Signals transmitted through other networks via network link 1478 and communication interface 1470 carry information to and from computer system 1400. Computer system 1400 may transmit and receive information, including program code, through networks 1480, 1490, among others, through network link 1478 and communication interface 1470. In one example using the Internet 1490, a server 1492 transmits program code for a particular application requested by a message sent from computer 1400 through the Internet 1490, the ISP equipment 1484, the local network 1480 and communication interface 1470. The received code may be executed by processor 1402 as it is received, or stored in storage device 1408 or other non-volatile storage device for later execution, or both. In this manner, computer system 1400 obtains application program code in the form of signals on a carrier wave.

[0140] Various forms of computer readable media may be involved in carrying one or more sequences of instructions and / or data to the processor 1402 for execution. For example, the instructions and data may initially be carried on a magnetic disk of a remote computer, such as the host 1482. The remote computer loads the instructions and data into its dynamic memory and transmits the instructions and data over a telephone line using a modem. A modem local to the computer system 1400 receives the instructions and data on the telephone line and uses an infrared transmitter to convert the instructions and data to an infrared carrier wave signal that serves as the network link 1478. An infrared detector, serving as the communications interface 1470, receives the instructions and data carried in the infrared signal and places information representing the instructions and data on the bus 1410. The bus 1410 carries the information to the memory 1404, and the processor 1402 retrieves and executes the instructions from the memory 1404 using a portion of the data transmitted with the instructions. The instructions and data received by memory 1404 may optionally be stored on storage device 1408 before or after execution by processor 1402.

[0141] FIG. 15 illustrates a chipset 1500 in which an embodiment of the present invention can be implemented. The chipset 1500 is programmed to perform one or more operations of the methods described herein and includes, for example, the processor and memory components described in connection with FIG. 14 included in one or more physical packages (e.g., chips). By way of example, the physical package includes an arrangement of one or more materials, components, and / or wires on a structural assembly (e.g., baseboard) to provide one or more properties such as physical strength, size conservation, and / or limited electrical interaction. It is contemplated that in certain embodiments, the chipset may be embodied in a single chip. The chipset 1500, or portions thereof, constitute a means for performing one or more operations of the methods described herein.

[0142] In one embodiment, the chipset 1500 includes a communication mechanism, such as a bus 1501, for communicating information between components of the chipset 1500. The processor 1503 is coupled to the bus 1501 to execute instructions and process information stored in, for example, the memory 1505. The processor 1503 may include one or more processing cores, each configured to perform independently. A multi-core processor allows for multiple processing within a single physical package. Examples of multi-core processors include two, four, eight or more processor cores. Alternatively or additionally, the processor 1503 may include one or more microprocessors configured in series via the bus 1501 to allow for independent execution of instructions, pipelining and multithreading. The processor 1503 may also include one or more specialized components for performing certain processing functions and tasks, such as one or more Digital Signal Processors (DSPs) 1507 or one or more ASICs 1509. The DSP 1507 is typically configured to process real-world signals (e.g., sound) in real time independently of the processor 1503. Similarly, the ASIC 1509 may be configured to perform specialized functions not easily performed by a general-purpose processor. Other specialized components for assisting in performing the inventive functions described herein include one or more Field Programmable Gate Arrays (FPGAs) (not shown), one or more controllers (not shown), or one or more other special purpose computer chips.

[0143] The processor 1503 and associated components are coupled to memory 1505 via bus 1501. The memory 1505 includes both dynamic memory (e.g., RAM, magnetic disk, writable optical disk, etc.) and static memory (e.g., ROM, CD-ROM, etc.) for storing executable instructions that, when executed, perform one or more operations of the methods described herein. The memory 1505 stores data associated with or generated by the execution of one or more steps of the methods described herein. 6. Modifications, Enhancements and Modifications

[0144] In the foregoing specification, the invention has been described with reference to specific embodiments. However, it will be apparent that various modifications and changes may be made without departing from the broader spirit and scope of the invention. The specification and drawings are therefore to be regarded in an illustrative and not a restrictive sense. Throughout this specification and claims, unless the context otherwise requires, the word "comprises" and variations thereof, such as "comprises" and "comprising", will be understood to mean the inclusion of a mentioned item, element or operation, or group of items, elements or operations, without excluding any other items, elements or operations, or group of items, elements or operations. Also, the indefinite article is intended to represent one or more items, elements or operations modified by the article.

[0145] Numerical ranges and parameters indicating broad ranges, even if approximate, are reported as precisely as possible, such as the numerical values ​​set forth in the specific non-limiting examples. However, any numerical value inherently contains certain errors necessarily resulting from the standard deviation found in their respective experimental measurements at the time of writing. Moreover, unless the context clearly dictates otherwise, numerical values ​​presented herein have an implied precision given by the lowest digit. Thus, a value of 1.1 implies a value from 1.05 to 1.15. The term "about" is used to indicate a broader range around a given value, and unless the context clearly dictates otherwise, implies a broader range around the lowest digit, such as "about 1.1" implies 1.0 to 1.2. If the lowest digit is unknown, the term "about" means two times. For example, "about X" implies a value in the range of 0.5X to 2X, such as about 100 implies a value in the range of 50 to 200. Moreover, all ranges disclosed herein should be understood to include any and all subranges subsumed therein. For example, a range "less than 10" for a positive parameter includes any and all subranges between (and including) a minimum value of 0 (Zero) and a maximum value of 10, i.e., any and all subranges having a maximum value of greater than or equal to 0 and a maximum value of less than or equal to 10, e.g., 1 to 4.

[0146] Some embodiments of the present invention are described herein in the context of binary, π / 2 (90 degree) phase encoding of a radio frequency signal modulated onto an optical signal. However, the embodiments are not limited to this context. For example, in some embodiments, other phase encodings using different phase differences (e.g., 30, 60 or 180 degrees) or encodings using three or more different phases are used. In other embodiments, other distance measurement modulation formats use time-separated in-phase and quadrature-phase local oscillator signals as the reference signal. Such modulation formats include, but are not limited to, "On Off Keying", "Frequency Shift Keying" and "Noise" waveforms. Although the embodiments are described in the context of a single optical beam and return to a single detector or detector pair, other embodiments may use any known scanning means, such as linear stepping or rotating optics, or may be scanned using a transmitter array or detector array or detector pair.

Claims

1. a laser for providing an optical signal; a modulator that modulates the optical signal from the laser to generate a modulated optical signal; an optical coupler that generates a reference optical signal related to the optical signal from the laser, transmits the modulated optical signal toward an object, and receives a returned optical signal in response to transmitting the modulated optical signal toward the object; an optical mixer that mixes the returned optical signal with a reference optical signal to generate a mixed optical signal, the reference optical signal being related to the optical signal from the laser; a photodetector that detects the mixed optical signal and generates an electrical signal; a processor that determines parameters of internal reflections of the returned optical signal from one or more optical components based on the electrical signal and the modulated optical signal, and operates a vehicle based on the parameters of the internal reflections.

2. 2. The system of claim 1, wherein the parameter is selected from at least one of phase, amplitude, frequency or angular frequency, and the modulator modulates the reference optical signal before mixing the returned optical signal with the reference optical signal, and the optical signal is modulated based on a phase sequence of the optical signal.

3. 3. The system of claim 2, wherein the optical coupler transmits the modulated optical signal toward the object for each of a first plurality of measurements and receives the returned optical signal for each of the first plurality of measurements.

4. 4. The system of claim 3, wherein the optical mixer generates the mixed optical signal for each of the first plurality of measurements, and the optical detector generates the electrical signal for each of the first plurality of measurements.

5. The system of claim 4 , wherein the parameters include a phase of the internal reflection for each of the first plurality of measurements.

6. 6. The system of claim 5, wherein the processor determines the parameters of the internal reflection based on a dot product of the modulated optical signal and the electrical signal circularly shifted by an expected range index for each of the first plurality of measurements.

7. The processor, determining an average of a second plurality of measurements of the electrical signal, the second plurality of measurements including one or more measurements, the average being determined based at least in part on the phase of the internal reflection for each of the second plurality of measurements; 6. The system of claim 5, further comprising: (i) determining the average of the second plurality of measurements; and (ii) determining a corrected return signal based at least in part on the electrical signal for at least one of the first plurality of measurements.

8. The system of claim 7 , wherein the processor operates the vehicle based on the corrected return signal.

9. modulating an optical signal from a laser to generate a modulated optical signal; transmitting the modulated optical signal towards an object; receiving a returned optical signal in response to the step of transmitting the modulated optical signal; mixing the returned optical signal with a reference optical signal to generate a mixed optical signal, the reference optical signal being related to the optical signal from the laser; detecting the mixed optical signal to generate an electrical signal; determining parameters of internal reflection of the returned optical signal from one or more optical components based on the electrical signal and the modulated optical signal; and actuating a vehicle based on the parameter of the internal reflection.

10. modulating the reference optical signal before mixing the returned optical signal with the reference optical signal; The parameters include phase, amplitude, frequency or angular frequency, and 10. The method of claim 9, wherein modulating the optical signal from the laser to generate the modulated optical signal comprises modulating the optical signal based on a phase sequence of the optical signal.

11. 11. The method of claim 10, wherein transmitting the modulated optical signal towards the object comprises transmitting the modulated optical signal for each of a first plurality of measurements, and receiving the returned optical signal comprises receiving the returned optical signal for each of the first plurality of measurements.

12. The method of claim 11 , wherein the mixed optical signal is generated for each of the first plurality of measurements and the electrical signal is generated for each of the first plurality of measurements.

13. The method of claim 12 , wherein the parameters include a phase of the internal reflection for each of the first plurality of measurements.

14. The step of determining the parameters of the internal reflections comprises: determining the parameter of the internal reflection based on a dot product of the modulated optical signal and the electrical signal circularly shifted by an expected range index for each of the first plurality of measurements; determining an average of a second plurality of measurements of the electrical signal, the second plurality including one or more measurements, the average being determined based at least in part on the phase of the internal reflection for each of the second plurality of measurements; 14. The method of claim 13, further comprising: (i) determining a corrected return signal based at least in part on the average of the second plurality of measurements; and (ii) the electrical signal for at least one of the first plurality of measurements.

15. The method of claim 14 further comprising operating the vehicle based on the corrected return signal.

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