Control apparatus of synchronous motor

Abstract

To provide a control apparatus of a synchronous motor that can secure the accuracy of phase difference θe even if the synchronous motor is suddenly accelerated or decelerated.SOLUTION: A control apparatus 1 of a synchronous motor 200 controls the synchronous motor 200 by using a phase (a rotor position estimation value θest) of a dc-axis, with a rotor position of the synchronous motor 200 being d-axis, an axis progressed from the d-axis by 90 degrees being q-axis, an axis with a phase difference between the d-axis and the θe being dc-axis, and an axis progressed from the dc-axis by 90 degrees being qc-axis. A processor of the control apparatus 1 of the synchronous motor 200 calculates the θe using voltages (voltage commands vdc, vqc) of the dc-axis and the qc-axis, an angular speed (an electric angular speed estimation value ω^) of the dc-axis, a characteristic value (a resistance value R and a statistic q-axis inductance Lq) of the synchronous motor 200, and magnetic fluxes (Φdc and Φqc) of the dc-axis and the qc-axis, without using an approximated value of the angular speed (an electric angular speed true value ω) of the d-axis.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to a control device for a synchronous motor.

Background Art

[0002] In electrified vehicles typified by hybrid vehicles and electric vehicles, a permanent magnet synchronous motor (hereinafter referred to as a synchronous motor) having a permanent magnet on the rotor is often used as a drive source for the vehicle. In order to drive and control such a synchronous motor, an inverter, which is a power converter, is generally used. And, as a method of controlling the torque of the synchronous motor by the inverter, current vector control is generally known. In order to perform current vector control, it is necessary to accurately grasp the magnetic pole position of the rotor of the synchronous motor. As one means for grasping this magnetic pole position, a resolver, which is a position sensor, is often provided in the synchronous motor.

[0003] However, providing a resolver in the synchronous motor has led to an increase in the number of system components, an increase in size, and an increase in cost.

[0004] In addition, the resolver has an output winding for outputting a position signal, and there is a risk of failure due to disconnection or short circuit of the winding, which also leads to a decrease in the reliability of the system.

[0005] Therefore, sensorless vector control is known as a technique for driving a synchronous motor without using a sensor such as a resolver.

[0006] Sensorless vector control obtains the rotor position information based on information such as the voltage applied to the synchronous motor and the current flowing due to the voltage, and performs current vector control. Since a position sensor such as a resolver is not required, it is possible to prevent the system from becoming larger, the cost from increasing, and the reliability from decreasing. The method for estimating the rotor position is roughly classified into a method based on the position dependence of the inductance of the synchronous motor and a method based on the induced voltage of the synchronous motor. In the low-speed region where the induced voltage of the synchronous motor is zero or small, the former method is used, and in the medium- and high-speed regions where the induced voltage is high, it is generally known to switch to the latter method to perform sensorless control.

[0007] As a sensorless control of a synchronous motor in the medium- and high-speed region, a method based on an extended induced voltage (Patent Document 1) is disclosed. Further, Patent Document 2 discloses a method based on magnetic flux that enables highly accurate position estimation regardless of the speed range of the synchronous motor.

Prior Art Documents

Patent Documents

[0008]

Patent Document 1

Patent Document 2

Summary of the Invention

Problems to be Solved by the Invention

[0009] In Patent Document 1, the rotor position of the synchronous motor is defined as the d-axis, and the axis orthogonal to the d-axis is defined as the q-axis. A coordinate system with a phase difference between these dq axes and θe is defined as the dc-qc axis, and sensorless control is performed by estimating the extended induced voltage to obtain the information of θe. However, there is a problem that the extended induced voltage cannot be accurately detected unless the speed of the synchronous motor is in the medium- and high-speed region.

[0010] On the other hand, in Patent Document 2, based on the voltage equation of the synchronous motor, an observer is configured to estimate the magnetic flux using the voltage command value, current detection value, and current estimated value of the synchronous motor, and the position is estimated using the magnetic flux estimated value obtained by the observer. In a synchronous motor equipped with a permanent magnet on the rotor, since the interlinking magnetic flux generated by the permanent magnet always occurs regardless of the speed, by utilizing this, sensorless position detection can be realized in a wide speed range.

[0011] However, when a current flows through the windings of the synchronous motor, a magnetic flux is generated by the current, so a phenomenon (magnetic saturation) occurs in which the magnetic flux linking the stator of the synchronous motor becomes over-dense and the inductance decreases. As a result, the pole arrangement of the observer changes, so there is a risk that the estimation performance (responsiveness and accuracy) of the observer will deteriorate. By reflecting the inductance change due to magnetic saturation in the estimation formula of the observer according to the current detection value, etc., it is possible to consider magnetic saturation, but since the pole arrangement calculation is performed in real time, there is a problem that the calculation becomes complicated.

[0012] In addition to the inductance change due to magnetic saturation, the inductance has a dynamic inductance and a static inductance. The former is represented by the slope of the magnetic flux near the current operating point, and the latter represents the slope of the magnetic flux between the current operating point and the origin. The extended induced voltage method represented by Patent Document 1 requires the use of quantities different from both the d-axis magnetic flux and the q-axis magnetic flux, such as the product of the static inductance of the q-axis and the d-axis current, and the product of the dynamic inductance of the d-axis and the q-axis current, and it is difficult to consider the static and dynamic inductances. Even in a configuration using an observer as in Patent Document 2, separately configuring the static and dynamic inductances causes a problem of complexity.

[0013] Furthermore, in both Patent Document 1 and Patent Document 2, it is necessary to assume that the speed of the dc axis (= estimated speed) is equal to the speed of the d axis (true speed). In regions where the assumption does not hold, such as during rapid acceleration or deceleration of the synchronous motor, a position estimation error may occur and the control system may become unstable.

[0014] An object of the present invention is to provide a control device for a synchronous motor that ensures the accuracy of the phase difference θe (axis error) even when the synchronous motor accelerates or decelerates rapidly.

Means for Solving the Problems

[0015] To achieve the above object, the present invention uses the position of the rotor of a synchronous motor as the d-axis, the axis advanced 90 degrees from the d-axis as the q-axis, the axis having a phase difference θe from the d-axis as the dc-axis, and the axis advanced 90 degrees from the dc-axis as the qc-axis, and is a control device for a synchronous motor that controls the synchronous motor using the phase of the dc-axis, and has a processor that calculates the phase difference θe using the voltages of the dc-axis and the qc-axis, the angular velocity of the dc-axis, the characteristic values of the synchronous motor, and the fluxes of the dc-axis and the qc-axis without using an approximate value of the angular velocity of the d-axis.

Effects of the Invention

[0016] According to the present invention, the accuracy of the phase difference θe (axis error) is ensured even when the synchronous motor accelerates or decelerates rapidly. Problems, configurations, and effects other than those described above will be clarified by the description of the following embodiments.

Brief Description of the Drawings

[0017]

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5

Figure 6

Figure 7

Figure 8

Figure 9

Figure 10

Embodiments for Carrying Out the Invention

[0018] Hereinafter, embodiments of the present invention will be described with reference to the drawings. The following description and drawings are examples for explaining the present invention, and for the sake of clarity of explanation, appropriate omissions and simplifications have been made. The present invention can also be implemented in various other forms. Unless otherwise particularly limited, each component may be singular or plural.

[0019] [First Embodiment] FIG. 1 is a configuration diagram of a drive device for a synchronous motor according to this embodiment. The drive device for the synchronous motor according to this embodiment has a position estimation technique that can consider changes in inductance due to magnetic saturation and static / dynamic inductance, and further does not require the assumption that the estimated speed is equal to the true speed.

[0020] The drive device for the synchronous motor includes a position estimation unit 100A, a synchronous motor 200, a power converter 300, a current command calculation unit 400, a current detection unit 500, a current control unit 600, and a coordinate inverse conversion unit 700A and a coordinate conversion unit 700B, and drives the synchronous motor 200. In the example of this embodiment, the position estimation unit 100A, the current command calculation unit 400, the current control unit 600, and the coordinate inverse conversion unit 700A and the coordinate conversion unit 700B are realized by the control device 1. The control device 1 is, for example, a microcomputer, and is composed of a memory (storage device), a CPU (processor), an input / output circuit (communication device), etc. The CPU functions as each unit by executing a program stored in the memory.

[0021] The synchronous motor 200 is a permanent magnet type synchronous motor, which is composed of a permanent magnet (ferromagnetic material) on the rotor and armature windings on the stator. In this embodiment, there is no position sensor for detecting the magnetic pole position of the synchronous motor 200. Instead, the position estimation unit 100A estimates the magnetic pole position.

[0022] The power converter 300 is, for example, an inverter. When driving the synchronous motor 200, it pulse-width modulates (PWM) the voltage commands Vu*, Vv*, Vw* from the coordinate inverse conversion unit to control the on / off of the semiconductor switch elements of the power converter. Thereby, voltages Vu, Vv, Vw are applied to the synchronous motor 200 to drive the synchronous motor 200.

[0023] The current command calculation unit 400 receives a torque command T* as a drive command for the synchronous motor 200 from an upper control device (not shown), calculates current commands idc* and iqc*, which are the target values of the current flowing through the synchronous motor 200, and transmits them to the current control unit 600.

[0024] The current detection unit 500 is composed of current sensors 500u, 500v, 500w that detect the three-phase current flowing through the synchronous motor 200. The current sensors 500u, 500v, 500w are respectively arranged in each phase of the synchronous motor 200.

[0025] The current detection unit 500 detects the three-phase currents Iu, Iv, Iw and transmits signals to the coordinate conversion unit 700B. Although an example is shown where the current sensors 500u, 500v, 500w are arranged in each of the three phases of the synchronous motor 200, taking advantage of the fact that the sum of the three-phase alternating currents is zero, they may be arranged in only two phases (for example, the u-phase and the v-phase). Also, a configuration may be adopted to obtain the three-phase current of the synchronous motor 200 from the current flowing through the DC bus (not shown) on the input side of the power converter 300. With these configurations, the number of current sensors can be reduced, and cost reduction can be achieved.

[0026] The current control unit 600 receives the current commands idc* and iqc* from the current command calculation unit and the current values idc and iqc from the coordinate conversion unit 700B, and adjusts and controls the voltage commands vdc and vqc so that idc* and idc, and iqc* and iqc match respectively. The configuration of the current control unit 600 can take various forms that generally combine known proportional-integral control, control for compensating the armature reaction of a synchronous motor, control for compensating the dead time of the power converter 300, and so on.

[0027] The inverse coordinate conversion unit 700A receives the voltage commands vdc and vqc, which are signals from the current control unit, and performs conversion processing based on the estimation result θest of the position estimation unit 100A to obtain the three-phase AC voltage commands Vu*, Vv*, and Vw*.

[0028] The coordinate conversion unit 700B receives the three-phase AC currents Iu, Iv, and Iw obtained from the current detection unit 500 and performs conversion processing based on the estimation result θest of the position estimation unit 100A to obtain the currents idc and iqc in the dc-qc coordinate system.

[0029] The position estimation unit 100A is a functional block that brings about the main effects in this embodiment. It receives vdc and vqc, which are output signals of the current control unit 600, and idc and iqc, which are output signals of the coordinate conversion unit 700B, and calculates the rotor position of the synchronous motor 200 as its estimated value θest based on these.

[0030] Next, the basic principle of the present invention will be explained. Equations (1) and (2) are voltage equations in the dc-qc coordinates using the extended induced voltage, which is a prior art.

[0031]

Equation

[0032]

Equation

[0033] However, R is the winding resistance of the synchronous motor, Ld is the static d-axis inductance of the synchronous motor, Lq is the static q-axis inductance of the synchronous motor, Ldh is the dynamic d-axis inductance of the synchronous motor, Lqh is the dynamic q-axis inductance of the synchronous motor, ω is the angular velocity of the d-axis (true value of the electrical angular velocity of the synchronous motor), ω^ is the angular velocity of the dc-axis (estimated value of the electrical angular velocity of the synchronous motor), Eex is the extended induced voltage, p is the differential operator, and p() represents the time derivative inside the parentheses.

[0034] As shown in Equation (1) and Equation (2), the extended induced voltage components of the dc-axis and qc-axis are represented by trigonometric functions using the axis error θe. Therefore, by calculating the extended induced voltage components of the dc-axis and qc-axis, it is possible to estimate the axis error θe.

[0035] However, Equation (1) and Equation (2) require ω, which is the true value of the electrical angular velocity. In sensorless control, since the true value of ω is unknown, it is necessary to estimate the extended induced voltage components of the dc-axis and qc-axis using the approximation that ω is equal to ω^, which is the estimated value of ω. In the steady state where the rotational speed of the synchronous motor is constant, this approximation holds. However, in the case of transient operations such as sudden changes in the rotational speed, this approximation does not hold, and thus there is a risk of generating an error in position estimation.

[0036] In addition, dynamic inductance and static inductance are mixed in Equation (1) and Equation (2). Considering these using the methods described in Patent Document 1 and Patent Document 2 in order to improve the position estimation accuracy will complicate the calculation.

[0037] Furthermore, generally, the multiplication of inductance and current results in magnetic flux. However, the second term of Equation (2) includes the product of the dynamic d-axis inductance and the qc-axis current, and the third term includes the product of the static q-axis inductance and the dc-axis current. Since these are products between axes that are orthogonal to each other, they are physical quantities different from the magnetic flux of the dc-axis or qc-axis. Therefore, it causes a problem that it is difficult to configure a magnetic flux observer as in Patent Document 2.

[0038] Therefore, in this embodiment, these are solved by the following means.

[0039] In the d-q coordinate system, which is the coordinate system of a synchronous motor, when using the d-axis and q-axis fluxes instead of the d-axis and q-axis currents, the following equations (3) and (4) are obtained.

[0040]

Number

[0041]

Number

[0042] Here, Φd: d-axis flux (flux generated by d-axis current and permanent magnet), Φq: q-axis flux (flux generated by q-axis current), Φm: interlinking flux by permanent magnet.

[0043] Equation (3) can be transformed into the form of the following equation (5).

[0044]

Number

[0045] Equation (5) is the transformation of equation (3) such that the coefficient of Φd in the first term on the right side of equation (3) becomes (R / Lq).

[0046] When the voltage equations of the d-axis and q-axis expressed based on the fluxes of equations (5) and (4) are transformed from the d-q coordinate system to the dc-qc coordinate system with a phase difference of θe, the following equations (6) and (7) are obtained.

[0047]

Number

[0048]

Number

[0049] As shown in Expressions (6) and (7), the terms of the axis error θe appear on the d - axis and q - axis respectively, and the axis error can be estimated in the same way as the conventional extended induced voltage.

[0050] In addition, since the term of the true value of the electrical angular velocity ω is not included and it is expressed only by the estimated value ω^, it is possible to accurately estimate the position even when transient speed fluctuations occur.

[0051] Furthermore, since the right - hand sides of Expressions (6) and (7) are expressed only using the magnetic fluxes including the information of the inductance (only the static inductance), it is not necessary to distinguish and consider the dynamic and static inductances, and it is possible to improve the position estimation accuracy and simplify the calculation.

[0052] From the relationship of Expressions (6) and (7), the axis error θe can be expressed using the following Expression (8).

[0053]

Equation

[0054] Note that arctan is the inverse tangent function. As the characteristics of the synchronous motor 200 required, there are two types: the resistance value R and the static q - axis inductance Lq. Since the d - axis inductance is not required, it is robust against the electrical constants of the synchronous motor 200.

[0055] Next, an embodiment using the basic principle of the present invention described above will be described with reference to FIG. 2. FIG. 2 is a configuration diagram of a position estimation unit 100A according to the first embodiment.

[0056] The position estimation unit 100A includes a dc - qc axis magnetic flux calculation unit 110, an axis error calculation unit 120, a speed calculation unit 130, and a position calculation unit 140.

[0057] The dc - qc axis magnetic flux calculation unit 110 calculates the dc - qc axis magnetic fluxes Φdc and Φqc using the dc - qc axis currents idc and iqc as input values.

[0058] The axis error calculation unit 120 calculates the axis error θe using the vdc and vqc which are the dc-qc axis voltages, the Φdc and Φqc which are the dc-qc axis fluxes, and the estimated value ω^ of the electrical angular velocity as input values.

[0059] The speed calculation unit 130 calculates the estimated value ω^ of the electrical angular velocity using the axis error θe as an input value.

[0060] The position calculation unit 140 calculates the estimated value θest of the rotor position of the synchronous motor 200 using the estimated value ω^ of the electrical angle as an input value.

[0061] Hereinafter, each functional block constituting the position estimation unit 100A will be described.

[0062] FIG. 3 shows the dc-qc axis flux calculation unit 110 in the position estimation unit 100A. The dc-qc axis flux calculation unit 110 includes a dc axis flux calculation unit 111 and a qc axis flux calculation unit 112.

[0063] The dc axis flux calculation unit 111 has the characteristics of the dc axis flux Φdc with respect to the dc axis and qc axis currents idc and iqc in a table or the like. According to the values of the dc axis and qc axis currents idc and iqc, the value of the dc axis flux Φdc is referred to and output.

[0064] The qc axis flux calculation unit 112 has the characteristics of the qc axis flux Φqc with respect to the dc axis and qc axis currents idc and iqc in a table or the like. According to the values of the dc axis and qc axis currents idc and iqc, the value of the qc axis flux Φqc is referred to and output.

[0065] The dc axis flux calculation unit 111 and the qc axis flux calculation unit 112 may configure a table using the characteristic analysis values of the synchronous motor 200 and the experimental data obtained in advance.

[0066] As described above, since the dc-axis flux calculation unit 111 and the qc-axis flux calculation unit 112 represent the characteristics of the flux with respect to the current, they include inductance information. Therefore, by using these fluxes, it is not necessary to distinguish between the dynamic and static inductances, and the calculation can be simplified and the position estimation accuracy can be improved.

[0067] The axis error calculation unit 120 calculates the axis error θe, which is the phase difference between the d-axis and the dc-axis, based on the above-described equation (8). In the present embodiment, the inputs to the axis error calculation unit 120 are the dc-axis and qc-axis voltages vdc, vqc, the dc-axis and qc-axis fluxes Φdc, Φqc, and the estimated speed ω^. This is to simplify the expression of equation (8) by omitting the time differentiation terms p(Φdc) and p(Φqc). As a result, the calculation can be simplified. That is, the axis error θe may be calculated using the following equation (9) instead of equation (8).

[0068]

Equation

[0069] However, since the time differentiation of the flux is not considered in equation (9), the dynamic inductance is not considered. This may cause a transient position estimation error, so it may be calculated as in equation (8). When calculating according to the configuration of equation (8), as shown in FIG. 4, a differential calculation unit 150 for differentiating the dc-axis and qc-axis fluxes Φdc, Φqc may be provided, and the results of the differential calculation may be transmitted to the axis error calculation unit 120 as pΦdc, pΦqc. Note that the difference from the position estimation unit 100A in FIG. 2 is the presence or absence of the differential calculation unit 150, so the same reference numerals are used in FIG. 4 for the blocks having the same functions as those in FIG. 2.

[0070] The calculation in the differential calculation unit 150 may be a pure time differentiation, or a filter may be provided for the differential value to avoid noise amplification. In that case, phase advance compensation may be performed to compensate for the delay due to the filter.

[0071] In addition to calculating the dc-axis flux Φdc and the qc-axis flux Φqc using the characteristic table for the current described above, it is also possible to perform the calculation based on the voltage equation. The voltage equations on the dc-qc axes can also be expressed by the following equations (10) and (11).

[0072]

Number

[0073]

Number

[0074] Therefore, from the relationships in equations (10) and (11), the dc-axis flux Φdc and the qc-axis flux Φqc can be obtained from the following equations (12) and (13).

[0075]

Number

[0076]

Number

[0077] Since the calculations of equations (12) and (13) are composed of only simple arithmetic operations, the computational load on the processor can be reduced compared to the case of providing the flux characteristic table for the current described above. Also, similar to the case of providing the table, the flux can be derived while considering the static and dynamic inductances. In this case, as shown in FIG. 6, the dc-qc axis flux calculation unit 110 is configured to calculate the dc-axis flux Φdc and the qc-axis flux Φqc by taking the dc-axis voltage vdc and the qc-axis voltage vqc and the dc-axis current idc and the qc-axis current iqc as inputs.

[0078] The axis error calculation unit 120 can also have the configuration shown in FIG. 7. Substituting Φqc of equation (13) into Φqc of equation (6), the term of the axis error θe in equation (6) can be expressed by the following equation (14).

[0079] [Number]

[0080] Similarly, substituting the Φdc in Equation (12) into the Φdc in Equation (7), the term of the shaft error θe in Equation (7) can be expressed by the following Equation (15).

[0081] [Number]

[0082] From the relationship between Equation (14) and Equation (15), the shaft error θe can be calculated by the following Equation (16).

[0083] [Number]

[0084] Therefore, the dc-axis voltage vdc and the qc-axis voltage vqc are not required for the calculation of the shaft error θe. As a result, an advantage is obtained in that it is less affected by the output voltage error caused by the dead time or the like in the power converter. Furthermore, the characteristics of the synchronous motor to be used are only the static q-axis inductance Lq, and the robustness with respect to the parameters of the synchronous motor can be enhanced. When using Equation (16), the calculations of the dc-axis magnetic flux Φdc and the qc-axis magnetic flux Φqc may be performed by the method using the above-described table or the method based on the voltage equation.

[0085] FIG. 5 is a diagram showing the configuration of the speed calculation unit 130 in the position estimation unit 100A. The speed calculation unit 130 includes a subtractor 131, a proportionality unit 132, an integrator 133, and an adder, with the shaft error θe as an input value. Further, a command value CMD for setting the shaft error θe to a predetermined value is set, and the estimated speed ω^ is calculated by making the shaft error θe follow CMD.

[0086] The subtractor 131 calculates the deviation between the command value CMD and the axis error θe, and transmits it as an input value to the proportional controller 132 and the integrator 133.

[0087] The proportional controller 132 amplifies the deviation between the command value CMD and the axis error θe, and transmits it to the adder 134.

[0088] The integrator 133 time-integrates and amplifies the deviation between the command value CMD and the axis error θe, and transmits it to the adder 134.

[0089] The adder 134 receives the output signals from the proportional controller 132 and the integrator 133 as input values, and calculates the speed estimated value ω^ by adding their respective outputs.

[0090] The speed calculation unit 130 performs follow-up control on θe to make the speed estimated value ω^ match the actual speed ω when CMD and θe match.

[0091] CMD is, for example, zero. If θe is made to follow zero with CMD being zero, the dc axis and the d axis will match, so it is possible to perform sensorless drive equivalent to control using resolver-based rotor position (= d axis) detection.

[0092] CMD may use a value other than zero. For example, there is a method of defining a coordinate system such that the d-axis and q-axis currents are always at maximum torque / current (MTPA: Maximum Torque Per Ampere). When driving the synchronous motor 200 in this coordinate system, a value that gives MTPA may be set for CMD, and CMD does not necessarily have to be constant. Alternatively, CMD may be set to a known coordinate system.

[0093] Also, the speed calculation unit 130 in this embodiment is configured to make the axis error θe follow CMD by proportional-integral control, but it does not necessarily have to be proportional-integral control. Various known technical configurations can be adopted, such as providing derivative control or feed-forward compensation.

[0094] The position calculation unit 140 integrates the estimated speed ω^ to calculate an estimated value θest of the rotor position of the synchronous motor 200.

[0095] In this embodiment, a dc-qc axis flux calculation unit 110 is provided in the position estimation unit 100A in the control device of the synchronous motor to calculate the dc-qc axis fluxes Φdc and Φqc. Further, the axis error calculation unit 120 is configured to receive as inputs the dc-qc axis voltages vdc and vqc, the dc-qc axis fluxes Φdc and Φqc, and the estimated speed ω^, and calculate the axis error θe from the voltage equation based only on the fluxes.

[0096] The dc-qc axis flux calculation unit is configured to have a relationship with the dc axis and qc axis currents idc and iqc and the dc axis and qc axis fluxes, so that it is possible to easily consider the inductance change due to magnetic saturation. Further, by providing the differential calculation unit 150, it is possible to consider not only the static inductance but also the dynamic inductance.

[0097] Furthermore, the axis error calculation unit 120 is configured to calculate the axis error θe based on the voltage equation in the dc-qc coordinates expressed by fluxes without using currents. As a result, considering the above-mentioned magnetic saturation and dynamic / static inductances, the axis error θe can be calculated without using the approximation that the true speed and the estimated speed are equal, which was necessary in the past. Therefore, it is possible to accurately estimate the position even when the characteristics of the synchronous motor change due to current changes or when the rotational speed suddenly changes.

[0098] Therefore, by using the control device of the synchronous motor in this embodiment as the drive motor of the electric vehicle, it is possible to realize sensorless positioning, reduce costs, and miniaturize the system size. Since the position estimation accuracy is improved, the control performance (such as torque accuracy and torque response) of the synchronous motor is improved, and a comfortable riding experience can be provided to the passengers.

[0099] [Second Embodiment] FIG. 8 is a configuration diagram of a drive device for a synchronous motor according to a second embodiment. The difference from the first embodiment is that there are a magnetic flux command calculation unit 800 instead of the current command calculation unit 400, a magnetic flux control unit 900 instead of the current control unit 600, a position estimation unit 100B, and the dc-qc axis magnetic flux calculation unit 110 in the position estimation unit 100A is provided at the subsequent stage of the coordinate conversion unit 700B.

[0100] The main function of the dc-qc axis magnetic flux calculation unit 110 is the same as that in the first embodiment. Taking the dc axis current idc and the qc axis current iqc obtained from the coordinate conversion unit 700B as inputs, it calculates the dc axis magnetic flux Φdc and the qc axis magnetic flux Φqc. Note that the calculation method of the dc-qc axis magnetic flux calculation unit 110 in FIG. 8 shows the configuration shown in FIG. 3 as an example in the same manner as in the first embodiment. Additionally, as another configuration, the dc axis voltage vdc and the qc axis voltage vqc can be added as inputs and FIGS. 6 and equations (12) and (13) can be used.

[0101] The magnetic flux command calculation unit 800 receives a torque command T* as a drive command for the synchronous motor 200 from an upper control device (not shown), calculates a dc axis magnetic flux command Φdc* and a qc axis magnetic flux command Φqc*, which are target values of the magnetic flux generated in the synchronous motor 200, and transmits them to the magnetic flux control unit 900.

[0102] The magnetic flux control unit 900 receives the magnetic flux commands Φdc*, Φqc* from the magnetic flux command calculation unit 800 and the magnetic fluxes Φdc, Φqc from the dc-qc axis magnetic flux calculation unit 110, and adjusts and controls the voltage commands vdc, vqc so that Φdc* and Φdc, Φqc* and Φqc respectively match. By controlling based on the magnetic flux, the dynamic inductance and the static inductance can be considered in the drive control, and the torque responsiveness and torque accuracy of the drive control of the synchronous motor 200 can be made higher. Also, the magnetic flux control unit 900 outputs p(Φdc) and p(Φqc), which are the differential values of Φdc and Φqc, based on the deviation between Φdc* and Φdc and the deviation between Φqc* and Φqc, which will be described later, and transmits them to the position estimation unit 100B.

[0103] The position estimation unit 100B receives Φdc and Φqc which are the outputs of the dc-qc axis flux calculation unit 110, and vdc, vqc, p(Φdc), and p(Φqc) which are the outputs of the flux control unit 900, and calculates the rotor position of the synchronous motor 200 as its estimated value θest based on these values.

[0104] In addition, the coordinate inverse conversion unit 700A, the coordinate conversion unit 700B, the power converter 300, and the synchronous motor 200 have the same configurations as those in the first embodiment.

[0105] FIG. 9 is a diagram showing the configuration of the flux control unit 900 of this embodiment. The flux control unit 900 includes a dc axis flux control unit 910 and a qc axis flux control unit 920. Since these have the same configuration, only the details of the dc axis flux control unit 910 are shown. The dc axis flux control unit 910 takes as inputs the dc axis flux command Φdc* and the dc axis flux Φdc, and outputs the dc axis voltage vdc and p(Φdc) which is the differential value of the dc axis flux. The dc axis flux control unit 910 includes a subtractor 911, a proportional controller 912, an integrator 913, an adder 914, and a differential value calculation means 915.

[0106] The subtractor 911 calculates the deviation between the dc axis flux command value Φdc* and the dc axis flux Φdc, and transmits it to the proportional controller 912, the integrator 913, and the differential value calculation means 915 respectively.

[0107] The proportional controller 912 amplifies the deviation between the dc axis flux command value Φdc* and the dc axis flux Φdc, and transmits it to the adder 914.

[0108] The integrator 913 time-integrates and amplifies the deviation between the dc axis flux command value Φdc* and the dc axis flux Φdc, and transmits it to the adder 914.

[0109] The adder 914 receives the output signals from the proportional controller 912 and the integrator 913 as input values, and calculates the dc axis voltage vdc by adding their respective outputs.

[0110] Incidentally, the response of the feedback value Φdc to the command value Φdc* is generally controlled to achieve a desired response by adjusting and designing the amplification gains of the proportional controller 912 and the integrator 913. For example, for the dc-axis flux command Φdc*, the dc-axis flux Φ is often designed with a first-order lag. At this time, the relationship between Φdc* and Φdc is expressed by the following equation (17).

[0111]

Equation

[0112] Here, s is the Laplace operator, and T is the response time constant of Φdc. By transforming equation (17), the differential value of Φdc can be calculated as follows.

[0113]

Equation

[0114] Since the Laplace operator s represents differentiation, the differential operation of Φdc can be performed by using equation (18). In this way, by obtaining the time derivative from the deviation between the command value and the feedback value, direct time differentiation is not performed, so that noise amplification and the like can be suppressed. In this manner, the differential value calculation means 915 is configured and output as the dc-axis flux differential value p(Φdc).

[0115] Note that although the dc-axis flux control unit 910 is shown as an example having a configuration that performs proportional-integral control, it may take various known forms such as adding a term for compensating the armature reaction of the synchronous motor 200.

[0116] FIG. 10 is a diagram showing the configuration of the position estimation unit 100B. In the present embodiment, it is different from the position estimation unit 100A in the first embodiment in that the dc-qc axis flux calculation unit 110 for controlling based on the flux is provided outside the position estimation unit, and there is no differential operation unit 150 because the flux differential value is calculated by the flux control unit.

[0117] For other points, they are the same as those in the first embodiment.

[0118] In the figure, the axis error is shown to be calculated based on the dc-axis voltage vdc, qc-axis voltage vqc, dc-axis magnetic flux Φdc, qc-axis magnetic flux Φqc, differential value p(Φdc) of the dc-axis magnetic flux, and differential value p(Φqc) of the qc-axis magnetic flux. However, various applications such as the configuration of FIG. 7 and Equation (16) are possible.

[0119] The main features of the above-described first and second embodiments can be summarized as follows.

[0120] The control device 1 of the synchronous motor 200 uses the d-axis as the rotor position of the synchronous motor 200, the axis advanced 90 degrees from the d-axis as the q-axis, the axis having a phase difference θe from the d-axis as the dc-axis, and the axis advanced 90 degrees from the dc-axis as the qc-axis, and controls the synchronous motor 200 using the phase of the dc-axis (estimated rotor position θest). The processor of the control device 1 of the synchronous motor 200 calculates the phase difference θe using the voltages of the dc-axis and qc-axis (voltage commands vdc, vqc), the angular velocity of the dc-axis (estimated electrical angular velocity ω^), the characteristic values of the synchronous motor 200 (resistance value R, static q-axis inductance Lq), and the magnetic fluxes of the dc-axis and qc-axis (Φdc, Φqc) without using an approximation of the true electrical angular velocity ω (angular velocity of the d-axis) (FIG. 2, position estimation unit 100A, Equations (8), (9)).

[0121] Since the approximation that the true electrical angular velocity ω (angular velocity of the d-axis) = estimated electrical angular velocity ω^ (angular velocity of the dc-axis) is not required in Equations (8) and (9), the accuracy of the phase difference θe (axis error) is ensured even when the synchronous motor undergoes rapid acceleration or deceleration.

[0122] Since the terms of the dynamic inductance are not present in Expressions (6) and (7), it is possible to easily calculate θe while taking into account the static and dynamic inductances. Since the voltage equations of Expressions (6) and (7) are expressed in terms of magnetic fluxes instead of currents, they are highly compatible with the control of the motor using magnetic flux command values (magnetic flux commands Φdc*, Φqc*). For example, the processor derives magnetic flux command values (magnetic flux commands Φdc*, Φqc*) from torque command values (torque command T*) or current command values or the like using a table (map) or a mathematical formula or the like, and performs feedback control.

[0123] Since the characteristic values (parameters) of the synchronous motor 200 used in Expressions (8) and (9) are only the resistance value R and the static q-axis inductance Lq, the capacity of the storage device (memory) for storing the characteristic values is reduced. As a result, the manufacturing cost of the control device 1 can be reduced.

[0124] The processor may further calculate the phase difference θe using the magnetic flux differential values (pΦdc, pΦqc) of the dc axis and the qc axis (FIG. 4, position estimation unit 100A, Expression (8)). Since the magnetic flux differential values of the dc axis and the qc axis are used in Expression (8), the accuracy of θe is ensured even when the magnetic flux changes while taking into account the dynamic inductance.

[0125] In the above-described first embodiment, the processor uses a table (map) that defines the correspondence between the currents of the dc axis and the qc axis (current commands idc* and iqc*) and the magnetic fluxes of the dc axis and the qc axis (Φdc, Φqc) to derive the magnetic fluxes of the dc axis and the qc axis (Φdc, Φqc) (FIG. 3, dc-qc axis magnetic flux calculation unit 110). Since the table shown in FIG. 3 has information on the slopes of the magnetic fluxes with respect to the currents (the slope of the magnetic flux in the vicinity of the current operating point and the slope of the magnetic flux between the current operating point and the origin), the magnetic fluxes of the dc axis and the qc axis (Φdc, Φqc) can be derived from the currents of the dc axis and the qc axis while taking into account the dynamic and static inductances.

[0126] On the one hand, the processor may calculate the magnetic fluxes (Φdc, Φqc) of the d-axis and q-axis based on the voltage equations (Equations (10) and (11)) of the voltages (voltage commands vdc, vqc) of the d-axis and q-axis represented by the magnetic fluxes (Φdc, Φqc) of the d-axis and q-axis and the characteristic value (resistance value R) of the synchronous motor 200 (Fig. 6, d-q axis magnetic flux calculation unit 110, Equations (12) and (13)). Since the calculations of Equations (12) and (13) are composed of only simple arithmetic operations, the calculation load on the processor can be suppressed.

[0127] In the above-described first embodiment, the differential calculation unit 150 calculates the magnetic flux differential values (pΦdc, pΦqc) by pure time differentiation or a filter. However, the processor may calculate the magnetic flux differential values (pΦdc, pΦqc) of the d-axis and q-axis using the differences between the magnetic flux command values (magnetic flux commands Φdc*, Φqc*) of the d-axis and q-axis, which are drive commands of the synchronous motor 200, and the magnetic fluxes (Φdc, Φqc) of the d-axis and q-axis (Fig. 9, magnetic flux control unit 900, Equation (18), etc.). Since direct time differentiation is not performed, amplification of noise and the like can be suppressed.

[0128] In the above-described first embodiment, the axis error calculation unit 120 calculates θe using Equation (8) or (9). However, the processor may calculate the phase difference θe using the product of the q-axis current (iqc) and the q-axis inductance (static q-axis inductance Lq), the product of the d-axis current (idc) and the q-axis inductance (static q-axis inductance Lq), and the magnetic fluxes (Φdc, Φqc) of the d-axis and q-axis (Fig. 7, axis error calculation unit 120, for example, the equation obtained by multiplying Lq to the denominator and numerator of the argument of the arctangent function in Equation (16)).

[0129] In other words, the processor calculates the phase difference θe using the currents (idc, iqc) on the dc axis and qc axis, the q-axis inductance (static q-axis inductance Lq), and the fluxes (Φdc, Φqc) on the dc axis and qc axis (FIG. 7, axis error calculation unit 120, Equation (16)). As a result, advantages such as being less affected by output voltage errors caused by dead time in the power converter can be obtained. Furthermore, the characteristics of the synchronous motor to be used are only the static q-axis inductance Lq, and the robustness against the parameters of the synchronous motor can be enhanced.

[0130] As shown in FIG. 8, the synchronous motor 200 may include a position sensor 200A (for example, a resolver) for detecting the rotor position of the synchronous motor 200. Then, the processor calculates an estimated value of the phase of the dc axis (estimated rotor position θest) from the phase difference θe (FIG. 10, position estimation unit 100B). During the period when the synchronous motor 200 is being driven, the processor compares the estimated value of the phase of the dc axis (estimated rotor position θest) with the rotor position θ detected by the position sensor, and determines whether there is an abnormality in the position sensor. For example, when the difference between the rotor position θ and the estimated rotor position θest is equal to or greater than a threshold value, the processor determines that an abnormality (failure) has occurred in the position sensor. Since the accuracy of the phase of the dc axis (estimated rotor position θest) calculated from the highly accurate θe is high, the determination accuracy of the abnormality (failure) of the position sensor can be improved.

[0131] Note that the present invention is not limited to the above-described embodiments, and includes various modifications. For example, the above-described embodiments have been described in detail for easy understanding of the present invention, and are not necessarily limited to those having all the configurations described. Also, a part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment. Further, for a part of the configuration of each embodiment, addition, deletion, or replacement with other configurations is possible.

[0132] In addition, each of the above-described configurations, functions, etc. may be realized in hardware by designing part or all of them, for example, by using an integrated circuit. Further, each of the above-described configurations, functions, etc. may be realized in software by a processor (microcomputer) interpreting and executing a program for realizing each function. Information such as a program, table, file, etc. for realizing each function can be stored in a memory, a recording device such as a hard disk or SSD (Solid State Drive), or a recording medium such as an IC card, SD card, or DVD.

[0133] Note that the embodiments of the present invention may also be the following aspects.

[0134] (C1). In a control device for a synchronous motor that uses the rotor position of the synchronous motor as the d-axis, the axis advanced 90 degrees from the d-axis as the q-axis, the axis having a phase difference with the d-axis of θe as the dc-axis, and the axis advanced 90 degrees from the dc-axis as the qc-axis, and controls the synchronous motor using the phase information of the dc-axis, θe is calculated using the voltages of the dc-axis and the qc-axis, the angular velocity of the dc-axis, the characteristic values of the synchronous motor, the fluxes of the dc-axis and the qc-axis, and the fluxes of the dc-axis and the qc-axis are calculated by a dc-qc axis flux calculation unit having a table based on the relationship between the dc-axis and qc-axis currents and the dc-axis and qc-axis fluxes. A control device for a synchronous motor characterized by this.

[0135] (C2). The control device for a synchronous motor is characterized in that θe is calculated using the voltages of the dc-axis and the qc-axis, the angular velocity of the dc-axis, the characteristic values of the synchronous motor, the fluxes of the dc-axis and the qc-axis, and the differential values of the fluxes of the dc-axis and the qc-axis.

[0136] (C3). The dc-qc axis flux calculation unit calculates the fluxes of the dc-axis and the qc-axis based on a voltage equation in which the voltages of the dc-axis and the qc-axis are represented by the fluxes of the dc-axis and the qc-axis and the characteristic values of the synchronous motor. A control device for a synchronous motor characterized by this.

[0137] (C4). The magnetic flux differential values of the dc axis and the qc axis are calculated using the differences between the magnetic flux command values of the dc axis and the qc axis, which are drive commands of the synchronous motor, and the magnetic fluxes of the dc axis and the qc axis. A control device for a synchronous motor characterized by this.

[0138] (C5). The θe is calculated using the product of the qc axis current and the q-axis inductance of the synchronous motor, the product of the dc axis current and the q-axis inductance, and the magnetic fluxes of the dc axis and the qc axis. A control device for a synchronous motor characterized by this.

[0139] (C6). In the control device for a synchronous motor according to any one of (C1) to (C5), the synchronous motor is provided with a position sensor for detecting the rotor position of the synchronous motor. When the synchronous motor is in operation, the phase information of the dc axis obtained from the θe is compared with the rotor position obtained by the position sensor to determine the presence or absence of an abnormality in the position sensor. A control device for a synchronous motor characterized by this.

[0140] (C1)-(C6) According to this, the dc-qc axis magnetic flux is calculated by a dc-qc axis magnetic flux calculation unit having the relationship between the dc axis and qc axis currents and the dc axis and qc axis magnetic fluxes. This dc-qc axis magnetic flux has a change in inductance due to magnetic saturation, and furthermore, by expressing it in terms of magnetic flux, the static and dynamic inductances can also be considered, and the position can be estimated with high accuracy in the case of sensorless. Also, the approximation that the speed of the d axis, which is the true speed, is equal to the speed of the dc axis, which is the speed estimation value, is not required, and the position can be estimated with high accuracy even when the rotational speed of the synchronous motor changes suddenly.

Explanation of symbols

[0141] 1, 2... Control device 100A, 100B... Position estimation unit 110... Dc-qc axis magnetic flux calculation unit 111... Dc axis magnetic flux calculation unit 112... Qc axis magnetic flux calculation unit 120... Axis error calculation unit 130... Speed calculation unit 131... Subtractor 132... Scaler 133... Integrator 134... Adder 140... Position calculation unit 150... Differential calculation unit 200... Synchronous motor 200A... Position sensor 300... Power converter 400... Current command calculation unit 500... Current detection unit 600... Current control unit 700A... Inverse coordinate conversion unit 700B... Coordinate conversion unit 800... Flux command calculation unit 900... Flux control unit

Claims

1. A control device for a synchronous motor that uses the position of the rotor of the synchronous motor as the d-axis, the axis advanced 90 degrees from the d-axis as the q-axis, the axis having a phase difference θe from the d-axis as the dc-axis, the axis advanced 90 degrees from the dc-axis as the qc-axis, and controls the synchronous motor using the phase of the dc-axis, comprising: A control device for a synchronous motor having a processor that calculates the phase difference θe using the voltages of the dc-axis and the qc-axis, the angular velocity of the dc-axis, the characteristic values of the synchronous motor, and the fluxes of the dc-axis and the qc-axis without using an approximate value of the angular velocity of the d-axis.

2. The control device for a synchronous motor according to claim 1, wherein: The processor: Further calculates the phase difference θe using the differential values of the fluxes of the dc-axis and the qc-axis. A control device for a synchronous motor, characterized by this.

3. The control device for a synchronous motor according to claim 1, wherein: The processor: Calculates the fluxes of the dc-axis and the qc-axis based on the voltage equations of the voltages of the dc-axis and the qc-axis represented by the fluxes of the dc-axis and the qc-axis and the characteristic values of the synchronous motor. A control device for a synchronous motor, characterized by this.

4. The control device for a synchronous motor according to claim 1, wherein: The processor: Calculates the differential values of the fluxes of the dc-axis and the qc-axis using the differences between the flux command values of the dc-axis and the qc-axis, which are the drive commands of the synchronous motor, and the fluxes of the dc-axis and the qc-axis. A control device for a synchronous motor, characterized by this.

5. The control device for a synchronous motor according to claim 1, wherein: The processor: Calculates the phase difference θe using the product of the qc-axis current and the q-axis inductance, the product of the dc-axis current and the q-axis inductance, and the fluxes of the dc-axis and the qc-axis. A control device for a synchronous motor, characterized by this.

6. The control device for a synchronous motor according to claim 1, wherein: The synchronous motor is provided with a position sensor for detecting the position of the rotor of the synchronous motor. The processor: Calculates an estimated value of the phase of the dc-axis from the phase difference θe. During the period when the synchronous motor is being driven, compares the estimated value of the phase of the dc-axis with the rotor position detected by the position sensor, and determines the presence or absence of an abnormality in the position sensor. A control device for a synchronous motor, characterized by this.

7. The control device for a synchronous motor according to claim 1, wherein: The processor: Deriving the magnetic fluxes of the dc axis and the qc axis by using a table that defines the correspondence between the currents of the dc axis and the qc axis and the magnetic fluxes of the dc axis and the qc axis A control device for a synchronous motor, characterized by the above.

8. A control device for a synchronous motor according to claim 1, wherein the processor calculates a phase difference θe by using the currents of the dc axis and the qc axis, a q-axis inductance, and the magnetic fluxes of the dc axis and the qc axis A control device for a synchronous motor, characterized by the above.

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