Current resonance type DC / DC converter

Abstract

To provide a current resonance type DC / DC converter which can reduce a loss in a step-op operation.SOLUTION: A current resonance type DC / DC converter 1A comprises: a main circuit unit 10A comprising a transformer Tr, a primary side circuit 11, a primary side resonance circuit 12, and a secondary side circuit 13A; and a control unit 20A. The control unit 20A sets an ON period of a switching element Q8 (Q6) on the secondary side circuit 13A different from a switching element Q6 (Q8) for generating short circuit current to the secondary side circuit 13A by being turned on to a period of logical OR between an ON period of a switching element Q1 (Q2) and an ON period obtained by performing phase shift of the ON period by a first predetermined phase angle in step-up control.SELECTED DRAWING: Figure 1

Description

Technical Field

[0001] The present invention relates to a current resonance type DC / DC converter.

Background Art

[0002] In recent years, in line with the popularization of electric vehicles, miniaturization, cost reduction, and high efficiency of charging devices for charging electric vehicle batteries and V2H (Vehicle to Home) devices for supplying the battery voltage of electric vehicles to loads have been demanded. Therefore, for the DC / DC converters used as the power supplies of these devices, current resonance type DC / DC converters (LLC type DC / DC converters and CLLC type DC / DC converters capable of bidirectional operation) with a small number of components and high efficiency are used.

[0003] On the other hand, the CHAdeMO standard, which is a charging standard for electric vehicles, requires a wide battery voltage range from 150 [V] to 450 [V]. For this reason, in charging devices and V2H devices for electric vehicles, it is necessary not only to step down but also to step up the DC voltage output by the current resonance type DC / DC converter according to the battery voltage of the electric vehicle. However, the LLC type or CLLC type current resonance type DC / DC converter has a problem that the output voltage range is narrow.

[0004] Fig. 15(A) shows the main circuit part of the LLC type DC / DC converter described in Non-Patent Document 1. As shown in the figure, the primary side circuit of the main circuit part has a full-bridge configuration with switching elements Q1 to Q4 in each arm. The secondary side circuit of the main circuit part has a circuit configuration in which the upper arm is diodes D5 and D7 and the lower arm is switching elements Q6 and Q8.

[0005] The control unit (not shown) of the LLC-type DC / DC converter described in Non-Patent Document 1, as shown in Fig. 15(B), phase-shifts the drive pulses of the switching elements Q2 and Q3 by a phase angle θ to obtain the drive pulse of the switching element Q6, and phase-shifts the drive pulses of the switching elements Q1 and Q4 by a phase angle θ to obtain the drive pulse of the switching element Q8, causing the main circuit section to perform a boosting (step-up) operation and a synchronous rectification operation. In this control method, the periods of modes 3 and 4 are the periods of the boosting operation, and the periods of modes 1 and 2 are the periods of the synchronous rectification operation. Therefore, it is possible to expect an expansion of the voltage range due to the boosting operation and an improvement in the power conversion efficiency due to the synchronous rectification operation.

[0006] Fig. 16(A) shows the current path of the main circuit section during the period of mode 3, and Fig. 16(B) shows the current path of the main circuit section during the period of mode 4. During the period of mode 3, since the switching element Q6 is on and the switching element Q8 is off, the short-circuit current during the boosting operation flows through the diode D8 connected in parallel to the current path of the switching element Q8. Therefore, there is a problem that the loss becomes larger compared to the case where the short-circuit current flows through the switching element Q8. During the period of mode 4, since the switching element Q8 is on and the switching element Q6 is off, the short-circuit current during the boosting operation flows through the diode D6 connected in parallel to the current path of the switching element Q6. Therefore, there is a problem that the loss becomes larger compared to the case where the short-circuit current flows through the switching element Q6.

[0007] Patent Document 1 describes a CLLC-type DC / DC converter capable of bidirectional operation. In the CLLC-type DC / DC converter described in Patent Document 1, when the output voltage is below a certain value with respect to the input voltage, the control unit fixes the switching frequency and performs phase-shift control between the primary-side circuit and the secondary-side circuit to cause a boosting operation. This control method also has the same problem as the control method described in Non-Patent Document 1.

Prior Art Documents

Patent Documents

[0008]

Patent Document 1

Non-Patent Document

[0009]

Non-Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0010] The present invention has been made in view of the above circumstances, and an object thereof is to provide a current resonant type DC / DC converter capable of reducing losses during boost operation.

Means for Solving the Problems

[0011] In order to solve the above problems, a current resonant type DC / DC converter according to an embodiment of the present invention includes a main circuit section and a control section, wherein the main circuit section includes a transformer including a primary winding and a secondary winding, a primary-side circuit in which a first leg and a second leg are connected in parallel, and each leg includes an upper arm and a lower arm, including a first resonant inductance and a first resonant capacitance, a primary-side resonant circuit in which the first resonant inductance and the first resonant capacitance are connected in series to the primary winding between a connection point of the upper arm and the lower arm of the first leg and a connection point of the upper arm and the lower arm of the second leg; a secondary-side circuit in which a third leg and a fourth leg are connected in parallel, each leg including an upper arm and a lower arm, and the secondary winding is connected between a connection point of the upper arm and the lower arm of the third leg and a connection point of the upper arm and the lower arm of the fourth leg; A current resonant type DC / DC converter comprising: the upper arm and the lower arm of each leg of the primary-side circuit include at least a switching element; one of the upper arm or the lower arm of each leg of the secondary-side circuit includes at least a switching element, and the other includes at least a diode; the control unit performs first boost control to cause the main circuit unit to perform a boost operation with the secondary-side circuit as an output side; the switching element of the primary-side circuit includes a first switching element that generates a resonant current in the primary-side circuit when turned on; the switching element of the secondary-side circuit includes a second switching element that generates a short-circuit current in the secondary-side circuit when turned on, and a third switching element provided in a current path of the short-circuit current; the control unit during the first boost control; performs first on-period control such that an on-period of the third switching element is a period of a logical sum of a first on-period which is an on-period of the first switching element and an on-period obtained by phase-shifting the first on-period by a predetermined first phase angle.

[0012] In this configuration, during the period when a large short-circuit current flows through the secondary circuit (the period of the boosting operation), both the second switching element and the third switching element are turned on with relatively simple control to form a short-circuit circuit. As a result, a short-circuit current flows through the second switching element and the third switching element, so that in this configuration, the loss can be reduced as compared with the case where the short-circuit current flows through the diode.

[0013] The current resonant type DC / DC converter includes a second resonant inductance and a second resonant capacitance, and a secondary-side resonant circuit in which the second resonant inductance and the second resonant capacitance are connected in series to the secondary winding between the connection point of the upper arm and the lower arm of the third leg and the connection point of the upper arm and the lower arm of the fourth leg is provided. The upper arm and the lower arm of each leg of the secondary-side circuit include at least a switching element. The control unit performs a second boosting control for causing the main circuit unit to perform a boosting operation with the primary-side circuit as an output side. The switching element of the secondary-side circuit includes a fourth switching element that generates a resonant current in the secondary-side circuit when turned on. The switching element of the primary-side circuit includes a fifth switching element that generates a short-circuit current in the primary-side circuit when turned on, and a sixth switching element provided in the current path of the short-circuit current. During the second boosting control, the control unit can be configured to perform a second on-period control in which the on-period of the sixth switching element is set to the logical sum period of the third on-period, which is the on-period of the fourth switching element, and the on-period obtained by phase-shifting the third on-period by a predetermined second phase angle.

[0014] In the current resonant type DC / DC converter, During the first boosting control, the control unit The secondary-side circuit is caused to perform a synchronous rectification operation, and when the current due to the synchronous rectification operation is equal to or less than a preset first set value, the third switching element can be turned off.

[0015] In the current resonant DC / DC converter, The control unit during the second boosting control The primary-side circuit is caused to perform a synchronous rectification operation, and when the current due to the synchronous rectification operation is equal to or less than a preset second set value, the sixth switching element can be turned off.

[0016] A current resonant DC / DC converter according to another embodiment of the present invention includes a main circuit section and a control section, The main circuit section includes a transformer including a primary winding and a secondary winding, a primary-side circuit in which a first leg and a second leg are connected in parallel, and each leg includes an upper arm and a lower arm, a primary-side resonant circuit including a first resonant inductance and a first resonant capacitance, and the first resonant inductance and the first resonant capacitance are connected in series to the primary winding between the connection points of the upper arm and the lower arm of the first leg and the connection points of the upper arm and the lower arm of the second leg, a secondary-side circuit in which a third leg and a fourth leg are connected in parallel, and each leg includes an upper arm and a lower arm, and the secondary winding is connected between the connection points of the upper arm and the lower arm of the third leg and the connection points of the upper arm and the lower arm of the fourth leg, A current resonant DC / DC converter comprising The upper arms and the lower arms of each leg of the primary-side circuit include at least switching elements, One of the upper arm or the lower arm of each leg of the secondary-side circuit includes at least a switching element, and the other includes at least a diode, The control unit performs first boosting control to cause the main circuit section to perform a boosting operation with the secondary-side circuit as an output side, The switching element of the primary-side circuit includes a first switching element A that generates a first resonance current in the primary-side circuit when it is turned on, and a first switching element B that generates a second resonance current in the primary-side circuit in a direction opposite to the first resonance current when it is turned on. The switching element of the secondary-side circuit includes a second switching element that generates a short-circuit current in the secondary-side circuit when it is turned on, and a third switching element provided in the current path of the short-circuit current. During the first boost control, the control unit calculates a logical product A between the on-period of the first switching element A and a period obtained by phase-shifting and inverting the on-period by a predetermined first phase angle, calculates a logical product B between the on-period of the first switching element B and a period obtained by phase-shifting and inverting the on-period by the first phase angle, and performs inverted on-period control in which the on-period of the third switching element is set to the period of the logical sum of the logical product A and the logical product B.

Advantages of the Invention

[0017] According to the present invention, a current resonance type DC / DC converter capable of reducing losses during boost operation can be provided.

Brief Description of the Drawings

[0018]

Figure 1

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Figure 10

Figure 11

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Figure 15

Figure 16

Embodiments for Carrying Out the Invention

[0019] Hereinafter, an embodiment of a current resonance type DC / DC converter according to the present invention will be described with reference to the accompanying drawings.

[0020] FIG. 1 shows a current resonance type DC / DC converter 1A according to an embodiment of the present invention. The current resonance type DC / DC converter 1A is an LLC type DC / DC converter, and includes a main circuit section 10A, a control section 20A, and terminals T1 to T4.

[0021] Terminals T1 and T2 are input terminals, and a DC input voltage V1 is input between terminals T1 and T2. Terminals T1 and T2 are connected to, for example, a PFC (power factor correction) circuit. Terminals T3 and T4 are output terminals, and a DC output voltage V2 is output between terminals T3 and T4. Terminals T3 and T4 are connected to, for example, a battery of an electric vehicle.

[0022] The main circuit section 10A includes a primary side drive circuit 11, a primary side resonance circuit 12, a transformer Tr, a secondary side rectification circuit 13A, and a capacitor C21. The primary side drive circuit 11 corresponds to the "primary side circuit" of the present invention. The secondary side rectification circuit 13A corresponds to the "secondary side circuit" of the present invention.

[0023] The primary side drive circuit 11 includes a first leg and a second leg connected in parallel, and each leg includes an upper arm and a lower arm connected in series, having a full-bridge circuit configuration. The upper arm of the first leg includes a switching element Q1, the lower arm of the first leg includes a switching element Q2, the upper arm of the second leg includes a switching element Q3, and the lower arm of the second leg includes a switching element Q4. Diodes D1 to D4 are connected in parallel in the reverse direction to the current paths of the switching elements Q1 to Q4, and capacitors C1 to C4 are connected in parallel.

[0024] The switching elements Q1 to Q4 can use, for example, power semiconductor switching elements capable of switching at high frequencies such as IGBTs (Insulated Gate Bipolar Transistors) and MOSFETs (Metal Oxide Semiconductor Field Effect Transistors). The same applies to the switching elements Q6, Q8, and the switching elements of each modification described later.

[0025] The collector (drain) terminals of the switching elements Q1 and Q3 are connected to the terminal T1, and the emitter (source) terminals of the switching elements Q2 and Q4 are connected to the terminal T2, and the input voltage V1 is applied to the primary side drive circuit 11. The connection point between the emitter (source) terminal of the switching element Q1 and the collector (drain) terminal of the switching element Q2, and the connection point between the emitter (source) terminal of the switching element Q3 and the collector (drain) terminal of the switching element Q4 are connected to the primary side resonance circuit 12.

[0026] The diodes D1 to D4 are freewheeling diodes and may be built-in diodes of the switching elements Q1 to Q4, external diodes, or both. The capacitors C1 to C4 are partial resonance capacitors and may be the parasitic capacitances of the switching elements Q1 to Q4, external capacitors, or both.

[0027] The primary side resonance circuit 12 includes a resonance inductance Lr1 and a resonance capacitance Cr1 connected to the terminals of the primary winding of the transformer Tr. The resonance inductance Lr1 and the resonance capacitance Cr1 form a series resonance circuit together with the excitation inductance of the transformer Tr. Therefore, the primary side drive circuit 11 becomes the drive circuit of the primary side resonance circuit 12.

[0028] In addition, if the resonant inductance Lr1 and the resonant capacitance Cr1 are connected to the primary winding of the transformer Tr to form a series resonant circuit, their arrangement is arbitrary. The resonant inductance Lr1 may be the leakage inductance of the transformer Tr, may be a coil having an individual core, or may be both. The resonant capacitance Cr1 may be composed of individual capacitors, may be the parasitic capacitance of the switching elements Q1 to Q4, or may be both. The exciting inductance of the transformer Tr is not shown in the figure assuming it is included in the primary winding of the transformer Tr, but it may be a coil having an individual core.

[0029] The transformer Tr is composed of one or more high-frequency isolation transformers. The primary winding of the transformer Tr is connected to the primary-side drive circuit 11 via the primary-side resonant circuit 12. The secondary winding of the transformer Tr is connected to the secondary-side rectifying circuit 13A.

[0030] The secondary-side rectifying circuit 13A includes a third leg and a fourth leg connected in parallel, and each leg includes an upper arm and a lower arm connected in series. The upper arm of the third leg includes a diode D5 (not including a switching element), the lower arm of the third leg includes a switching element Q6, the upper arm of the fourth leg includes a diode D7 (not including a switching element), and the lower arm of the fourth leg includes a switching element Q8. Diodes D6 and D8 are connected in parallel in the reverse direction in the current paths of the switching elements Q6 and Q8, and capacitors C6 and C8 are connected in parallel.

[0031] The cathodes of the diodes D5 and D7 are connected to the terminal T3, and the emitter (source) terminals of the switching elements Q6 and Q8 are connected to the terminal T4, so that the secondary-side rectifying circuit 13A outputs an output voltage V2 between the terminals T3 and T4. The connection point between the anode of the diode D5 and the collector (drain) terminal of the switching element Q6 and the connection point between the anode of the diode D7 and the collector (drain) terminal of the switching element Q8 are respectively connected to the terminals on both sides of the secondary winding of the transformer Tr.

[0032] Diodes D5 to D8 are diodes for rectification or reverse current prevention. Diodes D5 and D7 are external diodes. Diodes D6 and D8 may be built-in diodes of switching elements Q6 and Q8, external diodes, or both. Capacitors C6 and C8 may be parasitic capacitances of switching elements Q6 and Q8, external capacitors, or both.

[0033] Capacitor C21 is a capacitor for reducing output ripple and is connected between terminals T3 and T4.

[0034] Control unit 20A includes a processing unit that generates control signals (control signals Q1t to Q4t, Q6t, and Q8t described later) for performing on / off control of switching elements Q1 to Q4, Q6, and Q8, and a driving unit (not shown) that turns on / off switching elements Q1 to Q4, Q6, and Q8 based on the control signals. Control unit 20A may be composed of a digital circuit such as a microprocessor or a digital signal processor, an analog circuit, or a circuit combining a digital circuit and an analog circuit. Control unit 20A may further include a detection unit (not shown). The detection unit includes, for example, a detection circuit such as a current sensor and / or a voltage sensor that detects current values and / or voltage values necessary for the control of control unit 20A, and its peripheral circuits.

[0035] Control unit 20A monitors the input to the terminal T1, T2 side (primary side) (input current, input voltage V1, or input power) and / or the output from the terminal T3, T4 side (secondary side) (output current, output voltage V2, or output power), and performs on / off control of switching elements Q1 to Q4, Q6, and Q8 so that the output becomes a desired value.

[0036] Specifically, the control unit 20A performs frequency modulation control to modulate the driving frequency of the primary side drive circuit 11 (switching elements Q1 to Q4), and performs synchronous rectification control to synchronize the on-timing of the switching elements Q6 and Q8 of the secondary side rectification circuit 13A with the on-timing of the switching elements Q1 to Q4 of the primary side drive circuit 11. For example, when increasing the output, the control unit 20A decreases the driving frequency of the switching elements Q1 to Q4, and when decreasing the output, the control unit 20A increases the driving frequency of the switching elements Q1 to Q4.

[0037] Also, although details will be described later, the control unit 20A performs boost control (corresponding to the "first boost control" of the present invention) to cause the main circuit unit 10A to perform a boost operation. For example, as the output increases, the driving frequency of the switching elements Q1 to Q4 decreases, and when the driving frequency reaches a predetermined set frequency, the control unit 20A may start boost control. During boost control, in addition to the above frequency modulation control and synchronous rectification control, the control unit 20A performs phase shift control and on-period control (corresponding to the "first on-period control" of the present invention).

[0038] FIG. 2 shows an example of the control timing of the switching elements Q1 to Q4, Q6, and Q8. In FIG. 2, the control unit 20A drives the primary side drive circuit 11 (switching elements Q1 to Q4) by frequency modulation control, and drives the secondary side rectification circuit 13A (switching elements Q6 and Q8) by synchronous rectification control. The switching elements Q1 to Q4, Q6, and Q8 turn on when the control signals of the switching elements Q1 to Q4, Q6, and Q8 shown in FIG. 2 are at a high level (H), and turn off when at a low level (L) (the same applies to the following control timing diagrams).

[0039] The control unit 20A drives the switching elements Q6 and Q8 based on the control formula of the following formula (1).

Equation

[0040] The main circuit section 10A in this state operates with an input / output gain that depends on the drive frequency in the input / output voltage ratio. For example, when the input voltage V1 is constant, if the drive frequency of the switching elements Q1 to Q4 is decreased, the output voltage V2 increases, and if the drive frequency of the switching elements Q1 to Q4 is increased, the output voltage V2 decreases.

[0041] In FIG. 2, the control unit 20A turns on and off the switching elements Q6, Q2, and Q3 at the same timing and turns on and off the switching elements Q8, Q1, and Q4 at the same timing for the switching elements Q1 to Q4, Q6, and Q8 with the same drive frequency and the same duty ratio (e.g., 50%). The switching element Q1 and the switching element Q2 have a phase difference of 180° except for the dead time, and the switching element Q3 and the switching element Q4 also have a phase difference of 180° except for the dead time.

[0042] For simplicity of explanation, the description of the dead time is omitted in FIG. 2 (the same applies to the following control timing diagrams). That is, in actual control, there are a time for soft switching and a time for preventing through-current (e.g., the through-current prevention time of the switching elements Q1 and Q2) when turning on from off between the upper and lower arms of the same leg (e.g., when turning on the switching element Q1 from the turn-off of the switching element Q2, or when turning on the switching element Q2 from the turn-off of the switching element Q1). The duty ratio of the switching elements Q1 to Q4, Q6, and Q8 is decreased by the dead time amount from, for example, 50%, or the off-duty ratio of the switching elements Q1 to Q4, Q6, and Q8 is increased by the dead time amount from, for example, 50%.

[0043] Fig. 3 shows the current paths of each mode in the control timing diagram of Fig. 2. In Fig. 3, (A) is the current path diagram during Mode 1 in Fig. 2, and (B) is the current path diagram during Mode 2 in Fig. 2.

[0044] As shown in Fig. 3(A), in Mode 1, since the switching elements Q1 and Q4 are on, a resonant current flows through the primary side drive circuit 11 and the primary side resonant circuit 12, and a load current flows through the secondary side rectifier circuit 13A via the secondary winding of the transformer Tr. In the secondary side rectifier circuit 13A, since the lower arm (switching element Q8) of the fourth leg is on, the load current flows through the switching element Q8 and the upper arm (diode D5) of the third leg.

[0045] As shown in Fig. 3(B), in Mode 2, since the switching elements Q2 and Q3 are on, a resonant current in the direction opposite to that in Mode 1 flows through the primary side drive circuit 11 and the primary side resonant circuit 12, and a load current flows through the secondary side rectifier circuit 13A via the secondary winding of the transformer Tr. In the secondary side rectifier circuit 13A, since the lower arm (switching element Q6) of the third leg is on, the load current flows through the switching element Q6 and the upper arm (diode D7) of the fourth leg.

[0046] Fig. 4 shows an example of the control timing of the switching elements Q1 to Q4, Q6, and Q8 during boost control. The control unit 20A drives the primary side drive circuit 11 (switching elements Q1 to Q4) by frequency modulation control and drives the secondary side rectifier circuit 13A (switching elements Q6 and Q8) by phase shift control, synchronous rectification control, and on-period control. In Fig. 4, the periods of Mode 3a and Mode 4a are the periods of the boost operation, and the periods of Mode 1 and Mode 2 are the periods of the synchronous rectification operation.

[0047] The control unit 20A phase-shifts the on / off timings of the switching elements Q6 and Q8 by a predetermined phase angle θ (0° ≤ θ ≤ 180° when one cycle is 360°) with respect to the on / off timings of the switching elements Q1 to Q4 by means of phase shift control and synchronous rectification control. When the control unit 20A increases the output, it increases the phase angle θ, and when it decreases the output, it decreases the phase angle θ. The phase angle θ corresponds to the "first phase angle" of the present invention.

[0048] The control timing represented by the dashed line in FIG. 4 is the control timing of the conventional example shown in FIG. 15(B). The difference between the example of FIG. 4 and the conventional example is that in the example of FIG. 4, during the period of mode 3a, the on-timing of the switching element Q8 is the same as the on-timings of the switching elements Q1 and Q4, and during the period of mode 4a, the on-timing of the switching element Q6 is the same as the on-timings of the switching elements Q2 and Q3.

[0049] Furthermore, in the example of FIG. 4, the on-period of the switching element Q6 is the logical sum (OR) of the on-period of the switching element Q2 (corresponding to the "first on-period" of the present invention) and the period obtained by phase-shifting the on-period by the phase angle θ (corresponding to the "second on-period" of the present invention). Similarly, the on-period of the switching element Q8 is the logical sum (OR) of the on-period of the switching element Q1 (corresponding to the "first on-period" of the present invention) and the period obtained by phase-shifting the on-period by the above phase angle θ (corresponding to the "second on-period" of the present invention). That is, the control unit 20A sets the on-period of the switching element Q8 (Q6), which is different from the switching element Q6 (Q8) that causes a short-circuit current to flow by turning on in the secondary-side rectifier circuit 13A, as the logical sum (OR) of the on-period of the switching element Q1 (Q2) that causes a resonance current to flow by turning on in the primary-side drive circuit 11 and the period obtained by phase-shifting the on-period.

[0050] The control unit 20A drives the switching elements Q6 and Q8 based on the control formula of the following formula (2).

Equation

[0051] In Equation (2), Q1S is a drive pulse (control signal) obtained by phase-shifting the on / off timing of Q1t by a phase angle θ, and Q2S is a drive pulse (control signal) obtained by phase-shifting the on / off timing of Q2t by a phase angle θ. The control unit 20A calculates the logical sum of the control signal Q2t and the control signal Q2S to generate the control signal Q6t, and calculates the logical sum of the control signal Q1t and the control signal Q1S to generate the control signal Q8t. Note that Equation (1) corresponds to the case where the phase angle θ = 0° in Equation (2).

[0052] FIG. 5 shows the current paths of each mode in the control timing diagram of FIG. 4. In FIG. 5, (A) is a current path diagram during the period of mode 3a in FIG. 4, and (B) is a current path diagram during the period of mode 4a in FIG. 4.

[0053] As shown in FIG. 5(A), in mode 3a, since the switching element Q6 is on, a short-circuit circuit is formed in the secondary rectifier circuit 13A, and a large short-circuit current flows through the short-circuit circuit. Here, if the switching element Q8 is off, the short-circuit current flows through the diode D8, but in mode 3a, since the switching element Q8 is on, the short-circuit current flows through the switching element Q8. As a result, the loss can be reduced compared to the case where the short-circuit current flows through the diode D8. In mode 3a, the switching elements Q1 and Q4 correspond to the "first switching element" of the present invention, the switching element Q6 corresponds to the "second switching element" of the present invention, and the switching element Q8 corresponds to the "third switching element" of the present invention.

[0054] As shown in FIG. 5(B), in mode 4a, since the switching element Q8 is on, a short - circuit circuit is formed in the secondary - side rectifier circuit 13A, and a large - current short - circuit current flows through the short - circuit circuit. Here, if the switching element Q6 is off, the short - circuit current flows through the diode D6, but in mode 4a, since the switching element Q6 is on, the short - circuit current flows through the switching element Q6. As a result, the loss can be reduced compared with the case where the short - circuit current flows through the diode D6. In mode 4a, the switching elements Q2 and Q3 correspond to the "first switching element" of the present invention, the switching element Q8 corresponds to the "second switching element" of the present invention, and the switching element Q6 corresponds to the "third switching element" of the present invention.

[0055] As described above, during the boost control, the control unit 20A drives the secondary - side rectifier circuit 13A (switching elements Q6 and Q8) by phase - shift control, synchronous rectification control, and on - period control. In the on - period control, the control unit 20A sets the on - period of the switching element Q8 (Q6), which is different from the switching element Q6 (Q8) that allows a short - circuit current to flow through the secondary - side rectifier circuit 13A by turning on, to be the logical sum period of the on - period of the switching elements Q1 and Q4 (Q2 and Q3) that allow a resonance current to flow through the primary - side drive circuit 11 by turning on and the period obtained by phase - shifting the on - period by a phase angle θ. Thereby, the short - circuit current of the secondary - side rectifier circuit 13A can flow through the switching element Q8 (Q6) without passing through the diode D8 (D6). Therefore, in the current - resonance - type DC / DC converter 1A, the loss can be reduced and the power - conversion efficiency can be improved without additional components.

[0056] Furthermore, in the current - resonance - type DC / DC converter 1A, the control unit 20A calculates the logical sum of the control signal Q2t and the control signal Q2S to generate the control signal Q6t, and calculates the logical sum of the control signal Q1t and the control signal Q1S to generate the control signal Q8t. That is, in the current - resonance - type DC / DC converter 1A, by the relatively simple control process (the process using the control formula of Equation (2)) as described above, the on - period control can be executed to obtain the above - mentioned effects.

[0057] As described above, embodiments of the current resonance type DC / DC converter according to the present invention have been explained, but the present invention is not limited to the above embodiments.

[0058] [First Modified Example] Fig. 6 shows a current resonance type DC / DC converter 1B according to a first modified example of the present invention. The current resonance type DC / DC converter 1B is an LLC type DC / DC converter, and includes a main circuit section 10B, a control section 20B, and terminals T1 to T4.

[0059] The main circuit section 10B has the same configuration as the above embodiment, except that it includes a secondary rectifier circuit 13B instead of the secondary rectifier circuit 13A. The secondary rectifier circuit 13B includes a switching element Q5 in the upper arm of the third leg, a diode D6 in the lower arm of the third leg (not including the switching element Q6), a switching element Q7 in the upper arm of the fourth leg, and a diode D8 in the lower arm of the fourth leg (not including the switching element Q8). Diodes D5 and D7 are connected in parallel in the reverse direction in the current paths of the switching elements Q5 and Q7, and capacitors C5 and C7 are connected in parallel.

[0060] The control section 20B has the same configuration as the above embodiment, except that it controls the switching elements Q7 and Q5 instead of the switching elements Q6 and Q8. That is, the control timing of the control section 20B is obtained by replacing the control signals Q6t and Q8t in Figs. 2 and 4 with control signals Q7t and Q5t having the same waveform.

[0061] During step-up control, the control unit 20B drives the secondary-side rectifier circuit 13B (switching elements Q7, Q5) by means of phase-shift control, synchronous rectification control, and on-period control. In on-period control, the control unit 20B sets the on-period of the switching element Q5 (Q7), which is different from the switching element Q7 (Q5) that allows a short-circuit current to flow through the secondary-side rectifier circuit 13B when it is turned on, to be the logical sum of the on-period of the switching elements Q1, Q4 (Q2, Q3) that allow a resonant current to flow through the primary-side drive circuit 11 when they are turned on and a period obtained by phase-shifting the on-period by a phase angle θ. As a result, the short-circuit current of the secondary-side rectifier circuit 13B can flow through the switching element Q5 (Q7) without passing through the diode D5 (D7). Therefore, in the current-resonant DC / DC converter 1B, losses can be reduced and power conversion efficiency can be increased without additional components.

[0062] Furthermore, in the current-resonant DC / DC converter 1B, the control unit 20B calculates the logical sum of the control signal Q2t and the control signal Q2S to generate the control signal Q7t, and calculates the logical sum of the control signal Q1t and the control signal Q1S to generate the control signal Q5t. That is, in the current-resonant DC / DC converter 1B, on-period control can be executed by the relatively simple control process as described above to obtain the above effects.

[0063] [Second Modified Example] FIG. 7 shows a current-resonant DC / DC converter 1C according to the second modified example of the present invention. The current-resonant DC / DC converter 1C is an LLC-type DC / DC converter and includes a main circuit unit 10C, a control unit 20C, and terminals T1 to T4.

[0064] The main circuit section 10C has the same configuration as that of the above embodiment, except that it includes a secondary rectifier circuit 13C instead of the secondary rectifier circuit 13A. The secondary rectifier circuit 13C includes a switching element Q5 in the upper arm of the third leg, a switching element Q6 in the lower arm of the third leg, a switching element Q7 in the upper arm of the fourth leg, and a switching element Q8 in the lower arm of the fourth leg. Diodes D5 to D8 are connected in parallel in the reverse direction to the current paths of the switching elements Q5 to Q8, and capacitors C5 to C8 are connected in parallel.

[0065] The control section 20C has the same configuration as that of the above embodiment, except that it performs synchronous rectification control and phase shift control on the switching elements Q5 to Q8.

[0066] Fig. 8 shows an example of the control timing of the switching elements Q1 to Q8 during boost control. In Fig. 8, the control section 20C drives the primary side drive circuit 11 (switching elements Q1 to Q4) by frequency modulation control, and drives the secondary rectifier circuit 13C (switching elements Q5 to Q8) by phase shift control, synchronous rectification control, and on-period control. However, the on-period control is performed on the switching elements Q6 and Q8 of the lower arm. Note that, as in the first modification, the same effect can be obtained even if the on-period control is performed on the switching elements Q5 and Q7 of the upper arm.

[0067] During the period of mode 3c, the switching elements Q6 and Q8 are turned on, so the secondary rectifier circuit 13C is short-circuited and the short-circuit current flows through the switching elements Q6 and Q8. During the subsequent period of mode 1, the switching elements Q5 and Q8 are turned on, so synchronous rectification is performed and the load current flows through the switching elements Q5 and Q8. During the period of mode 4c, the switching elements Q6 and Q8 are turned on, so the secondary rectifier circuit 13C is short-circuited and the short-circuit current flows through the switching elements Q6 and Q8. During the subsequent period of mode 2, the switching elements Q6 and Q7 are turned on, so synchronous rectification is performed and the load current flows through the switching elements Q6 and Q7.

[0068] During step-up control, the control unit 20C drives the switching elements Q6 and Q8 based on the control formula of the above formula (2). Also, the on / off period of the switching element Q5 is a period obtained by phase-shifting the on / off periods of the switching elements Q1 and Q4 by a phase angle θ. The on / off period of the switching element Q7 is a period obtained by phase-shifting the on / off periods of the switching elements Q2 and Q3 by a phase angle θ.

[0069] In the current-resonant DC / DC converter 1C, in addition to the effects of the current-resonant DC / DC converter 1A according to the above embodiment, the load current during synchronous rectification control can be made to flow through the switching elements Q5 and Q7 without passing through the diodes D5 and D7. Therefore, in the current-resonant DC / DC converter 1C, the losses during step-up operation and synchronous rectification operation can be reduced.

[0070] [Third Modified Example] FIG. 9 shows a current-resonant DC / DC converter 1D according to a third modified example of the present invention. The current-resonant DC / DC converter 1D is a CLLC-type DC / DC converter in which the current-resonant DC / DC converter 1A is duplicated, and includes a main circuit section 10D, a control section 20D, and terminals T1 to T4.

[0071] The main circuit section 10D has the same configuration as that of the above embodiment, except that it includes a capacitor C11, includes a secondary resonant circuit 14, and includes a secondary rectifier circuit 13D instead of the secondary rectifier circuit 13A. The main circuit section 10D can switch between forward power transmission from the terminals T1 and T2 (primary side) to the terminals T3 and T4 (secondary side) and reverse power transmission from the terminals T3 and T4 (secondary side) to the terminals T1 and T2 (primary side) under the control of the control section 20D.

[0072] The capacitor C11 is a capacitor for reducing the output ripple during reverse power transmission and is connected between the terminals T1 and T2.

[0073] The secondary resonance circuit 14 has the same configuration as the primary resonance circuit 12. The resonance inductance Lr2 and the resonance capacitance Cr2 of the secondary resonance circuit 14 are connected to the terminals of the secondary winding of the transformer Tr, and together with the excitation inductance of the transformer Tr, they form a series resonance circuit. Therefore, the secondary rectifier circuit 13D serves as the drive circuit of the secondary resonance circuit 14 during reverse power transmission.

[0074] The secondary rectifier circuit 13D includes a switching element Q5 in the upper arm of the third leg, a switching element Q6 in the lower arm of the third leg, a switching element Q7 in the upper arm of the fourth leg, and a switching element Q8 in the lower arm of the fourth leg. Diodes D5 to D8 are connected in parallel in the reverse direction in the current paths of the switching elements Q5 to Q8, and capacitors C5 to C8 are connected in parallel. That is, the secondary rectifier circuit 13D has the same configuration as the secondary rectifier circuit 13C of the second modification example.

[0075] The control unit 20D has the same configuration as the above-described embodiment, the first modification example, and / or the second modification example, except that it performs control during reverse power transmission in the same way as during forward power transmission. During forward power transmission and step-up control, the control unit 20D may perform on-period control on the switching elements Q6 and Q8 as in the above-described embodiment, or may perform on-period control on the switching elements Q7 and Q5 as in the first modification example, or may further perform synchronous rectification control and phase shift control as in the second modification example. The same applies during reverse power transmission and step-up control. The control unit 20D performs the control that was performed on the secondary rectifier circuit 13D during forward power transmission on the primary drive circuit 11 during reverse power transmission, and performs the control that was performed on the primary drive circuit 11 during forward power transmission on the secondary rectifier circuit 13D during reverse power transmission. Note that the step-up control during reverse power transmission corresponds to the "second step-up control" of the present invention.

[0076] Fig. 10 shows an example of the control timing of the switching elements Q1 to Q8 during reverse power transmission and boost control. In Fig. 10, the control unit 20D drives the secondary-side rectifier circuit 13D (switching elements Q5 to Q8) by frequency modulation control, and drives the primary-side drive circuit 11 (switching elements Q1 to Q4) by phase shift control, synchronous rectification control, and on-period control (corresponding to the "second on-period control" of the present invention). Therefore, the primary-side drive circuit 11 becomes a boost / rectifier circuit during reverse power transmission. The control timing in Fig. 10 is obtained by swapping the primary-side control signal and the secondary-side control signal in Fig. 8 of Modification 2.

[0077] The control unit 20D phase-shifts the on / off timing of the switching elements Q1 to Q4 by a predetermined phase angle θ (0° ≤ θ ≤ 180° when one cycle is 360°) with respect to the on / off timing of the switching elements Q5 to Q8 by phase shift control and synchronous rectification control. The phase angle θ corresponds to the "second phase angle" of the present invention. Note that the second phase angle may be the same value as the first phase angle or a different value.

[0078] Furthermore, in the example of Fig. 10, the on-period of the switching element Q2 is the logical sum (OR) of the on-period of the switching element Q6 (corresponding to the "third on-period" of the present invention) and the period obtained by phase-shifting the on-period by the phase angle θ (corresponding to the "fourth on-period" of the present invention). Similarly, the on-period of the switching element Q4 is the logical sum (OR) of the on-period of the switching element Q5 (corresponding to the "third on-period" of the present invention) and the period obtained by phase-shifting the on-period by the above phase angle θ (corresponding to the "fourth on-period" of the present invention). That is, during reverse power transmission and boost control, the control unit 20D sets the on-period of the switching element Q4 (Q2), which is different from the switching element Q2 (Q4) that causes a short-circuit current to flow by turning on in the primary-side drive circuit 11, as the logical sum (OR) of the on-period of the switching element Q5 (Q6) that causes a resonance current to flow by turning on in the secondary-side rectifier circuit 13D and the period obtained by phase-shifting the on-period.

[0079] During reverse power transmission and boost control, the control unit 20D drives the switching elements Q2 and Q4 based on the control formula of the following formula (3).

Number

[0080] In formula (3), Q6S is a drive pulse (control signal) obtained by shifting the on / off timing of Q6t by a phase angle θ, and Q5S is a drive pulse (control signal) obtained by shifting the on / off timing of Q5t by a phase angle θ. The control unit 20D calculates the logical sum of the control signal Q6t and the control signal Q6S to generate the control signal Q2t, and calculates the logical sum of the control signal Q5t and the control signal Q5S to generate the control signal Q4t.

[0081] In mode 3c', since the switching element Q2 is on, a short-circuit circuit is formed in the primary-side drive circuit 11, and a large short-circuit current flows through the short-circuit circuit. Here, if the switching element Q4 is off, the short-circuit current flows through the diode D4, but in mode 3c', since the switching element Q4 is on, the short-circuit current flows through the switching element Q4. As a result, the loss can be reduced compared to the case where the short-circuit current flows through the diode D4. In mode 3c', the switching elements Q5 and Q8 correspond to the "fourth switching element" of the present invention, the switching element Q2 corresponds to the "fifth switching element" of the present invention, and the switching element Q4 corresponds to the "sixth switching element" of the present invention.

[0082] During the subsequent mode 1' period, since the switching elements Q1 and Q4 are on, synchronous rectification is performed, and the load current flows through the switching elements Q1 and Q4. As a result, the loss can be reduced even during the mode 1' period. Note that the on / off period of the switching element Q1 is a period obtained by shifting the on / off period of the switching elements Q5 and Q8 by a phase angle θ.

[0083] In mode 4c', since the switching element Q4 is on, a short - circuit circuit is formed in the primary - side drive circuit 11, and a large - current short - circuit current flows through the short - circuit circuit. Here, if the switching element Q2 is off, the short - circuit current flows through the diode D2. However, in mode 4c', since the switching element Q2 is on, the short - circuit current flows through the switching element Q2. As a result, the loss can be reduced compared with the case where the short - circuit current flows through the diode D2. In mode 4c', the switching elements Q6 and Q7 correspond to the "fourth switching element" of the present invention, the switching element Q4 corresponds to the "fifth switching element" of the present invention, and the switching element Q2 corresponds to the "sixth switching element" of the present invention.

[0084] During the subsequent mode 2', since the switching elements Q2 and Q3 are on, synchronous rectification is performed, and the load current flows through the switching elements Q2 and Q3. As a result, the loss can be reduced even during mode 2'. Note that the on - off period of the switching element Q3 is equal to the period in which the on - off periods of the switching elements Q6 and Q7 are phase - shifted by the phase angle θ.

[0085] Therefore, in the current - resonant - type DC / DC converter 1D, the losses during step - up operation and synchronous - rectification operation can be reduced both during forward - power transmission and during reverse - power transmission. Furthermore, in the current - resonant - type DC / DC converter 1D, both during forward - power transmission and during reverse - power transmission, by a relatively simple control process (the process using the control equations of equations (2) and (3)), the on - period control can be executed to obtain the above - mentioned effects.

[0086] [Fourth Modified Example] FIG. 11 shows a current - resonant - type DC / DC converter 1E according to the fourth modified example of the present invention. The current - resonant - type DC / DC converter 1E is an LLC - type DC / DC converter, and includes a main - circuit section 10A, a control section 20E, and terminals T1 to T4. The current - resonant - type DC / DC converter 1E has the same configuration as the above - mentioned embodiment except that it includes a control section 20E instead of the control section 20A.

[0087] The control unit 20E has the same configuration as that of the above embodiment, except that during the boost control, the off-period control is performed under a predetermined condition. The off-period control is a control for turning off a switching element Q8 (Q6) different from the switching element Q6 (Q8) that allows a short-circuit current to flow by turning on when the current due to the synchronous rectification operation of the secondary-side rectifier circuit 13A (switching elements Q6, Q8) is equal to or less than a preset first set value. The first set value is preferably set to a value that can stably detect the decrease in the current due to the synchronous rectification operation, for example, zero or a value close to zero.

[0088] FIG. 12 shows an example of the control timings of the switching elements Q1 to Q4, Q6, and Q8 during the boost control and the off-period control. In FIG. 12, the control unit 20E drives the primary-side drive circuit 11 (switching elements Q1 to Q4) by frequency modulation control, and drives the secondary-side rectifier circuit 13A (switching elements Q6, Q8) by phase shift control, synchronous rectification control, on-period control, and off-period control.

[0089] In FIG. 12, during the period of mode 5, the switching element Q8 is turned off by the off-period control, and during the period of mode 6, the switching element Q6 is turned off by the off-period control. The other periods are the same as those in FIG. 4 of the above embodiment.

[0090] The control unit 20E drives the switching elements Q6 and Q8 based on the control formula of the following formula (4). Formula (4) is obtained by taking into account the synchronous rectification condition (off-period control condition) of the switching element Q6 or the switching element Q8 in formula (2).

Equation

[0091] In Equation (4), I6 is a function that becomes 0 when the current value of the current flowing through the switching element Q6 due to the synchronous rectification operation is equal to or less than the first set value, and becomes 1 when the current value exceeds the first set value. When the function I6 is 0, the control signal Q6t turns off the switching element Q6. Similarly, I8 is a function that becomes 0 when the current value of the current flowing through the switching element Q8 due to the synchronous rectification operation is equal to or less than the first set value, and becomes 1 when the current value exceeds the first set value. When the function I8 is 0, the control signal Q8t turns off the switching element Q8.

[0092] The control unit 20E calculates the logical product (AND) of the logical sum of the control signal Q2t and the control signal Q2S and the function I6, and generates the control signal Q6t. Similarly, the control unit 20E calculates the logical product (AND) of the logical sum of the control signal Q1t and the control signal Q1S and the function I8, and generates the control signal Q8t.

[0093] Regarding the output of the determination result as to whether or not the current value is equal to or less than the first set value, the control unit 20E can adopt any known method. For example, the control unit 20E may detect the output voltage V2 using a current transformer, and compare and output the output voltage V2 and the voltage value corresponding to the first set value using a comparator. Alternatively, the control unit 20E may detect the voltage between the current paths of the switching element Q6 or the switching element Q8 (for example, the voltage between the collector (drain) terminal and the emitter (source) terminal of the switching element Q6 or Q8), and compare and output the detected voltage value and the voltage value corresponding to the first set value using a comparator. Alternatively, the control unit 20E may estimate the phase of the current flowing through the secondary rectifier circuit 13A from the resonance current flowing through the primary side drive circuit 11, calculate the current value, and output the determination result.

[0094] In the current resonance type DC / DC converter 1E, since the currents flowing through the switching elements Q8 and Q6 become zero or a value close to zero due to the off period control and are turned off, the direction of the current does not reverse, and the loss can be reduced.

[0095] Note that the off-period control is also applicable in other modified examples. For example, in the current resonance type DC / DC converter 1D according to the third modified example, the control unit 20D can perform off-period control both during forward power transmission and reverse power transmission. For example, when the current due to the synchronous rectification operation of the primary side drive circuit 11 is equal to or less than a preset second set value during reverse power transmission, the control unit 20D that performs off-period control turns off the switching element Q4 (Q2) different from the switching element Q2 (Q4) through which a short-circuit current flows in the primary side drive circuit 11.

[0096] [Fifth Modified Example] FIG. 13 shows a current resonance type DC / DC converter 1F according to the fifth modified example of the present invention. The current resonance type DC / DC converter 1F is an LLC type DC / DC converter, and includes a main circuit section 10A, a control unit 20F, and terminals T1 to T4. The current resonance type DC / DC converter 1F has the same configuration as the above-described embodiment except that it includes a control unit 20F instead of the control unit 20A.

[0097] The control unit 20F during boost control has the same configuration as the above-described embodiment except that it performs PWM control (corresponding to the "inverted on-period control" of the present invention). The control unit 20F during boost control may, for example, switch between the control mode that performs the control of the above-described embodiment and the control mode that performs the PWM control described in this modified example, or may use only the control mode that performs the PWM control described in this modified example.

[0098] FIG. 14 shows an example of the control timing of the switching elements Q1 to Q4, Q6, and Q8 during boost control. In FIG. 14, the control unit 20F drives the primary side drive circuit 11 (switching elements Q1 to Q4) by frequency modulation control and drives the secondary side rectification circuit 13A (switching elements Q6, Q8) by PWM control. In the control mode in which the control unit 20F performs PWM control, the control unit 20F does not perform phase shift control on the secondary side rectification circuit 13A (switching elements Q6, Q8).

[0099] In FIG. 14, the switching elements Q6 and Q8 are turned on only during the periods of modes 3a and 4a during the boost operation. That is, the switching elements Q6 and Q8 are turned on in accordance with the on-period when the on-period of the primary-side drive circuit 11 (switching elements Q1 to Q4) starts, and are turned off at the start of the synchronous rectification control (thereby, the synchronous rectification operation is not performed in the control mode that performs PWM control).

[0100] During the control mode that performs PWM control, the control unit 20F drives the switching elements Q6 and Q8 based on the control formula of the following formula (5).

Equation

[0101] In formula (5), since Q1S is a control signal obtained by phase-shifting the on / off timing of Q1t by the phase angle θ, Not(Q1S) is a control signal obtained by inverting the high level (H) and low level (L) of the control signal Q1S. Similarly, since Q2S is a control signal obtained by phase-shifting the on / off timing of Q2t by the phase angle θ, Not(Q2S) is a control signal obtained by inverting the high level (H) and low level (L) of the control signal Q2S.

[0102] The control unit 20F calculates the logical sum of the logical product of the control signal Q1t and the control signal Not(Q1S) (corresponding to the "logical product A" of the present invention) and the logical product of the control signal Q2t and the control signal Not(Q2S) (corresponding to the "logical product B" of the present invention), and generates the control signal Q6t and the control signal Q8t. Note that the switching element Q1 corresponds to the "first switching element A" of the present invention, and the switching element Q2 corresponds to the "first switching element B" of the present invention.

[0103] In the period of Mode 3a in FIG. 14 (the period of phase angle θ), the switching elements Q1 and Q4 are simultaneously turned on in the primary side drive circuit 11, and a resonance current flows. In the secondary side rectifier circuit 13A, the switching elements Q6 and Q8 are simultaneously turned on, and a large short-circuit current flows through the secondary side rectifier circuit 13A. In the subsequent period of Mode 5, since the switching elements Q6 and Q8 are off, the load current flows through the diode D5 of the third leg and the diode D8 of the fourth leg.

[0104] In the period of Mode 4a (the period of phase angle θ), the switching elements Q2 and Q3 are simultaneously turned on in the primary side drive circuit 11, and a resonance current flows in the opposite direction to that in Mode 3a. In the secondary side rectifier circuit 13A, the switching elements Q6 and Q8 are simultaneously turned on, and a large short-circuit current flows through the secondary side rectifier circuit 13A. In the subsequent period of Mode 6, since the switching elements Q6 and Q8 are off, the load current flows through the diode D7 of the fourth leg and the diode D6 of the third leg.

[0105] In the current resonance type DC / DC converter 1F according to the fifth modification example, the boost operation can be performed by simple PWM control without performing phase shift control, and the loss during the boost operation can be reduced.

[0106] Note that the control mode for performing PWM control is also applicable in other modification examples. For example, in the current resonance type DC / DC converter 1D according to the third modification example, the control unit 20D can perform the PWM control of the fifth modification example both during forward power transmission and during reverse power transmission. For example, the control unit 20D that performs PWM control (corresponding to the "inverted on-period control" of the present invention) during reverse power transmission turns on the switching elements Q2 and Q4 only during the periods of Modes 3c' and 4c' shown in FIG. 10.

[0107] [Other Modification Examples] The configurations of the above-described embodiments and the configurations of the above-described modification examples can be combined as appropriate.

Explanation of Reference Numerals

[0108] 1A~1F Current resonance type DC / DC converter Main circuit sections 10A to 10D 11 Primary-side drive circuit 12 Primary-side resonance circuit 13A to 13D Secondary-side rectifier circuits 14 Secondary-side resonance circuit Control sections 20A to 20F

Claims

1. A current resonance type DC / DC converter comprising a main circuit section and a control section, wherein the main circuit section comprises a transformer including a primary winding and a secondary winding, a primary side circuit in which a first leg and a second leg are connected in parallel, and each leg includes an upper arm and a lower arm, a primary side resonance circuit including a first resonance inductance and a first resonance capacitance, and the first resonance inductance and the first resonance capacitance are connected in series to the primary winding between a connection point of the upper arm and the lower arm of the first leg and a connection point of the upper arm and the lower arm of the second leg, a secondary side circuit in which a third leg and a fourth leg are connected in parallel, and each leg includes an upper arm and a lower arm, and the secondary winding is connected between a connection point of the upper arm and the lower arm of the third leg and a connection point of the upper arm and the lower arm of the fourth leg, wherein the upper arm and the lower arm of each leg of the primary side circuit include at least a switching element, wherein one of the upper arm or the lower arm of each leg of the secondary side circuit includes at least a switching element and the other includes at least a diode, wherein the control section performs first boost control for causing the main circuit section to perform a boost operation with the secondary side circuit as an output side, wherein the switching element of the primary side circuit includes a first switching element that generates a resonance current in the primary side circuit when turned on, wherein the switching element of the secondary side circuit includes a second switching element that generates a short-circuit current in the secondary side circuit when turned on and a third switching element provided in a current path of the short-circuit current, wherein the control section during the first boost control performs first on-period control in which the on-period of the third switching element is set to a logical sum period of a first on-period which is the on-period of the first switching element and an on-period obtained by phase-shifting the first on-period by a predetermined first phase angle. A current resonance type DC / DC converter characterized by the above.

2. A secondary side resonance circuit including a second resonance inductance and a second resonance capacitance, and the second resonance inductance and the second resonance capacitance are connected in series to the secondary winding between a connection point of the upper arm and the lower arm of the third leg and a connection point of the upper arm and the lower arm of the fourth leg. ​ The upper arm and the lower arm of each leg of the secondary side circuit include at least a switching element, The control unit performs second boost control to cause the main circuit unit to perform a boost operation with the primary side circuit as an output side, The switching element of the secondary side circuit includes a fourth switching element that generates a resonance current in the secondary side circuit when turned on, The switching element of the primary side circuit includes a fifth switching element that generates a short-circuit current in the primary side circuit when turned on, and a sixth switching element provided in the current path of the short-circuit current, During the second boost control, the control unit performs second on-period control such that the on-period of the sixth switching element is a period that is a logical sum of the third on-period, which is the on-period of the fourth switching element, and an on-period obtained by phase-shifting the third on-period by a predetermined second phase angle. The current resonance type DC / DC converter according to claim 1, characterized in that.

3. During the first boost control, the control unit causes the secondary side circuit to perform a synchronous rectification operation, and turns off the third switching element when the current due to the synchronous rectification operation is equal to or less than a preset first set value. The current resonance type DC / DC converter according to claim 1, characterized in that.

4. During the second boost control, the control unit causes the primary side circuit to perform a synchronous rectification operation, and turns off the sixth switching element when the current due to the synchronous rectification operation is equal to or less than a preset second set value. The current resonance type DC / DC converter according to claim 2, characterized in that.

5. Comprising a main circuit unit and a control unit, The main circuit unit includes a transformer including a primary winding and a secondary winding, a primary side circuit in which a first leg and a second leg are connected in parallel, and each leg includes an upper arm and a lower arm, a primary side resonance circuit including a first resonance inductance and a first resonance capacitance, and the first resonance inductance and the first resonance capacitance are connected in series to the primary winding between the connection points of the upper arm and the lower arm of the first leg and the connection points of the upper arm and the lower arm of the second leg. The third leg and the fourth leg are connected in parallel, each leg includes an upper arm and a lower arm, and a secondary side circuit to which the secondary winding is connected is provided between a connection point of the upper arm and the lower arm of the third leg and a connection point of the upper arm and the lower arm of the fourth leg. A current resonance type DC / DC converter comprising: The upper arm and the lower arm of each leg of the primary side circuit include at least a switching element. One of the upper arm or the lower arm of each leg of the secondary side circuit includes at least a switching element, and the other includes at least a diode. The control unit performs first boost control for causing the main circuit unit to perform a boost operation with the secondary side circuit as an output side. The switching elements of the primary side circuit include a first switching element A that generates a first resonance current in the primary side circuit when turned on, and a first switching element B that generates a second resonance current in the primary side circuit in a direction opposite to the first resonance current when turned on. The switching elements of the secondary side circuit include a second switching element that generates a short-circuit current in the secondary side circuit when turned on, and a third switching element provided in a current path of the short-circuit current. During the first boost control, the control unit calculates a logical product A between the on period of the first switching element A and a period obtained by phase-shifting and inverting the on period by a predetermined first phase angle, calculates a logical product B between the on period of the first switching element B and a period obtained by phase-shifting and inverting the on period by the first phase angle, and performs inverted on period control in which the on period of the third switching element is set to a period of a logical sum of the logical product A and the logical product B. A current resonance type DC / DC converter characterized by the above.

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