Power control circuit of direct ac-ac converter, and power control method

Abstract

To provide a power control method that is not required for setting a switching frequency in a high frequency band, that enables controlling high frequency power by a fixed control frequency, and is downsized in capacity of a power supply filter without becoming low-frequency pulsation according to a power supply current; and to provide a power control circuit thereof.SOLUTION: There is provided a high frequency power control circuit in an electromagnetic induction type non-contact energy transmission system composed of a bidirectional switch 2 and a resonant tank 3. One of a bidirectional switch 2 in a positive half-cycle of an input power supply is a main switch; the other is a sub switch; correspondence between the main switch and the sub switch is changed in a negative half-cycle; two operation patterns of a first mode in which the sub switch is turned off at all times and a second mode in which the sub switch is turned on at all times are alternately switched; control cycles of groups of the first mode and the second mode are fixed; and each of the cycles of the first mode and the second mode is adjusted according to a load to substantially linearly control a high frequency output power.SELECTED DRAWING: Figure 5

Description

[Technical Field]

[0001] The present invention relates to power control of a direct frequency conversion circuit (direct AC-AC converter) that can directly generate single-phase high-frequency alternating current from a commercial power source in a non-contact electromagnetic energy transmission system such as induction heating (IH) or wireless power transfer (WPT). [Background technology]

[0002] Both IH systems and WPT systems belong to the electromagnetic induction contactless energy transfer (IPT) system category, and research and development into these systems is intensifying as a power supply technology for home appliances, commercial consumer devices, information and communication terminals, electric vehicles (EVs), and other applications. The high-frequency power conversion inverter, a core component of an IPT system, typically consists of a rectifier stage, a boost PFC (power factor correction) stage, and a DC / HFAC (direct current / high-frequency alternating current) inverter (see Figure 16(1)). Aiming to simplify the system configuration, reduce size, and increase efficiency, a direct AC-AC converter (see Figure 16(2)) has been proposed, which consists of a resonant tank connected to a commercial power source and a bidirectional switch consisting of a power device. This converter can generate single-phase high-frequency AC directly from the commercial power source (see Patent Document 1 and Non-Patent Document 1).

[0003] In the case of this direct AC-AC converter, pulse frequency modulation (PFM) is the basis for controlling high-frequency power, but because audible noise (interference noise) is likely to occur due to deviations in operating frequency between loads such as multiple induction devices that operate at different frequencies, it is necessary to control power at a constant frequency (fixed frequency) to reduce interference noise between multiple loads. However, in an IPT system with a resonant circuit on the power transmission and reception sides, such as the magnetic resonance method, it is necessary to follow the resonant frequency with respect to coil misalignment and fluctuations in the gap length. In this magnetic resonance method, it is necessary to match the frequencies of the primary side (transmission side) and the secondary side (reception side). This is mainly due to the reasons of resonance conditions and maximum power transmission. The resonance condition means a state where the maximum power can be transmitted from the power transmission side to the power reception side. Maximum power transmission means that under the resonant frequency, the sharpness (Q value) of the resonance of the system becomes maximum (which means the energy loss is minimum), and by operating the primary side and the secondary side at the same frequency, maximum power transmission and system efficiency can be guaranteed. In addition, since the impedance characteristics are determined according to the load, it is necessary to set multiple PFM bands. Also, in the case of a household IH cooker, for example, according to the standards of the Radio Law of Japan, 90 kHz is the upper limit of the switching frequency and there is a limit to the output range, and there is a problem that it becomes difficult to fit within the frequency band defined by such standards. Furthermore, in the case of high output, there is also a problem that the soft commutation operation (soft switching) of the power semiconductor switch cannot be performed, leading to an increase in power loss and electromagnetic noise. In addition, when the load impedance fluctuates greatly and deviates from the soft switching range, there is also a problem that the circuit operation becomes unstable.

[0004] In view of the above problems, the present inventors have already proposed a power control circuit and method for a direct AC-AC converter that applies pulse density modulation (PDM) of a fixed-frequency pulse. The proposed power control method for a direct AC-AC converter is a power control circuit in a high-frequency power conversion system for induction heating composed of a resonant tank and a bidirectional switch. In the positive half-cycle of the input power supply, one of the bidirectional switches is the main switch and the other is the sub-switch. In the negative half-cycle, the correspondence between the main switch and the sub-switch is swapped, and the sub-switch is always turned off in the reversely blocking (RB) mode and always turned on in the reversely conducting (RC) mode. The ratio of the two operation patterns is adjusted according to the induction heating load. Thereby, while fixing the driving frequency of the bidirectional switch, the high-frequency output power is controlled almost linearly (see Patent Document 2). In this proposed method, the driving frequencies of the two active switches constituting the bidirectional switch are fixed (maintained constantly), the RB mode and the RC mode of the current are continuously operated respectively, and pulse density modulation (PDM) that modulates the ratio of the RB mode and the RC mode according to the load is applied. Thus, the ratio of the number of pulses of the two modes within a fixed control period is adjusted to control the output power.

[0005] The proposed power control method for a direct AC-AC converter has a plurality of problems described below. In the control period of the proposed direct AC-AC converter, since pulse density modulation (PDM) is applied, in order to ensure the control resolution of the output power, it is necessary to sufficiently lower the switching frequency of the RB mode and the RC mode with respect to the switching frequency. In this case, the switching frequency becomes a low-frequency pulsation in the power supply current, and there is a problem that the filter capacitance needs to be increased and the power supply filter becomes large. In addition, since it becomes magnetic noise when it enters the audible range (20 kHz or less), it is inevitably necessary to set the switching frequency in the high-frequency range (for example, 200 kHz or more), and there is a problem that the application range of the power control method is limited. In addition, the power control resolution is low, and discrete power control is inevitable.

Prior Art Documents

Patent Documents

[0006]

Patent Document 1

Patent Document 2

Non-Patent Documents

[0007]

Non-Patent Document 1

Summary of the Invention

Problems to be Solved by the Invention

[0008] As described above, in PDM, it is necessary to sufficiently lower the switching frequencies of the RB mode and the RC mode with respect to the switching frequency. However, the switching frequency becomes a low-frequency pulsation in the power supply current, which not only increases the capacitance of the power supply filter but also causes a problem of magnetic noise when it enters the audible frequency range. Therefore, it is inevitably necessary to set the switching frequency in the high-frequency range, and there is a problem that the application range of power control is limited.

[0009] In view of the above situation, an object of the present invention is to provide a power control method and a power control circuit capable of high-frequency power control with a fixed control frequency, not causing low-frequency pulsation in the power supply current, enabling downsizing of the power supply filter, and not requiring setting of the switching frequency in the high-frequency range, that is, having a wide selection range of the switching frequency.

Means for Solving the Problems

[0010] To solve the above problems, the power control circuit of the direct AC-AC converter of the present invention is a high-frequency power control circuit of an electromagnetic induction non-contact energy transmission system composed of a resonant tank and a bidirectional switch. In the positive half-cycle of the input power supply, one of the bidirectional switches is the main switch and the other is the auxiliary switch. In the negative half-cycle, the correspondence between the main switch and the auxiliary switch is swapped, and the auxiliary switch is always turned off in the first mode, and the second mode in which the auxiliary switch is always turned on. The two operation patterns are alternately switched, the control period of the set of the first and second modes is fixed, and each period of the first and second modes is adjusted according to the load to control the high-frequency output power almost linearly.

[0011] A pulse group in which the first and second modes are continuously arranged one by one is set as one set. Among them, while fixing the main switch off period in the first mode and the main switch off period in the second mode, the main switch on period in the first mode and the main switch on period in the second mode are adjusted to control the power at high speed. Specifically, the on-time ratio of one of the two modes is controlled. In this specification, it is called dual pulse-density modulation (DPRM).

[0012] In the power control circuit of the direct AC-AC converter of the present invention, it is preferable that the switch control unit controls the main switch on period in the first mode and the main switch on period in the second mode while fixing the main switch off period in the first mode and the main switch off period in the second mode in the pulse group in which the first and second modes are continuously arranged. Fixing the off period of the main switch is to achieve soft switching. Thereby, fixed frequency control can be realized according to the load fluctuation.

[0013] In the power control circuit of the direct AC-AC converter of the present invention, when the first mode and the second mode have the same period, there is a difference in the effective value of the main switch current. In controlling the main switch on-time period of the first mode and the main switch on-time period of the second mode, the increase amount of the positive current time integral of the main switch current due to the increase in the main switch on-time period of the second mode is larger than the decrease amount of the positive current time integral of the main switch conduction current due to the decrease in the main switch on-time period of the first mode. In the second mode where the auxiliary switch is always on, as will be described later, there is a regenerative current (negative part of the current) and the output power is low. However, in the combination of the first mode and the second mode, within a fixed period, when the second mode increases, only the positive current time integral value of the main switch current increases. Therefore, the increased current time integral of the second mode is larger than the decreased current time integral of the first mode. As a whole (as a combination of the first mode and the second mode), the output power can be increased or decreased, and the respective periods of the first mode and the second mode can be adjusted according to the load.

[0014] The power control circuit of the direct AC-AC converter of the present invention includes: 1) a mode ratio determination unit that determines the mode ratio in the control period of the first mode or the second mode of the bidirectional switch based on the deviation between the input current and the set current; 2) a switch control unit that controls the gate drive of the bidirectional switch based on the input power supply voltage, the resonance capacitor voltage, and the mode ratio; and 3) a switch drive unit that drives the gate of the bidirectional switch based on the drive signal output by the switch control unit. Details of the control logic of the power control circuit will be described in the embodiments below.

[0015] In the power control circuit of the direct AC-AC converter of the present invention, the control period is fixed to twice the average value of the switching periods of the first mode and the second mode. As a result, the control period of the pulse group combination in which the first mode and the second mode are continuous is fixed, and it becomes twice the switching period of the main switch (the frequency is 1 / 2). Consequently, higher-resolution power control can be achieved. Furthermore, since the power control period is twice the switching frequency, it can respond to load fluctuations faster than ten times or more that of the prior art. Therefore, it is expected that the responsiveness of the power control system will also be improved.

[0016] In the power control circuit of the direct AC-AC converter of the present invention, in the first mode, when the main switch of the bidirectional switch turns off, the resonance capacitor voltage rises, resonates with the inductance of the load, and the voltage of the main switch rises from zero to achieve zero-voltage turn-off. After the voltage of the bidirectional switch reaches the peak value, it drops and reverses to a negative voltage, which is applied to the antiparallel diode of the auxiliary switch, and both ends of the main switch maintain zero voltage. On the other hand, in the second mode, the auxiliary switch is turned on in accordance with the turn-off of the main switch, the resonance capacitor voltage rises, resonates with the inductance of the load, and the voltage of the main switch rises from zero to achieve zero-voltage turn-off. After the voltage of the bidirectional switch reaches the peak value and then drops to zero, at the same time, the current reverses through the auxiliary switch and the antiparallel diode of the main switch, and the auxiliary switch turns off in accordance with the turn-on timing of the main switch.

[0017] Next, the power control method of the direct AC-AC converter of the present invention will be described. The power control method of the direct AC-AC converter of the present invention is a power control method in a high-frequency power conversion circuit of an electromagnetic induction non-contact energy transmission system composed of a resonance tank and a bidirectional switch, including the steps of: in the positive half-cycle of the input power supply, one of the bidirectional switches is the main switch and the other is the auxiliary switch, and in the negative half-cycle, the correspondence between the main switch and the auxiliary switch is swapped; alternately switching between two operation patterns, namely the first mode in which the auxiliary switch is always off and the second mode in which the auxiliary switch is always on, and setting the control period of the combination of the first and second modes as a fixed period; and adjusting each period of the first mode and the second mode according to the load, so as to control the high-frequency output power substantially linearly.

[0018] In the step of adjusting each cycle of the first mode and the second mode in the power control method of the direct AC-AC converter of the present invention according to the load, it is preferable to fix the main switch off period of the first mode and the main switch off period of the second mode, and control the main switch on period of the first mode and the main switch on period of the second mode. This is because soft switching can be achieved and fixed frequency control can be realized according to the load fluctuation.

Advantages of the Invention

[0019] According to the present invention, the power control frequency can be constantly fixed at half of the switching frequency. Therefore, compared with pulse density modulation (PDM), the selection range of the switching frequency is widened, and it can be extended to a lower frequency range (however, it does not fall below the audible range). Also, since the control period is relatively shorter and the frequency is higher, in addition to improving the control responsiveness, the power control resolution can be improved. Furthermore, the reduction of the pulsation phenomenon included in the power supply current enables the miniaturization of the power supply filter.

Brief Description of the Drawings

[0020]

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5

Figure 6

Figure 7

Figure 8

Figure 9

Figure 10

Figure 11

Figure 12

Figure 13

Figure 14

Figure 15

Figure 16

Mode for Carrying Out the Invention

[0021] Hereinafter, an example of an embodiment of the present invention will be described in detail with reference to the drawings. Note that the scope of the present invention is not limited to the following examples and illustrated examples, and numerous changes and modifications are possible.

Example

[0022] The configuration of the main circuit of the direct AC-AC converter of the present invention is shown in FIG. 1. As shown in FIG. 1, the main circuit 1 is composed of a bidirectional switch 2 and a resonant tank 3 formed by a power device (power semiconductor) connected to a commercial power supply 4. The resonant tank 3 is shown in a magnetic coupling model, and its primary side is represented by an inductance L p and a resonant capacitor C r and its secondary side is represented by an inductance L s and a resistance R0. Further, the bidirectional switch 2 is composed of, for example, power transistors of SiC-MOSFETs connected in series in a drain connection and is driven by a gate. Diodes (D1, D2) are connected in parallel to each switch (S1, S2).

[0023] The control logic of the main circuit of the direct AC-AC converter system is shown in FIG. 2. The control logic switches the correspondence between the main switch and the auxiliary switch in the positive half cycle of the input power supply, with one of the bidirectional switches as the main switch and the other as the auxiliary switch, and reverses the correspondence between the main switch and the auxiliary switch in the negative half cycle. It alternately switches between two operation patterns: the reverse blocking (RB) mode in which the auxiliary switch is always off and the reverse conduction (RC) mode in which the auxiliary switch is always on. The control period of the pair of the RC mode and the RB mode is fixed, and each period of the RC mode and the RB mode is adjusted according to the load to control the high-frequency output power almost linearly.

[0024] The mode ratio determination unit 10 determines the control quantity ρ RC (the ratio of the RC mode in the fixed control period of the bidirectional switch 10) from the signal calculated through the error amplifier (proportional integral controller) 12 that takes the deviation 11 between the effective value of the input current and its set current (command value) as the input, through the limit circuit 13.

[0025] The switch control unit 20 controls the on period of the main switch in the RC mode and the on period of the main switch in the RB mode while fixing the off period of the main switch in the RC mode and the off period of the main switch in the RB mode in the pulse group that makes the pair of the RC mode and the RB mode continuous, based on the input power supply voltage v in and the resonance capacitor voltage v Cr and the mode ratio ρ RC .

[0026] First, a signal for determining the timing of the polarity determination of the input power supply voltage v in and the resonance capacitor voltage v Cr is selected through a multiplexer (MUX). That is, as shown in FIG. 5, using the signal detected from the input power supply voltage sensor 6, the comparator 21 determines the polarity of the input power supply voltage v in , and using the signal detected from the voltage sensor 7 of the resonance capacitor Cr, the comparators (22a, 22b) determine the resonance capacitor voltage v CrThe timing of polarity determination is determined by comparing the output signals of the comparator (22a) and the comparator (22b) with the input power supply voltage v in The resonant capacitor voltage v is selected by the multiplexer (MUX) 23 using the polarity judgment of Cr The end point of the off period in the RC mode and the end point of the off period in the RB mode are detected by the polarity determination timing signal. Then, after a small delay time is given by the delay circuit 24, the main switch (V in The on-time of the monostable vibrator is determined by the positive edge (rising edge) of the oscillator (monostable vibrator) 26, which determines the on-timing of the control variable ρ determined by the mode ratio determination unit 10. RC Based on this, the on-period of RC mode (ρ RC ) and RB mode on period (1-ρ RC ) is selected by a multiplexer (MUX) 27.

[0027] Here, the D flip-flop 25 receives the output signal of the delay circuit 24 as a CK (clock) input signal, and its rising edge is as shown in FIG. 11 (in FIG. 11, S DFF (25) is D flip-flop 25, S DFF (28) is a D flip-flop 28) synchronized with the rising edge of the main switch, and its rising timing is determined by the resonant capacitor voltage v Cr Polarity and input power voltage V in It is determined by the polarity of the CK input signal. In a D flip-flop, if the Q inverted output is connected to the D input terminal, it is possible to achieve frequency division by 2 (output frequency is 1 / 2 of the input frequency). In other words, it inverts at each rising edge of the CK input signal, and the control period for the two operating pattern sets of RC mode and RB mode is set to 2-fold the CK input signal, generating a Q output signal that selects the respective delay time depending on the RC mode or RB mode.

[0028] In the RC mode and the RB mode, the on-time from the turn-on timing needs to be set separately. In the subsequent logic of the D flip-flop 25, it is necessary to select the respective delay times according to the RC mode and the RB mode. The continuous set of the RC mode and the RB mode is a fixed control cycle. For example, if the moment of the first turn-on is in the RC mode, the signal connected to the input terminal of the monostable vibrator 26 will be the delay time of the RC mode. At the moment of the next turn-on, since it switches to the RB mode, the signal connected to the input terminal of the monostable vibrator 26 will be the delay time of the RB mode. That is, the frequency of the signal connected to the input terminal of this monostable vibrator 26 is half of the main switch frequency. In the RC mode, it is the delay time ρ of the RC mode RC , and in the RB mode, it is the delay time 1 - ρ of the RB mode RC . Therefore, the output of the MUX 27 becomes the signal connected to the input terminal of the monostable vibrator 26. The input frequency is half of the main switch frequency, and it is a PWM (Pulse Width Modulation) signal that is Low in the RC mode and High in the RB mode. Note that this signal is the output of the D flip-flop 25.

[0029] The main switch control signal SW main generates a synchronization signal from the zero-crossing (positive to negative) point of the resonant capacitor detected by the voltage sensor 7 of the resonant capacitor C r , and gives a slight delay time to the generated synchronization signal by the delay circuit 24 to determine the on-timing of the main switch. The pulse width is determined by the output signal from the above-mentioned oscillator (monostable vibrator) 26, is input to the MUX (29a, 29b), and selects between the RC mode and the RB mode via the switch drive unit 30. The CK input signal of the D flip-flop 28 is the inverted output signal from the oscillator (monostable vibrator) 26. The signal after dividing this by two is input to the MUX (29a, 29b), and selects between the RC mode and the RB mode via the switch drive unit 30.

[0030] Based on the output signal of the switch control unit 20, the switch driving unit 30 selects the switch (Q1, Q2) to be driven and drives the gate of the power transistor of the corresponding switch. Through the switch driving unit 30, the main switch Q1 and the sub-switch Q2 are turned on and off at high frequency, and a set of two operating modes, namely the RC mode and the RB mode, are repeated.

[0031] As shown in Fig. 3(1), in the direct AC-AC converter of the present invention, two operating patterns, namely the RB mode in which the sub-switch is always off and the RC mode in which the sub-switch is always on, alternate with each other, and a set of control periods of the RC mode and the RB mode is fixed. This is different from the proposed direct AC-AC converter in that, as shown in Fig. 3(2), in the proposed direct AC-AC converter, the periods of the RC mode and the RB mode are fixed, and in a certain control period, the output power is adjusted by adjusting the ratio of each mode. When the periods of the RC mode and the RB mode are fixed, comparing the RC mode and the RB mode with the same period, the power of the RB mode is higher than that of the RC mode. Therefore, when the periods of the RC mode and the RB mode are fixed and the ratio of each mode is adjusted in a certain control period, as shown in Fig. 3(2), when the ratio of the RC mode is large, the power is low, and when the ratio of the RB mode is large, the power is high.

[0032] On the other hand, as shown in Fig. 4, in the direct AC-AC converter of the present invention, a set of control periods of the RC mode and the RB mode is fixed, and the periods of the continuous RC mode and RB mode in that set are adjusted according to the load to adjust the output power. Here, when the ratio of the RC mode increases, the output power increases (high output), and conversely, when the ratio of the RB mode increases, the output power decreases (low output). This is because there is a regenerative current (negative component of the current) in the RC mode. When the RC mode increases, only the positive part of the current increases. Therefore, the current time integral when the RC mode increases is larger than the current time integral when the RB mode decreases. This will be described in detail later.

[0033] Next, the state transition diagrams and theoretical operation waveforms of the RB mode and the RC mode are shown in FIGS. 6 and 7, respectively, and the circuit operations of each are described below. When the main switch is turned off, the resonant tank (L P and C r ) becomes the resonant state RB1 of the RB mode or the resonant state RC1 of the RC mode.

[0034] <RB mode> In the RB mode, when the main switch is turned off, the voltage V r of the capacitor C Cr gradually increases and resonates with the equivalent inductance L r of the IH load. From this, the voltage of the main switch is the sum of V Cr and the power supply voltage V in , and gradually increases from zero to achieve ZVS turn-off. After the bidirectional switch voltage V SW reaches the peak value, it gradually decreases and reverses to a negative voltage. This negative voltage is applied to the anti-parallel diode of the auxiliary switch, and both ends of the main switch maintain zero voltage. Therefore, when a gate signal is applied, the main switch realizes zero-voltage turn-on. At this time, although a load current flows into the switch, due to the aforementioned zero-voltage turn-on, no switching loss occurs. Thereafter, the current linearly increases.

[0035] (1-1) State RB1: Resonant state positive half cycle When the main switch is turned off, the current of the IH coil L P flows into the capacitor C r connected in parallel to the IH load, and the Cr voltage v Cr (t) and the Lp current i Lp (t) resonate as follows.

[0036] [Equation]

[0037] (1-2) State RB2: Resonant state negative half cycle (before Q1 turn-on) Bidirectional switch V during the blocking period sw After resonating in the positive direction, it starts negative excitation after passing through the zero voltage crossing. At this time, the total voltage of V sw appears across the auxiliary switch.

[0038] (1 - 3) State RB3: Resonant state negative half cycle (after Q1 turns on) When the main switch S1 is turned on at time t2, Q1 obtains ZVS turn-on. During this period, the auxiliary switch S2 remains off and no current flows through the bidirectional switch.

[0039] (1 - 4) State RB4: Conduction period of the main switch After the negative polarity excitation of the voltage V sw of the switch Q1 ends, the voltage applied across the auxiliary switch reverses and starts conduction via S1 - D2. From this, L P has V in applied to it, and from Equation 2 below, i sw rises almost linearly. Equation 2 is used to describe the change in current while the current is conducting, which corresponds to the transient response of an equivalent RL circuit. Since the conduction time is very short and far from the steady state, the first two terms of the Taylor expansion are used here as approximate values. This approximation does not affect the subsequent analysis of the circuit.

[0040]

Number

[0041]

Number

[0042]

Number

[0043] <RC mode> On the other hand, in the RC mode, the auxiliary switch is turned on in accordance with the turn-off of the main switch. From this, the voltage V r of the capacitor CCr gradually increases, and the equivalent inductance L of the IH load r resonates with it. As a result, the voltage of the main switch is V Cr and the power supply voltage V in are added and gradually increase from zero to achieve ZVS turn-off. After the bidirectional switch voltage V SW reaches its peak, it gradually decreases. When it becomes zero, current flows backward through the antiparallel diode of the auxiliary switch and the main switch. During that time, when the gate of the main switch is driven, zero-voltage and zero-current turn-on is achieved. On the other hand, the auxiliary switch turns off in accordance with the on-timing of the main switch. The auxiliary switch has zero-voltage turn-on and zero-voltage turn-off.

[0044] (2-1) State RC1: Resonance section When the main switch S1 is turned off, the IH coil L P current flows into the capacitor C r connected in parallel with the IH load, and starts to resonate at L P -C r as shown in the above formula 1.

[0045] (2-2) State RC2: Before turn-on After the resonance voltage V sw increases and then reaches zero again during the cut-off period, the auxiliary switch automatically conducts. During the conduction period of the main switch, a constant voltage is applied to the IH coil L P , and i SW increases almost linearly from the above formula 2.

[0046] (2-3) State RC3: After turn-on When the main switch S1 is turned on during the period when both switches are conducting, ZVS turn-on is obtained.

[0047] (2-4) State RC4: Current i SW Positive period When the conduction current of both switches switches to the forward direction at time t3, it becomes a power supply section from the power supply, and the circuit operation returns to the non-resonance section and i SW increases linearly.

[0048] The main circuit operates in two operating patterns: the RB mode and the RC mode. In the RB mode, during the conduction period of the main switch (state RB4), the commercial power supply voltage is applied to L P and I sw increases almost linearly. When it turns off, the resonant tank enters the resonant state (states RB1, RB2, RB3). After the resonant voltage V sw draws an arc in the positive resonant state (state RB1) and passes through zero, when the main switch is turned on from the point when it passes through zero until it reaches zero again after being negatively excited, zero voltage soft switching (ZVS) is achieved. From the point when it reaches zero again, I sw returns to the state of the first conduction period, and one cycle of operation is completed.

[0049] On the other hand, in the RC mode, during the cut-off period (state RC1), after the resonant voltage V sw increases and first reaches zero, I sw increases almost linearly from negative (states RC2, RC3, RC4). To achieve ZVS, the main switch is turned on before it reaches 0A.

[0050] <Soft switching conditions> The operating waveform of V sw in Fig. 7 shows the ZVS achievement states of the main switch and the auxiliary switch. In the RC mode, during the cut-off period (state RC1), when the resonant voltage V SW reaches zero after rising, i SW increases linearly from negative (states RC2 to RC4). In this section, when the main switch is turned on, ZVS can be obtained. Furthermore, it is a prerequisite that the auxiliary switch is turned on during the period when the polarity of i SW is negative (states RC2, RC3). On the other hand, the necessary condition for achieving ZVS in the RB mode is to maintain the off state of the auxiliary switch during the period when V SW is negatively excited (states RB2, RB3).

[0051] <Regarding high-frequency power control> Two pulses each are within one fixed control period T Ctrl and the two modes are switched alternately. Here, if the time of the RC mode is defined as T RC the duty ratio of the RC mode is expressed by the following Equation 5.

[0052]

Equation

[0053] With the same conduction time, the effective values of the conduction currents in the RC mode and the RB mode are very different. That is, when adjusting the duty ratio ρ Ctrl of RC within the fixed control period T RC the effective value of the total conduction current during the control period changes, and thus the output power can be controlled. As shown in the circuit configuration of the mode ratio determination unit 10 in the power control circuit of FIG. 5, the deviation 11 between the input current i in detected by the current sensor and the set current i in * is calculated by the PI controller 12 to determine the mode ratio ρ RC of the RC mode. Note that as another power control method, the detected input current i in and the input voltage V in are used to calculate the output power P o and the deviation from the set power P set is calculated by the PI controller to determine the mode ratio ρ RC of the RC mode. Note that due to the constraints of the soft switching range, the RB mode has an upper limit, so for power control, the RC mode is adjusted and the RB mode is provided as its complement. That is, the master-slave relationship between the RC mode and the RB mode cannot be changed.

[0054] There are differences in the effective current values between the two operating modes of the RC mode and the RB mode, as shown in FIGS. 8 and 9. Based on this difference, the mode ratio of each mode in the combination of the RC mode and the RB mode is adjusted. The adjustment of the ratio between the RC mode and the RB mode is performed according to the output target value (set value) with a fixed control period. Here, the difference in the effective current values of the switch currents in the RC mode and the RB mode will be described. In a situation where the parameters of the circuit elements are fixed, the time integral of the switching current over one cycle is determined by two parameters (T sw *d and t sw ). T A *d is defined as the conduction time of the bidirectional switch within one cycle (d = conduction time of the one-cycle current / switching period T sw ). Also, t sw defines the current transient response waveform of the RL circuit that simplifies the circuit operation when the switch is on, and the value of t = 0 is set as the minimum value of the resonance current of the RLC circuit that simplifies the circuit operation when the switch is off. The magnitude of the current when the switching current conducts corresponds to the time of the above current transient response waveform as I A . A

[0055] Under normal parameter sets, the effective value I s of the conduction current's AC input current is expressed by the following formula. Here, a normal parameter set means that the capacitor resonance voltage does not exceed the maximum withstand voltage of the MOSFET transistor of the switch and there is an appropriate margin, and the current does not exceed the maximum current of the MOSFET. Figure 10 shows the visualization of this formula. According to Figure 10, it can be seen that there is an obvious difference in the magnitudes of the currents in the two modes of the RC mode and the RB mode. The d in Figure 10 is expressed as d = conduction time of the one-cycle current / switching period T sw , which is a concept close to the duty ratio.

[0056]

Number

[0057] <Regarding simulation results> The effectiveness of the power control circuit and method of the present invention is verified by simulation. The circuit parameter conditions are shown in Table 1 below.

[0058]

Table 1

[0059] Figure 12 shows the simulation results of the operating waveforms under the conditions of switching frequency f sw = 64 kHz, control cycle frequency f ctrl set to 32 kHz, and power supply frequency 60 Hz. From this result, it can be seen that the primary side coil current i LP and the resonant capacitor voltage v Cr both show high-frequency switching frequency components of the bidirectional switch along with the low-frequency envelope waves corresponding to the double frequency of the power supply frequency, indicating that direct AC-AC conversion is achieved.

[0060] Next, the waveforms of the simulation results at the switching frequency f sw = 64 [kHz] level are shown in Figure 13. Figures 13(1) to (4) show the simulation results of different operating waveforms with the mode ratio ρ RC being 0.4, 0.45, 0.5, and 0.6 respectively. As shown in the figure, in any of the different ρ RC (0.4, 0.45, 0.5, 0.6), the master-slave relationship of the two active switches Q1 and Q2 in the bidirectional switch is switched according to the polarity of the power supply voltage, and the RC mode and RB mode are achieved according to the control cycle frequency f ctrl (32 kHz). Since the main switch obtains soft switching (ZVS) and soft switching can be achieved during the switching between the two modes, it can be seen that ZVS of the bidirectional switch is achieved even for load fluctuations. Also, as ρ RC increases, the resonant voltage peak value in the RC mode decreases and its current conduction time expands. At the same time, the switch-off voltage in the RB mode increases and its current conduction time is shortened.

[0061] Figure 14 shows the harmonic analysis of the primary side coil current i RC when the mode ratio ρ LP is 0.6. Within the control cycle, the periods of the RC mode and RB mode change, but in the frequency spectrum, f sw and f ctrlSince only the components of

[0062] Control frequency f ctrl is fixed at 32 [kHz], and the power characteristics obtained by adjusting ρ RC are shown in Fig. 15. The graph shown in the figure shows the power characteristics when the mode ratio ρ RC of the two operation patterns of the RB mode and the RC mode is changed from 35% to 60%. Since there is an approximate linearity between the output power and ρ RC , it becomes easy to design the controller parameters of the power control circuit shown in Fig. 5, etc., and it can be seen that it is a control system suitable for high-frequency power control.

[0063] As described above, a power control circuit and a power control method capable of high-frequency power control with a fixed frequency in the direct AC-AC converter of the present invention are shown, and their effectiveness is shown by simulation.

Industrial Applicability

[0064] The present invention is useful for industrial, industrial, household, and medical IH devices, and for EV battery chargers, etc. as a WPT system.

Explanation of Signs

[0065] 1 Main circuit of direct AC-AC converter 2 Bidirectional switch 3 Resonant tank 4 Input power supply (commercial power supply) 5 Load 6 Input power supply voltage sensor 7 Resonant capacitor voltage sensor 8 Current sensor 10 Mode ratio determination unit 11 Differentiator 12 PI controller (proportional integral controller) 13 Limit circuit 20 Switch control unit 21, 22a, 22b Comparator 23, 27, 29a, 29b Multiplexer (Selection Circuit) 24 Delay Circuit 25, 28 D Flip-Flop 26 Oscillator (Monostable Vibrator) 30 Switch Driving Unit 31a, 31b Gate Driving Circuit

Claims

1. A high-frequency power control circuit for an electromagnetic induction non-contact energy transmission system composed of a resonant tank and a bidirectional switch, wherein in the positive half-cycle of the input power supply, one of the bidirectional switches is the main switch and the other is the auxiliary switch, and in the negative half-cycle, the correspondence between the main switch and the auxiliary switch is reversed, and the auxiliary switch is always turned off in the first mode, and the second mode in which the auxiliary switch is always turned on are alternately switched, the control period of the combination of the first and second modes is fixed, the periods of the first and second modes are adjusted according to the load, and the high-frequency output power is controlled substantially linearly. A power control circuit for a direct AC-AC converter.

2. In a pulse group in which the combinations of the first mode and the second mode are continuous, while fixing the main switch off period of the first mode and the main switch off period of the second mode, the main switch on period of the first mode and the main switch on period of the second mode are controlled. The power control circuit for a direct AC-AC converter according to claim 1, characterized in that.

3. The power control circuit for a direct AC-AC converter according to claim 1, characterized in that when the first mode and the second mode have the same period, there is a difference in the effective value of the main switch current.

4. In controlling the main switch on period of the first mode and the main switch on period of the second mode, The increase amount of the positive current time integral of the main switch current due to the increase of the main switch on period of the second mode is larger than the decrease amount of the positive current time integral of the main switch current due to the decrease of the main switch on period of the first mode. The power control circuit for a direct AC-AC converter according to claim 2, characterized in that.

5. The power control circuit for a direct AC-AC converter according to claim 2, characterized in that the control period is fixed to twice the average value of the switching periods of the first mode and the second mode.

6. A mode ratio determination unit that determines the mode ratio in the control period of the first mode or the second mode of the bidirectional switch based on the deviation between the input current and the set current, A switch control unit that controls the gate drive of the bidirectional switch based on the input power supply voltage, the resonant capacitor voltage, and the mode ratio, A switch drive unit that drives the gate of the bidirectional switch based on the drive signal output by the switch control unit, The power control circuit of the direct AC-AC converter according to claim 2, characterized by comprising

7. In the first mode, when the main switch of the bidirectional switch turns off, the capacitor voltage of the resonant tank rises and resonates with the inductance of the load. The voltage of the main switch rises from zero to zero-voltage turn-off. After the voltage of the bidirectional switch reaches the peak value, it drops and reverses to a negative voltage. The negative voltage is applied to the antiparallel diode of the auxiliary switch, and both ends of the main switch maintain zero voltage. In the second mode, the auxiliary switch is turned on in accordance with the turn-off of the main switch. The capacitor voltage of the resonant tank rises and resonates with the inductance of the load. The voltage of the main switch rises from zero to zero-voltage turn-off. After the voltage of the bidirectional switch reaches the peak value, it drops to zero, and at the same time, the current reverses through the auxiliary switch and the antiparallel diode of the main switch. The auxiliary switch turns off in accordance with the turn-on timing of the main switch. The power control circuit of the direct AC-AC converter according to any one of claims 1 to 6, characterized by the above.

8. A power control method in a high-frequency power conversion circuit of an electromagnetic induction type non-contact energy transmission system composed of a resonant tank and a bidirectional switch, A step of making one of the bidirectional switches the main switch and the other the auxiliary switch in the positive half-cycle of the input power supply, and swapping the correspondence between the main switch and the auxiliary switch in the negative half-cycle; A step of alternately switching between two operation patterns: a first mode in which the auxiliary switch is always off and a second mode in which the auxiliary switch is always on, and setting the control period of the set of the first and second modes as a fixed period; A step of adjusting each period of the first mode and the second mode according to the load; The power control method of the direct AC-AC converter, characterized by comprising and controlling the high-frequency output power substantially linearly.

9. In the step of adjusting each period of the first mode and the second mode according to the load, the off period of the main switch in the first mode and the off period of the main switch in the second mode are fixed, and the on period of the main switch in the first mode and the on period of the main switch in the second mode are controlled. The power control method of the direct AC-AC converter according to claim 7, characterized by the above.

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