Ultrasonic vibrator drive circuit

Abstract

To provide an ultrasonic vibrator drive circuit capable of improving high turn-on and turn-off speeds in the ultrasonic vibrator drive circuit using a voltage-driven transistor.SOLUTION: An ultrasonic vibrator drive circuit includes: a capacitor arranged in series between a secondary winding for a first switching element and the source of the first switching element; and a capacitor also arranged in series between the secondary winding of the second switching element and the source of the second switching element, thereby a gate bias voltage generated by the capacitors is added in series to a drive voltage generated by the secondary winding of the first switching element, and the gate bias voltage generated by the capacitors is also added in series to the drive voltage generated by the secondary winding of the second switching element.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] The present invention relates to an ultrasonic transducer drive circuit, and more particularly to an ultrasonic transducer drive circuit using a voltage-driven transistor that is devised to enable faster turn-on and turn-off. [Background technology]

[0002] A conventional ultrasonic transducer drive circuit has a configuration as shown in FIG. 23, for example. First, there is a series circuit 101 of a choke coil L2 and an ultrasonic vibrator UT, and an oscillation circuit 103 is connected to this series circuit 101. This oscillation circuit 103 imparts a resonant frequency to the ultrasonic vibrator UT to drive it.

[0003] The oscillator circuit 103 has the following configuration: First, there is a power transformer T2, which is composed of a primary winding N2P and a secondary winding N2S. A first switching element Q1 and a second switching element Q2 are provided to intermittently energize the primary winding N2P of the power transformer T2.

[0004] The first switching element Q1 and the second switching element Q2 are bipolar junction transistors (BJTs).

[0005] The first switching element Q1 is composed of a base B1, a collector C1, and an emitter E1, and the second switching element Q2 is also composed of a base B2, a collector C2, and an emitter E2.

[0006] Also, a DC power supply Vin is provided, and the positive side of this DC power supply Vin is connected to the collector C1 of the first switching element Q1, and the negative side is connected to the emitter E2 of the second switching element Q2.

[0007] A voltage detection unit 105 is provided to detect the voltage across the ultrasonic transducer UT. This voltage detection unit 105 is made up of three capacitors C5, C6, and C7.

[0008] There is provided a phase adjustment section 107 that adjusts the phase so that the voltage detected by the voltage detection section 105 is fed back positively. This phase detection section 107 is made up of a coil L3 and a capacitor C4.

[0009] There is provided a positive feedback transformer T1 that positively feeds back to the first switching element Q1 and the second switching element Q2 the voltage whose phase has been adjusted by the phase adjustment unit 107. The positive feedback transformer T1 is composed of a primary winding N1P, a secondary winding N1SA for the first switching element Q1, and a secondary winding N1SB for the second switching element Q2.

[0010] A resistor R3 and a diode D3 are connected in series between the base B1 of the first switching element Q1 and the secondary winding N1SA of the positive feedback transformer T1, and a capacitor C3 is connected in series between the secondary winding N1SA and the primary winding N2P of the power transformer T2.

[0011] A capacitor C10 is connected in parallel to the series circuit of the resistor R3 and diode D3. A resistor R10 is connected in parallel to the base B1 and collector C1 of the first switching element Q1. A diode D1 is connected in parallel to the emitter E1 and collector C1 of the first switching element Q1.

[0012] A resistor R4 and a diode D4 are connected in series between the base B2 of the second switching element Q2 and the secondary winding N1SB of the positive feedback transformer T1.

[0013] A capacitor C11 is connected in parallel to the series circuit of the diode D4 and resistor R4. A resistor R23 is connected in parallel to the base B2 and collector C2 of the second switching element Q2. A diode D2 is connected in parallel to the emitter E2 and collector C2 of the second switching element Q2.

[0014] For example, Patent Document 1 discloses the configuration of this type of ultrasonic transducer drive circuit.

[0015] Another conventional ultrasonic transducer drive circuit is shown in Fig. 24. In this example, voltage-driven transistors are used as the first switching element Q1 and the second switching element Q2. The first switching element Q1 is composed of a gate G1, a drain D1, and a source S1, and the second switching element Q2 is also composed of a gate G2, a drain D2, and a source S2.

[0016] Moreover, instead of the diode D3, capacitor C10, and resistor R3 on the side of the first switching element Q1 in the drive circuit shown in FIG. 23, Zener diodes ZN2 and ZN3, a capacitor C18, and a resistor R3 are used. Also, resistor R27 is used instead of resistor R10. Similarly, instead of the diode D4, capacitor 11, and resistor 4 on the second switching element Q2 side, Zener diodes ZN4 and ZN5, a capacitor C19, and a resistor R4 are used. Also, resistor R28 is used instead of resistor R23. The other configurations are the same as those of the ultrasonic transducer drive circuit shown in FIG. 23, and the same parts in the figure are given the same reference numerals and their explanations are omitted.

[0017] For example, Patent Document 2 discloses the configuration of this type of ultrasonic transducer drive circuit. [Prior art documents] [Patent documents]

[0018] [Patent Document 1] Japanese Patent Application Publication No. 7-265794 [Patent Document 2] Japanese Patent Application Publication No. 4-46562 Summary of the Invention [Problem to be solved by the invention]

[0019] The above conventional configuration has the following problems. First, in the case of the ultrasonic vibrator driving circuit shown in FIG. 23, the first switching element Q1 and the second switching element Q2 use bipolar transistors, which causes the problem that the on-voltage cannot be reduced and the conduction loss is large. Furthermore, since the first switching element Q1 and the second switching element Q2 use bipolar transistors, there is a problem in that the switching speed is slow and switching loss is large. Furthermore, since the first switching element Q1 and the second switching element Q2 use bipolar transistors, there is a problem in that the currents in the bases B1 and B2 are large, resulting in large driving losses. Furthermore, since the first switching element Q1 and the second switching element Q2 use bipolar transistors, there is a problem in that the current flowing through the coil L3 is large in order to supply current to the bases B1 and B2.

[0020] In contrast, in the case of the ultrasonic transducer driving circuit shown in FIG. 24, the first switching element Q1 and the second switching element Q2 use voltage-driven transistors, which solves the problems that arise when bipolar transistors are used, but there is another problem.

[0021] First, the drive voltages VT11 and VT12 from the positive feedback transformer T1 are sine waves. Also, because the first switching element Q1 and the second switching element Q2 use voltage-driven transistors, they have a gate threshold voltage Vth(ON) and will not turn on unless the gate voltage exceeds the gate threshold voltage Vth(ON). This poses the problem of high-speed switching being impossible and resulting in large switching losses.

[0022] Furthermore, in order to self-excite the circuit or to reduce the switching loss, the secondary voltages VT11 and VT12 of the positive feedback transformer T1 must be increased, which increases the current flowing through the Zener diodes ZN2, ZN3, ZN4, and ZN5, resulting in larger losses, and therefore the Zener diodes ZN2, ZN3, ZN4, and ZN5 becoming larger and more expensive.

[0023] Furthermore, because the currents of the Zener diodes ZN2, ZN3, ZN4, and ZN5 are supplied from the secondary side of the power transformer T2, there is a problem that the current through the coil L3 becomes large, and there is also a problem that the coil L3 becomes large and expensive.

[0024] The present invention has been made based on these points, and its purpose is to provide an ultrasonic transducer drive circuit that can speed up turn-on and turn-off in an ultrasonic transducer drive circuit that uses a voltage-driven transistor.

[0025] In order to solve the above problems, an ultrasonic vibrator drive circuit according to claim 1 of the present invention comprises a series circuit of an ultrasonic vibrator and a choke coil, and an oscillation circuit that drives the ultrasonic vibrator by applying the resonant frequency of the ultrasonic vibrator to the series circuit, wherein the oscillation circuit comprises a power transformer, first and second switching elements that intermittently energize a primary winding of the power transformer, a voltage detection unit that detects the voltage between both ends of the ultrasonic vibrator, a phase adjustment unit that adjusts the phase so that the voltage detected by the voltage detection unit is positively fed back, and a positive feedback transformer that positively feeds back the voltage whose phase has been adjusted by the phase adjustment unit to the first switching element and the second switching element, and the first switching element and the second switching element are voltage-driven transistors each having a gate, a drain, and a source. a capacitor is arranged in series between the secondary winding for the first switching element and the source of the first switching element, and a capacitor is also arranged in series between the secondary winding for the second switching element and the source of the second switching element, whereby a gate bias voltage generated by the capacitor is added in series to a drive voltage generated by the secondary winding for the first switching element, and the gate bias voltage generated by the capacitor is added in series to a drive voltage generated by the secondary winding for the second switching element. Furthermore, the ultrasonic vibrator driving circuit according to claim 2 is the ultrasonic vibrator driving circuit according to claim 1, characterized in that a resistor is arranged in series between the secondary winding for the first switching element and the gate of the first switching element, and a resistor is also arranged in series between the secondary winding for the second switching element and the gate of the second switching element. Furthermore, the ultrasonic vibrator driving circuit according to claim 3 is the ultrasonic vibrator driving circuit according to claim 2, characterized in that a series circuit of two Zener diodes is connected in parallel to the series circuit of the secondary winding for the first switching element, the capacitor, and the resistor, and a series circuit of two Zener diodes is connected in parallel to the series circuit of the secondary winding for the second switching element, the capacitor, and the resistor. [Effects of the Invention]

[0026] As described above, the ultrasonic vibrator drive circuit according to claim 1 of the present invention comprises a series circuit of an ultrasonic vibrator and a choke coil, and an oscillation circuit that drives the ultrasonic vibrator by applying the resonant frequency of the ultrasonic vibrator to the series circuit, wherein the oscillation circuit comprises a power transformer, a first switching element and a second switching element that intermittently energize the primary winding of the power transformer, a voltage detection unit that detects the voltage between both ends of the vibrator, a phase adjustment unit that adjusts the phase so that the voltage detected by the voltage detection unit is positively fed back, and a positive feedback transformer that positively feeds back the voltage whose phase has been adjusted by the phase adjustment unit to the first switching element and the second switching element, and the first switching element and the second switching element use voltage-driven transistors each having a gate, a drain, and a source. the positive feedback transformer is composed of a primary winding, a secondary winding for the first switching element, and a secondary winding for the second switching element, a capacitor is arranged in series between the secondary winding for the first switching element and the source of the first switching element, and a capacitor is also arranged in series between the secondary winding for the second switching element and the source of the second switching element, so that the gate bias voltage generated by the capacitor is added in series to the drive voltage generated by the secondary winding for the first switching element, and the gate bias voltage generated by the capacitor is added in series to the drive voltage generated by the secondary winding for the second switching element, thereby enabling faster turn-on and turn-off. Furthermore, according to the ultrasonic vibrator driving circuit of claim 2, in the ultrasonic vibrator driving circuit of claim 1, a resistor is arranged in series between the secondary winding for the first switching element and the gate of the first switching element, and a resistor is also arranged in series between the secondary winding for the second switching element and the gate of the second switching element, thereby enabling faster turn-on and turn-off. According to the ultrasonic vibrator drive circuit of claim 3, in the ultrasonic vibrator drive circuit of claim 2, a series circuit of two Zener diodes is connected in parallel to the series circuit of the secondary winding for the first switching element, the capacitor, and the resistor, and a series circuit of two Zener diodes is connected in parallel to the series circuit of the secondary winding for the second switching element, the capacitor, and the resistor, thereby enabling faster turn-on and turn-off. [Brief explanation of the drawings]

[0027] [Figure 1] FIG. 1 is a circuit diagram showing an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 2] FIG. 1 is a waveform diagram showing a simulation result of an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 3] FIG. 10 is a waveform diagram showing a simulation result of a conventional ultrasonic transducer drive circuit. [Figure 4] FIG. 1 is a waveform diagram showing a simulation result of an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 5] FIG. 10 is a waveform diagram showing a simulation result of a conventional ultrasonic transducer drive circuit. [Figure 6] FIG. 1 is a waveform diagram showing a simulation result of an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 7] FIG. 10 is a waveform diagram showing a simulation result of a conventional ultrasonic transducer drive circuit. [Figure 8]FIG. 1 is a waveform diagram showing a simulation result of an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 9] FIG. 10 is a waveform diagram showing a simulation result of a conventional ultrasonic transducer drive circuit. [Figure 10] FIG. 1 is a waveform diagram showing a simulation result of an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 11] FIG. 10 is a waveform diagram showing a simulation result of a conventional ultrasonic transducer drive circuit. [Figure 12] FIG. 1 is a waveform diagram showing a simulation result of an ultrasonic transducer drive circuit according to an embodiment of the present invention. [Figure 13] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 14] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 15] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 16] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 17] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 18] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 19] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 20] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 21] FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 22]FIG. 1 is a diagram showing an embodiment of the present invention, and is a partial circuit diagram for explaining an approximate analysis of the operation. [Figure 23] FIG. 1 is a circuit diagram showing a configuration of an ultrasonic transducer drive circuit according to a conventional example. [Figure 24] FIG. 1 is a circuit diagram showing a configuration of an ultrasonic transducer drive circuit according to a conventional example. DETAILED DESCRIPTION OF THE INVENTION

[0028] An embodiment of the present invention will be described below with reference to Figures 1 to 22. According to this embodiment, the ultrasonic transducer drive circuit has a configuration as shown in Figure 1. First, there is a series circuit 1 of a choke coil L2 and an ultrasonic transducer UT, and an oscillator circuit 3 is connected to this series circuit 1. This oscillator circuit 3 imparts a resonant frequency to the ultrasonic transducer UT to drive it.

[0029] The oscillator circuit 3 has the following configuration: First, there is a power transformer T2, which is composed of a primary winding N2P and a secondary winding N2S, and there are also provided a first switching element Q1 and a second switching element Q2 that intermittently energize the primary winding N2P of the power transformer T2.

[0030] In this embodiment, voltage-driven transistors are used as the first switching element Q1 and the second switching element Q2. Examples of voltage-driven transistors include an insulated gate bipolar transistor (IGBT), a silicon metal oxide semiconductor field effect transistor (Si-MOSFET), a silicon carbide metal oxide semiconductor field effect transistor (SiC-MOSFET), and a gallium nitride field effect transistor (Gan-FET). In this embodiment, a Si-MOSFET is used.

[0031] The first switching element Q1 is composed of a gate G1, a drain D1, and a source S1, and the second switching element Q2 is also composed of a gate G2, a drain D2, and a source S2.

[0032] A DC power supply Vin is also provided to supply input power Vin. The positive side of this DC power supply Vin is connected to the drain D1 of the first switching element Q1, and the negative side is connected to the source S2 of the second switching element Q2.

[0033] A voltage detection unit 5 is provided to detect the voltage across the ultrasonic transducer UT. This voltage detection unit 5 is made up of three capacitors C5, C6, and C7.

[0034] A phase adjustment section 7 is provided to adjust the phase so that the voltage detected by the voltage detection section 5 is fed back positively. The phase detection section 7 is made up of a coil L3 and a capacitor C4.

[0035] A positive feedback transformer T1 is provided that positively feeds back to the first switching element Q1 and the second switching element Q2 the voltage whose phase has been adjusted by the phase adjustment unit 7. The positive feedback transformer T1 is composed of a primary winding N1P, a secondary winding N1SA for the first switching element Q1, and a secondary winding N1SB for the second switching element Q2.

[0036] A gate drive resistor R3 is connected in series between the gate G1 of the first switching element Q1 and the secondary winding N1SA of the positive feedback transformer T1. Furthermore, a capacitor C18 and a capacitor C3 are connected in series between the secondary winding N1SA and the primary winding N2P of the power transformer T2. A bias voltage Vgs1 (Bias) is newly generated by the capacitor C18 and is added in series to the drive voltage VT11 generated by the secondary winding N1SA.

[0037] A series circuit of Zener diodes ZN2 and ZN3 is connected in parallel to the series circuit of the gate drive resistor R3, secondary winding N1SA, and capacitor C18. A starting resistor R27 is also connected.

[0038] A gate drive resistor R4 is connected in series between the gate G2 of the second switching element Q2 and the secondary winding N1SB of the positive feedback transformer T1. A capacitor C19 is connected in series between the secondary winding N1SB and the source S2 of the second switching element Q2. A bias voltage Vgs2 (Bias) is newly generated by the capacitor C19 and is added in series to the drive voltage VT12 generated by the secondary winding N1SB.

[0039] A series circuit of Zener diodes ZN4 and ZN5 is connected in parallel to the series circuit of the gate drive resistor R4, secondary winding N1SB, and capacitor C19. A starting resistor R28 is also connected.

[0040] The operation of the above configuration will now be described. First, the first switching element Q1 and the second switching element Q2 are switched in opposite phases (duty cycle 50%) using a half-bridge system. A drive voltage VT11 is supplied to the gate G1 of the first switching element Q1 via a secondary winding N1SA of a positive feedback transformer T1. Similarly, a drive voltage VT12 is supplied to the gate G2 of the second switching element Q2 via a secondary winding N1SB of the positive feedback transformer T1. The drive voltages VT11 and VT12 are sinusoidal voltages with opposite phases.

[0041] The drive voltage VT11 will now be described in detail. The drive voltage VT11 is a substantially sinusoidal voltage that utilizes the resonance of the secondary circuit of the power transformer T2. To make the drive voltage VT11 a voltage that can drive the first switching element Q1, the secondary voltage of the power transformer T2, the constant of the ultrasonic vibrator UT, the constant of the choke coil L2, and the constants of the capacitors C5, C6, and C7 are set to appropriate values.

[0042] Moreover, Zener diodes ZN2 and ZN3 are used to protect the gate G1 of the first switching element Q1 so that the sine wave peak voltage of the drive voltage VT11 does not exceed the allowable voltage of the gate G1.

[0043] Furthermore, the constants of the coil L3 and the capacitor C4 are set to a frequency and phase that allow zero current switching of the first switching element Q1. The above constants are not set individually but simultaneously.

[0044] The same is true for the drive voltage VT12, which is an approximately sinusoidal voltage because it utilizes the resonance of the secondary circuit of the power transformer T2. In order to obtain a voltage capable of driving the second switching element Q2 with the drive voltage VT12, the secondary voltage of the power transformer T2, the constant of the ultrasonic vibrator UT, the constant of the choke coil L2, and the constants of the capacitors C5, C6, and C7 are set to appropriate values.

[0045] Zener diodes ZN4 and ZN5 are used to protect the gate G2 of the second switching element Q1 so that the sine wave peak voltage of the drive voltage VT12 does not exceed the allowable voltage of the gate G2.

[0046] Furthermore, the constants of the coil L3 and the capacitor C4 are set to have a frequency and phase that allow zero current switching of the second switching element Q2. The above constants are not set individually but simultaneously.

[0047] Furthermore, the capacitor C18 generates a bias voltage Vgs1(Bias) equivalent to the gate threshold voltage Vgs1(ON). This generated bias voltage Vgs1(Bias) is a DC voltage and is added in series to the drive voltage VT11. The drive voltage V1 is obtained by adding the bias voltage Vgs1(Bias) in series to the drive voltage VT11.

[0048] The drive voltage VT11 is a sine wave voltage and the bias voltage Vgs1 (Bias) is a direct current, so the drive voltage V1 that drives the gate G1 of the first switching element Q1 has a waveform that is the sum of these. The bias voltage Vgs1 (Bias) is added to the drive voltage V1 from the zero crossing of the drive voltage VT11, and a sinusoidal current with the same frequency and phase as the drive voltage VT11 can be made to flow as the drain current Id of the first switching element Q1.

[0049] The same is true for the drive voltage VT12, where a bias voltage Vgs2(Bias) equivalent to the gate threshold voltage Vgs2(ON) is generated in the capacitor C19. This generated bias voltage Vgs2(Bias) is a DC voltage and is added in series to the drive voltage VT12. The drive voltage V2 is the result of adding the bias voltage Vgs2(Bias) in series to the drive voltage VT12.

[0050] The drive voltage VT12 is a sine wave voltage and the bias voltage Vgs2 (Bias) is a direct current, so the drive voltage V2 that drives the gate G2 of the second switching element Q2 has a waveform that is the sum of these. The bias voltage Vgs2 (Bias) is added to the drive voltage V2 from the zero crossing of the drive voltage VT12, and a sinusoidal current with the same frequency and phase as the drive voltage VT12 can be made to flow as the drain current Id of the first switching element Q2.

[0051] As described above, the present embodiment can provide the following effects. First, zero current switching is achieved, which reduces switching loss. Furthermore, the sine wave voltage of the drive voltage VT11 for the first switching element Q1 can be lowered, thereby reducing the current flowing through the Zener diodes ZN2 and ZN3 and reducing the loss in the Zener diodes ZN2 and ZN3. Similarly, the sine wave voltage of the drive voltage VT12 for the second switching element Q2 can be lowered, thereby reducing the current flowing through the Zener diodes ZN4 and ZN5 and reducing the losses in the Zener diodes ZN4 and ZN5. Furthermore, the loss of the Zener diodes ZN2, ZN3, ZN4, and ZN5 is supplied via the positive feedback transformer T1, i.e., the coil L3, so the current flowing through the coil L3 can be reduced, which results in the positive feedback transformer T1 and the coil L3 being smaller and less expensive. Furthermore, the coil L3 is a component with adjustable inductance, and since the current can be reduced, it is easy to obtain.

[0052] Next, the results of a simulation performed to confirm the above-mentioned effects will be explained using the first switching element Q1 as an example. This simulation was performed using the circuit simulation software "SCALE" from Smart Energy Laboratory Co., Ltd., by inputting the constants of each element. First, FIG. 2 shows a diagram (a1) showing the change over time of the drive voltage V1 in this embodiment, and a diagram (b1) showing the change over time of the drain current Id of the first switching element Q1. FIG. 3 also shows a line (a2) showing the change over time of the drive voltage VT11 in the conventional example shown in FIG. 24, and a line (b2) showing the change over time of the drain current Id of the first switching element Q1.

[0053] In the present embodiment shown in Figure 2, as shown in diagram (a1), the drive voltage V1 rises and falls sharply near the zero crossing. This is because the bias voltage Vgs1 (Bias) from the capacitor C18 is added in series to the drive voltage VT11 from the secondary winding N1SA of the positive feedback transformer T1, which speeds up turn-on and turn-off. As a result, the peak voltage of the drive voltage VT11 from the secondary winding N1SA of the positive feedback transformer T1 can be reduced.

[0054] In contrast, in the case of the conventional example shown in Figure 3, as shown in line (a2), the drive voltage VT11 does not rise or fall suddenly near the zero crossing. Therefore, in order to speed up turn-on and turn-off near the zero crossing, the slope of the drive voltage VT11 must be made steeper, and the peak value must be made larger.

[0055] Fig. 4 is a diagram (c1) showing the change over time of the current Iz of the Zener diodes Z2 and Z3 in this embodiment. Fig. 5 is a diagram (c2) showing the change over time of the current Iz of the Zener diodes Z2 and Z3 in the conventional example shown in Fig. 24.

[0056] In the present embodiment, the Zener diode current Iz is kept low and loss is small, as shown in Fig. 4. In contrast, in the conventional example, the peak value of the drive voltage VT11 must be increased by the secondary winding N1SA, which results in a high Zener diode current Iz and large loss. Although the Zener diode current Iz can be reduced by increasing the gate drive resistance R3, this increases the loss in the first switching element Q1.

[0057] FIG. 6 shows a diagram (a1) showing the change over time of the drive voltage V1 in this embodiment, a diagram (b1) showing the change over time of the drain current Id of the first switching element Q1, and a diagram (d1) showing the change over time of the drain voltage relative to the source of the first switching element Q1. Figure 7 shows a line (a2) showing the change in drive voltage V1 over time in the conventional example shown in Figure 24, a line (b2) showing the change in drain current Id of the first switching element Q1 over time, and a line (d2) showing the change in drain voltage relative to the source of the first switching element Q1.

[0058] As shown in Figure 6, in the case of this embodiment, the drive voltage V1 drops and rises sharply near the zero crossing, so turn-on and turn-off are smooth, loss is small, and no disturbance occurs in the diagram (d1).In contrast, as shown in Figure 7, in the case of the conventional example, the drive voltage VT11 has a slope near the zero crossing, so high-speed turn-on and turn-off are not possible, and the diagram (d2) is disturbed.

[0059] Figure 8 shows a part of Figure 6 with the time axis shifted, and Figure 9 shows a part of Figure 7 with the time axis shifted. The difference can be seen more clearly.

[0060] Figure 10 shows a different part of Figure 6 with the time axis shifted, and Figure 11 shows a different part of Figure 7 with the time axis shifted. The differences can be seen more clearly.

[0061] 12 shows a diagram (a1) showing the change over time of the drive voltage V1 in this embodiment, a diagram (e1) showing the change over time of the drive voltage VT11 generated by the secondary winding N1SA of the positive feedback transformer T1, and a diagram (f1) showing the change over time of the bias voltage Vgs1 (Bias) generated by the capacitor C18. The drive voltage V1 is the sum of the drive voltage VT11 and the bias voltage Vgs1 (Bias). Specifically, the bias voltage Vgs1 (Bias) is approximately 1 V.

[0062] The simulation was performed using the first switching element Q1 side as an example, but it is believed that the same applies to the second switching element Q2 side.

[0063] Next, the results of an approximate analysis of the operation will be explained. First, in the circuit diagram shown in FIG. 1, the first switching element Q1 and the second switching element Q2 have the same components and component constants that drive them, and their operations are also the same except for the fact that they are out of phase with each other by 180°. Therefore, the first switching element Q1 will be used as an example for explanation. First, the bias voltage Vgs1 (Bias) from the capacitor C18 is a substantially constant DC voltage that does not change polarity and is not affected by the load of the ultrasonic transducer UT. Furthermore, the drive voltage of the gate G1 of the first switching element Q1 can be adjusted by the input power supply voltage Vin, the gate drive resistor R3, and the magnitude of the starting resistor R27. However, in order to make the bias voltage Vgs1 (Bias) approximately DC, the following conditions must be met. First, the time constant of the starting resistor R27 and capacitor C18 must be sufficiently large relative to the resonance period (switching period) of the ultrasonic transducer UT. Next, the time constant of the gate drive resistor R3 and capacitor C18 must be sufficiently large with respect to the resonance period (switching period) of the ultrasonic transducer UT.

[0064] Based on these assumptions, first, in the drive circuit for the first switching element Q1 in the drive circuit shown in Figure 1, equivalent circuits are obtained for each of the "on-off" and "off-on" states of the first switching element Q1 and the second switching element Q2. From each equivalent circuit, the bias voltage Vgs1(Bias)on and bias voltage Vgs1(Bias)off due to capacitor C18 are obtained, and the bias voltage Vgs1(Bias) is obtained using the principle of superposition.

[0065] First, the case where the first switching element Q1 is "on" and the second switching element Q2 is "off" will be described. 13 is an equivalent circuit diagram that shows only the secondary winding N1SA of the positive feedback transformer T1 and the drive circuit for the first switching element Q1. In this equivalent circuit diagram, D1-S1 of the first switching element Q1 is shorted (ON), D2-S2 of the second switching element Q2 is open (OFF), the drive circuit for the second switching element Q2 is deleted, and the entire secondary-side circuit for phase negative feedback from capacitor C3 to the power transformer T2 is deleted.

[0066] The equivalent circuit diagram in FIG. 14 is obtained by rotating and moving the components in the equivalent circuit shown in FIG. 13 and rearranging the circuit. In the circuit diagram of Figure 14, the waveform of the drive voltage VT11 from the secondary winding N1SA of the positive feedback transformer T1 is first approximated by an ideal waveform. The secondary winding N1SA of the positive feedback transformer T1 is also used as a voltage source. This waveform is approximated by only a positive sine half-wave VT11on (peak value VT11pk) when the first switching element Q1 is "on." Moreover, the Zener diodes ZN2 and ZN3 are approximated by only the positive square wave Vgs1on (peak value Vgs1pk) when the first switching element Q1 is "on."

[0067] Under these conditions, the bias voltage Vgs1 (Bias) becomes approximately DC, and the capacitor C18 can be considered open. Also, it is sufficient to consider only the drive voltage VT11 due to the secondary winding N1SA of the positive feedback transformer T1 and the DC component of the bias voltage Vgs1 (Bias), i.e., VT11onDC and Vgs1onDC.

[0068] VT11onDC and Vgs1onDC are expressed by the following formulas (I) and (II). VT11onDC=VT11pk / π ---(I) Vgs1onDC=Vgs1pk / 2 ---(II) Moreover, Vgs1(Bias)on is expressed by the following formula (III). Vgs1(Bias)on =R27·(-VT11onDC+Vgs1onDC) / (R27+R3) =R27·(Vgs1pk / 2-VT11pk / π) / (R3+R27) ---(III)

[0069] Furthermore, the following equations (IV) and (V) hold based on the values ​​shown in Figures 17, 18, and 9. Figures 17, 18, and 19 are diagrams in which the values ​​are entered for the "on / off" cases of the first switching element Q1 and the second switching element Q2. R27*C18=470Ω*10μF=4.7ms>>Resonance period 1 / 30kHz=36μs---(IV) This time constant is sufficiently longer than the resonant period (switching period) 1 / 30 kHz = 36 μs. R3*C18=68Ω*10μF=680μs>>Resonance period 1 / 30kHz=36μs ---(V) This time constant is sufficiently longer than the resonant period (switching period) 1 / 30 kHz = 36 μs. In this way, the two conditions that are met are: "The time constant of the starting resistor R27 and capacitor C18 must be sufficiently large relative to the resonance period (switching period) of the ultrasonic vibrator UT" and "The time constant of the gate drive resistor R3 and capacitor C18 must be sufficiently large relative to the resonance period (switching period) of the ultrasonic vibrator UT." Therefore, Vgs1(Bias)on becomes the value shown in the following formula (VI) from formula (III). Vgs1(Bias)on=-1.749V---(VI)

[0070] Next, the case where the first switching element Q1 is "off" and the second switching element Q2 is "on" will be described. Figure 15 shows an equivalent circuit that retains only the secondary winding N1SA of the positive feedback transformer T1 and the drive circuit for the first switching element Q1. D1-S1 of the first switching element Q1 is open (off), and D2-S2 of the second switching element Q2 is short-circuited (on). In addition, the drive circuit for the second switching element Q2 has been deleted, and the entire secondary-side circuit for phase negative feedback from capacitor C3 to the power transformer T2 has been deleted.

[0071] The equivalent circuit diagram of FIG. 16 is obtained by rotating and moving the components in the equivalent circuit shown in FIG. 15 and rearranging the circuit. In the circuit diagram shown in Figure 16, first, the waveform of the drive voltage VT11 from the secondary winding N1SA of the positive feedback transformer T1 is approximated by an ideal waveform. The secondary winding N1SA of the positive feedback transformer T1 is used as the voltage source. This waveform is approximated by only the negative sine half-wave VT11off (peak value VT11pk) when the first switching element Q1 is "off." Moreover, the Zener diodes ZN2 and ZN3 are approximated by only the negative square wave Vgs1off (peak value Vgs1pk) when the first switching element Q1 is "off."

[0072] Under these conditions, the bias DC voltage Vgs1 (Bias) becomes almost DC, and the capacitor C18 can be considered open. Also, the drive voltage VT11 due to the secondary winding N1SA of the positive feedback transformer T1 and the DC component of the bias DC voltage Vgs1 (Bias), i.e., VT11offDC and Vgs1offDC, can be considered.

[0073] VT11offDC and Vgs1offDC are expressed by the following formulas (VII) and (VIII). VT11offDC=-VT11pk / π ---(VII) Vgs1offDC=-Vgs1pk / 2 ---(VIII) Therefore, Vgs1(Bias)off is expressed by the following equation (IX). Vgs1(Bias)off =R27·(-VT11offDC+Vgs1offDC-VinoffDC) / (R27+R3)+VinoffDC =R27·(-Vin / 2+(VT11pk / π-Vgs1pk / 2)) / (R3+R27)+Vin / 2 ---(IX)

[0074] Furthermore, the following equations (X) and (X1) are established based on the values ​​shown in Figures 20, 21, and 22. Figures 20, 21, and 22 are diagrams in which the values ​​are entered for the "off / on" cases of the first switching element Q1 and the second switching element Q2. R27*C18=470Ω*10μF=4.7ms>>Resonance period 1 / 30kHz=36μs---(X) This time constant is sufficiently longer than the resonant period (switching period) 1 / 30 kHz = 36 μs. R3*C18=68Ω*10μF=680μs>>Resonance period 1 / 30kHz=36μs ---(XI) This time constant is sufficiently longer than the resonant period (switching period) 1 / 30 kHz = 36 μs. In this way, the two conditions that are met are: "The time constant of the starting resistor R27 and capacitor C18 must be sufficiently large relative to the resonance period (switching period) of the ultrasonic vibrator UT" and "The time constant of the gate drive resistor R3 and capacitor C18 must be sufficiently large relative to the resonance period (switching period) of the ultrasonic vibrator UT." Therefore, Vgs1(Bias)·off is calculated from equation (III) to obtain the value shown in the following equation (XII). Vgs1(Bias)·off=2.697V---(XII)

[0075] Next, Vgs1(Bias) is expressed by the following formula (XIII). Vgs1(Bias) =Vgs1(Bias)·on+Vgs1(Bias)·off =-1.749V+2.697V =0.948V―――(XIII) This result is consistent with the simulation result, i.e., approximately 1V.

[0076] The present invention is not limited to the above-described embodiment, and the illustrated configuration is merely an example. First, in the above embodiment, a Si-MOSFET is used as an example of a voltage-driven transistor, but the present invention is not limited to this and other voltage-driven transistors may also be used. Additionally, the illustrated configuration is merely an example. [Industrial Applicability]

[0077] The present invention relates to an ultrasonic vibrator drive circuit, and in particular to an ultrasonic vibrator drive circuit using a voltage-driven transistor that is devised to enable faster turn-on and turn-off, and is suitable for ultrasonic vibrator drive circuits incorporated in vacuum cleaners, for example. [Explanation of symbols]

[0078] 1 Series Circuit 3 Oscillator Circuit 5 Voltage detection section 7 Phase adjustment section L2 choke coil UT ultrasonic transducer Q1 First switching element Q2 Second switching element T1 Positive feedback transformer N1P Primary winding of positive feedback transformer N1SA Positive feedback transformer secondary winding N1SB Positive feedback transformer secondary winding T2 Power transformer N2P power transformer primary winding N2S Power transformer secondary winding C18 capacitor C19 capacitor ZN2 Zener diode ZN3 Zener diode ZN4 Zener diode ZN5 Zener diode R3 resistance R4 resistance

Claims

1. a series circuit of an ultrasonic vibrator and a choke coil; an oscillation circuit that applies a resonance frequency of the ultrasonic wave oscillator to the series circuit to drive the ultrasonic wave oscillator; Equipped with The above oscillator circuit is A power transformer; a first switching element and a second switching element for intermittently energizing a primary winding of the power transformer; a voltage detection unit that detects a voltage between both ends of the vibrator; a phase adjustment unit that adjusts the phase so that the voltage detected by the voltage detection unit is fed back positively; a positive feedback transformer that positively feeds back the voltage whose phase has been adjusted by the phase adjustment unit to the first switching element and the second switching element; It is equipped with The first switching element and the second switching element use voltage-driven transistors each having a gate, a drain, and a source, the positive feedback transformer is composed of a primary winding, a secondary winding for the first switching element, and a secondary winding for the second switching element, a capacitor is arranged in series between the secondary winding for the first switching element and the source of the first switching element, and a capacitor is also arranged in series between the secondary winding for the second switching element and the source of the second switching element, As a result, the gate bias voltage generated by the capacitor is added in series to the drive voltage generated by the secondary winding for the first switching element, and the gate bias voltage generated by the capacitor is added in series to the drive voltage generated by the secondary winding for the second switching element.

2. 2. The ultrasonic transducer drive circuit according to claim 1, an ultrasonic vibrator drive circuit, characterized in that a resistor is arranged in series between a secondary winding for the first switching element and a gate of the first switching element, and a resistor is also arranged in series between a secondary winding for the second switching element and a gate of the second switching element.

3. 3. The ultrasonic transducer drive circuit according to claim 2, a series circuit of two Zener diodes is connected in parallel to a series circuit of the secondary winding for the first switching element, the capacitor, and the resistor; 10. An ultrasonic transducer drive circuit, comprising: a secondary winding for said second switching element; a series circuit of two Zener diodes connected in parallel to said series circuit of said capacitor and said resistor;

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