Digital carrier transmitter for high-speed radio

Abstract

To use a non-resonance method replacing modulated carrier wave radio to achieve narrowband, 1 Mbit per second digital radio communication in the short-wave bands, and faster communication at high-frequency bands.SOLUTION: In a communication system, a square wave input 310 from a frequency generator triggers a high-speed D flip-flop 320, and the flip-flop 320 has an output that switches a high-speed analog switch 330 at zero crossings. Thus, a transmitter enables the selection of a perfect and clear sine wave. The selected sine wave from the switch 330 is fed to a three-transistor linear amplifier 340. The output stage of the amplifier 340 consists of a high-voltage MOSFET 350, whose drain output pin is connected directly to a non-resonant antenna 360.SELECTED DRAWING: Figure 3

Description

[Technical Field]

[0001] The present invention relates to a digital carrier transmitter for high speed radio as an alternative to modulated carrier radio. [Background technology]

[0002] Modern communications encode and decode signals by modulating and demodulating a carrier wave. This basic technology has come a long way since the spark transmitter designs of 120 years ago. Carrier modulation is now used in everything from commercial broadcasting and satellite communications to the physical layer of the internet and even SETI (the search for extraterrestrial intelligence). These modes of communication require many repeated waves to encode a small amount of information.

[0003] Unfortunately, using carrier modulation to embed information in sidebands limits communication speeds. For example, amplitude shift keying ("ASK") switches a carrier on and off (or up and down). This digital control is severely limited by the availability of sidebands and is used for low-speed digital communications. Because of these complexities, ASK is, as the art has noted, "very bandwidth inefficient" and "unsuitable for high bit-rate data transmission." See, for example, https: / / www.watelectronics.com / what-is-an-amplitude-shift-key

[0004] These communication technologies are based on the pre-quantum physics of waves, rather than manipulating electromagnetic energy as discrete groups of photons. Thus, radio transmitters generate the same wave repeatedly and impress a signal onto it through modulation. In fact, radio wave energy is synonymous with "radio waves" in our lexicon.

[0005] This focus on waves is enhanced by the use of the Fourier transform, where the Fourier transform requires multiple waves to encode and decode each bit of information. Fourier transform theory teaches that the bandwidth of a signal increases as the number of modulating waves in a wave train decreases. As the length of the wave train approaches one wavelength, the electromagnetic signal expands to infinite bandwidth, making it unusable for communication.

[0006] Long Wave Trains Have Limitations Early pioneers of radio technology invented frequency filters to select and purify signals from newly discovered long, resonant wave trains. They discovered that in the process of filtering waveforms with tuned circuits, the filters generated transients that corrupted the first and last waves of the waveform's modulation changes (signals). This signal degradation occurs because resonant devices like filters temporarily store energy from each wave of the signal. The stored energy from each wave is later returned to the circuit by the filter. Thus, as the filter responds to information from signal changes, it must repeatedly ring up and down to a steady state, delaying communication. This renders the first and last waves useless each time the signal's wave train changes modulation.

[0007] The transient response of filters used in radios causes signal broadening and delay The problem of transient distortion due to resonance was recognized by early radio inventors. A. J. Starr, a research engineer for Marconi's wireless telegraphy in the 1930s, wrote in his textbook "The Problem of Transient Distortion" published by Sir Isaac Pitman & Sons, 1934, 1938: Electrical circuits and wave filters He explains this in his book "The Characteristics of a Filter." He points out that "the filter's properties are completely lost during transient conditions." Furthermore, such transient conditions can last for some time, as "many cycles pass through it before the current settles down to a very small steady-state value" (pp. 352-353).

[0008] D.G. Tucker in 1946 wrote in his book " Filter Transient Response" summarizes two "rules of thumb" on this topic. One is: "The rise time of a band-pass filter for an applied signal at a mid-band frequency is equal to the inverse of its bandwidth in cycles per second. For a low-pass filter, the build-up time of an applied DC signal is one-half the inverse of its cutoff frequency in cycles per second." The other is: "The peak amplitude of the transient produced by the sudden application or removal of a frequency outside the passband of a band-pass filter is proportional to its bandwidth and inversely proportional to the difference between the applied frequency and the mid-band frequency."

[0009] Various resonant filters have been invented, trading off frequency discrimination for smoothness of response to signal transients. Figure 1 compares the responses of three common filters: Bessel 10, Butterworth 20, and Chebyshev 30. The responses shown in this figure are normalized to a 1 Hz frequency cutoff on the x-axis at 40. Comparing these three filters in a second plot shows that signal discrimination (filter power) improves as transient behavior gets progressively worse. The steepest filter, Chebyshev 30, causes the greatest delay (plot 31), while progressively less steep filters (plots 21 and 11) cause less delay, as shown in the graph below. These prior art graphs show how all filters absorb energy from signal transitions passing through them and later remove it, with some smearing. Also, "All filters have time delay, this is an unavoidable truth." (Published by the American Radio Relay League) Experimental methods in RF design 3.32 2012 page.

[0010] Even so-called "pulse" communications, such as the "chirps" used in satellite communications, are actually wave trains of many cycles with distorted beginning and ending regions. See Figure 2, which shows the time response of a "pulse" transmission 210 used in satellite communications. This is often called a "chirp" because many photon cycles are combined in a time sequence, varying in frequency and bandwidth. The "pulse" bandwidth 220 in this example is approximately 15 MHz. The concept of a "pulse" in wireless communications, consisting of one or a few sinusoidal waves of energy, is unfamiliar to many practitioners.

[0011] Carrier modulation techniques utilize wave theory, where a signal frequency is mixed with a "carrier" wave train to create another wave train. This creates two sidebands of multiple frequencies that encode the signal information. The carrier is often removed, and one of the redundant sidebands is also removed. Unfortunately, as more information is added to a transmission, the sidebands occupy increasingly wider frequency bandwidths.

[0012] The century-old heterodyne technique, based on wave theory, does not take advantage of the new quantum physics perspective that considers individual photons. In fact, many practitioners do not even believe that photons are involved in wireless communication.

[0013] This application claims priority from U.S. Patent No. 63,638,444, filed April 25, 2024, entitled "Non-Resonant Wireless Transmitter for High-Speed ​​Communications," and U.S. Patent No. 63,640,237, filed April 30, 2024, entitled "Controlled Non-Resonant Wireless Reception of Photon Pulses," the contents of which are incorporated by reference. [Prior art documents] [Patent documents]

[0014] [Patent Document 1] U.S. Patent Application No. 2005 / 0201490 [Brief explanation of the drawings]

[0015] [Figure 1]Figure 1 is a comparison of three prior art filters. [Figure 2] Figure 2 shows the envelope shape of a typical pulse transmission. [Figure 3] FIG. 3 is a block diagram of an embodiment of a digital carrier wave. [Figure 4] FIG. 4 is a schematic diagram of an embodiment of a digital carrier wave. [Figure 5] FIG. 5 is a schematic diagram of a communication system using a digital carrier embodiment. [Figure 6] FIG. 6 is a diagram showing transmitted pulses and received pulses from a 4 MHz communication system. [Figure 7] FIG. 7 is a diagram showing a transmission pulse and a reception pulse in a 10 MHz communication system. [Figure 8] FIG. 8 is a block diagram of one embodiment of a transmitter. [Figure 9] FIG. 9 is a schematic embodiment of a harmonic rejection circuit. [Figure 10] FIG. 10 shows two block diagrams of an embodiment of sideband rejection. [Figure 11] FIG. 11 is a schematic diagram of an embodiment of sideband removal. [Figure 12] FIG. 12 shows data obtained from a sideband removal embodiment. [Figure 13] FIG. 13 shows an example of creating a digital carrier wave with sidebands using the above described apparatus. DETAILED DESCRIPTION OF THE INVENTION

[0016] A digital carrier wave transmitter can generate individual single sine waves, each one wavelength long, or half-sine wave pulses of photons instead of waves. The transmitter can group one or more photon pulse sine waves into a transmitted digital signal. For example, a one photon duration (i.e., one sine wave long) pulse from a 10-watt transmitter operating at 10 MHz (30-meter wavelength band) will generate approximately 1.5x10 pulses within the 100 ns time period of one wavelength. 20Contains 1.5x10 identical photons 20 Each pulse of photons is ideally perceived by the receiver as one bit of digital information.

[0017] Ideally, a distant antenna would intercept enough 10 MHz (30 m long) photons to generate a one wavelength (100 ns) bit pulse at the receiver. This was demonstrated by energizing metallic bond valence electrons on the antenna surface with partial or full-wave DC pulses. An oscilloscope attached to a non-resonant beverage antenna 3.5 wavelengths away detected enough photons to form a strong 15 mV pulse without filtering. See https: / / vixra.org / abs / 2310.0123.

[0018] Unfortunately, no receiver exists that conveniently detects a long sinusoidal pulse as a single bit of information. Without being bound by any theory, it is believed that a resonant device such as a filter or impedance matcher absorbs each pulse and later re-emits the absorbed energy by collapsing the wave function. The absorbed energy is emitted as a substantially identical wave function, but at integer multiples of the wave time. This complicates the separation of the individual photon pulses.

[0019] As a result, each frequency filter in the radio confuses the beginning and end of the modulation transition. However, by carefully selecting the minimum amount of resonance, a compromise hybrid radio can be created that minimizes resonance while still detecting bit signals as short as 3, 5, or 10 cycles. Preferably, the added resonance is at or near the critical resonance of the resonator to prevent ringing and minimize signal disturbances. Preferably, the resonator, such as a quartz crystal, an inductor-capacitor combination, or a surface acoustic wave filter, is configured to critically resonate by adding a resistor. In this embodiment, "at or near critical resonance" refers to a Q factor between 0.1 and 10, more preferably between 0.25 and 1. Preferably, a resistor is added in series with the quartz crystal or the inductor of the inductor-capacitor pair to prevent ringing and excessive resonance.

[0020] For an LC resonator, the reactance (Ω) can be calculated for a selected frequency. Any resistance added in series should be within five times this value, preferably within two times, and even more preferably less. While it may be difficult to calculate or measure the reactance of a given crystal resonator at a given frequency, for a typical AT-cut crystal, the added resistance is preferably 100-1000 ohms. Most preferably, this resistance is varied by a circuit that serves the need to add the minimum amount of crystal resonance necessary to remove the carrier signal component.

[0021] In one embodiment, the digital carrier transmitter and receiver are capable of sensing pulses, operating with continuous photon train segments of three photon sine wavelengths or less. In another embodiment, the length of the photon train is six sine waves or less, and in another embodiment, the length of the photon train is set to 11 sine waves or less, or a number between 10 and 100. It is best if the tuning circuit used to sense the continuous photon train has adjustable attenuation and an adjustable variable resistance.

[0022] As an example, a 4 megahertz transmission of a digital carrier with five cycles (i.e., a string length of five consecutive sine waves per bit) allows communication at this low frequency up to 800 kilobits per second. A 10 megahertz transmission with 10 cycles per bit similarly allows communication at 1 megabit per second. By allowing each bit to occupy multiple cycles, as in these examples, conventional tuned circuits can be operated near or at critical resonance, at the expense of the beginning and end of each bit signal.

[0023] One consideration is to monitor the relationship between bit size and the strength of the tuning or filter circuit, providing sufficient selectivity to obtain a readable signal. In one embodiment, the degree of resonance is sensed and a signal output corresponding to the ringing or amount of resonant energy in the circuit is controlled, either in strength or other quality, such as phase. The output signal triggers a resonance adjuster to increase the resistance of the resonant filter. In response to higher resonance or ringing levels, this feedback circuit reduces resonance by adding resistance to the resonant device, such as an LC (inductor + capacitor of equal impedance at the desired filter resonant frequency) filter or a crystal. For example, a JFET with its source and drain in series with the inductor or crystal can be activated, increasing its resistance in response to a signal that increases resonance. By increasing the resistance, the JFET increases the damping of the filter resonator, lowering the resonance.

[0024] Digital carrier wave transmission circuit A new digital carrier transmitter has been discovered that can transmit small multiples of successive sinusoidal photon cycles. Groups of bits of 5, 10, 50, or 100 cycles in length are transmitted at frequencies up to 10 megahertz. The transmitter has a 128-bit shift register, which can be manually programmed by the user to repeatedly transmit messages up to 128 bits long. Higher frequencies can be achieved by selecting faster chips. The response of the shift register and divide-by-10 counter chip used was limited to approximately 10 MHz.

[0025] No transmission line was used. A nonresonant antenna was attached directly to the output of the nonresonant transmitter's final stage. The antenna's resistance was adjusted to eliminate any slight residual resonance. For both the Beverage and "TEFV" (Terminal Fed Vertical) antennas, this was approximately 300 ohms. Nonresonant antennas have high impedance, typically 250-400 ohms, which, perhaps not coincidentally, resembles the natural impedance of free space, which is 377 ohms. In contrast, a 50 ohm transmitter / antenna system accepts a low-voltage, low-impedance input and resonates to a higher voltage during operation. Therefore, a transmitter directly matched to a high-impedance (70 ohms or greater) antenna must have a higher-voltage output stage to transmit the same power without benefiting from voltage-varying resonance.

[0026] High-voltage MOSFETs were selected for the transmitter output stage. Finally, linear amplification was implemented to more completely eliminate resonances. In this circuit, analog switches were used to select the desired portion of the square-wave reference signal train. By turning on the switches that controlled the passage of the sine-wave train, square waves of different durations were selected to select the desired signal duration (number of sine waves in the signal train) and timing. The selected portion was amplified in a linear amplifier stage without resonant coupling between stages.

[0027] Digital carrier transmitter for short pulse trains One embodiment of a non-resonant digital carrier transmitter consists of a sine wave oscillator input and a synchronous square wave input. See the block diagram in Figure 3 and the circuit diagram in Figure 4. A square wave input 310 from a frequency generator triggers a high-speed D flip-flop 320, which has an output that toggles a high-speed analog switch 330 at its zero crossings, thus allowing for the selection of a clean, unambiguous sine wave. The selected sine wave from switch 330 is fed to a three-transistor linear amplifier 340. The output stage of amplifier 340 consists of a high-voltage MOSFET 350, whose drain output pin is directly connected to a non-resonant antenna 360. The antenna / MOSFET, connected in series with a 300 volt power supply (not shown), presents a DC circuit path for electronic energy traveling through the antenna in only one direction. A version of this circuit, used to discover DC generation of radio waves for single-photon pulse detection, was previously described in U.S. Ser. No. 63 / 530,982, filed August 6, 2023, entitled "Digital Radio." The entire contents of this previously filed patent application, particularly the wireless circuits, drawings and concepts therein, are incorporated herein by reference.

[0028] The signal generator also supplied a synchronized sine wave 370, shown in Figure 3, at approximately 1 volt into 50 Ω. A square wave 310 synchronized with the sine wave triggers a flip-flop 320, which outputs a change in the zero-crossing data by turning a high-speed analog switch 330 on and off at the appropriate times. A separate square wave was convenient here because its phase and duty cycle could be easily adjusted to compensate for slight timing delays due to wiring inductance, transistor switching times, etc.

[0029] The square wave 310 also triggers a division by a 5, 10, 50, 100 counter 375, which also triggers a D flip-flop 320 at the zero crossings of the sine wave. The flip-flop 320 is triggered at a frequency 5, 10, 50, 100 times slower than the reference frequency, allowing 5, 10, 50, 100 sine long pulses to enter the amplifier chain 340. The preferred embodiment includes a single oscillator as a signal reference from which to prepare the synchronized square and sine waves.

[0030] The amplifier chain 340 in this embodiment is composed of Class A biased transistors. The first two transistors are bipolar low impedance transistors in a voltage follower configuration, providing a large amplification current to the gate of a high-voltage MOSFET 350. The drain of the MOSFET 350 is directly connected to a non-resonant antenna 360, which is driven by a 300 V DC power supply (not shown). To minimize resonant degradation of the signal pulse, no inductors or resonant filters are used in the signal path.

[0031] The D flip-flop 320 acts as a latch for the input data. In the example shown, a data input device 380, such as a keyer or computer output, is fed into a shift register 390. The shift register 390 is clocked by a time base divider 375, which determines the number of sine wave cycles allowed for each data bit. In this way, a Morse code keyer or other data stream can be input at a much slower rate than a square wave. In this way, the keyer, computer, or other memory device can control how many integral sine waves are sent with each data bit. A switchable endless loop option 392 allows for data reuse.

[0032] Figure 4 shows a circuit diagram of an embodiment. The upper right portion 410 is an amplifier chain. The upper left portion 420 comprises flip-flops and analog switches. The lower left portion 430 shows a time divider that determines the bit rate. The lower right portion 440 shows the connections of a shift register. Preferably, the shift register or other data buffer is interfaced to a computer via a serial or parallel interface.

[0033] The transmitter embodiments shown in Figures 3 and 4 were used in the communications system shown in Figure 5. Figure 5 shows, at the top left, the transmitter 510 DC coupled (i.e., without a coupling capacitor) to a non-resonant antenna 520. The antenna 520 may have a termination resistor 525 to limit reflections at the antenna.

[0034] The signal from this transmitter preferably radiates from the long axis of the non-resonant antenna 520 toward the end of the termination resistor 525, as depicted in Figure 5. Each bit of the communication signal in this embodiment consists of three sine waves, as shown by waveform 530. This three-sine energy travels toward another non-resonant antenna 540, which is connected to a receiver 550. Many non-resonant antennas have a termination resistor 560 located at the end of the long axis of such antenna. When using this embodiment, the resistor preferably has a resistance value of 100-2000 ohms, more preferably 200-500 ohms.

[0035] In high frequency embodiments above 500 MHz, the transmitter is preferably connected to a non-resonant diamond antenna. Receiver 550 receives signal energy from the antenna, preferably in the form of induced electrical sinusoidal pulses containing each of the three sinusoids of the originally transmitted signal. In one embodiment, half-sine signals are generated and received. In yet another embodiment, a positive-going half-sine wave is mixed with a negative-going half-sine wave to form a communication data stream of alternating ones and zeros. In the embodiment of FIG. 5, multiple sinusoids, such as the 3 shown here representing a bit, are detected as a single pulse via appropriate circuitry. For example, an AC-DC high-frequency circuit with fast diodes can rectify the multiple sinusoidal pulses into a DC pulse of appropriate length by selecting appropriate RC time constants for the components.

[0036] Non-resonant DC energy proportional to antenna loading The transmitter described in Figures 3 and 4 was used at 2-28MHz, with end-fed vee antennas (TEFVs) as the non-resonant antennas for both the transmitter and receiver. Each TEFV was 40 meters long and 10 meters maximum height and was constructed as described at https: / / vu3dxr.in / diy-terminated-end-fed-vee-antenna-tefv / . The first transmissions were performed using an oscilloscope attached to the receiving antenna. In later experiments with multiple photon sine wave transmission, single-tuned and double-tuned receivers were constructed and attached directly to the non-resonant receiving antenna. [Example]

[0037] Transmission of short sinusoidal photon pulse trains Early experiments (https: / / vixra.org / abs / 2310.0123 The first study (reported in "Direct Current Generation of Radio Photons from an Antenna") was done with a version of the transmitter that sent only one long sinusoidal pulse as a bit. In that study, an oscilloscope was used to detect the long single-photon pulse from the transmitter without worrying about sidebands, which was not practical.

[0038] To develop and test a more practical system, a compromise transmitter was created, using a short repeating sine wave to allow for limited resonance with a more flexible receiver. A resonant filter added to the receiver separated the transmitted signal from background signals, but destroyed the single sine wave signal.

[0039] The compromise system described here consists of a transmitter that transmits (for example) 5, 10, 50, or 100 successive photon sine wave pulses. The receiver can include a small amount of resonance to filter the transmitted signal, sacrificing one or a few photon sine cycles per bit. For example, bits five photon cycles long could be created at a frequency of 4 MHz (75 meters) at speeds up to 800 kbits per second.

[0040] The receiver was a compromise hybrid, combining an "LC" (inductor-capacitor resonant pair) legacy resonant filter with a linear amplifier chain. The receiver amplified and purified the signal from a non-resonant antenna. In other words, increasing the multiplexed bit length was traded off for accepting a somewhat limited filter resonance in order to obtain sufficient filtering of the received signal. This combination allowed digital signal transmission at 800 kilobits per second on the ham bands below 75 meters and 1 millibit per second on the 30-meter ham band. The receiver is described in pending patent application 63 / 640,237. Generally, the receiver consisted of two linear amplifier stages with one or two tuned circuits, each with an adjustable resistor in the LC circuit, to limit resonant ringing in the received signal by lowering the filter Q.

[0041] Transmission Bandwidth The signal input to the gate transistor of the output amplifier and the amplified power signal output from this output transistor to the antenna were examined with a high-speed oscilloscope, but no sidebands were observed. However, a frequency scan indicated the presence of sidebands. However, these sidebands could be reduced by the carrier signal comparison procedure exemplified in Example 3.

[0042] 10MHz transmission results A 10 MHz sine wave and a synchronized 10 MHz square wave were input to the transmitter. The MOSFET gate was adjusted to 4.4 volts. The signal output at the gate and the amplified and sampled signal at the antenna connection are shown in two diagrams in Figure 6. The top scope plot in Figure 6 shows signal 610 applied to the MOSFET gate. Signal 610 consists of 10 sine-wave alternating bit (1s and 0s) signals at approximately 0.8 volts. The wave peaks are spaced 100 ns apart along the x-axis. The y-axis shows relative signal strength. The bottom scope plot shows the resulting signal 620 at the antenna connection to the transmitter. Signal 620 consists of 10 long sine wave distortions at the antenna, with each sine clearly distinguishable. This distortion is due to the antenna's resonance and inductive characteristics and was minimized by adjusting the antenna terminal resistance to ground to eliminate the resonance.

[0043] Figure 7 shows a scope plot of the corresponding signal 710 recovered at the receiver. The x-axis is time, with major divisions at 500-nanosecond intervals. The y-axis is linear signal strength. The receiver is mounted on a non-resonant TEFV antenna. The received signal underwent two stages of Class A transistor amplification and one stage of simple LC filtering. As can be seen in this plot, the early cycles of each train of signal 710 lose some energy to the filter, and each 10-cycle train contains several trailing cycles, as expected, due to the filter contributing stored energy to the circuit. The scope plot of signal 710 shows three complete 01 bit trains 720, 730, and 740. The three consecutive 10-cycle long bits are easily distinguishable, separated by a 10-cycle long blank period. Each cycle is 100 ns long. This demonstrates that a 10 MHz transmitter with a simple LC receiver system can communicate at 1 mbit / s. [Example]

[0044] Eliminate harmonic impurities without resonant filtering of the signal No matter what kind of transmitter it is, the transmitted signal must be filtered out of its unwanted harmonics. Passing the signal through one or more resonant filters is standard procedure in radio technology to filter out harmonics. This is an absolute requirement for all existing commercial equipment. For recently manufactured amateur radio transmitters, FCC regulations require that harmonics from the transmitted signal be at least 43 dB below the transmitted fundamental frequency.

[0045] Most of the contaminating harmonic energy in a signal chain is found in the second and third harmonics. A digital subtraction technique has been discovered that can remove most of the energy contained in these harmonics, bringing their levels within the required range, without introducing resonance. In this embodiment, the signal is sampled early in the transmitter, at least before the final power output stage. One or more harmonics are removed from the sampled signal and then amplified. The amplified harmonics are added 180 degrees out of phase with the harmonics of the signal downstream from the sampled location, but before the reference sinusoidal train is chopped to incorporate the bit information.

[0046] In a preferred embodiment, the amplified reference signal is sampled after one or more stages of amplification. One or more harmonics are purified and returned by subtracting or adding their opposite phases, with the amount of returned energy established by sensing the harmonic contamination after the last stage, preferably at the antenna connection itself. This allows for the correction of contaminating harmonics by removing their effects before they are generated, since a correction signal to remove the contaminating harmonics (i.e., negative harmonics) is added before the harmonic contamination occurs. This means that at some stage in the transmitter, the signal contains a small amount of harmonic contamination energy that is 180 degrees out of phase with the normal contaminating harmonics. This proactive approach to anticipated harmonic contamination is effective because the fundamental frequency itself generates harmonics.

[0047] In one embodiment, as depicted in the block diagram of Figure 8, a reference signal 810 is sampled early in the signal chain before being modified into a short digital segment via a high-speed analog switch 820. This modified signal is amplified by a current amplifier 830, a voltage amplifier 840, and sent to an antenna 850 as previously described with reference to Figure 3.

[0048] In an embodiment, harmonic contaminants are removed by sampling reference signal 810 at an early stage. The sample passes through harmonic filter 820, which selects either the second or third harmonic. A controlled amount of the selected harmonic is returned via path 830 as an inverted waveform to form signal 840 before chopping into data bits. This allows for the removal of harmonics, including those generated later in the transmit signal path. Amplifiers 832 and 834 buffer and isolate signal 810 from the harmonics, preventing their interaction. Preferably, amplifier 836 isolates the sampled reference signal from the composite modified signal 840. Flip-flop 860 processes the data flow as previously described.

[0049] The harmonic filter in this embodiment is a bandpass type with a variable capacitor for tuning. The filtered signal is amplified and added to the normal signal 810 to form the modified signal 840, which is then split into digital segments. Figure 9 shows a more detailed circuit diagram of a typical bandpass filter. This filter design, along with the listed values ​​of the inductors and capacitors, was taken from qrp-labs. See https: / / qrp-labs.com / images / bpfkit / bpf3.pdf.

[0050] The two tunable capacitors 910 and 920 of the four-element bandpass filter 925 shown on the left side of Figure 9 were adjusted so that the generated harmonics were 180 degrees out of phase with the harmonic signals contaminating the signal chain. The potentiometer 930 on the lower right side of Figure 9 adjusted the correction signal strength by setting the amount of out-of-phase harmonic energy that was fed back into the signal chain. This feeding back of the out-of-phase harmonics into the sinusoidal signal occurred prior to selecting the individual bit segments in the square wave.

[0051] The phase adjustment capacitors 910 and 920 and signal strength potentiometer 930 were adjusted to minimize harmonics perceived at the antenna mounting point. Using the oscilloscope's Fourier function, the ratio of harmonic to fundamental energy was measured while making these adjustments on a continuous (unbroken) carrier test signal. These controls were adjusted to minimize the ratio of harmonics to fundamental energy seen on the scope. This is best done at 100% duty cycle (sine signal 100% on with no blank periods). This allows for compensation (removal) of harmonic distortion generated by all amplifier stages. Adding the inverse of the second harmonic at the beginning of the amplifier chain allows for compensation (removal) of harmonic distortion generated by all amplifier stages while also detecting the purity of the result after all amplification steps.

[0052] A second bandpass filter similarly samples and amplifies the third harmonic, allowing it to be subtracted from the sine signal chain. With proper care in Class A amplifier design, the higher harmonics, 4th, 5th, etc., do not require further purification, so two harmonic filter subtraction circuits can be used to achieve the required output signal purity.

[0053] In practice, a small amount of filtering may be necessary to clean up the signal. It has been found that with a well-tuned amplifier bias and low power settings, such as 5 watts, this transmitter can transmit signals of acceptable harmonic quality without harmonic control. In one embodiment, a small amount of resonance is used to further purify the transmitted signal by feeding the signal into a resonant antenna or by adding an LC filter before the antenna. The latter is particularly useful when transmitting short bit segments of 5, 10, 25, 100, or more sine cycles per bit at a time, but not when transmitting only a single sine signal per bit.

[0054] This technique, using the circuit of Figure 9, was applied to the 8 MHz second harmonic of a 4 MHz signal, reducing the second harmonic from 35 db below the fundamental to 43 db below. [Example]

[0055] Eliminating sidebands while avoiding resonant smearing of digital carrier signals Digital carrier wave technology encodes information onto a carrier wave and does not use sidebands for communication. To eliminate sidebands, many technologies have proposed the use of resonant filters. However, such filters have been found to destroy the integrity of the carrier signal, even when their passband or cutoff frequency is far from the carrier frequency. This has prevented the use of conventional filters to eliminate sidebands.

[0056] An indirect subtraction technique has now been discovered that removes the sideband energy without destroying the integrity of the carrier. Two embodiments are shown in Figure 10. The top half of Figure 10 is a block diagram illustrating an input signal 1010 consisting of a strong carrier with strong, wide sidebands. The input signal 1010 enters two paths. In the upper path 1020, the input signal is amplified by amplifier 1030, which outputs a buffered signal to amplifier 1050. This buffered output signal is electrically connected to crystal 1040, which removes resonant energy from the carrier frequency portion of the signal passing through amplifier 1050. Crystal 1040 removes a portion of the carrier frequency, leaving a usable sideband-enhanced signal for amplifier 1050. Amplifier 1050 passes the buffered, reduced carrier signal from upper path 1020 to signal comparator 1070.

[0057] Meanwhile, signal 1010 also enters lower path 1025 and is amplified by amplifier 1035. Amplifier 1035 passes the buffered signal from lower path 1025 to signal comparator 1070. Signal comparator 1070 compares the signals produced by the two paths and outputs carrier-enriched (sideband-removed) signal 1090. In this embodiment, signal comparator 1070 subtracts the upper path signal from the lower path signal.

[0058] The upper signal path produces a carrier-attenuated signal that is in phase with the original signal 1010 because each amplifier 1030 and 1050 inverts the signal by 180 degrees. However, the lower signal path 1025 inverts the original signal 1010 by 180 degrees only once via amplifier 1035. When added together, their common waveform patterns cancel out.

[0059] The bottom half of Figure 10 is a block diagram that removes the carrier energy with the help of a crystal 1042, and then also removes the sidebands by adding them back to the signal. As with the top half, the input signal 1012 consists of the same strong carrier with strong, wide sidebands.

[0060] Input signal 1012 enters two paths. In the upper path 1022, the input signal is amplified by amplifier 1032, which outputs a buffered signal to amplifier 1052. Crystal 1042 is connected to the negative feedback of amplifier 1052. In an inverting transistor amplifier embodiment (e.g., common-emitter configuration), the crystal is connected between the collector and emitter (or drain and gate), creating negative feedback at the crystal's resonant frequency. This selectively reduces the amplification of the carrier wave by amplifier 1052. Amplifier 1052 then passes the reduced carrier signal from upper path 1022 to signal comparator 1072. In a preferred embodiment, an additional non-inverting amplifier, such as a voltage follower, is inserted between amplifier 1052 and signal comparator 1072 to buffer the signal.

[0061] The lower path of this lower block diagram is the same as the upper block diagram: signal 1012 enters lower path 1027, is amplified by amplifier 1035, and passed to signal comparator 1072, which subtracts the upper path signal from the lower path signal.

[0062] Figure 11 shows a circuit implementation of the block diagram in the top half of Figure 10. This circuit accepts a digital carrier signal 1110 and creates two identical paths for the signal by feeding them into linear amplifiers 1120 and 1130, respectively, which output a carrier-purified signal 1200. The output of linear amplifier 1120 in the top path is exposed to crystal 1150 fitted with damping resistor 1155, which removes carrier-frequency energy. This sideband-purified signal is buffered by linear amplifier 1125.

[0063] The sideband purified signal from linear amplifier 1125 is adjusted by resistor 1170 to equalize the signal strength of the sideband signals in the two paths so that they cancel each other out. The signals from both paths enter amplifier 1190 via summing resistor 1180. The upper amplifiers 1120 and 1125 invert the signal twice, while the lower path amplifier 1130 inverts the signal once, allowing them to subtract from each other.

[0064] Damping resistor 1155 is adjusted to reduce resonance, preferably below critical damping where the crystal will not continue to oscillate without energy input. In a preferred embodiment, a differential amplifier replaces amplifier 1190, comparing the two signals from amplifiers 1130 and 1125 and outputting an enriched carrier signal. Other methods of removing the upper path carrier can be determined by those skilled in the art, such as using a controlled resonant LC circuit in place of the crystal. Two or more sideband removal circuits may be employed in series. Each amplifier in this example is an MMBT3904 NPN transistor biased for common-emitter linear operation.

[0065] The key to this technique is to shift the phase of the two signals by 180 degrees. We found that adding even one transistor to the upper path limits the purification capability by approximately 10 ns of phase shift. In one embodiment, the transistors in the long path have higher frequency response than the transistors in the short path. Preferably, the gain-bandwidth product of one or more of the transistors in the long path is at least twice the gain-bandwidth product of one or more of the transistors in the short path. The higher-frequency transistors pass signals faster than the lower-frequency transistors. Those skilled in the art can select and test different transistor combinations to equalize the transit times. While this was not a significant issue at 4 MHz, such differences must be considered at higher frequencies. In one embodiment, a small inductor, approximately 0.1–5 nanohenries, is added to the short path to ameliorate this issue.

[0066] Figure 12 shows spectrum analyzer data obtained by operating the circuit in Figure 11. The digital carrier signal was 3.9996 MHz and a 4 MHz crystal was used. Even when the crystal was operating at 3.9996 MHz (minimum reactance) and with a damping resistor of 400 ohms or greater, the effective passband (roll-off) sensitivity bandwidth of the crystal operation was approximately 200 Hz, so the digital carrier signal had to be lowered by 400 Hz. The digital carrier signal was a random pattern consisting of 10 sine wave segments (1 bits) separated by 10 sine wave long pauses (0 bits).

[0067] The output of a spectrum analyzer shown here is a scan from 3MHz to 5MHz. Each vertical bar on the display represents a 200kHz frequency separation.

[0068] The top plot 1201 shows a scan from a spectrum analyzer of the unfiltered signal. The bottom plot 1291 is the scan after filtering, with resistor 1155 adjusted to what is considered optimal: 400 ohms.

[0069] The top plot 1201 shows a 3.9995 MHz digital carrier 1210. There is a lot of energy in the wide sideband signal 1220 between 3.7 MHz and 4.3 MHz. The bottom plot 1291 shows the same digital carrier 1230 at 3.9995 MHz with a much reduced sideband signal 1240.

[0070] Several 4.000 MHz quartz crystals were compared, all exhibiting a sharp resonant peak even with damping resistors. The carrier frequency had to be adjusted over a narrow range, from about 3.9996 MHz to 3.9997 MHz. Filtering resulted in a bandpass of about 200 Hz, allowing for the broadcast of a narrow-bandwidth filtered digital carrier signal. Bandwidths below 500 Hz and even below 200 Hz are achievable with this technology if temperature compensation is used to correct for frequency drift.

[0071] A wider filter bandpass may be required to accommodate multiple closely spaced signals. In one embodiment, two or more crystals may be connected in parallel to widen the bandpass to 500 Hz, 1 kHz, or even wider in the transmitter, receiver, or both. The term bandwidth used here is familiar to radio experts. It can be measured as the -3 dB power point on either side of the carrier frequency. [Example]

[0072] Sideband elimination by selective suppression with inverter resonators. It has been discovered that sidebands can be selectively removed by using a resonator such as a quartz crystal in parallel resonant mode and connecting it to the negative feedback of an inverting amplifier. This works particularly well with transistor amplifier circuits, which invert the amplified signal. Figure 13 shows an example of creating a digital carrier with sidebands using the above-described device. This complex signal is applied to an inverting amplifier, such as a common-emitter or common-source transistor amplifier. The quartz crystal is chosen to have a nominal resonant frequency just below the digital carrier frequency to be filtered.

[0073] This technique is useful in any application or circuit where you want to remove a narrowband signal from nearby signals. It is desirable that the circuitry other than the crystal be resonance-free. That is, the signal enters the inverting amplifier and exits without any resonance at the connections. It has been found that using conventional interstage coupling, impedance couplers, or resonant filters with this inverting amplifier does not work due to excessive crystal resonance. Preferably, the input and output connections to the inverting amplifier, as shown in Figure 13, are resonance-free and have a series resistance of preferably 100-1000 ohms or more.

[0074] FIG. 13 shows an inverting circuit 1340 based on a transistor 1341. A sinusoidal reference 1310 creates a carrier signal that is manipulated by circuit 1330. In this example, circuit 1330 uses a synchronous square wave 1320 to generate 0s and 1s on the carrier. The output of manipulation circuit 1330 is a digital (interrupted) carrier with sidebands. This sideband-rich signal enters inverting amplifier circuit 1340 via resistor 1343. Transistor 1341 is preferably biased for an amplification ratio of at least 10, more preferably at least 50. The base bias is not shown, and emitter (or source) resistor 1344 is preferably bypassed with a capacitor, as is customary. The inverted amplified signal 1350 is then taken from the collector at the junction with resistor 1345. Those skilled in the art can adapt virtually any inverting amplifier to replace amplifier 1340, as long as no extraneous resonant source is connected that would affect resonator 1342. Other parallel resonant sources, such as a parallel connected inductor and capacitor chosen to have equal resonance at the transmit frequency, can be used for selective suppression. The "resonator" in this embodiment does not swing to resonance with the carrier frequency, but instead allows negative feedback for signals on either side of the carrier frequency while minimizing the effect on the carrier.

[0075] In one embodiment, two quartz crystals are used in parallel to widen the bandpass of the selective extinction, while in another embodiment, two selective extinction filters are used in series to improve sideband rejection. This sideband filter was built with a 4.000 MHz crystal and successfully reduced the sidebands by at least 10 times, typically 100 times, as seen on a spectrum analyzer of the input and output signals. The parallel resonance peak of the 4 MHz crystal is at 4.001 MHz, and the degree of purification was controlled by varying the transmit frequency over a 500 Hz range. In effect, this frequency was used to tune the filter. Adding capacitance (up to 50 pF) in series or parallel with the AT-cut crystal was useful for optimizing this circuit for a given frequency.

[0076] The techniques and principles explored in this research apply to all communications at all wavelengths. This project focused on simple on / off operation of sinusoidal signals at low ham band frequencies between 2-21 MHz. These results point the way to higher photon pulse transmission rates for faster communications in narrower bandwidths.

[0077] Embodiments contemplate the use of a wide variety of antennas, reference signal generation techniques, and frequencies. For example, for transmissions at frequencies above 400 MHz, small non-resonant diamond antennas are preferred.

[0078] The above describes in detail the embodiments of the present invention with reference to the drawings, but the specific configuration is not limited to the embodiments, and designs within the scope of the claims do not deviate from the essential spirit of the present invention. [Industrial Applicability]

[0079] By eliminating resonance from the transmit signal path and using zero signal crossings for transmit and receive detection, a high data rate digital carrier transmitter can be provided. [Explanation of symbols]

[0080] 10 Vessel 20 Butterworth 30 Chebyshev 11,21 Filter 31 Chebyshev 30 delay 210 Pulse Transmission 220 pulse bandwidth 310 Square wave input 320 High-Speed ​​D Flip-Flop 330 High-Speed ​​Analog Switch 340 Transistor Linear Amplifier 350 High Voltage MOSFET 360 Non-resonant antenna 370 Synchronous Sine Wave 375 Time Base Divider 380 Data Entry Device 390 Shift Register 392 Endless Loop Option 410 Amplifier Chain 420 Flip-Flops and Analog Switches 430 Time division divider to determine bit rate 440 Shift register connection 510 Transmitter 520, 540 Non-resonant antenna 525 Termination Resistor 530 waveform 550 receiver 560 terminating resistor 610 signal 620 Result Signal 710 signal 720,730,740 01 bit string 810 Reference Signal 820 High-Speed ​​Analog Switch / Harmonic Filter 830 Current Amplifier / Path 832,834,836 Amplifier 832,834 Amplifier 840 Voltage Amplifier / Correction Signal 850 Antenna 860 flip-flops 910,920 Phase adjustment capacitor 925 4-element bandpass filter 930 potentiometer 1010,1012 input signal 1020,1022 Route 1025,1027 Lower route 1030,1032,1035,1050,1052 amplifiers 1040,1042 crystal oscillator 1070,1072 signal comparator 1090 signal 1110 digital carrier signal 1120,1125 Linear amplifier / upper amplifier 1130 Linear Amplifier / Lower Path Amplifier 1150 crystal oscillator 1155 Damping Resistor 1170,1180 Resistance 1190 Amplifier 1200 Carrier Purification Signal 1201,1291 plots 1210,1230 digital carrier 1220,1240 sideband signal 1310 sine reference 1320 synchronous square wave 1330 circuits 1340 Inverting circuit, amplifier 1341 Transistor 1342 resonator 1343,1345 Resistance 1344 emitter (or source) resistor 1350 signal

Claims

1. frequency generator, an input for accepting a digital message to be sent; Non-resonant antennas, and a circuit connected to the frequency generator for receiving a digital message from the input, generating one or more sinusoidal pulses of a determined sinusoidal wavelength from the frequency generator, and outputting the sinusoidal pulses as direct current to the antenna; A digital carrier wave transmitter consisting of:

2. 10. The digital carrier wave transmitter of claim 1, further comprising a zero-crossing detector that detects a zero-crossing time of the reference frequency and switches the output transmit signal at the zero-crossing.

3. 10. The digital carrier transmitter of claim 1, wherein the circuit is non-resonant.

4. 10. The digital carrier transmitter of claim 1, further comprising a sideband rejection circuit that enriches sinusoidal pulses at the digital carrier frequency without using a resonant filter to absorb sidebands.

5. 5. The digital carrier transmitter of claim 4, wherein the sideband rejection circuit generates a signal for transmission having a bandwidth of less than 200 Hz.

6. 10. The digital carrier transmitter of claim 1, wherein the transmitter is transmission line free.

7. a radio signal generator having a frequency output; Non-resonant antennas, and a circuit for receiving the frequency output and generating a discrete integer long wavelength portion or a discrete half integer long wavelength portion of the wavelength of the radio signal reference as a direct current to a non-resonant antenna; A direct current radio transmitter that transmits each data bit as a long, discrete pause or pulse of electromagnetic energy of up to 20 sine waves.

8. 8. The DC radio transmitter of claim 7, further comprising a data input for accepting bit data that triggers circuitry to select an integer copy or half copy of the wavelength of the radio signal reference that is output as DC to the non-resonant antenna.

9. 8. The DC radio transmitter of claim 7, wherein the transmitter does not have a resonant filter with a Q factor greater than 10 for removing harmonics or sidebands of the signal being broadcast.

10. 8. The digital carrier transmitter of claim 7, further comprising a sideband rejection circuit that enriches the sinusoidal pulses of the digital carrier.

11. 11. The digital carrier transmitter of claim 10, wherein the sideband rejection circuit generates a signal for transmission having a bandwidth of less than 500 Hz.

12. 8. The digital carrier transmitter of claim 7, wherein the transmitter is transmission line free.

13. High frequency signals, At least one linear amplifier that amplifies the radio frequency signal and outputs a direct current to a connected non-resonant antenna; and a zero-crossing detector for selecting an integral number or half of a sine wave from the high frequency signal; A digital carrier wave radio transmitter, wherein the non-resonant antenna receives direct current in the form of integral or half pulses of a selected sine wave.

14. 14. The digital carrier radio transmitter of claim 13, wherein the radio transmitter simultaneously transmits two signals that are 180 degrees out of phase with each other.

15. Further, the harmonic suppressor circuit includes: a harmonic filter for purifying the harmonics of a high frequency signal to produce a harmonically enriched signal; and 14. The digital carrier radio transmitter of claim 13, further comprising a differential amplifier or summing circuit for comparing said harmonically enriched signal with a radio frequency signal and producing a comparison signal having reduced harmonic energy than said radio frequency signal.

16. 14. The digital carrier transmitter of claim 13, further comprising a sideband rejection circuit that enriches the sinusoidal pulses relative to the digital carrier frequency by selectively suppressing the sidebands with an inverting amplifier that includes negative feedback of the sidebands.

17. 17. The digital carrier transmitter of claim 16, wherein the sideband rejection circuit purifies sinusoidal pulses at the digital carrier frequency having a bandwidth less than 500 Hz.

18. 14. The digital carrier transmitter of claim 13, wherein the transmitter is transmission line free.

19. 14. The digital carrier wave transmitter of claim 13, wherein the transmitter transmits less than 20 sinusoidal wavelengths of discrete pulses of electromagnetic energy for each bit of signal information.

20. 10. The digital carrier wave transmitter of claim 1, wherein a DC power supply is electrically connected to the transmitter, said connection providing zero voltage to the antenna and positive voltage to ground at the transmitter output, said power supply conducting DC electronic energy transfer from the transmitter to the antenna.

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