Motor control device for electric compressor

Abstract

To provide a motor control device of an electric compressor capable of suppressing inaccurate position estimation of a motor (rotor) due to superimposition of a harmonic voltage on a voltage command.SOLUTION: A motor control device 100 includes a PWM calculator 24 that generates a signal for controlling a motor 210 of an electric compressor 200 based on an addition value obtained by adding a harmonic voltage command value for generating a harmonic current to a voltage command value for driving the motor 210, a position error estimator 29 that calculates an estimation error Δ θ of a rotor position using the harmonic current, a harmonic amplitude adjuster 23 that adjusts an amplitude Vh of the harmonic voltage command value, and a position estimator 30 that estimates the rotor position using the estimation error Δ θ calculated by the position error estimator 29. The position error estimator 29 calculates the estimation error Δ θ using the γ - axis harmonic current and the δ - axis harmonic current. The harmonic amplitude adjuster 23 increases the amplitude Vh as the input voltage Vdc increases.SELECTED DRAWING: Figure 5

Description

[Technical Field]

[0001] The present disclosure relates to a motor control device for an electric compressor. [Background technology]

[0002] For example, Japanese Patent Laid-Open Publication No. 8-308286 (Patent Document 1) discloses detecting the operation of a motor based on an induced electromotive force induced in the motor.

[0003] Japanese Patent Laid-Open Publication No. 7-245981 (Patent Document 2) discloses a system that detects the magnetic pole position of a motor using a d-axis voltage command value on which an alternating voltage is superimposed. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] Japanese Patent Application Publication No. 8-308286 [Patent Document 2] Japanese Patent Application Publication No. 7-245981 Summary of the Invention [Problem to be solved by the invention]

[0005] When driving a motor for a relatively large-capacity electric compressor, the motor may frequently switch between on and off (ON / OFF operation) when the refrigerant flow rate of the electric compressor is low. In this case, the electric compressor may be damaged due to an oil shortage or the like. Therefore, in order to avoid this ON / OFF operation, it is possible to control the motor at a relatively low speed.

[0006] However, since the induced electromotive force of a motor is small in the low speed region, there is a drawback in that the position of the motor cannot be accurately estimated using the method described in Patent Document 1.

[0007] In contrast, the technique of Patent Document 2 makes it possible to estimate the position of the motor (rotor) relatively accurately even in low-speed regions. However, by adding a high-frequency alternating voltage (harmonic voltage) to the voltage command, a high-frequency alternating current (harmonic current) is superimposed on the AC current flowing through the motor. This causes irregular timing at which the AC current switches between positive and negative, which can result in inaccurate dead-time correction. In this case, there is a risk that the motor position will be estimated inaccurately.

[0008] An object of the present technology is to provide a motor control device for an electric compressor that can suppress inaccuracies in motor (rotor) position estimation caused by superimposing harmonic voltages on a voltage command. [Means for solving the problem]

[0009] According to one aspect of the present disclosure, there is provided a motor control device for an electric compressor, the motor control device including: a signal generating unit that generates a signal for controlling the motor based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current used to estimate a rotor position of the motor to a driving voltage command value for driving the motor of the electric compressor; a position error estimator that calculates an estimation error of the rotor position using the harmonic current; a position estimator that estimates the rotor position using the estimation error calculated by the position error estimator; and a harmonic amplitude adjuster that adjusts the voltage amplitude, which is the amplitude of the harmonic voltage command value. The position error estimator calculates the estimation error using a γ-axis harmonic current that is a harmonic current on the γ-axis, which is an estimation axis of the d-axis, and a δ-axis harmonic current that is a harmonic current on the δ-axis, which is an estimation axis of the q-axis. The harmonic amplitude adjuster adjusts the voltage amplitude based on an input voltage to the harmonic amplitude adjuster, and makes the voltage amplitude larger when the input voltage is at a first voltage value than when the input voltage is at a second voltage value smaller than the first voltage value.

[0010] In the motor control device for an electric compressor according to one aspect of the present disclosure, as described above, the position error estimator calculates the position error using the γ-axis harmonic current and the δ-axis harmonic current. The harmonic amplitude adjuster increases the voltage amplitude when the input voltage is at a first voltage value compared to the voltage amplitude when the input voltage is at a second voltage value that is smaller than the first voltage value. When calculating the estimation error using the γ-axis harmonic current and the δ-axis harmonic current, the estimation error increases as the input voltage to the harmonic amplitude adjuster increases, and decreases as the voltage amplitude increases. Therefore, by increasing the voltage amplitude when the input voltage is at the first voltage value compared to the voltage amplitude when the input voltage is at a second voltage value that is smaller than the first voltage value, it is possible to prevent an increase in the estimation error caused by the input voltage increasing from the second voltage value to the first voltage value. This prevents inaccurate motor (rotor) position estimation when harmonic voltages are superimposed on the voltage command.

[0011] The position error estimator may calculate the estimation error based on the ratio between the amplitude of the γ-axis harmonic current and the amplitude of the δ-axis harmonic current. Here, the amplitudes of the γ-axis harmonic current and the δ-axis harmonic current are each proportional to the voltage amplitude. Therefore, the ratio is a value independent of the voltage amplitude. Therefore, the estimation error can be calculated regardless of the voltage amplitude.

[0012] The harmonic amplitude adjuster may increase the voltage amplitude as the input voltage increases. As described above, the estimation error increases as the input voltage to the harmonic amplitude adjuster increases, and decreases as the voltage amplitude increases. Therefore, by increasing the voltage amplitude as the input voltage increases, it is possible to effectively suppress an increase in the estimation error.

[0013] The harmonic amplitude adjuster may perform a process of reducing the voltage amplitude when the absolute value of the AC current output from the power converter, which outputs an AC voltage to the motor and performs dead time correction, is smaller than a predetermined value. Here, reducing the voltage amplitude reduces the amplitude of the harmonic current. Therefore, by performing a process of reducing the voltage amplitude when the absolute value of the AC current output from the power converter to the motor is smaller than a predetermined value, it is possible to prevent harmonic currents with large amplitudes from being superimposed on the AC current when the AC current value is near zero. As a result, it is possible to prevent frequent positive and negative switching of the AC current on which harmonic currents are superimposed, thereby preventing inaccurate dead time correction.

[0014] The harmonic amplitude adjuster may execute a process of reducing the voltage amplitude when the amplitude of the AC current output from the power converter, which outputs an AC voltage to the motor and performs dead time correction, to the motor is smaller than a predetermined value. Here, reducing the voltage amplitude reduces the amplitude of the harmonic current. Therefore, it is possible to prevent harmonic currents with relatively large amplitudes from being superimposed on AC currents with relatively small amplitudes. As a result, it is possible to prevent frequent switching between positive and negative for the AC currents with superimposed harmonic currents, thereby preventing inaccurate dead time correction. [Effects of the Invention]

[0015] According to the present technology, it is possible to prevent the position estimation of the motor (rotor) from becoming inaccurate due to the superposition of harmonic voltages on the voltage command. [Brief explanation of the drawings]

[0016] [Figure 1] 1 is a block diagram showing a configuration of a motor system according to an embodiment; [Figure 2] FIG. 2 is a diagram illustrating a detailed configuration of a motor system according to an embodiment. [Figure 3] FIG. 2 is a diagram showing the d, q axes and the γ, δ axes. [Figure 4]FIG. 2 is a block diagram showing a detailed configuration of a controller according to an embodiment. [Figure 5] FIG. 10 is a diagram illustrating the relationship between an input voltage Vdc and an amplitude Vh according to an embodiment. [Figure 6] FIG. 10 is a diagram illustrating the relationship between amplitude Aδ of harmonic current and estimation error Δθ according to one embodiment. [Figure 7] FIG. 2 illustrates an AC current with superimposed harmonic currents according to one embodiment. [Figure 8] FIG. 10 is a diagram illustrating an AC current on which a harmonic current is superimposed according to a modified example of an embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0017] DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described in detail with reference to the accompanying drawings, in which the same or corresponding parts are designated by the same reference numerals and will not be described repeatedly.

[0018] FIG. 1 is a diagram showing the overall configuration of a motor system 1 including a motor control device 100 for an electric compressor 200 according to this embodiment. The motor system 1 is mounted on, for example, an electric vehicle. However, the use of the motor system 1 is not limited to vehicle applications. The motor system 1 may also be used in a stationary system (for example, an air conditioning system). The motor system 1 includes the motor control device 100, a motor 210 for the electric compressor 200, a power source 300, and a main controller 400.

[0019] The power source 300 supplies power to the motor control device 100. The power source 300 is, for example, a DC power source (DC system) such as a storage battery or a solar cell. The power source 300 may also be an AC power source (AC system). In the case of an AC power source, a rectifier must be provided to convert AC to DC.

[0020] The motor control device 100 drives (controls) the motor 210. The motor control device 100 includes a power conversion unit 10 and a controller 20. The power conversion unit 10 performs power conversion operations on the power supplied from the power source 300. The controller 20 controls the power conversion unit 10 in accordance with control commands from a main controller 400. The control commands from the main controller 400 to the controller 20 include a speed command Fm* (a command related to the angular acceleration of the motor 210). The power conversion unit 10 is an example of a "power converter" in the present disclosure.

[0021] Motor 210 is a three-phase AC rotating electric machine or a three-phase brushless DC rotating electric machine, such as an IPM (Interior Permanent Magnet) motor. Motor 210 is not provided with a position sensor (resolver) that detects the position of rotor 211 (described later, FIG. 3). Therefore, motor control device 100 executes sensorless control of motor 210.

[0022] Fig. 2 is a diagram showing an example of the configuration of the motor system 1. However, in Fig. 2, the main controller 400 (Fig. 1) is not shown.

[0023] In this example, the power source 300 is a storage battery. The power source 300 outputs DC power to the power conversion unit 10 via DC terminals Tp and Tn of the power conversion unit 10. The power source 300 is provided with a monitoring unit (including a voltage sensor, a current sensor, etc.) 310 that monitors the state of the power source 300. The monitoring unit 310 outputs the monitored voltage, current, etc. to the controller 20.

[0024] Power conversion unit 10 converts DC power (DC voltage) from power source 300 into AC power (AC voltage) in accordance with a control command from controller 20, and outputs the AC power (AC voltage) to motor 210. More specifically, power conversion unit 10 includes, for example, a voltage sensor 12 and an inverter 13.

[0025] The voltage sensor 12 detects the voltage between the power line PL and the power line NL, and outputs the detected voltage to the controller 20.

[0026] Inverter 13 is, for example, a two-level three-phase full-bridge circuit. In accordance with a control command from controller 20, inverter 13 converts DC power between power lines PL and NL into AC power and outputs the AC power (AC voltage) to AC terminals Tu, Tv, and Tw. In this example, inverter 13 includes six switching elements Q1 to Q6 and six freewheel diodes D1 to D6. Each of switching elements Q1 to Q6 is a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor), an IGBT (Insulated Gate Bipolar Transistor), a bipolar transistor, or the like. The freewheel diodes D1 to D6 are connected in antiparallel to switching elements Q1 to Q6, respectively. Switching elements Q1 and Q2 are connected in series to form a U-phase arm of the full-bridge circuit. Switching elements Q3 and Q4 are connected in series to form a V-phase arm of the full-bridge circuit. Switching elements Q5 and Q6 are connected in series to form a W-phase arm of the full-bridge circuit. The U-phase arm, V-phase arm, and W-phase arm are connected to AC terminals Tu, Tv, and Tw, respectively. Each phase arm is connected between power line PL and power line NL. When MOSFETs are used as switching elements Q1 to Q6, parasitic diodes of the MOSFETs serve as freewheel diodes D1 to D6.

[0027] The motor 210 includes a rotor 211 (FIG. 3) having a permanent magnet and a stator 212 around which coils are wound. In this example, the stator 212 has a U-phase coil, a V-phase coil, and a W-phase coil. One end of each phase coil is star-connected to the neutral point. The other end of each phase coil is connected to a connection point of the switching elements of each phase arm of the inverter 13.

[0028] The motor 210 is provided with current sensors 213 and 214. The current sensor 213 detects a V-phase current Iv flowing through the motor 210. The current sensor 214 detects a W-phase current Iw flowing through the motor 210. Each of the current sensors 213 and 214 outputs the detected current to the controller 20. Note that the U-phase current Iu and the V-phase current Iv, or the U-phase current Iu and the W-phase current Iw, may be output to the controller 20.

[0029] The controller 20 controls the inverter 13 based on a speed command Fm* from the main controller 400 (FIG. 1) and detection results from various sensors (such as the monitoring unit 310, the voltage sensor 12, and the current sensors 213 and 214). For example, the controller 20 outputs a switching signal SW to each of six switching elements Q1 to Q6 included in the inverter 13. The switching signal SW (FIG. 1) is typically a PWM (Pulse Width Modulation) signal.

[0030] The controller 20 includes, as its main components, a processor 201 and a memory 202. The processor 201 includes processing circuitry such as a central processing unit (CPU) and a microprocessing unit (MPU). The memory 202 includes volatile storage devices such as dynamic random access memory (DRAM) and static random access memory (SRAM), and nonvolatile storage devices such as a hard disk drive (HDD), a solid state drive (SSD), and a flash memory. The memory 202 stores a system program including an operating system (OS), a control program including computer-readable code, and various parameters for controlling the power conversion operation of the power conversion unit 10. The processor 201 performs various arithmetic operations by reading the system program, the control program, and the parameters, and then loading and executing them in the memory 202. The arithmetic operations performed by the controller 20 may be performed by an application-specific integrated circuit (ASIC), a field-programmable gate array (FPGA), or the like.

[0031] In inverter 13, a dead time is provided to prevent the upper arm switching elements (Q1, Q3, Q5) and the lower arm switching elements (Q2, Q4, Q6) in each phase from being turned on simultaneously. Furthermore, dead time correction is performed in inverter 13 to prevent a voltage error (a voltage error between a PWM command and the inverter output voltage) from occurring due to the dead time. If the dead time correction is incorrect, an unintended excessive or insufficient voltage will be applied to motor 210. This voltage error causes a ripple in the current flowing through motor 210, and this current ripple also causes a ripple in the estimated position of rotor 211 (estimated position ripple).

[0032] It is not essential that controller 20 and main controller 400 of motor control device 100 are provided separately. Controller 20 may be configured to calculate speed command Fm* by itself.

[0033] <Calculating the angle difference> 3 is a diagram illustrating the relationship between the magnetic pole position of rotor 211 and coordinate axes while motor 210 is in operation. As shown in FIG. 3, the d-axis is an axis extending from rotation axis C of rotor 211 toward the N-pole of rotor 211. The d-axis rotates counterclockwise at angular velocity ω of rotor 211. The q-axis is an axis orthogonal to the d-axis (an axis extending in a direction 90 degrees electrical angle ahead of the d-axis).

[0034] When performing sensorless control of the motor 210, it is difficult for the controller 20 to accurately determine the d-axis and q-axis of the rotor 211. Therefore, a γ-δ rotating coordinate system is used instead of a dq rotating coordinate system defined by the d-axis and q-axis. The γ-δ rotating coordinate system is defined by the γ-axis and δ-axis, which are estimated d-axis and q-axis. The γ-axis is an axis extending from the rotation axis C toward the estimated north pole of the rotor 211. The δ-axis is an axis orthogonal to the γ-axis (an axis extending in a direction 90 electrical degrees ahead of the γ-axis).

[0035] The γ-axis current and δ-axis current in the γ-δ rotating coordinate system are written as Iγ and Iδ, respectively. The γ-axis current command and δ-axis current command required to generate torque in the motor 210 are written as Iγ* and Iδ*, respectively. The γ-axis current Iγ is a current used to generate a magnetic field in the motor 210. The δ-axis current Iδ is a current corresponding to the torque of the motor 210. The controller 20 sets the δ-axis current command Iδ* to zero and sets the γ-axis current command Iγ* to a variable value, thereby preventing the motor 210 from generating torque and generating a magnetic field at a specified position.

[0036] Hereinafter, the angular difference between the γ axis and the d axis (γ-δ rotating coordinate system with respect to the dq rotating coordinate system) will be referred to as the “estimation error Δθ.” The d-axis self-inductance and q-axis self-inductance of the coil of stator 212 will be referred to as Ld and Lq, respectively.

[0037] <Function block> 4 is a functional block diagram of controller 20 in this embodiment. Controller 20 includes speed control unit 21, current control unit 22, harmonic amplitude regulator 23, PWM calculator 24, coordinate converter 25, BSF (Band Stop Filter) 26, BPF (Band Pass Filter) 27, amplitude estimator 28, position error estimator 29, and position estimator 30. Controller 20 also includes subtractors 31 to 33, a multiplier 34, and an adder 35. Note that PWM calculator 24 is an example of the "signal generator" of the present disclosure.

[0038] Subtractor 31 calculates an angular velocity error by subtracting the current estimated velocity (angular velocity) Fm output from position estimator 30 from velocity command Fm* input to motor control device 100 from main controller 400 (FIG. 1), for example.

[0039] The speed control unit 21 generates a torque command value such that the angular velocity error input from the subtractor 31 approaches 0, and generates a γ-axis current command Iγ* and a δ-axis current command Iδ* for generating the generated torque command value according to PI control. The speed control unit 21 outputs the γ-axis current command Iγ* and the δ-axis current command Iδ* to subtractors 32 and 33, respectively.

[0040] The subtractor 32 calculates a γ-axis current deviation ΔIγ (=Iγ*−Iγ), which is the deviation between the γ-axis current Iγ from the coordinate converter 25, from which harmonic components have been removed by passing through the BSF 26, and the γ-axis current command Iγ* from the speed control unit 21, and outputs the γ-axis current deviation ΔIγ to the current control unit 22.

[0041] The subtractor 33 calculates a δ-axis current deviation ΔIδ (=Iδ*−Iδ), which is the deviation between the δ-axis current Iδ from the coordinate converter 25, from which harmonic components have been removed by passing through the BSF 26, and the δ-axis current command Iδ* from the speed control unit 21, and outputs the δ-axis current deviation ΔIδ to the current control unit 22.

[0042] The current control unit 22 performs a proportional-plus-integral (PI) calculation on the γ-axis current deviation ΔIγ from the subtractor 32, and outputs the calculation result as a γ-axis voltage command Vγ* to the adder 35. The current control unit 22 performs a proportional-plus-integral (PI) calculation on the δ-axis current deviation ΔIδ from the subtractor 33, and outputs the calculation result as a δ-axis voltage command Vδ* to the PWM calculator 24. Each of the γ-axis voltage command Vγ* and the δ-axis voltage command Vδ* is a "driving voltage command value for driving a motor" in the present disclosure.

[0043] The voltage of the power source 300 is input to the harmonic amplitude adjuster 23 as an input voltage Vdc, for example, from the monitoring unit 310 (FIG. 2). The harmonic amplitude adjuster 23 calculates (adjusts) the amplitude Vh of the harmonic voltage command value based on the input voltage Vdc. The harmonic voltage command value is used to generate a harmonic current used to estimate the rotor position of the motor 210. The amplitude Vh is an example of the "voltage amplitude" in the present disclosure.

[0044] The multiplier 34 multiplies the amplitude Vh input from the harmonic amplitude adjuster 23 by the harmonic angular frequency ω h and time t (ω h cosine value of t (cosω h t). The multiplied value calculated by the multiplier 34 is added to the γ-axis voltage command Vγ* by the adder 35. This makes it possible to suppress torque pulsation caused by adding the multiplied value (harmonic component) to the γ-axis voltage command that is unrelated to the torque of the motor 210. The added value (Vγ*+Vhcosω h t) is input to the PWM calculator 24. The multiplied value may be added to the δ-axis voltage command Vδ*.

[0045] The PWM calculator 24 converts the γ-axis voltage command Vγ* and the δ-axis voltage command Vδ* on the dq(γδ) two-phase coordinate system into a U-phase voltage command, a V-phase voltage command, and a W-phase voltage command on the UVW three-phase coordinate system in accordance with a known coordinate transformation formula (dq two-phase → UVW three-phase transformation formula (inverse Park transformation formula)) using the estimated position Hm of the rotor 211 input from the position estimator 30. The PWM calculator 24 further generates a switching signal SW from the voltage commands for the three phases. More specifically, the PWM calculator 24 generates a PWM signal as the switching signal SW based on a comparison between the voltage commands for each phase and a predetermined carrier wave. The PWM calculator 24 outputs the generated switching signal SW to the power conversion unit 10 (inverter 13, FIG. 2).

[0046] The coordinate converter 25 calculates the γ-axis current Iγ and the δ-axis current Iδ based on the V-phase current Iv detected by the current sensor 213 and the W-phase current Iw detected by the current sensor 214. The coordinate converter 25 calculates the γ-axis current Iγ and the δ-axis current Iδ in accordance with a known coordinate transformation formula (UVW three-phase to dq two-phase transformation formula (Park transformation formula)) using the estimated position Hm of the rotor 211 input from the position estimator 30. The coordinate converter 25 outputs the γ-axis current Iγ and the δ-axis current Iδ to the BSF 26 and the BPF 27, respectively.

[0047] The BPF 27 removes sub-harmonic components from each of the input γ-axis current Iγ and δ-axis current Iδ. The BPF 27 extracts the γ-axis harmonic current iγ′ and the δ-axis harmonic current iδ′ from which the sub-harmonic components have been removed, and outputs the γ-axis harmonic current iγ′ and the δ-axis harmonic current iδ′ to the amplitude estimator 28.

[0048] Here, the γ-axis harmonic current iγ' and the δ-axis harmonic current iδ' are expressed by the following equations (1) and (2), respectively. Note that L0 is (Ld+Lq) / 2, and L1 is (Lq-Ld) / 2. Also, Δθ represents the difference (estimation error) between the estimated angle of the rotor 211 and the actual angle of the rotor 211. iγ'=Vh(L0+L1cos2Δθ)sinω h t / ω h (L02 -L1 2 )···(1) iδ'=Vh(L1sin2Δθ)sinω h t / ω h (L0 2 -L1 2 )···(2)

[0049] The amplitude estimator 28 estimates (calculates) the amplitude Aγ of the γ-axis harmonic current iγ′ from the input γ-axis harmonic current iγ′. The amplitude estimator 28 estimates (calculates) the amplitude Aδ of the δ-axis harmonic current iδ′ from the input δ-axis harmonic current iδ′. The amplitude estimator 28 outputs the amplitude Aγ and the amplitude Aδ to the position error estimator 29.

[0050] Position error estimator 29 calculates estimated error Δθ using amplitudes Aγ and Aδ input from amplitude estimator 28. Specifically, position error estimator 29 calculates estimated error Δθ using the following equations (3) and (4). Note that equation (4) is an approximation equation when it is assumed that Δθ can be approximated to 0. Position error estimator 29 outputs the calculated estimated error Δθ to position estimator 30. Note that Δθ can be approximated to 0 means that the absolute value of Δθ can be assumed to be sufficiently small. Aδ / Aγ=(L1sin2Δθ) / (L0+L1cos2Δθ)···(3) tan -1 [{(L0+L1) / L1}(Aδ / Aγ)] / 2≒Δθ···(4)

[0051] The position estimator 30 uses the estimated error Δθ input from the position error estimator 29 to calculate the current estimated speed Fm of the rotor 211 and the current estimated position Hm of the rotor 211. Specifically, the position estimator 30 calculates the estimated speed Fm and the estimated position Hm that cause the estimated error Δθ to converge to zero by PI control. The position estimator 30 outputs the estimated speed Fm to the subtractor 31. The position estimator 30 outputs the estimated position Hm to each of the PWM calculator 24 and the coordinate converter 25.

[0052] In conventional motor control devices, the timing at which the AC current switches between positive and negative becomes irregular due to harmonic currents being superimposed on the AC current, which can result in inaccurate dead time correction.

[0053] Therefore, in this embodiment, as shown in Fig. 5, the harmonic amplitude adjuster 23 increases the amplitude Vh as the input voltage Vdc increases. For example, the harmonic amplitude adjuster 23 linearly increases the amplitude Vh in proportion to the input voltage Vdc. Alternatively, the amplitude Vh may be increased exponentially in proportion to the input voltage Vdc.

[0054] As can be seen from the above equations (1) and (2), the amplitude Aγ of the γ-axis harmonic current iγ' and the amplitude Aδ of the δ-axis harmonic current iδ' each increase as the amplitude Vh increases. When Δθ can be approximated to 0, the amplitude Aγ becomes sufficiently larger than the amplitude Aδ (Aγ >> Aδ). Therefore, when the amplitude Aγ and the amplitude Aδ increase by increasing the amplitude Vh, the slope of the estimation error Δθ relative to the amplitude Aδ decreases based on the above equation (4) (see Figure 6). In other words, the fluctuation of the estimation error Δθ (estimated position ripple) relative to the fluctuation of the amplitude Aδ caused by the δ-axis current ripple (current ripple due to failure of dead-time compensation) decreases. Note that Δθ can be approximated to 0 means that Δθ can be assumed to be sufficiently small.

[0055] Furthermore, the voltage error caused by the failure of dead time correction is proportional to the pulse height (input to the inverter 13) based on the input voltage Vdc. Therefore, the voltage error is proportional to the input voltage Vdc. Therefore, the estimated position ripple increases in proportion to the input voltage Vdc.

[0056] As described above, the estimated position ripple increases in proportion to the input voltage Vdc and decreases in proportion to the amplitude Vh. Therefore, as shown in Fig. 5, by controlling the amplitude Vh to increase as the input voltage Vdc increases, it is possible to prevent the estimated position ripple from increasing (by the amount that the estimated position ripple caused by the amplitude Vh decreases). In a region where the input voltage Vdc is relatively low, the estimated position ripple caused by the input voltage Vdc is relatively small, so even if the amplitude Vh is set relatively small, the estimated position ripple is prevented from becoming excessively large.

[0057] Furthermore, in a region where the input voltage Vdc is relatively low, there is a relatively small upper limit to the voltage that can be output by the inverter 13. Therefore, by making the amplitude Vh relatively small in a region where the input voltage Vdc is relatively low, it is possible to prevent the harmonic superimposed voltage from being limited to the above upper limit.

[0058] Furthermore, since the amplitude Vh is relatively small in the region where the input voltage Vdc is relatively low, it is possible to reduce noise caused by harmonic currents and to prevent the efficiency of the motor 210 from deteriorating due to harmonic currents.

[0059] Furthermore, as shown in FIG. 7, the harmonic amplitude regulator 23 performs a process of reducing the amplitude Vh when the absolute value of the AC current (U-phase current Iu is shown as a representative in FIG. 7) output from the inverter 13 (FIG. 2) to the motor 210 is smaller than the threshold value Ith1. FIG. 7 illustrates how the amplitude of the harmonic current superimposed on the AC current is reduced by reducing the amplitude Vh. This reduces the amplitude of the harmonic current superimposed on the AC current, as shown in FIG. 7. As a result, it is possible to suppress the occurrence of zero crossings of harmonics. Note that the threshold value Ith1 is an example of a "predetermined value" in the present disclosure. The threshold value Ith1 is a value close to 0. For example, the threshold value Ith1 may be a value substantially equal to the (maximum value of) the amplitude of the harmonic current before the process of reducing the amplitude Vh.

[0060] Furthermore, harmonic amplitude regulator 23 may reduce amplitude Vh by a preset percentage (for example, 50%) when the absolute value of the AC current becomes smaller than threshold value Ith1. Note that harmonic amplitude regulator 23 may determine the percentage using a map, table, or the like, depending on the current amplitude Vh, etc. Also, while FIG. 7 has been described based on U-phase current Iu, the same applies to W-phase current Iw and V-phase current Iv.

[0061] The harmonic amplitude adjuster 23 may perform the processing of Fig. 7 based on the γ-axis harmonic current iγ and the δ-axis harmonic current iδ output from the coordinate converter 25 and the estimated position Hm output from the position estimator 30. The harmonic amplitude adjuster 23 may also perform the processing of Fig. 7 based on the V-phase current Iv detected by the current sensor 213 and the W-phase current Iw detected by the current sensor 214. The processing of reducing the amplitude Vh shown in Fig. 7 is not essential.

[0062] As described above, in this embodiment, the harmonic amplitude regulator 23 increases the amplitude Vh as the input voltage Vdc to the harmonic amplitude regulator 23 increases. As a result, while the ripple of the estimation error Δθ increases due to an increase in the input voltage Vdc, the ripple of the estimation error Δθ decreases due to an increase in the amplitude Vh, so that it is possible to suppress an increase in the (total) estimation error Δθ. As a result, it is possible to suppress an increase in torque pulsation due to an increase in the estimation error Δθ (estimated position ripple).

[0063] Furthermore, in this embodiment, the estimated error Δθ is a value that does not depend on the amplitude Vh based on the above equation (4), so the estimated error Δθ can be accurately calculated no matter how the amplitude Vh is changed.

[0064] <Modification> In the above embodiment, an example has been described in which the amplitude Vh is reduced when the absolute value of the AC current output to the motor 210 is smaller than the threshold value Ith1, but the present disclosure is not limited to this. For example, as shown in FIG. 8, the harmonic amplitude regulator 23 may perform a process to reduce the amplitude Vh when the amplitude of the AC current (U-phase current Iu in FIG. 8) output from the inverter 13 (FIG. 2) to the motor 210 is smaller than the threshold value Ith2. For example, the threshold value Ith2 may be a value substantially equal to (the maximum value of) the amplitude of the harmonic current before the process to reduce the amplitude Vh. The threshold value Ith2 is an example of a "predetermined value" in the present disclosure.

[0065] As a result, when the motor 210 is under a low load and the amplitude of the AC current is small, the amplitude Vh is reduced, thereby reducing the amplitude of the harmonic current, making it possible to prevent the occurrence of zero crossings of harmonics. Fig. 8 shows that the amplitude of the harmonic current superimposed on the AC current is smaller due to the reduction in amplitude Vh when the amplitude of the AC current is smaller than the threshold value Ith2 (solid line) than when the amplitude of the AC current is equal to or greater than the threshold value Ith2 (dashed line).

[0066] For example, the harmonic amplitude regulator 23 may reduce the amplitude Vh by a preset percentage (e.g., 50%) when the amplitude of the AC current becomes smaller than the threshold value Ith2. The harmonic amplitude regulator 23 may determine the percentage using a map, a table, or the like depending on the current amplitude Vh, etc. Also, while FIG. 8 has been described based on the U-phase current Iu, the same applies to the W-phase current Iw and the V-phase current Iv. The processing in FIG. 8 may be performed based on the detection values ​​of the current sensors 213, 214, or may be performed based on the γ-axis current command Iγ*, the δ-axis current command Iδ*, etc.

[0067] In the above embodiment, the amplitude Vh increases as the input voltage Vdc increases, but the present disclosure is not limited to this. For example, the amplitude Vh may be constant at a first voltage when the input voltage Vdc is equal to or less than a threshold, and may be constant at a second voltage higher than the first voltage when the input voltage Vdc is greater than the threshold.

[0068] The configurations described in the above embodiment and the various modifications described above may be implemented in any combination.

[0069] The embodiments disclosed herein should be considered to be illustrative in all respects and not restrictive. The scope of the present disclosure is defined by the claims, not by the description of the above embodiments, and is intended to include all modifications within the meaning and scope of the claims. [Explanation of symbols]

[0070] 10 power conversion unit (power converter), 23 harmonic amplitude regulator, 24 PWM calculator (signal generation unit), 29 position error estimator, 30 position estimator, 100 motor control device, 200 electric compressor, 210 motor, Aγ amplitude (amplitude of γ-axis harmonic current), Aδ amplitude (amplitude of δ-axis harmonic current), Ith1 threshold (predetermined value), Ith2 threshold (predetermined value), Vh amplitude (voltage amplitude).

Claims

1. a signal generating unit that generates a signal to control the motor based on a sum obtained by adding a harmonic voltage command value for generating a harmonic current used to estimate a rotor position of the motor to a driving voltage command value for driving the motor of the electric compressor; a position error estimator that calculates an estimated error in the rotor position using the harmonic current; a position estimator that estimates the rotor position using the estimated error calculated by the position error estimator; a harmonic amplitude adjuster that adjusts a voltage amplitude that is the amplitude of the harmonic voltage command value, the position error estimator calculates the estimation error using a γ-axis harmonic current, which is the harmonic current in a γ-axis that is an estimated axis of the d-axis, and a δ-axis harmonic current, which is the harmonic current in a δ-axis that is an estimated axis of the q-axis; The harmonic amplitude adjuster comprises: adjusting the voltage amplitude based on an input voltage to the harmonic amplitude adjuster; A motor control device for an electric compressor, wherein the voltage amplitude when the input voltage is a first voltage value is made larger than the voltage amplitude when the input voltage is a second voltage value smaller than the first voltage value.

2. 2. The motor control device for an electric compressor according to claim 1, wherein the position error estimator calculates the estimation error based on a ratio between an amplitude of the γ-axis harmonic current and an amplitude of the δ-axis harmonic current.

3. 3. The motor control device for an electric compressor according to claim 1, wherein the harmonic amplitude regulator increases the voltage amplitude as the input voltage increases.

4. a power converter that outputs an AC voltage to the motor performs dead time correction; 3. The motor control device for an electric compressor according to claim 1, wherein the harmonic amplitude adjuster executes a process of reducing the voltage amplitude when an absolute value of the AC current output from the power converter to the motor is smaller than a predetermined value.

5. a power converter that outputs an AC voltage to the motor performs dead time correction; 3. The motor control device for an electric compressor according to claim 1, wherein the harmonic amplitude regulator executes a process to reduce the voltage amplitude when the amplitude of the AC current output from the power converter to the motor is smaller than a predetermined value.

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