Semiconductor devices, communication devices, imaging systems and radar devices

Abstract

To provide a technology advantageous for communication using terahertz waves. [Solution] The system includes a reference signal oscillator that generates a first signal at a first frequency, an injection-locked oscillator that generates a second signal at a second frequency in the terahertz band in synchronization with the first signal, and a modulation unit that generates a fourth signal at a fourth frequency from the second signal and a third signal at a third frequency that corresponds to a signal supplied from a baseband circuit.

Description

[Technical Field]

[0001] The present invention relates to a semiconductor device, a communication device, an imaging system, and a radar device. [Background technology]

[0002] Development of oscillators and detectors for wireless communication systems using terahertz waves is underway. Patent Document 1 discloses a phased array radio that transmits and receives signals at frequencies of 300 GHz or higher. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Japanese Patent Publication No. 2023-010120 [Non-patent literature]

[0004] [Non-Patent Document 1] M.Asada et.al., “Resonant Tunneling Diodes for Sub-Terahertz and Terahertz Oscillators”, Jpn.J.Appl.Phys., Vol.47, No.6(2008), pp.4375-4384 [Non-patent document 2] M.Asada et.al., “Theoretical analysis of coupled oscillator array using resonant tunneling diodes in subterahertz and terahertz range”, J.Appl.Phys., Vol. 103, 124514 (2008) Summary of the Invention [Problem to be solved by the invention]

[0005] Generally, as the frequency increases, the wavelength becomes shorter, which reduces the size of the antenna, but the circuit scale becomes larger due to increased harmonic loss. In a phased array antenna, the antenna spacing must be less than the wavelength to suppress grating lobes. In order to achieve smaller antennas and array configurations, the oscillator circuit and modulator circuit must also be made smaller.

[0006] An object of the present invention is to provide a technique that is advantageous for communication using terahertz waves. [Means for solving the problem]

[0007] In view of the above problems, a semiconductor device according to an embodiment of the present invention is characterized by comprising: a reference signal oscillator that generates a first signal at a first frequency; an injection-locked oscillator that generates a second signal at a second frequency in the terahertz band in synchronization with the first signal; and a modulation unit that generates a fourth signal at a fourth frequency from the second signal and a third signal at a third frequency corresponding to a signal supplied from a baseband circuit. [Effects of the Invention]

[0008] According to the present invention, it is possible to provide a technique that is advantageous for communication using terahertz waves. [Brief explanation of the drawings]

[0009] [Figure 1] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 2] FIG. 4 is a diagram showing the injection locking characteristics of the RTD oscillator of the present embodiment. [Figure 3] 1A and 1B are a top view and a cross-sectional view showing a configuration example of a semiconductor device according to an embodiment of the present invention. [Figure 4] FIG. 2 is a diagram showing an example of the configuration of a filter circuit according to the present embodiment. [Figure 5] FIG. 2 is a diagram showing an example of an equivalent circuit of the semiconductor device according to the embodiment. [Figure 6] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 7]FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 8] FIG. 2 is a diagram showing an example of an equivalent circuit of the semiconductor device according to the embodiment. [Figure 9] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 10] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 11] FIG. 1 is a diagram showing an example of the configuration of a semiconductor device according to an embodiment of the present invention. [Figure 12] FIG. 1 is a diagram showing an example of the configuration of a communication system and an imaging system using the semiconductor device of the present embodiment. DETAILED DESCRIPTION OF THE INVENTION

[0010] Hereinafter, embodiments will be described in detail with reference to the accompanying drawings. Note that the following embodiments do not limit the scope of the invention claimed. Although multiple features are described in the embodiments, not all of these multiple features are necessarily essential to the invention, and multiple features may be combined arbitrarily. Furthermore, in the accompanying drawings, the same reference numerals are used to designate the same or similar components, and redundant explanations will be omitted.

[0011] Semiconductor devices according to embodiments of the present disclosure will be described with reference to FIGS. 1 to 12. In the following description, a case where the semiconductor device is used as a transmitter of a communication system (communication device) will be described. However, this is not limiting, and the semiconductor device of this embodiment can also be used as a receiver of a communication system (communication device). Furthermore, in this disclosure, terahertz waves refer to electromagnetic waves in a frequency range of 100 GHz or more and 10 THz or less, more specifically, 300 GHz or more and 3 THz or less. In the description of each embodiment, description of configurations that are the same as those in other embodiments may be omitted. Furthermore, the embodiments can be modified or combined as appropriate.

[0012] A semiconductor device 100 according to a first embodiment of the present disclosure will be described with reference to Fig. 1(a) to Fig. 5. Fig. 1(a) is a conceptual diagram showing the wiring of functional blocks in the semiconductor device 100 according to this embodiment. Fig. 1(b) is a diagram showing an outline of a resonant tunneling diode (RTD) resonating unit 102 in the semiconductor device 100 according to this embodiment. Fig. 1(c) is a diagram showing a further outline of the functional blocks in the semiconductor device 100 according to this embodiment.

[0013] 1(a), the semiconductor device 100 includes a reference signal oscillator 101, an RTD resonator 102, a mixer 103, and an antenna 104. The semiconductor device 100 is configured to emit or receive terahertz waves TW from the antenna 104. The terahertz waves TW are generated on the semiconductor device 100 and are radio signals (RF signals) for wireless communication with a transceiver (not shown).

[0014] The conceptual diagram in FIG. 1(a) schematically illustrates the functional blocks described above and shows the connections between the components. The semiconductor device 100 has a semiconductor structure for generating terahertz waves TW. This semiconductor structure uses an RTD oscillator 110 including a resonant tunneling diode (RTD). The semiconductor device 100 includes an RTD resonator 102 that combines the RTD oscillator 110 with a resonant structure. The RTD resonator 102 is an injection-locked oscillator connected to a reference signal oscillator 101. The RTD resonator 102 receives a reference signal 108 with a frequency f1 from the reference signal oscillator 101. The RTD resonator 102 is an injection-locked oscillator that synchronizes with the reference signal 108 to generate a signal with a frequency f2 in the terahertz band. The terahertz band output signal from the RTD resonator 102 is input to a mixer 103 as a local oscillator (LO) signal 106.

[0015] The mixer 103 is connected to an IF connection port 121 for inputting and outputting a data signal 105, which is an Intermediate Frequency (IF) signal with a frequency f3. The data signal 105 can be supplied, for example, from or to a baseband circuit (not shown). The mixer 103 is also connected to an antenna 104 for inputting and outputting an RF signal with a frequency f4. In this embodiment, the mixer 103 is arranged as a modulation / demodulation unit (which can be a modulation unit when used as a transmitter and a demodulation unit when used as a receiver).

[0016] Mixer 103 has a function of mixing signals of different frequencies. Specifically, mixer 103 mixes the frequencies of LO signal 106 and data signal 105 to synthesize RF signal 107 of frequency f4 in the terahertz band, and inputs and outputs the RF signal 107. When semiconductor device 100 functions as a transmitter, mixer 103 receives input of LO signal 106 and data signal 105 and outputs RF signal 107. When semiconductor device 100 functions as a receiver, mixer 103 receives input of LO signal 106 and RF signal 107 and outputs data signal 105.

[0017] When the semiconductor device 100 functions as a transmitter, the antenna 104 radiates the RF signal 107 output from the mixer 103 into space as a terahertz wave TW. When the semiconductor device 100 functions as a receiver, the antenna 104 receives the terahertz wave TW propagating through space, and the signal received by the antenna 104 is supplied to the mixer 103 as the RF signal 107.

[0018] The RTD resonating unit 102 included in the semiconductor device 100 will be described using FIG. 1(b). The RTD oscillator 110 arranged in the RTD resonating unit 102 is a negative resistance element having negative resistance, and is used as a high-frequency source for the RTD resonating unit 102. The RTD resonating unit 102 includes the RTD oscillator 110, a resonant conductor 111 that forms a resonant structure, and a GND conductor 116 (details will be described later using FIG. 3(b), etc.). A reference signal 108 with a frequency f1 is input to the RTD resonating unit 102 from a reference signal oscillator 101. The RTD resonating unit 102 can self-oscillate at a frequency f2 even when used alone, but the reference signal oscillator 101 is used as a master oscillator, and the RTD resonating unit 102 operates as a slave. As a result, the RTD resonating unit 102 oscillates in a state where it is injection-locked to the reference signal 108 of the reference signal oscillator 101, and the oscillation frequency and phase noise of the slave RTD resonating unit 102 follow the accuracy of the master reference signal oscillator 101. In this way, the RTD resonating unit 102 behaves as an injection-locked oscillator. This method makes it possible to stabilize the oscillation frequency of the RTD resonating unit 102 (LO) and reduce phase noise. The reference signal oscillator 101 is a wave source for synchronizing the timing of oscillation at the terahertz frequency f2 of the RTD resonating unit 102. Therefore, the frequency f1 of the reference signal 108 may be a subharmonic frequency that is 1 / N (N is a natural number) times the frequency f2 of the terahertz wave oscillation of the RTD resonating unit 102. In this embodiment, the frequency f1 will be described as 1 / 2 (N=2) of f2.

[0019] The power injected from the reference signal oscillator 101 to the RTD resonator 102 is the output P RTDThe RTD oscillator 110 may have a current difference and a voltage difference, respectively, between the current peak and the current valley in the negative resistance region of the RTD oscillator 110. There is no limit to the number of RTD oscillators 110 arranged in the RTD resonating unit 102, and the power sum of the number of RTD oscillators 110 arranged may be injected from the reference signal oscillator 101 to the RTD resonating unit 102. Therefore, the output of the reference signal oscillator 101 is (the number of RTD oscillators 110 arranged in the RTD resonating unit 102) × P RTD In addition, the output may be greater than the transmission loss from the reference signal oscillator 101 to each RTD oscillator 110 arranged in the RTD resonating unit 102. In this embodiment, the number of RTD oscillators 110 arranged in the RTD resonating unit 102 will be described as one.

[0020] The output from the reference signal oscillator 101 is P RTD Specifically, even if the injected signal is a small signal of about 1 / 10,000 of the output of one RTD oscillator 110, the RTD resonating unit 102 may synchronize with the signal supplied from the reference signal oscillator 101. Specifically, when r=5×10 -4 Injection locking is possible with a small signal up to about 1000 MHz. Here, r is called the injection ratio, and V inj / V osc The injection ratio r is expressed as the voltage amplitude (V inj ) and the voltage amplitude of the oscillation signal (V osc ) is the voltage amplitude ratio between P and P. As the injection ratio r decreases, the effect of reducing phase noise due to injection locking weakens, but it is possible to adjust it to the level of phase noise required for the semiconductor device 100. Therefore, in a circuit to which the present disclosure is applied, P RTD Even when using a reference signal oscillator 101 having an output smaller than ((3 / 16)cosωτ ×ΔIΔV), it is possible to synchronize the RTD resonating unit 102 while maintaining the frequency accuracy output by the reference signal oscillator 101. Therefore, the signal strength output by the reference signal oscillator 101 is PRTD It can be set without being heavily dependent on the value of

[0021] A bias voltage is supplied to the RTD resonating unit 102 from a bias unit (not shown). The bias unit supplies power to the bias pattern 112, and the bias voltage is supplied to the resonant conductor 111 through a bias supply via 113. The bias unit supplies the power required to drive the RTD oscillator 110 and adjusts the bias voltage applied to the RTD oscillator 110. The bias voltage is selected from voltages that fall within the negative differential resistance region of the RTD oscillator 110 and is applied from the bias unit via the bias pattern 112. An AC shunt 115 may also be connected to the bias pattern 112. The AC shunt 115 functions as a filter that shorts out signals in a specific frequency band by connecting to GND (e.g., GND conductor 116) using resistance and capacitance. The AC shunt 115 may be disposed in various locations on the bias pattern 112 or in the bias unit as appropriate to suppress parasitic oscillations, which are oscillations of the RTD resonating unit 102 at frequencies other than the specified frequency.

[0022] The AC shunt 115 may be any filter that suppresses parasitic oscillation, and other configurations such as a DC shunt may be used instead. The bias power supply that drives the RTD and is applied from the bias unit to the RTD oscillator 110 is DC. Therefore, the bias unit and bias pattern 112 may be connected directly or via a low-pass filter (LPF) or the like.

[0023] The RTD resonator 102 includes a microstrip line with a specific impedance for oscillating in the terahertz band, which is the characteristic of the RTD oscillator 110. The resonator structure using the microstrip line of the RTD oscillator 110 is composed of a resonator conductor 111, which is a line-shaped upper conductor, a GND conductor 116, which is a wide lower conductor, and a dielectric 141 arranged between the resonator conductor 111 and the GND conductor 116. A detailed layer structure will be described later in conjunction with the explanation of FIG. 3(b). The characteristic impedance of the resonator conductor 111 depends on the thickness and material of the dielectric 141. The thickness of the dielectric 141 is typically designed to be sufficiently thinner than the self-oscillation wavelength of the RTD resonator 102. It is known that the thicker the dielectric 141, the higher the characteristic impedance, and the thinner the dielectric 141, the lower the characteristic impedance. It is also known that the smaller the dielectric constant of a material, the higher the characteristic impedance, and the larger the dielectric constant, the lower the characteristic impedance.

[0024] The frequency f2 of the terahertz waves oscillated from the RTD resonating unit 102 can be determined as the resonant frequency of an all-parallel resonant circuit that combines the reactance of the resonant conductor 111 and the RTD oscillator 110. Specifically, from the description of the equivalent circuit of the oscillator in Non-Patent Document 1, for the resonant circuit that combines the admittance of the RTD oscillator 110 and the resonant conductor 111, the frequency that satisfies the amplitude condition of Equation (1) and the phase condition of Equation (2) can be determined as the oscillation frequency f2. Re[YRTD]+Re[YOSC]≦0 (1) Im[YRTD]+Im[YOSC]=0 (2) Here, Re[YRTD] is the real part of the admittance of the RTD oscillator 110 and has a negative value. Re[YOSC] indicates the real part of the admittance of the resonant conductor 111. Im[YRTD] indicates the imaginary part of the admittance of the RTD oscillator 110. Im[YOSC] indicates the imaginary part of the admittance of the resonant conductor 111.

[0025] The resonant conductor 111 has a line length (resonator length) indicated by an arrow 118 in FIG. 1(a) that is equal to or shorter than the effective wavelength λ of the LO signal oscillated in the RTD resonant section 102. f2λ is set to be a multiple of 1 / 2. f2 is the effective wavelength in the dielectric 141 of the terahertz wave oscillated in the RTD resonator 102, the wavelength of the terahertz wave in a vacuum is λ0, and the relative dielectric constant of the dielectric 141 is ε r1 Then λ f2 =λ0×ε r1 -1 / 2 Specifically, in this embodiment, λ f2 is 0.35 THz, and the wavelength λ0 in vacuum is 0.86 mm. Since the relative dielectric constant of the resonator is about 2, λ f2 The electrical length of the resonant conductor 111 is 0.215 mm.

[0026] As a result, the resonant conductor 111 resonates at a frequency in the terahertz band, and a resonant electric field is generated as a standing wave within the resonant conductor 111. The resonant electric field is generated along the direction of arrow 118, and forms antinodes (points of maximum amplitude) of the electric field at both ends of the resonant conductor 111 and nodes (points of zero amplitude) of the electric field at the center of the resonant conductor 111.

[0027] In addition, the multiple RTD oscillators 110 are arranged at λ f2 When the RTD oscillators 110 are arranged on a resonant conductor 111 of λ / 2, they are arranged at positions facing each other with a node of the resonant electric field at the center. For example, when two RTD oscillators 110 are mutually injection locked with their phases reversed (anti-phase), the two RTD oscillators oscillate in push-pull mode. Also, when two RTD oscillators 110 are mutually injection locked with their phases inverted, the two RTD oscillators oscillate in push-push mode. Specifically, the self-oscillation frequency f2 of the RTD resonating unit 102 can be determined by considering the mutual injection locking in the configuration in which two individual RTD resonating units are coupled, as disclosed in Non-Patent Document 2. The thickness of the dielectric is λ f2 / 10 or more and λ f2 It may be less than / 3.

[0028] The resonant conductor 111 may include an open stub 114 extending in the direction of an arrow 117 shown in FIG. 1(b). One end of the open stub 114 is connected to the center of the resonant conductor 111, and the other end is open. The length of the open stub 114 indicated by the arrow 117 is equal to the effective wavelength λ of the LO signal. f2 / 4, and plays a role in fixing the node of the standing wave oscillating within the resonant conductor 111 to the center of the resonant conductor 111.

[0029] The width of the connection portion of the bias pattern 112 connected to the bias supply via 113 is smaller (thinner) than the width in the direction of the arrow 118 in which the standing wave of the resonant conductor 111 occurs. f2 1 / 10 or less (λ f2 / 10 or less). This is because arranging the bias supply via 113 and the bias pattern 112 in a size and position that does not interfere with the resonant electric field in the resonant conductor 111 is suitable for improving resonance efficiency. In addition, the bias supply via 113 is arranged in the center, which is a node of the standing wave of the resonant conductor 111. By making the connection point of the bias supply via 113 a node of the standing wave in the resonant conductor 111, the impedance of the oscillation signal at the connection point is maximized. Therefore, it is possible to minimize loss caused by leakage of the oscillation signal in the resonant conductor 111 from the bias supply via 113 to the bias line. The bias unit is adjusted as appropriate to efficiently synchronize the reference signal oscillator 101 and the RTD resonant unit 102.

[0030] Furthermore, the RTD resonating unit 102 self-oscillates at a frequency in the terahertz band due to the wiring impedance determined by the structures of the RTD oscillator 110, resonant conductor 111, and open stub 114. This stub structure may be a spiral inductor or interdigital capacitor structure that can be configured with a microstrip line, and self-oscillation can be achieved at any frequency by appropriately designing the inductance and capacitance.

[0031] The connections of the semiconductor device 100 will be described using FIG. 1(c). In the configuration shown in FIG. 1(c), the reference signal oscillator 101 and a phase adjuster 145 constitute a synchronization signal source 144. However, this is not limiting, and the phase adjuster 145 does not have to be provided. The phase adjuster 145 has a function of adjusting the phase of the reference signal 108 output from the reference signal oscillator 101, and the phase-adjusted reference signal 108 is output from the synchronization signal source 144. The synchronization signal source 144 (reference signal oscillator 101) is electrically connected to the RTD resonating unit 102 via a low-pass filter (LPF) 146. The reference signal 108 output from the synchronization signal source 144 (reference signal oscillator 101) is supplied to the RTD resonating unit 102 through the LPF 146. The frequency f1 of the reference signal 108 is upconverted to a frequency f2 of a terahertz wave by injection-locked oscillation into the RTD resonating unit 102. The upconverted signal is output from the RTD resonating unit 102 as an LO signal 106. The RTD resonating unit 102 is electrically connected to a mixer 103, which is a modulation unit, via an intermediate frequency bandpass filter (MPF) 147. The LO signal 106 output from the RTD resonating unit 102 is input to a synchronization signal source 144 (reference signal oscillator 101) attenuated by an LPF 146, and is input to the mixer 103 through the MPF 147. The MPF 147 is also called a band pass filter (BPF). The mixer 103 is connected to an antenna 104 via a high frequency bandpass filter (HPF) 148. The frequency f2 of the LO signal 106 is mixed in the mixer 103 with a data signal 105 corresponding to a signal supplied from a baseband circuit (not shown), and is modulated to a frequency f4 of a terahertz wave. The modulated signal is output from the mixer 103 as an RF signal 107. The input of the RF signal 107 output from the mixer 103 to the RTD resonating unit 102 is reduced by the MPF 147, and the RF signal 107 is input to the antenna 104 through the HPF 148. The frequency f2 of the LO signal 106 is higher than the frequency f1 of the reference signal 108. Furthermore, the frequency f4 of the RF signal 107 is a frequency equal to or higher than the frequency of the LO signal 106.Therefore, MPF 147 may pass signals with lower frequencies than the signals passed by HPF 148, and LPF 146 may pass signals with lower frequencies than the signals passed by MPF 147.

[0032] Next, injection locking between the reference signal oscillator 101 and the RTD resonating unit 102 will be described using Figures 2(a) to 2(c). Figure 2(a) is a graph with the bias voltage applied to the RTD oscillator 110 on the horizontal axis and the frequency oscillated by the RTD resonating unit 102 on the vertical axis. The process of supplying a reference signal from the reference signal oscillator 101 to the RTD resonating unit 102 and stabilizing the frequency of the RTD resonating unit 102 is called injection locking. In this embodiment, the RTD resonating unit 102 uses the harmonic components of the signal output by the reference signal oscillator 101. A solid line 201 in the graph plots the correlation between the oscillation frequency and bias voltage of the RTD resonating unit 102 when a reference signal is supplied from the reference signal oscillator 101, i.e., when injection locked. In this embodiment, a reference signal of 0.175 THz, which is a subharmonic frequency half that of 0.35 THz, is supplied from the reference signal oscillator 101 to the RTD resonating unit 102. The dashed line 202 shown in FIG. 2(a) plots the correlation between the oscillation frequency and bias voltage in the RTD resonating unit 102 when injection locking is not achieved. Here, the frequency described in this embodiment is just an example, and the frequency of the signal output by the reference signal oscillator 101 is not limited to the above example as long as it is the same as that of the RTD resonating unit 102 or has a harmonic component.

[0033] First, regarding the self-oscillation frequency of the RTD oscillator 110, the dashed line 202 in FIG. 2(a) shows that the RTD oscillator 110 oscillates at 0.35 THz when the bias voltage is 0.7 V. It can be seen that the self-oscillation frequency decreases when the bias voltage is reduced, and increases when the bias voltage is increased. Next, the solid line 201 shows the case when injection locking is performed. When the bias voltage is 0.7 V, the oscillation frequency of the RTD oscillator 110 is 0.35 THz, the same as the frequency indicated by the dashed line 202 when injection locking is not performed. Even when the bias voltage is reduced to 0.69 V, the oscillation frequency remains at 0.35 THz. Similarly, even when the bias voltage is increased to 0.71 V, the oscillation frequency of the RTD oscillator 110 remains at 0.35 THz. This is a phenomenon in which the oscillation frequency of the RTD resonator 102 is locked by injection locking from the reference signal oscillator 101. In this way, the oscillation frequency of the RTD resonating unit 102 is locked, and the change from a locked state to an unlocked state is called unlocking. The graph in Figure 2(a) shows that the region in which the oscillation frequency of the RTD resonating unit 102 is locked is the region where the bias voltage is between 0.69 V and 0.71 V, which is called the locking range. Within the locking range, the RTD resonating unit 102, which is the injection-locked oscillator, has its oscillation frequency locked depending on the frequency of the reference signal oscillator 101. It is also known that the stronger the signal supplied from the reference signal oscillator 101 to the RTD resonating unit 102, the wider the bias voltage range becomes, and conversely, the weaker the signal, the narrower the voltage range becomes. Figure 2(a) illustrates an example in which unlocking occurs with a 0.01 V change. However, the bias voltage value and change amount are not limited to this; finer or coarser changes (e.g., 0.1 V) are also possible. These can be set arbitrarily depending on the configuration of the RTD, RTD resonating unit 102, and bias unit used.

[0034] FIG. 2(b) is a graph showing the bias voltage applied to the RTD oscillator 110 on the horizontal axis and the phase difference between the injected reference signal of the reference signal oscillator 101 and the signal of the RTD resonating unit 102 on the vertical axis when injection locking is achieved. At 0.7 V, the phase difference is 0°. This indicates a state in which the injection locking frequency and the self-oscillation frequency match, so the phase of the signal of the reference signal oscillator 101 and the signal of the RTD resonating unit 102 match (frequency f1 × N of the reference signal 108 = oscillation frequency f2 of the RTD resonating unit 102). When the bias voltage applied to the RTD resonating unit 102 decreases, the phase shifts in the -90° direction, and when the bias voltage increases, the phase shifts in the +90° direction. This phenomenon is understood to occur because the difference between the self-oscillation frequencies of the reference signal oscillator 101 and the RTD resonating unit 102 causes the frequency difference between the two signals to appear as a phase difference when locked. This phase difference can be calculated using a theoretical formula: f RTD is the self-oscillation frequency of the RTD resonating unit 102, and f is the frequency at which the RTD resonating unit 102 oscillates due to injection locking, φ=sin -1 (Q√(P o / P i )×(f RTD -f) / (f RTD )) ··· (3) where φ is the oscillation phase of the RTD resonating unit 102, and P i is the oscillation power. P o is the oscillation power of the harmonic components of the reference signal input from the reference signal oscillator 101 and synchronized, and Q is the Q value of the RTD resonating unit 102. The sharper the frequency spectrum, the higher the Q value, and is used as an output index of the oscillation circuit. In this embodiment, f RTD is twice the frequency f1 of the reference signal, and f is the frequency of oscillation by injection locking, i.e., the frequency f2 of the LO signal output from the RTD resonating unit 102. As can be seen from equation (3), this phase difference varies within the range of -90° to +90°. In other words, the phase difference within the locking range varies within the range of -90° to +90°, and the width of the locking range varies depending on the ratio between the oscillation power and the injection power of the RTD resonating unit 102, which is the injection-locked oscillator.

[0035] FIG. 2(c) is a graph plotting the frequency f1 of the reference signal 108 output by the reference signal oscillator 101 and the frequency f2 of the LO signal 106 output by the RTD resonator 102 on the horizontal axis and the oscillation intensity of each on the vertical axis. The dotted line 203 in FIG. 2(c) is a graph plotting the frequency f1 of the reference signal 108 output by the reference signal oscillator 101. Examining the dotted line 203 reveals that the reference signal oscillator 101 oscillates at a subharmonic frequency of 0.175 THz, which is half the frequency of 0.35 THz. The solid line 201 is a graph plotting the frequency f2 of the LO signal 106 output by the RTD resonator 102 when there is an input from the reference signal oscillator 101, i.e., when injection locking is performed. The dashed line 202 is a graph plotting the self-oscillation frequency output by the RTD resonator 102 when injection locking is not performed. These frequencies are merely examples, and the frequency of the signal output by reference signal oscillator 101 is not limited to these examples as long as it is the same as or a harmonic component of RTD resonating unit 102. The signal peak of reference signal 108 output by reference signal oscillator 101 is at 0.175 THz, which matches the injection locking frequency and 0.175 THz, which is a subharmonic frequency that is half the self-oscillation frequency of 0.35 THz when a bias voltage of 0.7 V is applied to RTD resonating unit 102.

[0036] When the RTD resonating unit 102 is not injection locked, i.e., when it is unlocked, it self-oscillates with a relatively low Q value, as indicated by the dashed line 202. Therefore, the spectrum contains a reference signal peak with an intensity of approximately 10 dB, as indicated by the dotted line 203, and a non-injection locked peak with an intensity of approximately 30 dB, as indicated by the dashed line 202. When the RTD resonating unit 102 is injection locked (locked), the reference signal peak, as indicated by the dotted line 203, and the injection locked peak, as indicated by the solid line 201, corresponding to the reference signal peak's Q value, are generated. The injection locked spectrum peak is 50 dB, higher than the 30 dB peak in the asynchronous state. The higher the Q value of the reference signal, the higher the frequency accuracy of the injection locked RTD resonating unit 102, resulting in oscillation with a high Q value. This frequency accuracy refers to the width of the spectrum 3 dB below the peak of the solid line 201, which indicates the injection locked spectrum. This is one of the effects of injection locking, and it also has the effect of improving the frequency stability of the reference signal and making it less susceptible to frequency fluctuations due to external factors. From the relationship with the locking range described above, it can be seen that in order to obtain a stable injection-locked spectrum, it is necessary to supply a reference signal from reference signal oscillator 101, which outputs a reference signal with a high Q-factor spectrum, to RTD resonating unit 102 and lock RTD resonating unit 102. The above spectrum intensity is an example, and can be set arbitrarily depending on the locking range of RTD resonating unit 102.

[0037] 3A is a top view showing an example of the configuration of the semiconductor device 100 of this embodiment. In FIG. 3A, the configuration from the LO port 120 for supplying the reference signal 108 to the RTD resonating unit 102 to the antenna 104 is shown.

[0038] Although not shown in FIG. 3( a), the above-mentioned reference signal oscillator 101 is an oscillator that outputs reference signal 108 at subharmonic frequency f2 (0.175 THz). Reference signal oscillator 101 may be, for example, a general phase-locked loop (PLL) circuit. The PLL circuit is configured to oscillate, at a desired high frequency, a low-frequency oscillation signal supplied from a source of an even lower frequency than reference signal 108, such as a high-precision low-frequency oscillation source using a crystal oscillator. Reference signal 108 output from reference signal oscillator 101 is input via LO port 120.

[0039] LO port 120 may be, for example, a three-terminal connection using a coplanar waveguide. In a coplanar waveguide, two parallel GND lines are arranged on either side of a single signal line, sandwiching the signal line. If reference signal oscillator 101 is arranged on a semiconductor substrate other than the semiconductor substrate on which the components shown in FIG. 3(a) are arranged, these three wiring patterns (GND line, signal line, GND line) are arranged in parallel and connected to LO port 120 on the semiconductor substrate on which RTD resonator 102 is arranged. The connection may be made using bumps such as a flip chip, or may be made by wire bonding from a common interposer substrate. LO port 120 is provided with a three-terminal pad for connection, and a signal input to the substrate is transmitted and converted to a microstrip line. The reference signal is a high-frequency signal of several GHz or more, which is higher than the signals used in ordinary electrical circuits. Therefore, a circuit configuration used for microwaves, millimeter waves, etc. may be used for the connection at LO port 120.

[0040] The microstrip line connected to the LO port 120 is connected to the RTD resonator 102. The RTD resonator 102 is an injection-locked oscillator that receives a reference signal, which is a master signal output from the reference signal oscillator 101, and oscillates in synchronization at 0.35 THz, a frequency in the terahertz band. The RTD oscillator 110 is connected to the resonator 111. The resonator 111 is connected to the mixer 103. In this embodiment, the resonator 111 is T-shaped, but other shapes are also possible. For example, the resonator 111 may be configured with a resonant structure including resistance R, inductance L, and capacitance C that resonates the oscillation signal. Furthermore, the number of RTD oscillators 110 arranged on the resonator 111 is not limited to one, and two or more may be arranged. By arranging multiple RTD oscillators 110 and generating a negative resistance sufficient to compensate for the resonance of the resonator 111 and the loss in the circuit, oscillation in the terahertz band is possible.

[0041] The output of the RTD resonating unit 102 is input to a mixer 103, which is a modulation unit. In this embodiment, a diode-based mixer 103 is used. In this embodiment, the mixer 103 includes an LO filter 122, an IF filter 123, a diode unit 125, and an RF filter 124.

[0042] The LO filter 122 is designed to pass the LO signal 106 of frequency f2 and blocks the data signal 105 of frequency f3 and the RF signal 107 of frequency f4. The LO filter 122 is the above-mentioned MPF 147 that passes only the frequency f2. The LO filter 122 is configured to supply the LO signal 106 from the RTD resonator 102 to the mixer 103 and suppresses backflow of the RF signal 107 and data signal 105 from the mixer 103 to the RTD resonator 102. The LO filter 122 is used to prevent degradation of the quality of the LO signal. Because the RTD resonator 102 is an injection-locked oscillator, if a signal backflow occurs, it is injection-locked by the backflowing signal, reducing the accuracy of the frequency f2 of the LO signal 106. Therefore, the bandwidth of the LO filter 122 (MPF 147) can be designed to achieve a narrow-band filter of approximately frequency f2 ±5 GHz. The LO filter 122 may be designed to have a narrow band with the same frequency accuracy as the LO signal output from the RTD resonating unit 102, and the band may be at most about frequency f2±f3.

[0043] The data signal 105 is supplied to the semiconductor device 100 from the IF connection port 121. IF is often used in the heterodyne system to remove noise components from the data signal to be modulated. When modulating the data signal directly, as in the direct conversion system, the data signal can be used directly. The data signal input from the IF connection port 121 is a data signal modulated at frequency f3, and is combined with the LO signal through the IF filter 123. The IF filter 123 acts as an LPF that separates high-frequency signals such as the LO signal and RF signal from the IF signal line.

[0044] The mixer 103 of this embodiment is a single-diode mixer, with the cathode of the diode unit 125 connected to GND. The mixer 103 mixes the LO signal of frequency f2 and the IF signal of frequency f3 input to the diode unit 125 using the nonlinearity of the diode unit 125, converting the frequency to an RF signal. As a result of mixing, the diode unit 125 outputs a composite wave of frequency f4. Frequency f4 is a composite wave containing the difference and sum frequencies of f2±f3. In particular, in the heterodyne system, the higher frequency f2+f3 can be selectively used. In this case, the difference frequency component f2−f3 is filtered using an MPF ​​or HPF located at the output end of the mixer. Similarly, in this configuration, only the sum frequency component f2+f3 of f4 is filtered by the RF filter 124 at the output end of the mixer 103, and is output from the antenna 104.

[0045] The RF signal generated by the mixer 103 passes through the RF filter 124 and is supplied to the antenna 104. The RF filter 124 is the HPF 148 described above, and serves to match the impedance of the circuit connecting the mixer 103 and the antenna 104. The RF filter 124 is a component necessary for reducing reflected waves to the mixer and for efficient RF signal transmission to the antenna 104. In this embodiment, a single-ended mixer is used as the mixer 103, but it is sufficient if the mixer has the configuration of an element known as a mixer. The mixer 103 may be configured, for example, by a single-balanced mixer using multiple diodes, a double-balanced mixer, or a bidirectional diode mixer also known as a harmonic mixer. It is also possible to appropriately arrange a mixer using a transistor or the like.

[0046] The antenna 104 includes an antenna connection line 143 and an antenna conductor 140. The mixer 103 is electrically connected to the antenna connection line 143. The modulated RF signal is supplied from the mixer 103 to the antenna conductor 140 via the antenna connection line 143. The antenna connection line 143 may have a bent meander stripline configuration in order to reduce the circuit area of ​​the semiconductor device 100. The antenna 104 includes a GND conductor 116, an antenna conductor 140, and a dielectric 142 disposed therebetween (described later with reference to FIG. 3(c)), and is a planar patch antenna that radiates terahertz waves vertically upward from the GND conductor 116. The size of the radiating element is determined so that the frequency f4 of the radiated terahertz waves is equal to or greater than the effective wavelength λ of the electromagnetic waves propagating through the radiating element in the direction in which the resonant electric field is generated as a standing wave. f4 λ taking into account f4 The length of the antenna connection line 143 can be set to about 1 / 2. In this embodiment, the direction in which the standing wave is generated is parallel to the dashed line A-A' shown in FIG. 3(a). By adjusting the connection position between the antenna connection line 143 and the antenna conductor 140 to an appropriate position inside the antenna conductor 140, the impedances of the antenna conductor 140 and the mixer 103 can be matched, and RF signals can be transmitted efficiently.

[0047] Next, the cross-sectional structure of the semiconductor device 100 will be described using FIGS. 3(b) and 3(c). FIG. 3(b) is a cross-sectional view taken along line A-A' in FIG. 3(a), and FIG. 3(c) is a cross-sectional view taken along line B-B' in FIG. 3(a). The semiconductor device 100, from the LO port 120 to the antenna conductor 140, is formed on a semiconductor substrate 130 using a compound semiconductor such as InP. As shown in FIGS. 3(b) and 3(c), dielectric layers 131-134 are disposed on the semiconductor substrate 130, and conductors are disposed between or on the dielectric layers 131-134, thereby constituting the semiconductor device 100 from the LO port 120 to the antenna conductor 140. Here, each of the dielectric layers 131-134 may be a single layer or may be composed of multiple layers. When composed of multiple layers, each layer may be composed of a dielectric material having the same composition, or dielectric materials having different compositions may be stacked.

[0048] A GND conductor 116 is disposed on the semiconductor substrate 130. The GND conductor 116 is electrically connected to the RTD oscillator 110 and the diode unit 125. A Schottky electrode 138, which is a conductor, and a bias pattern 112 are disposed on the upper surface of the dielectric layer 131. A dielectric layer 132 is disposed on the dielectric layer 131, and a resonant conductor 111, which is a conductor of the RTD resonating unit 102, is disposed on the upper surface of the dielectric layer 132. A dielectric layer 133 is disposed on the upper surface of the dielectric layer 132, and microstrip lines constituting the LO port 120, LO filter 122, RF filter 124, and antenna 104 are disposed on the upper surface of the dielectric layer 133. In addition, in the RTD resonating unit 102, a via 136 is disposed in a vertical structure penetrating the dielectric layer 132 to electrically connect the RTD oscillator 110 to the resonant conductor 111 above. Similarly, a via 113 for bias supply is arranged in a vertical structure penetrating the dielectric layer 132 to electrically connect the bias pattern 112 to the upper resonant conductor 111. In the diode section 125, a via 137 is arranged in a vertical structure penetrating the dielectric layers 131 and 132 to electrically connect the Schottky electrode 138 to the conductor of the upper microstrip line.

[0049] The GND conductor 116, RTD oscillator 110, and diode section 125 are composed of carrier-doped semiconductors on an InP semiconductor substrate 130. The GND conductor 116 is composed of a layer in which carriers are highly doped on the InP substrate, and the RTD oscillator 110 functions as a resonant tunneling diode due to the presence of an internal carrier-doped barrier layer 135. The diode section 125 is also Schottky-connected between a diode layer 139, which is a relatively highly conductive semiconductor located directly below the barrier layer 135, and a Schottky electrode 138, and functions as a Schottky barrier diode.

[0050] A part of the microstrip line, which is a wiring pattern, overlaps the upper part of the resonant conductor 111, sandwiching the dielectric layer 133. This is called a Metal Insulator Metal (MIM) structure, and acts as a capacitance, electrically connecting the AC component by AC coupling (capacitive coupling).

[0051] The resonant conductor 111 constituting the resonant structure of the RTD resonant unit 102 is a microstrip line resonator. The resonant structure using the microstrip line of the RTD resonant unit 102, which is an injection-locked oscillator, is configured to include a GND conductor 116, which is a wide lower conductor, a resonant conductor 111, which is a line-shaped upper conductor, and a dielectric 141 arranged between the lower conductor and the upper conductor. The dielectric 141 is configured to include a dielectric layer 131 and a dielectric layer 132.

[0052] The antenna 104 includes a GND conductor 116, an antenna conductor 140, and a dielectric 142 disposed between the GND conductor 116 and the antenna conductor 140. The antenna 104 is a planar patch antenna that radiates terahertz waves vertically above the GND conductor. The dielectric 142 includes a dielectric layer 131, a dielectric layer 132, and a dielectric layer 133.

[0053] 4(a) to 4(f) are examples of filter circuits used in this embodiment. FIG. 4(a) is an example of a bandpass filter whose frequency band can be designed by adjusting the width of the microstrip line, and can be used as the above-mentioned IF filter 123. The width of the line between the input port 410 and the output port 411 varies. Wide line portions 412, 414, and 416 can be regarded as capacitances inserted between GND. Furthermore, narrow line portions 413 and 415 can be regarded as inductances connected in series to the input and output ports. Arrows 421a to 421e shown in FIG. 4(a) indicate the line lengths of the respective lines, λ f3 It is set at / 8. f3 represents the effective wavelength of the data signal 105 passing through this filter.

[0054] Fig. 4(b) is an equivalent circuit diagram of the filter shown in Fig. 4(a). In this embodiment, the resonant structure functions as an IF filter 123. In this case, the impedance of the line is set so that capacitance C1 = 25 fF, C2 = 30 fF, and inductance L1 = 80 pH. These values ​​can be configured to maintain a low impedance in the IF signal band while attenuating signals to less than one-hundredth (-20 dB) at frequencies in the terahertz band.

[0055] In order to form such a filter on the same semiconductor substrate as the RTD resonating unit 102, the size of the filter must be such that the inductance and capacitance can be formed by the semiconductor process. Furthermore, the length of the line must be set to match the frequency of the LO signal to be cut off, and must be less than λ, the effective wavelength of the LO signal. f2 / 8. With this setting, IF filter 123 functions as an LPF, passing low-frequency IF signals in the range of several GHz to several tens of GHz, and blocking high-frequency signals (LO signals and RF signals). This makes it possible to prevent deterioration of signal quality in the IF signal circuit, increase in loss of the LO signal, and the like.

[0056] Fig. 4(c) shows an example configuration of the LO filter 122 shown in Fig. 3(a). Between the input port 425 and the output port 426, there is a capacitive coupling section 427, two capacitance sections 428 and 430, and an inductance section 429 arranged between the capacitance sections 428 and 430. The capacitive coupling section 427 is capacitively coupled (AC coupled) by overlapping the conductor of the microstrip line with an insulator. This provides DC isolation between the input port 425 and the output port 426.

[0057] The capacitive coupling unit 427 must pass the LO signal and block the IF signal, which is in a low frequency band. The IF signal reaching the RTD resonating unit 102 can adversely affect the oscillation stabilization of the RTD resonating unit 102, which is an injection-locked oscillator. Therefore, the strength of the IF signal may be lower than the strength of the oscillation signal generated by the RTD resonating unit 102, for example, to one-hundredth or less. Therefore, the capacitive coupling unit 427 and the LO filter are configured to reduce the impedance in the IF signal frequency band (several GHz to several tens of GHz) to about one-tenth, or even one-hundredth or less, of the frequency band in which the RTD resonating unit 102 oscillates. Specifically, the capacitance of the capacitive coupling unit 427 is about 20 fF, and the capacitive coupling unit 427 can be configured to be considered conductive in the frequency band of the LO signal generated by the RTD resonating unit 102. The capacitive coupling unit 427 can also be provided at the connection between the resonant conductor 111 and the LO filter 122.

[0058] The capacitance section 428, inductance section 429, and capacitance section 430 of the LO filter 122, together with the capacitive coupling section 427, constitute a bandpass filter (BPF) using LC resonance. This filter is primarily configured as a narrowband filter for passing the LO signal's oscillation frequency of 0.35 THz. Narrowing the band of the LO filter serves to reduce the return of the IF signal from the data signal 105 and signals of multiple frequencies generated by the mixer 103 to the RTD resonating section 102. Here, by determining the line width so that the capacitance components of the capacitance sections 428 and 430 are approximately 50 to 100 fF and the inductance component of the inductance section 429 is approximately several pH, an MPF ​​tailored to the LO signal frequency f2 can be formed.

[0059] 4(d) shows a wideband BPF and an example configuration of the RF filter 124 shown in FIG. 3(a). Capacitive coupling units 433 and 435 are connected in series between an input port 431 and an output port 432. Short stubs 434 and 436 are arranged between the two capacitive coupling units 433 and 435, and between the capacitive coupling unit 435 and the output port 432, respectively. The capacitive coupling units 433 and 435 are configured such that a capacitance component due to capacitive coupling and an inductance component of a line are connected in series. Furthermore, the short stubs 434 and 436 can be used as inductance components because they have an inductance component between the short stubs 434 and 436 and the GND conductor 116 by adjusting the length of the arrow 117. The RF filter 124 transmits the RF signal output by the mixer 103 to the antenna 104, while cutting out unwanted signals incident from the antenna 104 and removing unwanted harmonic components such as intermodulation products generated in the mixer 103. Furthermore, by adjusting the impedance between the upstream mixer 103 and the downstream antenna 104, transmission loss is reduced. The capacitance components of the capacitive coupling units 433 and 435 are approximately 1 to 30 fF, and the inductance components of the signals flowing through the capacitive coupling units 433 and 435 are approximately several fF to several tens of fF. Furthermore, the inductance components of the short stubs 434 and 436 are approximately 10 to 30 pF. Using these values, the filter is designed to resonate in the RF frequency band. By connecting multiple similar components, a BPF with a wide passband of several tens of GHz or more can be configured.

[0060] Another possible filter configuration is the pattern shown in FIG. 4(e). Coupling of inductance component L and capacitance component C can be considered at various points with respect to a T-shaped portion 418 extending from a line 417. FIG. 4(f) shows a filter configuration using an open stub 419 and a radial stub 420. By appropriately setting the length, size, and position on the microstrip line from input port 410 to output port 411 of the stub, filtering can be performed according to the frequency band of the passing signal. In this embodiment, too, the impedance can be adjusted for a specific frequency by appropriately changing the width of the stub or pattern, the position on the line, and the like, and it is also possible to adjust the impedance before and after the filter.

[0061] An equivalent circuit of the semiconductor device 100 in this embodiment will be described with reference to FIG. 5. The RTD resonating unit 102 is connected to a bias power supply in a bias unit 500, and a drive voltage (bias voltage) is applied to the RTD resonating unit 102. The RTD resonating unit 102 includes an RTD oscillator 110, a resonant conductor 111 represented by impedance Z2, and the capacitance impedance of the RTD, all connected in parallel. The reference signal oscillator 101 is located on the far left in FIG. 5. The reference signal oscillator 101 is connected to a transmission line 501. The transmission line 501 is an equivalent circuit of a microstrip line that constitutes a connection between the LO port 120 and a connection section that transmits a reference signal that synchronizes the RTD resonating unit 102. The transmission line 501 has series components of parasitic inductances L1 and L2 and a transmission line impedance Z1, and parallel components of parasitic capacitances C1 and C2 and parasitic resistances R1 and R2.

[0062] The transmission line 501 and the RTD resonating unit 102 are connected via a capacitance C3. The capacitance C3 is electrically connected by AC coupling so as to pass the reference signal 108 having the frequency f1 of the reference signal oscillator 101. Specifically, the capacitance C3 can have a capacitance of about tens to hundreds of fF as a low-pass filter for passing a signal of 100 GHz to several hundred GHz output from the reference signal oscillator 101.

[0063] The RTD resonating unit 102 is electrically connected to the mixer 103. The internal configuration of the mixer 103 will be described. In the mixer 103, the RTD resonating unit 102 is connected to the diode unit 125 via an LO filter 122 having an impedance Z4. Furthermore, the IF connection port 121 is connected to the diode unit 125 via an IF filter 123 having an impedance Z5. The diode unit 125 is electrically connected to the antenna 104 via an RF filter 124 shown with an impedance Z6.

[0064] The frequency f1 of the reference signal 108 output from the reference signal oscillator 101 is set to half the frequency f2 of the LO signal output from the RTD resonator 102. The capacitance C3 can be configured to pass the frequency f1 of the reference signal and block the frequency f2 of the LO signal. As described above, the capacitance C3 can have a capacitance of approximately tens to hundreds of fF as a low-pass filter for passing a signal of 100 to several hundred GHz from the reference signal oscillator. For example, the capacitance C3 can be AC ​​coupling in which an insulator is sandwiched between the resonant conductor 111 of the RTD resonator 102 and the microstrip line of the transmission path 501. The impedances Z4 and Z5 are circuits that perform filtering and impedance matching and are arranged on the signal line connecting the RTD resonator 102 and the mixer 103. The impedances Z4 and Z5 are arranged to efficiently input and output the data signal, which is an IF signal from the IF connection port 121, and the LO signal input from the RTD resonator to the mixer 103. The various filters are configured to pass desired frequencies and block unwanted frequencies, and are connected using filters consisting of capacitance, fan-shaped radial stubs, microstrip lines, etc. Furthermore, a planar antenna such as a patch antenna is used as the antenna 104, and the antenna 104 (antenna conductor 140) is connected to the line at a position offset from the end of the patch antenna to achieve impedance matching with the line. The RF signal output from the RTD resonator 102 and modulated by the mixer 103 forms a standing wave on the antenna 104 and is radiated into space. A filter circuit or the like may be appropriately arranged on the stripline between the antenna 104 and the impedance Z6.

[0065] It is also possible to appropriately insert a filter circuit or the like around the RTD resonating unit 102. Here, the wavelength of the reference signal output by the reference signal oscillator 101 is λ f1 , the wavelength of the LO signal generated in the RTD resonator 102 is λ f2 For example, a λ f1A short stub of λ / 4 may be provided. By providing an appropriate short stub on the input side of the RTD resonating unit 102, the frequency f1 of the reference signal is ignored by the short stub, so that only the reference signal can be transmitted. On the other hand, an LO signal with a frequency twice that of the reference signal oscillated in the RTD resonating unit 102 has a frequency of λ f1 The length of / 4 is λ f2 / 2, it is shorted to GND, and therefore it is possible to prevent the LO signal from flowing back to the reference signal oscillator 101 side. Furthermore, λ is connected to the output side of the RTD resonator 102, the side connected to the modulation / demodulation unit (which can be a modulation unit when used as a transmitter, and a demodulation unit when used as a receiver). f1 A λ / 4 open stub may be arranged in the terahertz band. This creates a filter that operates in the opposite way to the short stub, preventing the transmission of the reference signal frequency f1 and allowing only the LO oscillation signal frequency f2, which is twice the frequency, to pass. The stub filter configuration can be based on technology used in microwave and millimeter-wave circuits, so it can be scaled and arranged appropriately to match terahertz band frequencies.

[0066] The reference signal oscillator 101, mixer 103, antenna 104, and associated peripheral circuits of the semiconductor device 100 described in this embodiment do not necessarily need to be arranged on the same semiconductor substrate. For example, a semiconductor device can be configured in which multiple components are integrated on separate semiconductor substrates. In this case, the semiconductor device 100 can be realized by connecting multiple semiconductor substrates using wire bonding, flip chip, or the like during the semiconductor substrate mounting process. The semiconductor device 100 can also be manufactured by bonding and combining semiconductor substrates in a semiconductor process, or by integrating multiple semiconductor substrates into a single package. Furthermore, the semiconductor device 100 may employ an Antenna In Package (AiP) configuration in which an antenna and semiconductor substrates are integrated into a single package.

[0067] A balun, a conversion circuit, a matching circuit, etc. may be appropriately arranged at the boundary between multiple semiconductor substrates. Furthermore, in the above-described configuration, no active elements are arranged in the path between the RTD resonator 102, which is the injection-locked oscillator, and the mixer 103, which is the modulator. However, this is not limiting. In addition to the above-described components, the peripheral circuit may also include amplifier circuits such as a power amplifier (PA) or a low-noise amplifier (LNA), phase adjustment circuits such as a phase shifter, and switches. The semiconductor substrate may be based on silicon or a compound semiconductor such as indium phosphide (InP) or gallium arsenide (GaAs). For example, the reference signal oscillator 101 may be formed on a silicon substrate, and the LO port 120 and subsequent stages (the antenna 104 side) shown in FIG. 3(a) may be formed on a compound semiconductor substrate. In this case, a silicon substrate on which reference signal oscillator 101 and the like are formed and a compound semiconductor substrate on which RTD resonating units 102a, 102b and the like are formed may be stacked to form semiconductor device 100. The configuration and mounting form of the semiconductor substrate of semiconductor device 100 to which the present invention can be applied are not limited to those described above. In either case, it is sufficient that the output signal of RTD resonating unit 102, which functions as an injection-locked oscillator based on the reference signal supplied from reference signal oscillator 101, can have a modulated data signal. In this way, it is possible to configure semiconductor device 100 to which the present disclosure is applied, which can realize a transceiver circuit using RTD resonating unit 102.

[0068] In the configuration of this embodiment, the use of the RTD resonator 102 makes it possible to efficiently oscillate a terahertz wave oscillation signal in a small area. Furthermore, by arranging the RTD resonator 102 near a circuit such as a mixer, it is possible to minimize losses in transmission lines, etc. In other words, the semiconductor device 100 can be made smaller and its high-frequency characteristics can be improved, thereby improving signal quality.

[0069] Next, a semiconductor device 600 according to a second embodiment of the present disclosure will be described with reference to FIGS. 6(a) and 6(b). FIG. 6(a) is a conceptual diagram illustrating the wiring of functional blocks in the semiconductor device 600 according to this embodiment. FIG. 6(b) is a diagram illustrating an outline of a resonant tunneling diode (RTD) resonator 102 and a modulation / demodulation unit 6100 in the semiconductor device 600 according to this embodiment. Unlike the configuration using the mixer 103 described above, the modulation / demodulation unit 6100 is connected to the RTD resonator 102 and outputs an RF signal 607 having a frequency f2. An FM-modulated signal 602 having a center frequency f3 is used as a data signal to be input to the modulation / demodulation unit 6100. The modulation / demodulation unit 6100 functions as a modulation unit when used as a transmitter, and as a demodulation unit when used as a receiver.

[0070] As in the above-described embodiment, the reference signal oscillator 101 is an oscillator that outputs a reference signal 108 of frequency f1. The frequency f1 of the reference signal 108 is a subharmonic frequency of the oscillation frequency of the RTD resonating unit 102. When the reference signal is input, the RTD resonating unit 102 outputs an LO signal 106 of frequency f2 that has a higher Q value than self-oscillation by subharmonic injection locking.

[0071] The modulation / demodulation unit 6100 used in this embodiment will be described with reference to Fig. 6(b). The modulation / demodulation unit 6100 is configured using a varactor diode mixer, and the varactor diode is used as an impedance variable element. The modulation / demodulation unit 6100 receives a variable impedance element having an effective wavelength λ f2 Depending on f2Impedances Z20 and Z21, which are quarter-wave lines, are connected in series. The terminal of impedance Z20 opposite the terminal connected to impedance Z21 is open and functions as an open stub. The terminal of impedance Z21 opposite the terminal connected to impedance Z20 is connected to resonant conductor 111 via AC coupling (capacitive coupling). A variable impedance circuit 6101 is connected to the connection between Z20 and Z21. In variable impedance circuit 6101, a capacitor C20, an impedance Z22, and a varactor diode serving as variable capacitance Cv are connected between GND and a node connecting impedance Z20 and impedance Z21. A signal line that supplies an FM modulated signal 602 as a data signal to adjust the capacitance of variable capacitance Cv passes through inductor L20 and is connected to the node connecting impedance Z22 and variable capacitance Cv.

[0072] The modulation / demodulation unit 6100 is connected to the RTD resonating unit 102 and can change the impedance of the resonant conductor 111. By changing the impedance of the resonant conductor 111 in accordance with the FM-modulated signal 602, the RTD resonating unit 102 can output an RF signal with varying oscillation phase and intensity. The RF signal is a signal obtained by combining (superimposing) the LO signal 106 and the FM-modulated signal 602, and the RF signal has the same frequency f2 as the LO signal. The variable impedance element is not limited to a varactor diode. A variable impedance circuit using a variable capacitor, a variable resistor, or a transistor may also be used as the variable impedance element. Therefore, the RTD resonating unit 102 may be modulated not only by amplitude but also by frequency or phase. The other circuit configurations of the semiconductor device 600 may be the same as those of the above-described embodiment, and therefore will not be described here. As in the above-described embodiment, an RF signal is input to the antenna 104, and terahertz waves TW of frequency f4 are radiated into space.

[0073] By applying this embodiment, the RTD resonating unit 102 and the modulation unit can be integrated, and the scale of the terahertz wave generating circuit and modulation circuit can be reduced, which means that the semiconductor device 600 can be further miniaturized.

[0074] Next, a semiconductor device 900 according to a third embodiment of the present disclosure will be described with reference to FIGS. 7(a) to 7(d), 8(a), and 8(b). FIG. 7(a) is a conceptual diagram illustrating the wiring of functional blocks in the semiconductor device 900 according to this embodiment. FIG. 7(b) is a diagram illustrating an outline of an RTD resonating unit 902 in the semiconductor device 900 according to this embodiment. FIG. 7(c) is a cross-sectional view taken along line A-A' in FIG. 7(b). FIG. 7(d) is a cross-sectional view taken along line B-B' in FIG. 7(b). FIG. 8(a) is a circuit diagram of the semiconductor device 900, and FIG. 8(b) is a modified example of the circuit diagram shown in FIG. 8(a). Unlike the above-described embodiments, the semiconductor device 900 has two RTD oscillators 110a and 110b arranged in the RTD resonating unit 902. Furthermore, unlike the above-described embodiments, the semiconductor device 900 superimposes a bias and an IF signal in the bias pattern 112 and supplies the superimposed signal to the RTD resonating unit 902, so that the RTD resonating unit 902, which is an injection-locked oscillator, also functions as a modulator.

[0075] Reference signal 108 of frequency f1 output from reference signal oscillator 101 is input to RTD resonating unit 902. RTD resonating unit 902 is injection locked by reference signal 108 and outputs RF signal 107 of frequency f4. Data signal 105 is also input to RTD resonating unit 902 from bias pattern 112, and amplitude modulation according to data signal 105 is synthesized (superimposed) on the amplitude of RF signal 107.

[0076] The RTD resonating unit 902 will be described in more detail using FIG. 7(b). In this embodiment, the RTD resonating unit 902 also functions as a modulation / demodulation unit (it can be a modulation unit when used as a transmitter, and a demodulation unit when used as a receiver). A data signal 105 for modulation is supplied to the bias pattern 112. The bias voltage supplied to the RTD resonating unit 902 is a direct current (DC) voltage that drives the RTD oscillators 110a and 110b, and is supplied via the bias pattern 112 as described above. A signal is supplied from the IF connection port 121 as an alternating current (AC) component to be modulated. For example, as shown in FIG. 7(b), the data signal 105 may be supplied as an AM signal using a single signal line, with the voltage amplitude of DC and AC superimposed as data. Alternatively, as will be described later, the bias (DC component) and the data signal (AC component) may be supplied separately to the bias pattern 112.

[0077] The resonant conductor 111, which is the upper conductor of the resonant structure of the RTD resonant unit 102, has a line length (resonator length) indicated by an arrow 118 shown in FIG. 7(b) that is equal to or longer than the wavelength λ of the signal oscillating in the RTD resonant unit 102. eff For λ eff Specifically, the wavelength λ0 in a vacuum of 0.5 THz is 0.6 mm, and the relative dielectric constant on the resonator is about 2, so λ eff The electrical length of λ / 2 is 0.15 mm. This line length may be used as a reference for design. As a result, the resonant conductor 111 resonates at a frequency in the terahertz band, and a resonant electric field is generated as a standing wave within the resonant conductor 111. The resonant electric field is generated in the direction of arrow 118, with antinodes (points of maximum amplitude) of the electric field at both ends of the resonant conductor 111 and nodes (points of zero amplitude) of the electric field at the center of the resonant conductor 111. In addition, λ eff Two RTD oscillators 110a and 110b are arranged on a resonant conductor 111 of 1 / 2 at positions facing each other with a node of the resonant electric field at the center. In this configuration, the two RTD oscillators 110a and 110b oscillate in push-pull mode. The push-pull mode is an oscillation mode in which the two RTD oscillators 110a and 110b arranged on the resonant conductor 111 oscillate in sync with their phases reversed (out of phase).

[0078] The layer structure of the semiconductor device 900 will be described with reference to FIGS. 7(c) and 7(d). As shown in FIG. 7(d), the bias pattern 112 is connected to the resonant conductor 111 through a via 113 for supplying bias. The bias pattern 112 may be arranged on a different layer from the resonant conductor 111. As shown in FIG. 7(c), the resonant conductor 111 is connected to the RTD oscillators 110a and 110b through vias 136a and 136b. As a result, the RTD oscillators 110a and 110b share the bias voltage and oscillate with the resonant conductor 111 sandwiching the dielectric layer 132. Furthermore, the amplitude of the LO signal output by the RTD resonating unit 902 can be changed by changing the AC component of the bias, and the RTD resonating unit 902 performs amplitude modulation. Here, the RTD oscillators 110a and 110b in the RTD resonating unit 902 have the characteristic that the amplitude of the signal they output changes depending on the bias voltage applied. The modulation / demodulation unit that modulates the oscillation signal of the RTD resonating unit 902 using the data signal 105 is not limited to this embodiment, and may be configured to form and connect a bias T circuit to the bias unit and bias pattern 112. The data signal line 904 may also be electrically connected directly to the resonant conductor 111, rather than to the bias pattern 112. In this case, the semiconductor device 900 can also be modulated by an impedance adjustment circuit that changes the impedance of the resonant conductor 111 using the connected data signal line. A variable impedance element such as a varactor diode can be used for the impedance adjustment circuit. In this case, the semiconductor device 900 can be modulated by controlling the impedance using the external data signal 105. The modulated signal is emitted into space from the connected antenna 104.

[0079] FIG. 8(a) is a circuit diagram of a semiconductor device 900, and FIG. 8(b) is a modified example of the circuit shown in FIG. 8(a). Unlike the first embodiment, the circuit constituting the mixer 103 is eliminated, and the RTD resonator 902 and modulator are integrated, thereby reducing the circuit size for generating and modulating terahertz waves. As shown in FIG. 8(a), in this embodiment, a port 905 to which a reference signal oscillator 101 is supplied is connected to the RTD resonator 102 via a capacitive coupling unit 906. The capacitive coupling unit 906 is a DC cut filter that prevents a DC bias from being input to the reference signal oscillator 101. A DC shunt 907 is also connected to suppress unnecessary parasitic oscillations occurring in the RTD resonator 102. The RTD resonator 902 includes two RTD oscillators 110a and 110b, with the DC shunt 907 connected between them. The connection of the DC shunt 907 is not limited to within the RTD resonator 102.

[0080] 8(b) shows a modified example in which the input terminal for the data signal 105 and the bias input terminal are separated, and an AC shunt 908 is connected to suppress parasitic oscillation in the RTD resonating unit 102. A capacitive coupling unit 906a is disposed at the connection between the port 905 and the RTD resonating unit 102. Similarly, a capacitive coupling unit 906b is disposed at the connection between the input terminal for the data signal 105 and the RTD resonating unit 102. The capacitive coupling units 906a and 906b are used as DC cut filters.

[0081] By applying this embodiment, the RTD resonating unit 902 and the modulation unit can be integrated, and the scale of the terahertz wave generating circuit and modulation circuit can be reduced, which means that the semiconductor device 900 can be made smaller.

[0082] Next, a semiconductor device 1000 according to a fourth embodiment of the present disclosure will be described with reference to FIGS. 9(a) to 9(d). FIG. 9(a) is an equivalent circuit diagram of the semiconductor device 1000 according to this embodiment. FIG. 9(b) is a top view showing a configuration example of the semiconductor device 1000 according to this embodiment. FIG. 9(c) is a cross-sectional view taken along line A-A' in FIG. 9(b). FIG. 9(d) is a cross-sectional view taken along line B-B' in FIG. 9(b). The semiconductor device 1000 according to this embodiment does not include a mixer circuit. Furthermore, this embodiment uses a patch antenna 1004 as an antenna, and the patch antenna 1004 is connected to the RTD oscillators 110a and 110b as a resonant conductor constituting the resonant structure of the RTD resonating unit 1002. In other words, the patch antenna 1004 is an active antenna that is integrated with the RTD resonating unit 102, which is an injection-locked oscillator, and that oscillates and radiates terahertz waves. It can be said that the patch antenna 1004 constitutes a part of the RTD resonating unit 102. In the above points, this embodiment differs from the first embodiment described above.

[0083] The RTD resonating unit 1002 of this embodiment will be described with reference to FIG. 9(a). As in the above-described embodiments, a reference signal having a frequency f1 is supplied to the RTD resonating unit 1002 from the reference signal oscillator 101. A data signal having a frequency f3 is also input to the RTD resonating unit 1002. The RTD resonating unit 1002 modulates the LO signal output from the RTD resonating unit 1002 with the data signal, and outputs an RF signal having a frequency f4. The RTD resonating unit 1002 includes a patch antenna 1004, RTD oscillators 110a and 110b, an AC shunt 908, and a capacitive coupling unit 906. A bias voltage is applied to the RTD resonating unit 1002 via a bias pattern 112, and the RTD resonating unit 1002 oscillates terahertz waves. An IF signal (data signal) is also input to the RTD resonating unit 1002 via a data signal line 1005, and the RTD resonating unit 1002 performs modulation, oscillation, and radiation. The functions of the RTD oscillators 110a and 110b, the capacitive coupling section 906, and the AC shunt 908 are the same as those described above.

[0084] 9(b), the connection relationships will be described, focusing on the patch antenna 1004. The patch antenna 1004 is connected to the RTD oscillators 110a and 110b. The patch antenna 1004 is also connected to the bias pattern 112 through vias 113a and 113b. The patch antenna 1004 is further connected to the conductor pattern 1006 and the data signal line 1005 via capacitive coupling. The AC shunt 908 is connected to the bias pattern 112 through a via 1034. The bias pattern 112 is connected to the DC input. The conductor pattern 1006 is connected to the reference signal oscillator 101. The data signal line 1005 is connected to the IF input. In other words, the RTD resonator 1002 as a whole is a three-terminal device, and is a semiconductor device that receives inputs from the DC input, the reference signal oscillator 101, and the IF input, and oscillates, modulates, and emits terahertz waves.

[0085] The resonant structure of the RTD resonating unit 102 is a microstrip resonator, and includes a GND conductor 116, which is a wide lower conductor, a patch antenna 1004, which is an upper conductor, and a dielectric (dielectric layers 131-133) disposed between the lower and upper conductors. The GND conductor 116 is also called a reflector layer. The RTD oscillators 110a and 110b are electrically connected to the upper and lower conductors. The RTD oscillators 110a and 110b are connected to the GND conductor 116, which is the lower conductor, and are connected to the patch antenna 1004, which is the upper conductor, via vias 136a and 136b. In this embodiment, the vias 136a and 136b have a resistivity of 1×10 -6 Materials with a resistivity of Ω·m or less may be used. Specific materials for the vias 136a and 136b may include metals and metal compounds such as Ag, Au, Cu, W, Ni, Cr, Ti, Al, AuIn alloys, and TiN. Each component, such as the lower electrode, upper electrode, and vias, is formed from conductive materials including metals and semiconductors with high carrier concentrations, depending on the applicable process.

[0086] An insulating layer such as a passivation layer 1033 for protecting the semiconductor device 1000 may be disposed on the patch antenna 1004 and the dielectric layer 133. A bias pattern 112 is disposed between the dielectric layer 132 and the dielectric layer 133, and the bias pattern 112 is connected to the patch antenna 1004 through vias 113a and 113b for bias supply. These structures are formed on the semiconductor substrate 130 using a semiconductor process.

[0087] The patch antenna 1004 of the upper conductor has a line length (resonator length) indicated by an arrow 1016 in FIG. 9(b) that is equal to the wavelength λ of the signal oscillated in the RTD resonating unit 1002. eff For λ eff Specifically, the wavelength λ0 in a vacuum of 0.5 THz is 0.6 mm, and the relative dielectric constant on the resonator is about 2, so λ eff The electrical length of λ / 2 is 0.15 mm. This line length may be used as a reference for design. As a result, the patch antenna 1004 resonates at a frequency in the terahertz band, and a resonant electric field is generated as a standing wave within the patch antenna 1004. The resonant electric field is generated in the direction of the arrow 1016, with antinodes (points of maximum amplitude) of the electric field at both ends of the patch antenna 1004 and nodes (points of zero amplitude) of the electric field at the center of the patch antenna 1004. In addition, λ eff Two RTD oscillators 110a and 110b are arranged on the 1 / 2 patch antenna 1004 at positions facing each other with the node of the resonant electric field at the center. In this configuration, the two RTD oscillators 110a and 110b oscillate in push-pull mode. The push-pull mode is an oscillation mode in which the two RTD oscillators 110a and 110b arranged on the resonant conductor 111 oscillate in sync with their phases reversed (out of phase).

[0088] Furthermore, the width of the connection portion of bias pattern 112 connected to bias supply vias 113a and 113b is smaller (thinner) than the width of arrow 1016 where a standing wave occurs in patch antenna 1004. This width may be 1 / 10 or less of the effective wavelength λ (λ / 10 or less) of the terahertz band signal standing in patch antenna 1004. This is because, in order to improve resonance efficiency, it is appropriate to arrange bias supply via 113 and bias pattern 112 in a size and position that does not interfere with the resonant electric field in patch antenna 1004. In addition, bias supply vias 113a and 113b are arranged in the center, which is a node of the standing wave in patch antenna 1004. By making the connection point of bias supply vias 113a and 113b the node of the standing wave in patch antenna 1004, the impedance of the oscillation signal at the connection point is maximized. Therefore, it is possible to minimize loss caused by leakage of the oscillation signal in patch antenna 1004 to the bias line from bias supply vias 113a and 113b. The bias unit is appropriately adjusted to efficiently synchronize reference signal oscillator 101 and RTD resonator 102.

[0089] Here, the number of RTD oscillators 110 arranged in the RTD resonating unit 102 may be one, or may be three or more. The more RTD oscillators 110 there are, the stronger the oscillation intensity becomes, but it is necessary to address issues such as power consumption, variations between elements, and suppression of parasitic oscillation. Furthermore, the patch antenna 1004 is not limited to a rectangular patch antenna as shown in FIG. 9(b), and a slot antenna, a dipole antenna, a loop antenna, or various other planar antennas may be used.

[0090] The conductor pattern 1006 and the data signal line 1005 are conductors called microstrip lines. If the width of this microstrip line is too large, the resonance characteristics of the patch antenna 1004 may deteriorate and the radiation efficiency may decrease due to an increase in parasitic capacitance. Therefore, the width of the microstrip line is set to a size that does not interfere with the resonant electric field, typically, a size that does not exceed the oscillation frequency f that exists in the RTD resonant portion 1002. THzThe width of the conductor pattern 1006 and the data signal line 1005 may be small enough not to increase the series resistance, for example, to about twice the skin depth. To reduce the series resistance to a value not exceeding 1 Ω, the width of the conductor pattern 1006 and the data signal line 1005 may typically be in the range of 0.1 μm to 20 μm.

[0091] The bias pattern 112 is also electrically connected to the AC shunt section 1036. The AC shunt section 1036 uses an MIM structure as shown in FIG. 9(b). Specifically, the conductor layer of the GND conductor 116 and the MIM conductor pattern 1035 are connected by capacitive coupling via the dielectric layer 131. The GND conductor 116 is connected to the GND at a position (not shown) on the semiconductor substrate 130.

[0092] Changes in the AC component of the bias voltage change the amplitude of the signal oscillated by the RTD resonating unit 102, which is injection-locked to the reference signal from the reference signal oscillator 101. This results in amplitude modulation. The RTD oscillators 110a and 110b in the RTD resonating unit 102 have the characteristic that the amplitude of the signal they output changes depending on the bias voltage applied. In this embodiment, modulation is performed using the AC shunt 1036 provided in the RTD resonating unit 1002 and the characteristics of the RTD oscillators 110a and 110b, whose signal output, such as amplitude, changes depending on the bias voltage. The RTD resonating unit 102 synchronizes with the reference signal from the reference signal oscillator 101 and modulates in response to changes in the bias voltage from the bias pattern 112. Therefore, the injection power of the reference signal from the reference signal oscillator 101 must be strong enough to prevent the lock from being lost due to modulation. Specifically, in the graph of FIG. 2(a), the bias voltage in the locking range was 0.69 [V] to 0.71 [V]. Therefore, the amplitude of the data signal is set so that the amplitude of the bias voltage superimposed on the data signal is 0.7±0.01 V. Because this locking range can be controlled by the injected power, it is necessary to appropriately set the locking range of the reference signal and the amplitude of the data signal.

[0093] The configuration for modulating the oscillation signals of the RTD oscillators 110a and 110b with a data signal is not limited to this embodiment. For example, a general bias-T circuit can be formed and connected to the bias pattern 112. The bias-T circuit can support similar modulation by connecting its direct current (DC) side to a bias power supply, its alternating current (AC) side to a data signal, and its AC / DC output side to the RTD resonator 1002. The data signal line 1005 can also be electrically connected to the bias layer via AC coupling. In this case, the connection can be made in a manner that creates different impedances or has capacitance components in two frequency bands: the LO frequency band oscillated by the RTD resonator 102 and the IF frequency band of the data signal. Modulation can also be achieved by an impedance adjustment unit, such as a data signal line connected to the patch antenna 1004 to change the impedance of the patch antenna 1004. The impedance adjustment unit can control and change the impedance using an external data signal by using an impedance variable element such as a varactor diode.

[0094] By applying this embodiment, the RTD resonating unit 1002 and the modulator can be integrated. Furthermore, the patch antenna 1004 also functions as a resonant conductor for the RTD resonating unit 1002. This makes it possible to reduce the scale of the terahertz wave generating circuit and the modulator circuit. In other words, the semiconductor device 1000 can be made smaller.

[0095] Next, semiconductor devices 700 and 7000 according to a fifth embodiment of the present disclosure will be described with reference to FIGS. 10(a) to 10(c) and FIGS. 11(a) and 11(b). FIG. 10(a) is a conceptual diagram illustrating the wiring of functional blocks of the semiconductor device 700 according to this embodiment. The semiconductor device 700 includes, in addition to the configuration of the semiconductor device 100 according to the first embodiment, a phase adjuster 701 for adjusting the phase of a reference signal between the reference signal oscillator 101 and the RTD resonator 102, which is the injection-locked oscillator. The semiconductor device 700 according to this embodiment can control the phase of the terahertz wave TW emitted from the semiconductor device 700 by using the phase adjuster 701. In this embodiment, a reference signal 710 having a frequency f1 is supplied from the reference signal oscillator 101 to the phase adjuster 701. The phase adjuster 701 supplies a phase-controlled reference signal 709 having a frequency f1 to the RTD resonator 102.

[0096] Phase adjuster 701 can have a function of delaying the phase of reference signal 710 by a set amount. In the configuration shown in Fig. 10(b), reference signal 710 output from reference signal oscillator 101 is set as the phase reference (0°), and the phase can be shifted by π / 4 (90°).

[0097] The phase adjuster 701 has three terminals: an input port 703, an output port 704, and an output port 705. For example, a line 707 on the output port 705 side is connected to a line 706 on the output port 704 side, and the line 707 is connected to an effective wavelength λ of the input reference signal. f1 For λ f1 10(b), when reference signal 710 passes through line 706, it is output from output port 704 as reference signal 709 whose phase is delayed by 90° from reference signal 710. When reference signal 710 passes through line 707, it is output from output port 705 as reference signal 709 whose phase is delayed by 180° from reference signal 710. This makes it possible to obtain reference signal 709 whose phase is 90° different from reference signal 710 input from input port 703.

[0098] The input port 703 is connected to the reference signal oscillator 101. By switching the switch 708 arranged in the phase adjuster 701, the output port 704 or the output port 705 is connected to the RTD resonating unit 102. The phase adjuster 701 has a plurality of paths with different amounts of phase change of the reference signal 710, thereby performing phase control on the reference signal 710 and outputting the reference signal 709. The combination of the output ports 704, 705 and the lines 706, 707 is not limited to two, and may be arranged in the phase adjuster 701 according to the number of phases to be set, thereby enabling even finer phase control.

[0099] Furthermore, phase control is possible by using, for example, a transistor for switch 708 of phase adjuster 701, inputting a control signal to the transistor, and selecting a line based on the control signal. The above configuration makes it possible to control the phase delay according to the control signal, and semiconductor device 700 enables external time control and time synchronization between the transmitting side and the receiving side.

[0100] The reference signal 709 output from the phase adjuster 701 is input to the RTD resonating unit 102. The RTD resonating unit 102 performs injection-locked oscillation in accordance with the reference signal 709, thereby outputting an LO signal 106 of frequency f2 in the terahertz band, which has a phase difference set by the phase adjuster 701. The LO signal 106 is input to the mixer 103, which outputs an RF signal 107. The RF signal 107, like the LO signal, has a phase difference set by the phase adjuster 701. The RF signal 107 is radiated into space from the antenna 104 as a terahertz wave TW. This terahertz wave TW also has a phase difference set by the phase adjuster 701.

[0101] By arranging multiple semiconductor devices 700 in parallel and emitting terahertz waves TW that have been phase-locked or phase-controlled between adjacent antennas, it is possible to combine the terahertz waves TW. The radiation angle of the combined terahertz waves TW changes depending on the controlled phase. This phenomenon is called beamforming. The principle of beamforming will now be described. As an example, a configuration including three semiconductor devices 700 as shown in FIG. 10(c) will be described.

[0102] When three antennas 104 having the same distance d between them are oscillated with the same phase difference φ=φ1=φ3, the radiation angle is determined by the following equation (4). sinθ=(λ / d)×φ / 2π ··· (4) This radiation angle can be changed by the antenna shape, antenna type, and array arrangement.

[0103] In this embodiment, by controlling each phase adjuster 701 provided in multiple semiconductor devices 700 to set the same phase difference, terahertz waves TW having the same phase difference are emitted between the multiple semiconductor devices 700, and the emission angle is controlled.

[0104] FIG. 11(a) shows a semiconductor device 7000 having multiple active antennas AA, each of which includes an RTD resonating unit 7002 that is an injection-locked oscillator and a patch antenna. FIG. 11(a) shows a block diagram of three of n active antennas AA (n is a natural number). The active antennas AA include an RTD resonating unit 7002 that also functions as a modulator and a patch antenna 1004, and may have a configuration similar to that of the RTD resonating unit 1002 shown in FIG. 9(b) described above. A phase adjuster 701 is arranged between the reference signal oscillator 101 and the RTD resonating unit 7002. The active antennas AA1 to AAn each have a similar configuration, and a common data signal line 7005 is connected to these active antennas.

[0105] The data signal line 7005 is connected to the RTD resonating unit 7002. A reference signal 710 supplied from the reference signal oscillator 101 is input to the RTD resonating unit 7002 as a phase-adjusted reference signal 709 via a phase adjuster 701. Although not shown in FIG. 11(a), the multiple active antennas AA may include an active antenna AA to which the reference signal 710 is supplied to the RTD resonating unit 7002 without passing through the phase adjuster 701. The active antennas AA1 to AAn are configured to receive the same data signal, and therefore can radiate the same RF signal as a terahertz wave TW into space.

[0106] FIG. 11(b) is a top view of the semiconductor device 7000. The semiconductor device 7000 forms an antenna array, with four active antennas AA1 to AA4 arranged in a 2×2 matrix. Each of these active antennas AA1 to AA4 radiates an RF signal represented by a terahertz wave TW into space. The number of active antennas AA is not limited to four, and the active antennas may be arranged in a 3×3 matrix or a 2×4 matrix. A transmitter to which the present disclosure is applied can be configured with M×N active antennas (M and N are natural numbers). The RF signals radiated from these active antennas are signals oscillated by an RTD resonator 7002 synchronized with a reference signal 709 (710) supplied from a reference signal oscillator 101. This allows the RF signals of the active antennas AA1 to AA4 to be synchronized with high precision. This allows terahertz waves of higher intensity to be radiated into space than a single active antenna. This improves the gain of the antenna array.

[0107] Furthermore, adjusting the phase of the reference signal 710 is not limited to using the phase adjuster 701. In this embodiment, a common bias pattern 112 is used to supply power to the RTD resonating unit 7002 arranged in each active antenna AA. However, a bias voltage may be supplied to the RTD resonating unit 7002 arranged in each active antenna AA using an individual bias pattern. In this case, overcrowding of the wiring pattern can be avoided by forming a through via in a semiconductor substrate or the like and stacking it on a CMOS integrated circuit. As shown in FIG. 2(b), changing the bias voltage can change the phase of the RTD resonating unit within the locking range. In other words, the phase of each active antenna AA can be changed depending on the frequency difference between the actual oscillation frequency locked by the reference signal of the reference signal oscillator 101 and the self-oscillation frequency determined by the bias voltage of the RTD resonating unit 7002. For example, the frequency of self-oscillation of the RTD resonator 7002, which is an injection-locked oscillator arranged in the active antenna AA1, and the frequency of self-oscillation of the RTD resonator 7002, which is an injection-locked oscillator arranged in the active antenna AA2, may be made different from each other. This makes it possible to perform beamforming in active antenna arrays arranged on one-dimensional and two-dimensional planes (two-dimensional arrays).

[0108] In the semiconductor devices 700 and 7000 of the present embodiment, the positional accuracy of the distance d between the antennas 104 and 1104 can be set with an accuracy sufficiently small in terms of the wavelength of the terahertz band, thereby improving the accuracy of the radiation angle. Furthermore, if the control signal is a data signal, it can have a phase modulation function and a communication function. The arrangement and order of the components in each of the above-described embodiments are one aspect for realizing the present disclosure, and the connection configuration to which the present disclosure can be applied is not limited to this embodiment.

[0109] Other embodiments Although the embodiments of the present invention have been described above, the present invention is not limited to these embodiments, and various modifications and changes are possible within the scope of the gist of the present invention.

[0110] For example, while the above-described embodiment describes a transmitter in a wireless communication device as an example, the application is not limited thereto and may be, for example, a radar device. FMCW radar requires constant frequency sweeping. The output frequency of the reference signal oscillator 101 may be swept using the circuit of the present disclosure, as shown in FIG. 1, and an oscillation signal may be generated and emitted by the RTD resonator 102. In this case, problems may arise, such as an increase in the frequency sweep width causing injection locking to be lost or a change in the phase of the oscillation signal from the RTD resonator 102 due to a frequency difference with self-oscillation. Frequency and phase instability can be suppressed by appropriately adjusting the bias voltage supplied to the RTD resonator 102 according to the frequency of the signal output by the reference signal oscillator 101.

[0111] Although the above-described embodiment uses a square patch antenna as the terahertz wave antenna, the shape of the antenna is not limited to this. For example, a rectangular, triangular, polygonal, circular, or elliptical patch conductor, a planar antenna such as a loop antenna, a log-periodic antenna, or a Vivaldi antenna, or a horn antenna may be used as the antenna.

[0112] In addition, although one antenna is associated with one RTD resonator in the above description, connections such as 1:N, N:1, or N:N (N is a natural number) may also be used. Additionally, although the structure of the RTD resonator has been described using a laminated structure, the present invention is not limited to this. In other words, the above discussion can also be applied to oscillator devices that do not use a laminated structure.

[0113] In addition, the following combinations may be used as materials for the RTD oscillator. GaAs / AlGaAs and GaAs / AlAs, In GaAs / GaAs / AlAs InGaAs / InAlAs, InGaAs / AlAs, In formed on InP substrate GaAs / AlGaAsSb InAs / AlAsSb and InAs / AlSb grown on InAs substrates ·SiGe / SiGe formed on a Si substrate The above-mentioned structure and materials can be appropriately selected depending on the desired frequency and the like.

[0114] Next, a case where the semiconductor device of each of the above-described embodiments is applied to a terahertz camera system (imaging system) will be described. The following description will be made with reference to FIG. 12(a). The terahertz camera system 1100 includes a transmitter 1101 that emits terahertz waves TW and a receiver 1102 that detects the terahertz waves TW. The terahertz camera system 1100 further includes a controller 1103 that controls the operation of the transmitter 1101 and the receiver 1102 based on an external signal, and processes or externally outputs an image based on the detected terahertz waves. The semiconductor device of each of the embodiments may be the transmitter 1101 or the receiver 1102. The terahertz waves emitted from the transmitter 1101 are reflected by a subject 1105 (object) and detected by the receiver 1102. A camera system including such a transmitter 1101 and receiver 1102 may also be called an active reflection imaging camera system. The camera system including the transmitter 1101 and the receiver 1102 can also be applied to a transmission imaging system in which the transmitter 1101 and the receiver 1102 are opposed to each other and a subject is placed between them to observe terahertz waves that have passed through the subject. Furthermore, in a passive camera system in which the transmitter 1101 is not provided, the semiconductor device of each of the above-described embodiments can be used as the receiver 1102. Furthermore, by using the semiconductor device of each of the embodiments capable of beamforming, it is possible to improve the detection sensitivity of the camera system and obtain high-quality images.

[0115] Furthermore, a case where any of the devices according to the above-described embodiments is applied to a terahertz communication system (communication device) will be described below. The following description will be made with reference to FIG. 12(b). The semiconductor device can be used as a component of the communication system. Possible communication systems include a simple ASK system, a superheterodyne system, and a direct conversion system. A superheterodyne communication system includes, for example, an antenna 1200, an amplifier 1201, a mixer 1202, a filter 1203, a mixer 1204, a converter 1205, a digital baseband modulator / demodulator 1206, and local oscillators 1207 and 1208. The RTD resonator 102 of the above-described embodiment can be used as the local oscillator 1207, the antenna 104 as the antenna 1200, and the mixer 103 of FIG. 5 as the mixer 1202. In the receiver, terahertz waves received via antenna 1200 are converted into an intermediate frequency signal by mixer 1202, then converted into a baseband signal by mixer 1204, and then converted from an analog waveform to a digital waveform by converter 1205. The digital waveform is then demodulated at baseband to obtain a communication signal. In the transmitter, the communication signal is modulated and then converted from a digital waveform to an analog waveform by converter 1205, then frequency converted via mixers 1204 and 1202, and output as terahertz waves from antenna 1200. The direct conversion communication system includes antenna 1200, amplifier 1211, mixer 1212, modulator / demodulator 1213, and local oscillator 1214. In the direct conversion system, during reception, the mixer 1212 directly converts the received terahertz waves into a baseband signal, and during transmission, the mixer 1212 converts the baseband signal to be transmitted into a terahertz signal. Other configurations are similar to those of the superheterodyne system. The devices according to the above-described embodiments can perform beamforming of terahertz waves by electrically controlling semiconductor devices. This allows for alignment of radio waves between the transmitter and receiver.Therefore, by using the semiconductor device of each embodiment capable of beamforming, wireless quality such as signal-to-noise ratio can be improved in a communication system, and large-capacity information can be transmitted over a wide coverage area at low cost. Furthermore, if frequency sweeping can be performed using the same local oscillator for transmission and reception, it is possible to measure distance from delay and phase information of a transmitted signal (radiated wave) and a reflected wave as an FMCW radar device. By applying the semiconductor device of the present disclosure, wireless quality such as signal-to-noise ratio can be improved, and ranging accuracy can be improved, all at low cost.

[0116] The disclosure of this specification includes the following semiconductor devices, communication devices, imaging systems, and radar devices.

[0117] (Item 1) a reference signal oscillator generating a first signal at a first frequency; an injection-locked oscillator that generates a second signal having a second frequency in the terahertz band in synchronization with the first signal; a modulation unit that generates a fourth signal at a fourth frequency from the second signal and a third signal at a third frequency corresponding to a signal supplied from a baseband circuit; A semiconductor device comprising:

[0118] (Item 2) the second frequency is higher than the first frequency, 2. The semiconductor device according to item 1, wherein the fourth frequency is equal to or higher than the second frequency.

[0119] (Item 3) 3. The semiconductor device according to claim 1, wherein the third frequency is higher than the first frequency.

[0120] (Item 4) 4. The semiconductor device according to any one of items 1 to 3, wherein the third signal is an IF signal or an FM modulated signal.

[0121] (Item 5) 5. The semiconductor device according to any one of items 1 to 4, wherein the injection-locked oscillator includes a resonant tunneling diode.

[0122] (Item 6) 6. The semiconductor device according to any one of items 1 to 5, wherein the injection-locked oscillator comprises a microstrip line having a specific impedance for oscillating in the terahertz band.

[0123] (Item 7) 7. The semiconductor device according to item 6, wherein the resonant structure using the microstrip line of the injection-locked oscillator includes a resonant conductor, a bottom conductor, and a dielectric disposed between the resonant conductor and the bottom conductor.

[0124] (Item 8) 8. The semiconductor device according to item 7, wherein the resonant conductor comprises a λ / 4 open stub.

[0125] (Item 9) a bias pattern for supplying an electrode to the injection-locked oscillator is disposed on the dielectric; the bias pattern is connected to the resonator conductor through a via disposed in the dielectric; Item 9. The semiconductor device according to item 7 or 8, wherein the width of the connection portion of the bias pattern connected to the via is smaller than the width of the resonant conductor in the direction in which a standing wave is generated.

[0126] (Item 10) 10. The semiconductor device according to item 9, wherein an AC shunt is connected to the bias pattern.

[0127] (Item 11) The third signal is supplied to the bias pattern; Item 11. The semiconductor device according to item 9 or 10, characterized in that the injection-locked oscillator functions as the modulation unit by supplying the third signal superimposed on a bias voltage from the bias pattern to the injection-locked oscillator.

[0128] (Item 12) 12. The semiconductor device of claim 11, wherein the third signal is amplitude modulated.

[0129] (Item 13) 11. The semiconductor device according to item 9 or 10, wherein the modulation section includes an impedance variable circuit for changing the impedance of the resonant conductor in response to the third signal.

[0130] (Item 14) 11. The semiconductor device according to any one of items 1 to 10, wherein the modulation section includes a mixer.

[0131] (Item 15) Item 15. The semiconductor device according to item 14, wherein the modulation section comprises a microstrip line having a specific impedance.

[0132] (Item 16) 16. The semiconductor device according to item 14 or 15, wherein a signal line for supplying the third signal is electrically connected between the injection-locked oscillator and the modulation section.

[0133] (Item 17) 17. The semiconductor device according to any one of items 14 to 16, characterized in that a first filter is disposed between the injection-locked oscillator and the modulation section to pass the second signal and to reduce the input of the fourth signal from the modulation section to the injection-locked oscillator.

[0134] (Item 18) 18. The semiconductor device according to any one of items 1 to 17, further comprising an antenna for radiating the fourth signal into space.

[0135] (Item 19) a second filter is disposed between the injection-locked oscillator and the antenna; a third filter is disposed between the reference signal oscillator and the injection-locked oscillator; Item 19. The semiconductor device of item 18, wherein the third filter passes signals having a lower frequency than signals passed by the second filter.

[0136] (Item 20) further comprising an antenna for radiating the fourth signal into space; a second filter is disposed between the injection-locked oscillator and the antenna; a third filter is disposed between the reference signal oscillator and the injection-locked oscillator; the first filter passes signals having a lower frequency than signals passed by the second filter; Item 18. The semiconductor device according to item 17, wherein the third filter passes signals having a lower frequency than the signals passed by the first filter.

[0137] (Item 21) 21. The semiconductor device of any one of items 18 to 20, wherein the antenna includes a patch antenna.

[0138] (Item 22) 22. The semiconductor device according to item 21, wherein the patch antenna forms part of the injection-locked oscillator.

[0139] (Item 23) 23. The semiconductor device according to any one of items 18 to 22, wherein a plurality of active antennas each including the injection-locked oscillator and the antenna are arranged in a two-dimensional array.

[0140] (Item 24) the plurality of active antennas includes a first active antenna and a second active antenna; 24. The semiconductor device according to item 23, characterized in that the frequency of self-oscillation of the injection-locked oscillator arranged in the first active antenna and the frequency of self-oscillation of the injection-locked oscillator arranged in the second active antenna operate so as to be different from each other.

[0141] (Item 25) 25. The semiconductor device according to any one of items 1 to 24, wherein the first frequency is a subharmonic frequency of the second frequency.

[0142] (Item 26) further comprising a first semiconductor substrate on which the reference signal oscillator is disposed and a second semiconductor substrate on which the injection-locked oscillator is disposed; 26. The semiconductor device according to any one of items 1 to 25, wherein the first semiconductor substrate and the second semiconductor substrate are at least partially stacked.

[0143] (Item 27) 27. The semiconductor device according to any one of items 1 to 26, characterized in that a phase adjuster for adjusting the phase of the first signal is arranged between the reference signal oscillator and the injection-locked oscillator.

[0144] (Item 28) 28. The semiconductor device of claim 27, wherein the phase adjuster delays the phase of the first signal.

[0145] (Item 29) 29. The semiconductor device according to item 27 or 28, wherein the phase adjuster has a plurality of paths with different amounts of change in the phase of the first signal.

[0146] (Item 30) A transmitter including the semiconductor device according to any one of items 1 to 29; a receiving section for detecting a signal emitted from the transmitting section.

[0147] (Item 31) a transmitter that includes the semiconductor device according to any one of items 1 to 29 and emits a signal toward a subject; a receiving section that detects the signal reflected from or transmitted through the subject.

[0148] (Item 32) A transmitter including the semiconductor device according to any one of items 1 to 29; a receiving unit that detects a wave that is emitted from the transmitting unit and reflected by an object, A radar device that measures distance from the radiation wave emitted from the transmitting unit and the reflected wave.

[0149] The invention is not limited to the above-described embodiments, and various changes and modifications can be made without departing from the spirit and scope of the invention. Accordingly, the following claims are appended to apprise the public of the scope of the invention. [Explanation of symbols]

[0150] 100, 600, 700, 900, 1000, 7000: Semiconductor device, 101: Reference signal oscillator

Claims

1. a reference signal oscillator generating a first signal at a first frequency; an injection-locked oscillator that generates a second signal having a second frequency in the terahertz band in synchronization with the first signal; a modulation unit that generates a fourth signal at a fourth frequency from the second signal and a third signal at a third frequency corresponding to a signal supplied from a baseband circuit; A semiconductor device comprising:

2. the second frequency is higher than the first frequency, 2. The semiconductor device according to claim 1, wherein the fourth frequency is equal to or greater than the second frequency.

3. 2. The semiconductor device according to claim 1, wherein the third frequency is higher than the first frequency.

4. 2. The semiconductor device according to claim 1, wherein the third signal is an IF signal or an FM modulated signal.

5. 10. The semiconductor device of claim 1, wherein the injection-locked oscillator comprises a resonant tunneling diode.

6. 2. The semiconductor device according to claim 1, wherein the injection-locked oscillator comprises a microstrip line having a specific impedance for oscillating in the terahertz band.

7. 7. The semiconductor device according to claim 6, wherein the resonant structure using the microstrip line of the injection-locked oscillator includes a resonant conductor, a bottom conductor, and a dielectric disposed between the resonant conductor and the bottom conductor.

8. 8. The semiconductor device according to claim 7, wherein the resonant conductor comprises a λ / 4 open stub.

9. a bias pattern for supplying an electrode to the injection-locked oscillator is disposed on the dielectric; the bias pattern is connected to the resonator conductor through a via disposed in the dielectric; 8. The semiconductor device according to claim 7, wherein a width of a connection portion of the bias pattern that connects to the via is smaller than a width of the resonant conductor in a direction in which a standing wave is generated.

10. 10. The semiconductor device according to claim 9, wherein an AC shunt is connected to the bias pattern.

11. The third signal is supplied to the bias pattern; 10. The semiconductor device according to claim 9, wherein the third signal superimposed on a bias voltage is supplied from the bias pattern to the injection-locked oscillator, whereby the injection-locked oscillator functions as the modulation section.

12. 12. The semiconductor device of claim 11, wherein the third signal is amplitude modulated.

13. 10. The semiconductor device according to claim 9, wherein the modulation section includes an impedance variable circuit for changing the impedance of the resonant conductor in response to the third signal.

14. 2. The semiconductor device according to claim 1, wherein the modulation section includes a mixer.

15. 15. The semiconductor device of claim 14, wherein the modulation section comprises a microstrip line having a specific impedance.

16. 15. The semiconductor device according to claim 14, wherein a signal line for supplying the third signal is electrically connected between the injection-locked oscillator and the modulation section.

17. 15. The semiconductor device according to claim 14, wherein a first filter is disposed between the injection-locked oscillator and the modulation section to pass the second signal and to reduce the input of the fourth signal from the modulation section to the injection-locked oscillator.

18. 2. The semiconductor device according to claim 1, further comprising an antenna for radiating the fourth signal into space.

19. a second filter is disposed between the injection-locked oscillator and the antenna; a third filter is disposed between the reference signal oscillator and the injection-locked oscillator; 20. The semiconductor device of claim 18, wherein the third filter passes signals having a lower frequency than signals passed by the second filter.

20. further comprising an antenna for radiating the fourth signal into space; a second filter is disposed between the injection-locked oscillator and the antenna; a third filter is disposed between the reference signal oscillator and the injection-locked oscillator; the first filter passes signals having a lower frequency than signals passed by the second filter; 18. The semiconductor device of claim 17, wherein the third filter passes signals having a lower frequency than signals passed by the first filter.

21. 20. The semiconductor device of claim 18, wherein the antenna comprises a patch antenna.

22. 22. The semiconductor device of claim 21, wherein the patch antenna forms part of the injection-locked oscillator.

23. 20. The semiconductor device according to claim 18, wherein a plurality of active antennas each including the injection-locked oscillator and the antenna are arranged in a two-dimensional array.

24. the plurality of active antennas includes a first active antenna and a second active antenna; 24. The semiconductor device of claim 23, wherein the frequency of self-oscillation of the injection-locked oscillator arranged in the first active antenna and the frequency of self-oscillation of the injection-locked oscillator arranged in the second active antenna operate to be different from each other.

25. 2. The semiconductor device according to claim 1, wherein the first frequency is a subharmonic frequency of the second frequency.

26. further comprising a first semiconductor substrate on which the reference signal oscillator is disposed and a second semiconductor substrate on which the injection-locked oscillator is disposed; The semiconductor device according to claim 1 , wherein the first semiconductor substrate and the second semiconductor substrate are at least partially stacked.

27. 2. The semiconductor device according to claim 1, further comprising a phase adjuster disposed between the reference signal oscillator and the injection-locked oscillator for adjusting the phase of the first signal.

28. 28. The semiconductor device of claim 27, wherein the phase adjuster delays the phase of the first signal.

29. 28. The semiconductor device according to claim 27, wherein the phase adjuster includes a plurality of paths each having a different amount of change in the phase of the first signal.

30. a transmitter comprising the semiconductor device of any one of claims 1 to 29; a receiving section for detecting a signal emitted from the transmitting section.

31. a transmitter comprising the semiconductor device according to any one of claims 1 to 29, for emitting a signal toward a subject; a receiving section that detects the signal reflected from or transmitted through the subject.

32. a transmitter comprising the semiconductor device of any one of claims 1 to 29; a receiving unit that detects a wave that is emitted from the transmitting unit and reflected by an object, A radar device that measures distance from the radiation wave emitted from the transmitting unit and the reflected wave.

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