Radar Equipment

Abstract

A radar device according to the disclosed technique is an FMCW type or high-speed chirp type radar device comprising: a beat signal generation unit (8) that generates an I-axis local oscillation signal and a Q-axis local oscillation signal from a local oscillation signal that is a real signal, generates an I-axis beat signal by mixing the I-axis local oscillation signal and a received signal, and generates a Q-axis beat signal by mixing the Q-axis local oscillation signal and the received signal; and a signal processing unit (16) that performs signal processing on I-axis digital data and Q-axis digital data obtained by sampling the I-axis beat signal and the Q-axis beat signal, wherein the signal processing unit (16) generates complex digital data from the I-axis digital data and the Q-axis digital data, performs two-dimensional FFT on the complex digital data, and measures the range and Doppler velocity of the target to be observed on the basis of the property that the analytic signal does not have a negative frequency component.

Description

[Technical field]

[0001] The present disclosure relates to a radar device. [Background technology]

[0002] 2. Description of the Related Art Radar devices mounted on vehicles are known. Furthermore, with regard to radar devices mounted on vehicles, techniques are known for suppressing radio wave interference between radar signals of other vehicles.

[0003] For example, Patent Document 1 describes a technology that includes a camera that captures an area including the direction in which a radar signal is transmitted, and that makes the transmission section of the vehicle itself different from the transmission section of other vehicles based on the illumination state of the lights of the other vehicles contained in the image captured by the camera. Patent Document 1 also discloses that the high-speed chirp method is advantageous for separating and detecting multiple targets. [Prior art documents] [Patent documents]

[0004] [Patent Document 1] JP 2016-224024 A Summary of the Invention [Problem to be solved by the invention]

[0005] In the conventional radar device exemplified in Patent Document 1, in order to suppress radio interference, it is necessary to provide a period during which the radar signal is not transmitted and observe electromagnetic noise.

[0006] The disclosed technique aims to provide a radar device that can suppress interference caused by electromagnetic noise without providing a so-called "radar radiation quiet period" during which radar signal transmission is stopped and electromagnetic noise is observed. [Means for solving the problem]

[0007] A radar device according to the disclosed technique is an FMCW or high-speed chirp radar device, and includes a beat signal generation unit that generates an I-axis local oscillation signal and a Q-axis local oscillation signal from a local oscillation signal, which is a real signal, mixes the I-axis local oscillation signal with a received signal to generate an I-axis beat signal, and mixes the Q-axis local oscillation signal with the received signal to generate a Q-axis beat signal, and a signal processing unit that performs signal processing on I-axis digital data and Q-axis digital data obtained by sampling the I-axis beat signal and the Q-axis beat signal, and the signal processing unit generates complex digital data from the I-axis digital data and Q-axis digital data, performs FFT on the complex digital data, and measures the range and Doppler velocity of the target based on the property that the analytic signal does not have negative frequency components. and an amplitude / phase calculation unit that performs range FFT on each of the I-axis digital data and the Q-axis digital data, and a cancellation constant calculation unit that calculates a cancellation constant for canceling components caused by electromagnetic noise for each range bin based on a processing result of the amplitude / phase calculation unit. It is something. Effect of the Invention

[0008] Since the radar device according to the disclosed technique has the above-described configuration, it is possible to suppress interference caused by electromagnetic noise without providing radar radiation pause periods. [Brief description of the drawings]

[0009] [Figure 1] FIG. 1 is a block diagram showing components of a radar device according to the first embodiment. [Diagram 2] FIG. 2 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the first embodiment. [Diagram 3] FIG. 3 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the first embodiment. [Figure 4] FIG. 4 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the first embodiment. [Diagram 5] FIG. 5 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the second embodiment. [Figure 6] FIG. 6 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the second embodiment. [Figure 7] FIG. 7 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the second embodiment. [Figure 8] FIG. 8 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the third embodiment. [Figure 9] FIG. 9 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the third embodiment. [Figure 10] FIG. 10 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the third embodiment. [Figure 11] FIG. 11 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the fourth embodiment. [Figure 12] FIG. 12 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the fourth embodiment. [Figure 13] FIG. 13 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the fourth embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0010] In this specification, the name "XX Department" represents each component unit when the radar device according to the disclosed technology is divided into components. In other words, the name "XX Department" in this specification does not represent a business organizational division such as an agency or a company, nor does it represent a group of people who participate in club activities or circle activities. The means and methods shown in this specification are centered on the radar device, which is a machine, and are not intended to be centered on humans. In other words, the means and methods shown in this specification do not fall under methods that utilize only artificial arrangements.

[0011] Embodiment 1 Fig. 1 is a block diagram showing components of a radar device according to embodiment 1. As shown in Fig. 1, the radar device according to embodiment 1 includes a radar signal output unit 1, a transceiver unit 4, a beat signal generator 8, an I-axis ADC 14, a Q-axis ADC 15, and a signal processor 16. The radar signal output unit 1 includes a control unit 2 and a signal source 3 . The transceiver unit 4 includes a distributor 5, a transmitting antenna 6, and a receiving antenna . The beat signal generating section 8 includes a 90-degree phase shifter 9, an I-axis frequency mixing section 10, a Q-axis frequency mixing section 11, an I-axis filter section 12, and a Q-axis filter section 13. In the radar device according to the first embodiment, the functional blocks are connected together as shown in FIG.

[0012] Radar signal output section 1 The radar signal output unit 1 is a component that outputs a radar signal. The radar signal output by the radar signal output unit 1 is a signal of the FMCW (Frequency Modulated-Continuous Wave) system or the fast chirp (FCM) system. The fast chirp system performs modulation at a period that is overwhelmingly shorter than the modulation period of the FMCW system, and only either the frequency increase modulation or the frequency decrease modulation is used. That is, in the fast chirp system, one waveform of a transmission wave whose frequency changes in a sawtooth waveform becomes one chirp (for example, see the graph shown in the upper part of FIG. 4). In the fast chirp system, since the modulation period is very short, the frequency change due to the Doppler effect can be considered to be negligibly small. Furthermore, the fast chirp system has attracted much attention in recent years because it can solve the malfunction caused by the pairing process in the FMCW system. In any case, the radar device according to the disclosed technology is a CW (Continuous Wave) radar, not a pulse radar. In general terms, the radar signal output from the radar signal output unit 1 is a frequency modulated signal whose frequency changes over time, and is output intermittently and repeatedly. As shown in Fig. 1, the radar signal output from the radar signal output unit 1 is sent to the distribution unit 5 of the transmission / reception unit 4.

[0013] <<Control unit 2 in radar signal output unit 1>> The control unit 2 in the radar signal output unit 1 is a component that generates a control signal. The control signal generated by the control unit 2 determines, for example, the output timing of the radar signal. As shown in Fig. 1, the control signal output from the control unit 2 is sent to a signal source 3 and a signal processing unit 16.

[0014] Signal source 3 in radar signal output section 1 The signal source 3 in the radar signal output unit 1 is a component that serves as a generating source of a radar signal. As described above, the radar signal generated from the signal source 3 is sent to the distribution unit 5 of the transmission / reception unit 4.

[0015] <Transmitter / receiver 4> The transceiver 4 is a component that includes a transmission system for radar signals and a reception system for signals reflected from a target to be observed. As described above, the transceiver 4 includes the distributor 5, the transmitting antenna 6, and the receiving antenna 7.

[0016] <<Distribution unit 5 in transmitting / receiving unit 4>> The distributor 5 in the transceiver 4 is a component that distributes the radar signal into a transmission signal and a reference signal. In this specification, the radar signal for the transmission signal is also referred to as a "radar signal." In addition, in this specification, the radar signal for the reference signal is referred to as a "local oscillation signal." The radar signal for transmission is sent to a transmitting antenna 6 . The radar signal for the reference signal, that is, the local oscillation signal, is sent to an I-axis frequency mixer 10 and, via a 90-degree phase shifter 9, to a Q-axis frequency mixer 11.

[0017] <<Transmitting antenna 6 in transmitting / receiving unit 4>> The transmitting antenna 6 in the transmitting / receiving unit 4 is an antenna that radiates a radar signal into space such as the atmosphere.

[0018] <<Receiving antenna 7 in transmitting / receiving unit 4>> The receiving antenna 7 in the transceiver 4 is an antenna that receives the radar signal reflected by the observation target. In this specification, the radar signal reflected by the receiving antenna 7 is simply referred to as the "received signal." As shown in Fig. 1, the signal received by the receiving antenna 7 is sent to an I-axis frequency mixer 10 and a Q-axis frequency mixer 11.

[0019] Beat signal generating unit 8 The beat signal generating unit 8 is a component that generates a beat signal. A beat signal is a signal generated by mixing a local oscillation signal and a received signal. The beat frequency, which is the frequency of the beat signal, contains information on the distance to the target and the relative speed of the target. Taking into account the radar irradiation direction, the distance to the target gives the relative position of the target as seen from the radar device. When the radar signal output from the radar signal output unit 1 alternately uses both up-chirp and down-chirp, the beat frequency (f up ) and the beat frequency (f down ) Two pieces of information are obtained. If the radar signal output from the radar signal output unit 1 is of the high-speed chirp type, only one beat frequency is obtained. One of the technical features of the radar device according to the disclosed technology is that it performs so-called IQ conversion on a beat signal, which is a real signal, to generate a complex signal consisting of an I-axis beat signal and a Q-axis beat signal. The I-axis (in-phase axis) is what is called in-phase. The Q-axis (quadrature axis) is orthogonal-phase. Among complex signals, those that do not have negative frequency components are called analytic signals.

[0020] In the technical field of radar, a detection method generally called quadrature detection or IQ detection is known. In quadrature detection, two oscillators, a local oscillator (LO) having high frequency stability and a coherent oscillator (CO), are used. The received signal is first down-converted to an intermediate frequency (IF, Intermediate Frequency, hereinafter referred to as "IF frequency") band near the difference component by a signal from a local oscillator (corresponding to the signal source 3 in the disclosed technology) and a mixer (corresponding to the I-axis frequency mixing unit 10 in the disclosed technology). The down-converted signal then passes through an amplifier and a BPF (corresponding to the I-axis filter unit 12 in the disclosed technology) designed near the IF frequency band. The above operation is called frequency conversion or heterodyne detection. The in-phase and quadrature components of the received signal are then extracted by mixing the in-phase and quadrature components with a coherent oscillator (homodyne detection). The radar device according to the disclosed technique may use two oscillators, a local oscillator (LO) and a coherent oscillator (CO), to perform heterodyne detection and homodyne detection. The disclosed technique obtains not only amplitude information but also phase information from a received signal that has been converted into a complex signal.

[0021] <90-degree phase shifter 9 in beat signal generating unit 8> The 90-degree phase shifter 9 in the beat signal generating unit 8 is a component that imparts a phase difference (phase lead or phase delay) of 90 degrees to the local oscillation signal. The purpose of imparting a phase difference of 90 degrees to the local oscillation signal is to generate an analytic signal of the local oscillation signal. If the I axis is considered to be the real axis in the complex plane and the Q axis is considered to be the imaginary axis in the complex plane, the Q axis is 90 degrees in phase with respect to the I axis. For simplicity, in this specification, it is assumed that the 90-degree phase shifter 9 imparts a phase lead of 90 degrees. That is, the 90-degree phase shifter 9 receives an I axis local oscillation signal (hereinafter referred to as an "I axis local oscillation signal") as an input and outputs a Q axis local oscillation signal (hereinafter referred to as a "Q axis local oscillation signal"). The 90-degree phase shifter 9 may be realized as a Hilbert filter. Since the angular frequency of a chirp signal changes over time, it is difficult to imagine the operation of "advancing the phase by 90 degrees." For example, a chirp signal can be expressed in complex numbers as follows: TIFF0007607845000001.tif24166 Here, j represents the imaginary unit and A is the amplitude of the chirp signal. g shown in formula (1) chirp The real part of (t) can be considered to be a real signal and a local oscillation signal. The "adding a 90 degree phase lead" intended by the present disclosure can be expressed as g chirp (t) to g chirp The aim is to generate the imaginary part of (t).

[0022] It is not essential whether the 90-degree phase shifter 9 imparts a phase lead or a phase delay of 90 degrees to the locally oscillated signal. The radar device according to the disclosed technique may consider the local oscillation signal, which is a real signal, as the Q-axis, and create an I-axis signal using the 90-degree phase shifter 9. When the 90-degree phase shifter 9 imparts a phase delay of 90 degrees, the input to the 90-degree phase shifter 9 is the Q-axis local oscillation signal, and the output of the 90-degree phase shifter 9 is the I-axis local oscillation signal.

[0023] <I-Axis Frequency Mixing Section 10 in Beat Signal Generating Section 8> The I-axis frequency mixer 10 in the beat signal generator 8 is a component that mixes a locally oscillated signal and a received signal. In the I-axis frequency mixer 10, an I-axis beat signal is generated. The I-axis beat signal generated by the I-axis frequency mixer 10 is sent to the I-axis filter 12 .

[0024] Q-Axis Frequency Mixing Unit 11 in Beat Signal Generating Unit 8 The Q-axis frequency mixer 11 in the beat signal generator 8 is a component that mixes a locally oscillated signal delayed by a phase of 90 degrees with a received signal. In the Q-axis frequency mixer 11, a Q-axis beat signal is generated. The I-axis beat signal generated by the Q-axis frequency mixer 11 is sent to the Q-axis filter unit 13 .

[0025] <<I-Axis Filter Section 12 in Beat Signal Generator 8>> The I-axis filter unit 12 in the beat signal generating unit 8 is a filter for the I-axis beat signal. Specifically, the I-axis filter unit 12 is an LPF (Low Pass Filter) or a BPF (Band Pass Filter). The I-axis filter unit 12 is used to suppress unnecessary components such as spurious signals from the I-axis beat signal immediately after it is generated in the I-axis frequency mixing unit 10. Spurious signals are mainly high-frequency signals, and are frequency components that are not intended in the design and are included in AC signals.

[0026] Q-axis filter section 13 in beat signal generating section 8 The Q-axis filter unit 13 in the beat signal generating unit 8 is a filter for the Q-axis beat signal. Like the I-axis filter unit 12, the Q-axis filter unit 13 is specifically an LPF (Low Pass Filter) or BPF (Band Pass Filter). The Q-axis filter unit 13 is used to suppress unnecessary components such as spurious from the Q-axis beat signal immediately after it is generated in the Q-axis frequency mixing unit 11.

[0027] {ADC14 for I axis and ADC15 for Q axis} Specifically, the I-axis ADC 14 and the Q-axis ADC 15 are analog-to-digital converters. The I-axis ADC 14 converts the I-axis beat signal, which is an analog signal, into I-axis digital data. The I-axis digital data is represented as follows. TIFF0007607845000002.tif13166 Here, in formula (2), i k is a real number, but strictly speaking, it is a quantized real number, for example, a double type or a float type. k in the formula (2) is a sampling number, and ranges from 1 to N _smpl Takes an integer up to The Q-axis ADC 15 converts the Q-axis beat signal, which is an analog signal, into Q-axis digital data. The Q-axis digital data is expressed as follows. TIFF0007607845000003.tif11166 Here, in formula (3), q k Although it is shown that k is a real number, strictly speaking, it is a quantized real number, for example, a double type or a float type. k in Equation (3) is also a sampling number. The I-axis digital data and the Q-axis digital data are collectively referred to as IQ data. The IQ data is sent to the signal processing unit 16.

[0028] Signal Processing Unit 16 The signal processing unit 16 is a component that performs signal processing to calculate the distance to the target and the relative speed of the target. Signal processing unit 16 can identify the period during which a radar signal is being output from radar signal output unit 1 by referring to the control signal sent from control unit 2. In this specification, the period identified as the period during which a radar signal is being output by radar signal output unit 1 is referred to as a "specific period."

[0029] FIG. 2 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the first embodiment. As shown in FIG. 2 , the signal processing unit 16 in the radar device of embodiment 1 includes a spectrum calculation unit 1610, a distance / velocity spectrum calculation unit 1620, an electromagnetic noise spectrum calculation unit 1625, a distance / velocity information calculation unit 1630, an electromagnetic noise information calculation unit 1635, and a detection processing unit 1650. In the signal processing unit 16 in the radar device according to the first embodiment, each functional block is connected as shown in FIG.

[0030] Fig. 3 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the first embodiment. As shown in Fig. 3, the processing steps performed by the signal processing unit 16 include ST11, ST12, ST13, ST14, ST15, and ST16. Details of each processing step will become clear from the description below.

[0031] <<Spectrum Calculation Unit 1610 in Signal Processing Unit 16>> The spectrum calculation unit 1610 in the signal processing unit 16 is a component that performs a Fourier transform in the distance direction (hereinafter referred to as a "range Fourier transform") and calculates a frequency spectrum (ST11 shown in FIG. 3). The range Fourier transform is sometimes called a first Fourier transform because it is the first Fourier transform performed.

[0032] The spectrum calculation section 1610 uses the digital data in a specific period to generate the complex digital data given below. TIFF0007607845000004.tif13166 In the formula (4), k is also the sampling number.

[0033] More specifically, the spectrum calculation unit 1610 performs a range Fourier transform on the complex digital data shown in Equation 4. The result obtained by the range Fourier transform is called a frequency spectrum. The data obtained as a result of the range Fourier transform is complex data in the frequency domain. In an ideal noise-free case, the frequency at which the peak occurs in the frequency domain is the beat frequency. The peak in the frequency domain (spectral peak) is also a complex number, and the Doppler frequency can be calculated from the phase information of this spectral peak. In order to calculate the Doppler frequency from the phase information of the spectral peak, a Doppler Fourier transform, which will be described later, is performed. The beat signal is repeatedly generated, and the spectrum calculation section 1610 performs a range Fourier transform each time. The multiple frequency spectra calculated by the spectrum calculation unit 1610 are sent to a distance / velocity spectrum calculation unit 1620 and an electromagnetic noise spectrum calculation unit 1625 .

[0034] <<Distance velocity spectrum calculation unit 1620 in signal processing unit 16>> The distance velocity spectrum calculation unit 1620 in the signal processing unit 16 is a component that performs a Fourier transform in the relative velocity direction (hereinafter referred to as a "Doppler Fourier transform") and calculates a distance velocity spectrum (ST12 shown in FIG. 3). The Doppler Fourier transform is sometimes called a second Fourier transform because it is a second Fourier transform. The distance velocity spectrum calculation unit 1620 performs a Doppler Fourier transform on the positive frequency region (hereinafter referred to as "positive region frequency spectrum data") of the frequency spectrum data. That is, the result obtained by performing a Doppler Fourier transform on the positive region frequency spectrum data is referred to as a distance velocity spectrum. Incidentally, the fast Fourier transform is called FFT (Fast Fourier Transform). The radar device according to the disclosed technique may perform the range Fourier transform and the Doppler Fourier transform in the form of a range FFT and a Doppler FFT. The operation of performing both the range FFT and the Doppler FFT is also called a two-dimensional FFT because the information obtained is two-dimensional (see Figs. 4, 7, and 10). As shown in FIG. 2, the distance and speed spectrum is sent to the distance and speed information calculation unit 1630.

[0035] <Electromagnetic noise spectrum calculation unit 1625 in signal processing unit 16> The electromagnetic noise spectrum calculation unit 1625 in the signal processing unit 16 is a component that performs a Doppler Fourier transform and calculates the electromagnetic noise spectrum (ST13 shown in FIG. 3). The electromagnetic noise spectrum calculation unit 1625 performs a Doppler Fourier transform on the negative frequency region (hereinafter referred to as "negative region frequency spectrum data") of the frequency spectrum data. That is, the result obtained by performing a Doppler Fourier transform on the negative region frequency spectrum data is called an electromagnetic noise spectrum. As shown in FIG. 2, the electromagnetic noise spectrum is sent to the electromagnetic noise information calculation unit 1635 .

[0036] <<Distance / Speed ​​Information Calculation Unit 1630 in Signal Processing Unit 16>> The distance / speed information calculation unit 1630 in the signal processing unit 16 is a component that calculates the distance to the target and the relative speed of the target based on the distance / speed spectrum (ST14 shown in FIG. 3). More specifically, the distance / velocity information calculation unit 1630 detects a peak value of the distance / velocity spectrum and calculates a beat frequency and a Doppler frequency based on the peak value. The beat frequency gives the range to the target, and the Doppler frequency gives the Doppler velocity of the target. The beat frequency and Doppler frequency information calculated in distance and velocity information calculation section 1630 , or information on the distance to the target and the Doppler velocity of the target, is sent to detection processing section 1650 .

[0037] Electromagnetic Noise Information Calculation Unit 1635 in Signal Processing Unit 16 The electromagnetic noise information calculation unit 1635 in the signal processing unit 16 is a component that calculates the frequency and Doppler frequency originating from the electromagnetic noise based on the electromagnetic noise spectrum (ST15 shown in FIG. 3). More specifically, the electromagnetic noise information calculation unit 1635 detects the peak value of the electromagnetic noise spectrum, and calculates the frequency and Doppler frequency originating from the electromagnetic noise based on the peak value. Information on the frequency and Doppler frequency originating from electromagnetic noise calculated in the electromagnetic noise information calculation unit 1635 is sent to the detection processing unit 1650.

[0038] <Detection Processing Unit 1650 in Signal Processing Unit 16> The detection processing unit 1650 in the signal processing unit 16 is a component that suppresses the influence of electromagnetic noise and detects a plausible relative position and relative velocity of the target (ST16 shown in FIG. 3). As shown in FIG. 2, the process performed by the detection processing unit 1650 is based on information sent from the distance / speed information calculation unit 1630 and information sent from the electromagnetic noise information calculation unit 1635.

[0039] FIG. 4 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the first embodiment.

[0040] In the graph shown in the upper part of Figure 4, {L O (1), L O (2),…,L O{K} is a local oscillator signal. In the graph shown in the upper part of FIG. 4, the horizontal axis represents time and the vertical axis represents frequency. In FIG. 4, a down-chirp is shown as an example of the local oscillator signal. The sweep time of one chirp signal is represented by "T" and is on the order of μs (microseconds). The frequency band of the chirp signal is represented by "BW". In the graph shown in the upper part of Figure 4, {R X (1), R X (2),…,R X (K)} is the received signal. In FIG. 4, K is a chirp number that identifies which chirp signal it is.

[0041] In the graph shown in the upper part of Fig. 4, electromagnetic noise is indicated by a dashed line. For simplicity, in this specification, it is assumed that the electromagnetic noise is a continuous wave with a constant frequency. It is also assumed that the electromagnetic noise directly enters the I-axis ADC 14 and the Q-axis ADC 15. It is also assumed that the electromagnetic noise entering the I-axis ADC 14 and the electromagnetic noise entering the Q-axis ADC 15 are not correlated with each other. In general, since the I-axis ADC 14 and the Q-axis ADC 15 are arranged in different positions on the board, it can be assumed that there is no correlation between the entering noises.

[0042] 4, the multiple rectangles labeled "signal acquisition timing" represent periods within the specific period described above, and are periods during which the beat frequency can be acquired. The signal processing unit 16 acquires a signal at this signal acquisition timing.

[0043] The three grid graphs shown in the right column of Fig. 4 are graphs showing the results of the above-mentioned two-dimensional FFT. In this specification, the grid graphs showing the results of the two-dimensional FFT are referred to as "two-dimensional FFT grid diagrams." In the two-dimensional FFT grid diagram shown in Figure 4, the vertical axis is the beat frequency (distance) and the horizontal axis is the Doppler frequency (relative velocity). Note that there are also graphs showing the results of two-dimensional FFT that take the beat frequency on the horizontal axis and the Doppler frequency on the vertical axis. For the sake of simplicity, in the two-dimensional FFT lattice diagram illustrated in FIG. 4, one location corresponding to the observation target and one location corresponding to electromagnetic noise (false detection) are shaded in.

[0044] In Figure 4, the part marked "FFT(1)" represents the range FFT. The beat frequency (F sb_r ) satisfies the following relation: TIFF0007607845000005.tif13166 Here, Δf is the frequency difference between the upper and lower limits of the frequency band (BW) (also called the "maximum frequency deviation width"), R is the range, c is the speed of light, and T is the sweep time (or chirp period). In addition, in formula (5), it is assumed that the modulation period is very short, and no terms related to the Doppler frequency are included. sb_r In the subscript sb_r in ", sb is the initial letter of signal beat, and r is the initial letter of range.

[0045] In Figure 4, under the section labeled "FFT(1)", there are _smpl The diagram shows a vertical rectangle with a size of 1×. Each rectangle has three filled areas. The filled areas represent the positions of the peaks in the frequency spectrum. That is, in the example of Figure 4, there are three frequency spectrum peaks. In each rectangle, the second filled area from the top is the beat frequency (F sb_r )

[0046] As mentioned above, the analytic signal does not have any negative frequency components, and a complex signal related to an ideal beat signal that does not contain noise is the analytic signal. In the vertical rectangle shown in Figure 4, the upper half (numbered from 1 to N _smpl / 2) represents the positive frequency region. Also, in the vertically long rectangle, the lower half (numbered (N_smpl / 2)+1 to N _smpl The negative frequency domain is represented by the dashed rectangle. The center of the rectangle is the position where the beat frequency is 0. In the time domain, the sampling numbers are 1 to N in chronological order. _smpl In the frequency domain, the numbers are numbered from 1 to N, going from the more positive frequencies to the more negative frequencies. _smpl The date shall be attached. The fourth position counting up from the center of the rectangle is shown as being filled in, but this is intended to indicate a spectral peak resulting from a reflected wave caused only by the target. A spectral peak resulting from a reflected wave caused only by the target does not have negative frequency components, since it appears as a frequency analysis result of the analytic signal. Therefore, in FIG. 4, the fourth position counting up from the center of the rectangle is filled in, but the fourth position counting down from the center of the rectangle is not filled in. This indicates that, according to the procedure or method of the disclosed technology, the signal reflected from the target does not have a spectral peak in the negative frequency region.

[0047] When a real signal, not a complex signal, is Fourier transformed, spectral peaks appear symmetrically not only in the positive frequency domain but also in the negative frequency domain. Therefore, for example, if electromagnetic noise enters either the I-axis ADC 14 or the Q-axis ADC 15, a Fourier transform of this electromagnetic noise signal will produce spectral peaks in both the positive and negative frequency domains. In the vertically long rectangle shown in FIG. 4, the top and bottom peaks at symmetrical positions represent spectral peaks caused by electromagnetic noise.

[0048] In the example shown in FIG. 4, beat signals are acquired at K signal acquisition timings for K consecutive chirp signals, and K range FFTs are performed.

[0049] 4, the part indicated as "FFT(2)" represents the Doppler FFT performed by the distance velocity spectrum calculation unit 1620. sb_v ) satisfies the following relation: TIFF0007607845000006.tif13166 Here, f represents the center frequency of the local oscillator signal, and v represents the relative velocity of the target as seen from the radar device. Strictly speaking, v represents the velocity component in the radar radiation direction of the relative velocity of the target as seen from the radar device. In general, the velocity component of an object's velocity that causes the Doppler effect is called the Doppler velocity. Therefore, v in equation (6) is the Doppler velocity of the target. Note that "F sb_v In the subscript sb_v in “signal beat”, sb stands for signal beat, and v stands for velocity.

[0050] 4, the top one represents the result of the Doppler FFT performed by the distance velocity spectrum calculation unit 1620. In this two-dimensional FFT lattice diagram, the Doppler frequency of the observation object (i.e., the target) is illustrated as 0, and the Doppler frequency of the electromagnetic noise (false detection) is illustrated as the value corresponding to the second square to the right of 0.

[0051] The part indicated as "FFT(3)" in Fig. 4 represents the Doppler FFT performed by the electromagnetic noise spectrum calculation unit 1625. As illustrated in Fig. 4, the electromagnetic noise spectrum calculation unit 1625 may perform a process of inverting the sign of the data relating to the negative frequency domain obtained by the range FFT to positive, and then perform the Doppler FFT.

[0052] 4, the second from the top represents the result of the Doppler FFT performed by the electromagnetic noise spectrum calculation unit 1625. This two-dimensional FFT lattice diagram also illustrates that the Doppler frequency of the electromagnetic noise (false detection) is the value corresponding to the second square to the right from 0.

[0053] 4 can be said to be obtained by subtracting the second 2D FFT lattice diagram from the top from the top 2D FFT lattice diagram. The third 2D FFT lattice diagram from the top represents information obtained as a result of processing by the detection processing unit 1650.

[0054] One of the technical features of the radar device according to the first embodiment is that it includes a 90-degree phase shifter 9 that imparts a phase difference (phase lead or phase lag) of 90 degrees to the locally oscillated signal. With this configuration, the radar device according to the first embodiment generates an I-axis beat signal and a Q-axis beat signal. From another point of view, the technical feature of the radar device according to the first embodiment is that signal processing is performed by applying the principle that "the analytic signal has no negative frequency components."

[0055] As described above, the radar device according to the first embodiment has an advantage that it does not require a radar radiation pause period for observing only electromagnetic noise.

[0056] Embodiment 2 The radar device according to the second embodiment is a modified example of the radar device according to the disclosed technique. In the second embodiment, unless otherwise specified, the same reference symbols as those used in the first embodiment are used. In the second embodiment, descriptions that overlap with those in the first embodiment are omitted as appropriate.

[0057] FIG. 5 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the second embodiment. 5 with FIG. 2 (Embodiment 1), it can be seen that signal processing unit 16 according to embodiment 2 includes electromagnetic noise spectrum calculation unit 1625B as a component instead of electromagnetic noise spectrum calculation unit 1625. Information from distance / speed information calculation unit 1630 is input to electromagnetic noise spectrum calculation unit 1625B.

[0058] FIG. 6 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the second embodiment. Comparing FIG. 6 with FIG. 3 (first embodiment), it can be seen that the signal processing unit 16 according to the second embodiment executes ST21 after ST14, instead of ST13.

[0059] FIG. 7 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the second embodiment. Comparing Fig. 7 with Fig. 4 (Embodiment 1), it can be seen that the electromagnetic noise spectrum calculation unit 1625B according to Embodiment 2 performs Doppler FFT only on specific data having a spectral peak, not on the entire negative frequency domain. The specific data having a spectral peak is specifically data relating to a negative beat frequency corresponding to the beat frequency of a spectral peak occurring in the positive frequency domain. In the example of Fig. 7, the beat frequency of the spectrum peak occurring in the positive frequency region is the fourth square counting up from the center and the eleventh square counting up from the center. Therefore, the electromagnetic noise spectrum calculation unit 1625B performs a process of inverting the sign to positive for the data related to the negative frequency region obtained by the range FFT, and performs a Doppler FFT only for the fourth square counting up from the origin and the eleventh square counting up from the center. In other words, the electromagnetic noise spectrum calculation unit 1625B according to the second embodiment performs a Doppler FFT only for the necessary range (ST21 shown in Fig. 6).

[0060] The radar device according to the second embodiment has the same technical features as the radar device according to the first embodiment, and further has the point that the electromagnetic noise spectrum calculation unit 1625B performs Doppler FFT only in a required range.

[0061] As described above, the radar device according to the second embodiment has the effect of minimizing the number of times Doppler FFT is performed, in addition to the effect described in the first embodiment.

[0062] Embodiment 3 The radar device according to the third embodiment is a modified example of the radar device according to the disclosed technique. In the third embodiment, unless otherwise specified, the same reference numerals as those used in the previous embodiments are used. In the third embodiment, descriptions that overlap with those in the previous embodiments are omitted as appropriate.

[0063] The technical feature unique to the radar device of embodiment 3 is, simply put, that it determines whether the I-axis beat signal and the Q-axis beat signal that are actually sampled and measured are the real part and the imaginary part of an ideal analytic signal.

[0064] FIG. 8 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the third embodiment. As shown in FIG. 8, the signal processing unit 16 in the radar device according to the third embodiment includes a spectrum calculation unit 1610, a distance / velocity spectrum calculation unit 1620B, a distance / velocity information calculation unit 1640, a detection processing unit 1650B, an amplitude / phase calculation unit 1660, and a cancellation constant calculation unit 1670. In the signal processing unit 16 in the radar device according to the third embodiment, each functional block is connected as shown in FIG.

[0065] Fig. 9 is a flowchart showing processing steps performed by the signal processor 16 in the radar device according to the embodiment 3. As shown in Fig. 9, the processing steps performed by the signal processor 16 according to the embodiment 3 include ST11, ST31, ST32, ST33, ST34, and ST35. Details of each processing step will become clear from the description below.

[0066] Fig. 10 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the embodiment 3. As shown in Fig. 10, the signal processing unit 16 in the radar device according to the embodiment 3 performs a range FFT on only the I-axis beat signal (shown as "FFT(I)" in Fig. 10) and a range FFT on only the Q-axis beat signal (shown as "FFT(Q)" in Fig. 10).

[0067] <<Amplitude / Phase Calculation Unit 1660 in Signal Processing Unit 16>> The amplitude and phase calculation unit 1660 in the signal processing unit 16 is a component that performs range FFT on each of the I-axis beat signal and the Q-axis beat signal, and calculates the amplitude ratio and phase difference for each range bin. The result of the range FFT for the I-axis beat signal is expressed, for example, as follows: TIFF0007607845000007.tif16166 Here, the script F represents the Fourier transform operation. Also, as mentioned above, the left side of the time domain equation (7) is the sampling number from 1 to N in chronological order. _smpl On the other hand, the right-hand side of equation (7), which is the frequency domain, is numbered from 1 to N in the frequency direction from +∞ to -∞. _smpl The frequency domain numbers 1 to N _smpl Up to is also a number that identifies the range bin. Similarly, the range FFT result for the Q-axis beat signal is expressed as follows: TIFF0007607845000008.tif17166

[0068] The amplitude ratio and phase difference for each range bin calculated by the amplitude / phase calculation section 1660 are expressed as follows: TIFF0007607845000009.tif24166 Equation (9) is the amplitude ratio and phase difference of the I-axis beat signal seen from the Q-axis beat signal. The symbol of the absolute value appearing in Equation (9) represents the magnitude of the complex number (the distance from the origin in the complex plane). Also, the symbol representing the angle appearing in Equation (9) represents the argument of the complex number. Note that in FIG. 10, the magnitude of the complex number is represented by "A_", and the argument of the complex number is represented by "θ_". Also, in the example of FIG. 10, the arrows indicate that the amplitude ratio and phase difference for each range bin are calculated at the timing of first signal acquisition, but the disclosed technology is not limited to this. The amplitude-phase calculation unit 1660 according to the third embodiment may perform statistical calculations (for example, calculating the average value or median value) based on information obtained from a plurality of beat signals to calculate the amplitude ratio and phase difference for each range bin.

[0069] If the I-axis beat signal and Q-axis beat signal actually sampled and measured are the real and imaginary parts of an ideal analytic signal, the amplitude ratio and phase difference for each range bin shown in equation (9) will be 1 for all range bins and -90 degrees for all range bins. The amplitude ratio and phase difference for each range bin are sent to the cancellation constant calculation section 1670 .

[0070] <<Cancellation constant calculation unit 1670 in signal processing unit 16>> The cancellation constant calculation section 1670 in the signal processing section 16 is a component that calculates a cancellation constant (weight) for canceling the component caused by the electromagnetic noise for each range bin. Whether the I-axis beat signal and the Q-axis beat signal actually sampled and measured are the real part and the imaginary part of an ideal analytic signal can be determined, for example, by the following formula. TIFF0007607845000010.tif22166 Here, ε (epsilon) in equation (10) is a threshold value that determines how much error is allowed. k is a cancellation constant (weight) for canceling the components caused by electromagnetic noise. For simplicity, N _smpl is an even number. When the I-axis beat signal and the Q-axis beat signal actually sampled and measured are close to the real and imaginary parts of the ideal analytic signal, the norm condition given on the right side of equation (10) is satisfied, and W k On the other hand, when the I-axis beat signal and the Q-axis beat signal actually sampled and measured are far from the real and imaginary parts of the ideal analytic signal, the norm condition given on the right side of equation (10) is not satisfied, and W k will be 1. The norm condition shown in formula (10) is simply expressed as I k And, Q k We are comparing the result of multiplying by -j with . Q k Multiplying by -j gives Q k The phase of is delayed by 90 degrees to the original I k This is equivalent to converting it into a form that can be compared to

[0071] The meaning of Equation (10) becomes clear when applying the ideal analytic signal shown below. TIFF0007607845000011.tif13166 When the ideal analytic signal shown in equation (11) is Fourier transformed with cos(ω0t) as the fundamental wave, when the angular frequency is ω0, the I axis becomes 1+0j and the Q axis becomes 0+j. Then, when the kth range in equation (10) corresponds to ω0, the condition equation for the norm given on the right hand side of equation (10) is calculated as follows: TIFF0007607845000012.tif31166 Thus, in the case of an ideal analytic signal, the norm condition given on the right hand side of equation (10) is satisfied.

[0072] The cancellation constant (weight, W k ) does not have to be a binary value of 0 or 1 as shown in Equation (10). The cancellation constant (weight, W k ) may take on values ​​other than binary, for example as given in the following equation: TIFF0007607845000013.tif39166 As mentioned above, electromagnetic noise may enter either the I-axis ADC 14 or the Q-axis ADC 15. At a certain k, it cannot be said that the Fourier transform result will never be 0 on an axis that is not affected by electromagnetic noise. Equation (13) expresses the 2-norm given by Equation (10) as I k 2-norm of Q k In equation (13), in order to avoid division by zero, I k When dividing by the 2-norm of Q k Two ways are shown: dividing by the 2-norm of . Equation (13) expresses the normalized cancellation constant (weight, W k ), but the disclosed technology is not limited to this. The radar device according to the disclosed technology uses a non-normalized cancellation constant (weight, W k ) may also be used.

[0073] The radar device according to the disclosed technique may directly extract range bins related to the target by using the norm condition given by Equation (10). TIFF0007607845000014.tif22166 T given by equation (14) k It can be said that the range bin where is 1 is the range bin related to the target and not the interference caused by electromagnetic noise (see the 2D FFT grid diagram in Figure 10).

[0074] <<Distance Velocity Spectrum Calculation Unit 1620B in Signal Processing Unit 16>> Distance velocity spectrum calculation unit 1620B in signal processing unit 16 is a component that performs Doppler Fourier transform and calculates a distance velocity spectrum (ST33 shown in FIG. 9) in the same manner as distance velocity spectrum calculation unit 1620. However, unlike distance velocity spectrum calculation unit 1620, distance velocity spectrum calculation unit 1620B performs Doppler Fourier transform on T given by Equation (14). k Only range bins where is 1 are required. In addition, as shown in FIG. 10, the distance velocity spectrum calculation unit 1620B calculates the cancellation constant (weight, W k ) may be used to eliminate the effects of electromagnetic noise, followed by a Doppler Fourier transform.

[0075] As described above, the technical feature unique to the radar device of embodiment 3 is that it uses the conditional norm equation given on the right side of equation (10) to determine whether the I-axis beat signal and the Q-axis beat signal actually sampled and measured are the real part and the imaginary part of the analytic signal.

[0076] By having such technical features, the radar device according to the third embodiment has the effect of eliminating the influence of interference due to electromagnetic noise from the result of the two-dimensional FFT, in addition to the effects described in the first and second embodiments.

[0077] Embodiment 4 The radar device according to the fourth embodiment is a modified example of the radar device according to the disclosed technique. In the fourth embodiment, unless otherwise specified, the same reference numerals as those used in the previous embodiments are used. In the fourth embodiment, descriptions that overlap with those in the previous embodiments are omitted as appropriate.

[0078] FIG. 11 is a block diagram showing a detailed configuration of the signal processing unit 16 in the radar device according to the fourth embodiment. 11 with FIG. 8 (Embodiment 3), it can be seen that signal processing unit 16 according to embodiment 4 includes a canceling constant calculation unit 1670B as a component instead of canceling constant calculation unit 1670. Information from spectrum calculation unit 1610 is input to canceling constant calculation unit 1670B.

[0079] FIG. 12 is a flowchart showing processing steps performed by the signal processing unit 16 in the radar device according to the fourth embodiment. Comparing FIG. 12 with FIG. 9 (third embodiment), it can be seen that the signal processing unit 16 according to the fourth embodiment performs ST41 instead of ST32.

[0080] Fig. 13 is a diagram for explaining the processing contents performed by the signal processing unit 16 in the radar device according to the fourth embodiment. More specifically, the graph shown in Fig. 13 represents the frequency spectrum calculated by the spectrum calculation unit 1610. The horizontal axis in the graph represents the range proportional to the beat frequency (shown as "Distance" in Fig. 13) in units of [m]. The vertical axis in the graph represents the relative power of the spectrum (shown as "Relative Power" in Fig. 13). In the graph shown in Fig. 13, the dashed line represents the positive region frequency spectrum data, and the solid line represents the negative region frequency spectrum data with the sign of the frequency axis inverted. In the example of Fig. 13, the spectrum peak appearing in the vicinity of a distance of 10 [m] appears only in the positive region frequency spectrum data, and is an example of a peak caused by a target. In contrast, the spectrum peak appearing in the vicinity of a distance of 50 [m] appears in both the positive region frequency spectrum data and the negative region frequency spectrum data, and is an example of a peak caused by electromagnetic noise. Also, in the example of Fig. 13, the spectrum peak appearing in only the negative region frequency spectrum data in the vicinity of a distance of 40 [m] is an example of a secondary peak.

[0081] The spectrum calculation section 1610 calculates a frequency spectrum by performing range FFT on the complex digital data shown in Equation (4), and the frequency spectrum can be expressed, for example, as follows. TIFF0007607845000015.tif11166 Here, the left side of equation (15) is complex digital data in the time domain, and the subscripts on the right bottom are numbered from 1 to N in order of time. _smpl The right-hand side of equation (15) is the frequency spectrum {S1, ..., S N_sampl}, but the numbers in the bottom right subscript go from 1 to N, going from the more positive frequency side to the more negative frequency side. _smpl The date shall be attached.

[0082] <<Cancellation constant calculation unit 1670B in signal processing unit 16>> Similar to the cancellation constant calculation section 1670, the cancellation constant calculation section 1670B in the signal processing section 16 is a component that calculates a cancellation constant (weight) for canceling the component caused by the electromagnetic noise for each range bin. The cancellation constant calculation unit 1670B calculates the cancellation constant (weight, W) based on the following conditional expression instead of the conditional expression shown in Equation (10): k ) may be calculated. TIFF0007607845000016.tif22166 The epsilon appearing in the conditional expression of Equation (16) is a threshold value represented by the dashed line labeled "decision threshold" in the graph of FIG.

[0083] A technical feature unique to the radar device according to the fourth embodiment is that the magnitude of the frequency spectrum acquired by the range FFT is compared with a threshold value (see Equation (16)).

[0084] By having such technical features, the radar device according to the fourth embodiment achieves the same effects as those described in the previous embodiments.

[0085] Embodiment 5. The radar device according to the fifth embodiment is a modified example of the radar device according to the disclosed technique. In the fifth embodiment, unless otherwise specified, the same reference numerals as those used in the previous embodiments are used. In the fifth embodiment, descriptions that overlap with those in the previous embodiments are omitted as appropriate.

[0086] Of the frequency spectrum obtained by performing a range Fourier transform on complex digital data, a peak signal caused by electromagnetic noise is divided into a positive frequency domain and a negative frequency domain and is given by the following formula. TIFF0007607845000017.tif13166 However, P on the left side of equation (17) + represents the peak signal in the positive frequency domain, A represents the amplitude ratio between the I signal and the Q signal, θ1 represents the initial phase of the I signal, and θ2 represents the initial phase of the Q signal. TIFF0007607845000018.tif15166 However, P on the left side of equation (18) - is the peak signal in the negative frequency domain.

[0087] The disclosed technology uses P given by Equation (18) -By multiplying it by a cancellation constant (C) which is the complex conjugate of + ) may be cancelled. The conditional expression that the cancellation constant (C) must satisfy is given as follows: TIFF0007607845000019.tif30166 Here, the accent symbol bar in equation (19) represents a complex conjugate.

[0088] The technical feature unique to the radar device according to the fifth embodiment is that the peak signal (P - ) is multiplied by a cancellation constant (C) given by equation (19).

[0089] By having such technical features, the radar device according to the fifth embodiment achieves the same effects as those described in the previous embodiments. [Industrial Applicability]

[0090] INDUSTRIAL APPLICABILITY A radar device according to the disclosed technique can be applied to, for example, an in-vehicle millimeter wave radar and has industrial applicability. [Explanation of symbols]

[0091] 1 radar signal output unit, 2 control unit, 3 signal source, 4 transmitting / receiving unit, 5 distribution unit, 6 transmitting antenna, 7 receiving antenna, 8 beat signal generation unit, 9 90 degree phase shifter, 10 I-axis frequency mixing unit, 11 Q-axis frequency mixing unit, 12 I-axis filter unit, 13 Q-axis filter unit, 14 I-axis ADC, 15 Q-axis ADC, 16 signal processing unit, 1610 spectrum calculation unit, 1620, 1620B distance / speed spectrum calculation unit, 1625, 1625B electromagnetic noise spectrum calculation unit, 1630 distance / speed information calculation unit, 1635 electromagnetic noise information calculation unit, 1640 distance / speed information calculation unit, 1650, 1650B detection processing unit, 1660 amplitude / phase calculation unit, 1670, 1670B cancellation constant calculation unit.

Claims

1. A radar device using an FMCW system or a high-speed chirp system, a beat signal generating unit that generates an I-axis local oscillation signal and a Q-axis local oscillation signal from a local oscillation signal that is a real signal, mixes the I-axis local oscillation signal with a received signal to generate an I-axis beat signal, and mixes the Q-axis local oscillation signal with the received signal to generate a Q-axis beat signal; a signal processing unit that performs signal processing on I-axis digital data and Q-axis digital data obtained by sampling the I-axis beat signal and the Q-axis beat signal, the signal processing unit generates complex digital data from the I-axis digital data and the Q-axis digital data, performs an FFT on the complex digital data, and measures the range and Doppler velocity of an object to be observed based on the property that an analytic signal does not have negative frequency components; an amplitude / phase calculation unit that performs a range FFT on each of the I-axis digital data and the Q-axis digital data; The radar device further includes a cancellation constant calculation unit that calculates a cancellation constant for canceling a component caused by electromagnetic noise for each range bin based on a processing result of the amplitude / phase calculation unit.

2. The cancellation constant is W is given by ｋ That is, However, 1 to N ＿ｓｍｐｌ / 2 are range bin numbers related to the positive frequency region, epsilon is a threshold value that determines an allowable error, I k is the I-axis beat signal actually sampled and measured, and Q k is the Q-axis beat signal actually sampled and measured. The radar device according to claim 1 .

3. The cancellation constant is W is given by ｋ That is, However, 1 to N ＿ｓｍｐｌ / 2 are range bin numbers relating to the positive frequency region, I k is the I-axis beat signal actually sampled and measured, and Q k is the Q-axis beat signal actually sampled and measured. The radar device according to claim 1 .

4. The cancellation constant is W is given by ｋ That is, However, 1 to N ＿ｓｍｐｌ / 2 are range bin numbers related to the positive frequency domain, and S ｋ is a frequency spectrum obtained by performing a Fourier transform on the complex digital data, and epsilon is a threshold value that is compared with the magnitude of the frequency spectrum obtained by performing a Fourier transform on the complex digital data. The radar device according to claim 1 .

5. The cancellation constant is C is given by However, P ＋ is the peak signal in the positive frequency domain, and P － is a peak signal in the negative frequency domain, A is an amplitude ratio between the I-axis beat signal and the Q-axis beat signal, θ 1 is an initial phase of the I-axis beat signal, and θ 2 is an initial phase of the Q-axis beat signal, The cancellation constant is P － By multiplying by the complex conjugate of ＋ Cancel The radar device according to claim 1 .

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