Power Conversion Equipment

Abstract

In a DAB circuit constituted by a first bridge (10) and a second bridge (20), a boost circuit is shared by connecting reactors (L1, L2) between a storage battery (1) and the first bridge (10). In each of the legs (11, 12) of the first bridge (10), in order to increase the absolute values of reactor currents (L1, L2), the on-period of one switching element which is one of switching elements (Q11H, Q12H) on the high-voltage side and switching elements (Q11L, Q12L) on the low-voltage side is set, after which the on-period of the other switching element is set. The length of the on-period of the one switching element is set according to the control command value of a DC voltage V1. The length of the on-period of the other switching element is set to a length of time until the absolute values of the reactor currents (L1, L2) that have increased during the on-period of the one switching element return to zero.

Description

[Technical field]

[0001] The present disclosure relates to a power conversion device. [Background technology]

[0002] Power conversion devices that can bidirectionally exchange power with a power grid or a load, etc., using storage batteries in electric vehicles, etc., are in use. In such power conversion devices, it is desirable to ensure insulation between the storage battery and other devices, and to be able to charge and discharge the storage battery with high power conversion efficiency over a wide input voltage range from low output to high output in order to accommodate various storage batteries.

[0003] As a technology for charging and discharging a storage battery with efficient power conversion from low output to high output over a wide input voltage range using a bidirectional isolated converter, which is a type of power conversion device, a circuit configuration integrating a multi-phase boost circuit and a DAB (Dual Active Bridge) circuit is disclosed in "Design Considerations for PPS Controlled Current-Fed DAB Converter to Achieve Full Load Range ZVS with Low Inductor RMS Current" (Non-Patent Document 1). [Prior art documents] [Non-patent literature]

[0004] [Non-Patent Document 1] Jing Guo et al., "Design Considerations for PPS Controlled Current-Fed DAB Converter to Achieve Full Load Range ZVS with Low Inductor RMS Current", 2020 IEEE Energy Conversion Congress and Exposition pp. 5971-5975, October 30, 2020 Summary of the Invention [Problem to be solved by the invention]

[0005] In the CF (Current Fed)-DAB circuit described in Non-Patent Document 1, which integrates a multiphase boost circuit and a DAB circuit, the primary bridge of the DAB circuit and the bridge of the multiphase boost circuit are integrated, so there is a concern that the ripple of the reactor current in the boost circuit unit will go back and forth between the positive and negative electrodes at low output, resulting in large conduction losses. On the other hand, in the boost circuit, the conduction losses can be reduced by operating in discontinuous current mode so that the ripple of the reactor current does not go back and forth between the positive and negative electrodes at low output.

[0006] However, in the CF-DAB circuit of Non-Patent Document 1, when the switching of the high-voltage side power device of the boost circuit section is stopped during the discharge operation of the primary side storage battery to realize the current discontinuous mode, the switching of the high-voltage side power device of the primary side bridge of the DAB circuit section is also stopped. As a result, power transmission operation during the discharge of the storage battery is not possible during the period when the switching is stopped.

[0007] Similarly, in the CF-DAB circuit of Non-Patent Document 1, when the switching of the low-voltage side power device of the boost circuit is stopped during the charging operation of the primary side storage battery to realize the current discontinuous mode, the switching of the low-voltage side power device of the primary side bridge of the DAB circuit is also stopped. As a result, power transmission operation when charging the storage battery is not possible during the switching stop period.

[0008] On the other hand, in Non-Patent Document 1, flow The problem of power transfer operation being impossible due to the introduction of discontinuous mode is not taken into consideration.

[0009] The present disclosure has been made to solve the problems described above, and a purpose of the present disclosure is to improve power conversion performance in a power conversion device having a common configuration between the primary bridge of a DAB circuit and a boost circuit, by controlling the reactor current for boosting in discontinuous current mode and ensuring a power transmission period by the DAB circuit. [Means for solving the problem]

[0010] According to an aspect of the present disclosure, a power conversion device is provided. The power conversion device for performing DC voltage conversion includes a first bridge having a plurality of first legs, a second bridge having a plurality of second legs, a transformer connected between the first bridge and the second bridge, a plurality of reactors, and a control circuit for controlling on / off of each switching element of each of the plurality of first legs and the second legs. Each of the plurality of first legs has a high-voltage side switching element and a low-voltage side switching element connected in series between a first power line on the high voltage side and a second power line on the low voltage side via an intermediate node. Each of the plurality of second legs has a high-voltage side switching element and a low-voltage side switching element connected in series between a third power line on the high voltage side and a fourth power line on the low voltage side via an intermediate node. The transformer has a primary winding and a secondary winding. The primary winding is connected to an intermediate node of the plurality of first legs. The secondary winding is connected to an intermediate node of the second legs and is magnetically coupled to the primary winding. The multiple reactors are respectively connected between a rechargeable DC power source and intermediate nodes of the first legs. The control circuit, in the first bridge, controls on / off of each switching period of each switching element of each first leg in accordance with a first control command value for controlling a first DC voltage between the first and second power lines, so as to provide an on-period of one switching element of the high-voltage side and low-voltage side switching elements of each first leg for increasing an absolute value of a reactor current flowing through each of the multiple reactors, and to provide an on-period of the other switching element of the high-voltage side and low-voltage side switching elements until the absolute value of the reactor current returns to zero after the on-period of one switching element ends. The control circuit reflects a second control command value for controlling a second DC voltage between the third and fourth power lines in the second bridge, and controls on / off of each switching cycle of each switching element in each second leg so that the on-period lengths of the high-voltage side switching elements and the low-voltage side switching elements between the multiple first legs and the multiple second legs are equivalent. Effect of the Invention

[0011] According to the present disclosure, in a power conversion device having a common configuration of the primary bridge of a DAB circuit and a boost circuit, the reactor current for boosting can be controlled in discontinuous current mode and a power transmission period by the DAB circuit can be ensured, thereby improving power conversion performance. [Brief description of the drawings]

[0012] [Figure 1] 1 is a circuit diagram illustrating a circuit configuration of a power conversion device according to an embodiment of the present invention. [Diagram 2] 4 is a block diagram illustrating a control operation during a discharging operation of the power conversion device according to the first embodiment. FIG. [Diagram 3] FIG. 4 is a simulation waveform diagram illustrating an example of an operation during a discharging operation of the power conversion device according to the first embodiment. [Figure 4] 4 is a block diagram illustrating a control operation during a charging operation of the power conversion device according to the first embodiment. FIG. [Diagram 5] FIG. 4 is a simulation waveform diagram illustrating an example of an operation during a charging operation of the power conversion device according to the first embodiment. [Figure 6] FIG. 11 is a block diagram illustrating a control operation during a discharging operation of a power conversion device according to a modification of the first embodiment. [Figure 7] FIG. 11 is a block diagram illustrating a control operation during a charging operation of a power conversion device according to a modification of the first embodiment. [Figure 8] FIG. 11 is a block diagram illustrating a control operation during a discharging operation of the power conversion device according to the second embodiment. [Figure 9] FIG. 11 is a simulation waveform diagram illustrating an example of an operation during a discharging operation of the power conversion device according to the second embodiment. [Figure 10] FIG. 11 is a circuit diagram illustrating a circuit configuration of a power conversion device according to a first example of a third embodiment. [Figure 11] FIG. 13 is a circuit diagram illustrating a circuit configuration of a power conversion device according to a second example of the third embodiment. [Figure 12] 12 is a block diagram for explaining an example of a control operation (discharging operation) of the power conversion device shown in FIG. 11. FIG. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0013] Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. In the following, the same or corresponding parts in the drawings are denoted by the same reference characters, and their description will not be repeated in principle.

[0014] Embodiment 1 (Circuit configuration) As shown in FIG. 1, a power conversion device 100 according to this embodiment is connected between a storage battery 1 and a load 2, and performs bidirectional DC voltage conversion while electrically insulating the storage battery 1 and the load 2 by a transformer 30.

[0015] The power conversion device 100 includes a capacitor C1 arranged on the primary side, a capacitor C2 arranged on the secondary side, reactors L1 and L2 for a boost function, a reactor LT, a first bridge 10, a second bridge 20, a transformer 30 having a primary winding 30p and a secondary winding 30s, a control calculation unit 50, and a gate signal generation unit 60. The main circuit configuration of the power conversion device 100 is similar to that of the CF-DAB circuit of Non-Patent Document 1.

[0016] The first bridge 10 includes a leg 11 configured with semiconductor switching elements (hereinafter also simply referred to as "switching elements") Q11H and Q11L, and a leg 12 configured with switching elements Q12H and Q12L. In this embodiment, each switching element may be configured with an IGBT (Insulated Gate Bipolar Transistor) and a MOSFET (Metal-oxide-Semiconductor Field-effect Transistor), etc. Each switching element includes an anti-parallel diode (freewheel diode) for forming a freewheel path, either as an external element or as a built-in element.

[0017] Switching elements Q11H and Q11L constituting leg 11 are connected in series between power line PL1 on the high voltage side of the primary side and power line NL1 on the low voltage side via node N11. Switching elements Q12H and Q12L constituting leg 12 are connected in series between power line PL1 and power line NL1 via node N12. Legs 11 and 12 correspond to an embodiment of a "first leg", and each of nodes N11 and N12 corresponds to an "intermediate node". Moreover, power line PL1 corresponds to an embodiment of a "first power line", and power line NL1 corresponds to an embodiment of a "second power line".

[0018] In the configuration example of FIG. 1, the capacitor C1 is connected between the power line PL1 and the power line NL1. That is, the voltage of the capacitor C1 is equal to the DC voltage between the power lines PL1 and NL1. The storage battery 1 is connected between the node Ni and the power line NL1. The reactor L1 is connected between the node Ni (i.e., the positive electrode of the storage battery 1) and the node N11 of the leg 11. The reactor L2 is connected between the node Ni and the node N12 of the leg 12. The reactor LT is connected in series with the primary winding 30p of the transformer 30 between the nodes N11 and N12. The reactors L1 and L2 correspond to one example of "plurality of reactors".

[0019] The reactor LT may be formed by a leakage inductance of the magnetic coupling between the primary winding 30p and the secondary winding 30s. The reactors L1 and L2 may be formed by sharing magnetic components in order to reduce the number of components. For example, the reactors L1 and L2 can be formed by a loosely coupled inductor in which two coils are wound around a common magnetic core. Sharing the reactors L1 and L2 also has the effect of increasing the impedance value of the reactors L1 and L2 as viewed from the voltage VTrp between the nodes N11 and N12, which corresponds to the output end voltage of the first bridge 10.

[0020] As shown by the arrows in FIG. 1, for the reactor current IL1 flowing through the reactor L1 and the reactor current IL2 flowing through the reactor L2, the discharging direction of the storage battery 1 is defined as the positive direction (positive current) and the charging direction of the storage battery 1 is defined as the negative direction (negative current).

[0021] The second bridge 20 includes a leg 21 composed of switching elements Q21H and Q21L, and a leg 22 composed of switching elements Q22H and Q22L. The switching elements Q21H and Q21L constituting the leg 21 are connected in series between the power line PL2 on the high voltage side of the secondary side and the power line NL2 on the low voltage side via a node N21. The switching elements Q22H and Q22L constituting the leg 22 are connected in series between the power line PL2 and the power line NL2 via a node N22. The legs 21 and 22 correspond to an embodiment of the "second leg", and each of the nodes N21 and N22 corresponds to an "intermediate node". Moreover, the power line PL2 corresponds to an embodiment of the "third power line", and the power line NL2 corresponds to an embodiment of the "fourth power line".

[0022] Hereinafter, switching elements Q11L, Q12L, Q21L, and Q22L will also be referred to as "low-voltage side switching elements," and switching elements Q11H, Q12H, Q21H, and Q22H will also be referred to as "high-voltage side switching elements."

[0023] The load 2 is connected to the power lines PL2 and NL2. The capacitor C2 is connected in parallel with the load 2 between the power lines PL2 and NL2. The secondary winding 30s of the transformer 30 is connected between the nodes N21 and N22. The voltage VTrs between the nodes N21 and N22 corresponds to the output end voltage of the second bridge 20.

[0024] In the power conversion device 100, voltage control is performed to maintain the DC voltage V1 of the capacitor C1 and the DC voltage V2 of the capacitor C2 at voltage command values ​​VREF1 and VREF2 while the storage battery 1 is being charged or discharged. The voltage command value VREF1 is a command value for controlling the DC voltage between the power lines PL1 and NL1, and the voltage command value VREF2 is a command value for controlling the DC voltage between the power lines PL2 and NL2. In the first embodiment, the DC voltage V1 and the DC voltage V2 correspond to the "first DC voltage" and the "second DC voltage", respectively.

[0025] Although the charge / discharge power of the storage battery 1 is not directly instructed, when the DC voltage V2 of the load 2 is lower than the voltage command value VREF2, the above voltage control is performed in conjunction with a power transfer operation from the storage battery 1 to the load 2 (hereinafter also referred to as a discharging operation), so that the storage battery generates discharge power. On the other hand, when the DC voltage V2 is higher than the voltage command value VREF2, the above voltage control is performed in conjunction with a power transfer operation from the load 2 to the storage battery 1 (hereinafter also referred to as a charging operation), so that the storage battery 1 generates charge power.

[0026] The control calculation unit 50 calculates a first control command value REF1 and a second control command value REF2 for controlling the DC voltages V1 and V2 detected by voltage sensors (not shown) provided corresponding to the capacitors C1 and C2 to voltage command values ​​VREF1 and VREF2, respectively. The control calculation unit 50 further receives an output voltage (hereinafter referred to as battery voltage) VBAT of the storage battery 1 detected by a voltage sensor (not shown) provided in the storage battery 1.

[0027] The gate signal generating unit 60 generates gate signals S11H, S11L, S12H, and S12L for the first bridge 10 and gate signals S21H, S21L, S22H, and S22L for the second bridge 20 based on the first control command value REF1 and the second control command value REF2 calculated by the control calculation unit 50. The switching elements Q11H, Q11L, Q12H, and Q12L of the first bridge 10 are on-off controlled (switching controlled) according to the gate signals S11H, S11L, S12H, and S12L, respectively. Similarly, the switching elements Q21H, Q21L, Q22H, and Q22L are on-off controlled (switching controlled) according to the gate signals S21H, S21L, S22H, and S22L, respectively. Specifically, each of switching elements Q11L, Q12L, Q21L, Q22L, Q11H, Q12H, Q21H, and Q22H is turned on during an H level period of the corresponding gate signal and turned off during an L level period in accordance with gate signals S11L, S12L, S21L, S22L, S11H, S12H, S21H, and S22H, respectively. As will be described below, the control calculation unit 50 and the gate signal generation unit 60 constitute one embodiment of a "control circuit."

[0028] (Control and circuit operation during discharge operation) First, examples of control and operational waveforms during the discharging operation of the power conversion device 100 will be described with reference to FIGS.

[0029] FIG. 2 shows a block diagram for explaining a control operation during a discharging operation of the power conversion device according to the first embodiment.

[0030] 2, control calculation unit 50 has deviation calculation units 52 and 56, controllers 54 and 58, and a limiter 55. Deviation calculation unit 52 calculates a voltage deviation ΔV1 of DC voltage V1 (detection value) with respect to a voltage command value VREF1 (ΔV1=VREF1-V1). Similarly, deviation calculation unit 52 calculates a voltage deviation ΔV2 of DC voltage V2 (detection value) with respect to a voltage command value VREF2 (ΔV2=VREF2-V2).

[0031] The controller 54 generates the first control command value REF1 by a predetermined control calculation for the voltage deviation ΔV1. For example, the controller 54 generates the first control command value REF1 by proportional (P) control in which the voltage deviation ΔV1 is multiplied by a control gain Ka. The first control command value REF1 is limited by a limiter 55 so that the minimum value becomes 0. That is, when the output value of the controller 54 is a negative value, VREF1 is set to 0. As a result, when the DC voltage V1 is higher than the voltage command value VREF1, REF1 is set to 0.

[0032] The controller 58 generates the second control command value REF2 by a predetermined control calculation for the voltage deviation ΔV2. For example, the controller 58 generates the second control command value REF2 by proportional (P) control, which multiplies the voltage deviation ΔV2 by a control gain Kb. Note that the controllers 54 and 58 are not limited to proportional (P) control, and can be configured to perform any control calculation, such as proportional-integral (PI) control.

[0033] The gate signal generating section 60 includes carrier wave generators CG11, CG12, CG21, and CG22, comparators CMP11, CMP12, CMP21, and CMP22, a duty ratio calculation section 70, and logic sections LG11, LG12, LG21, and LG22.

[0034] During a discharge operation, the gate signal generating unit 60 generates low-voltage side gate signals S11L, S12L, S21L, and S22L with the first control command value REF1 from the control calculation unit 50 as the low-voltage side duty ratio DL. The low-voltage side duty ratio DL is defined as the time ratio of the on period to the switching period of the low-voltage side switching elements Q11L, Q12L, Q21L, and Q22L.

[0035] During the discharging operation, low-voltage side gate signals S11L, S12L, S21L, and S22L are generated by carrier wave generators CG11, CG12, CG21, and CG22 and comparators CMP11, CMP12, CMP21, and CMP22.

[0036] The carrier wave generators CG11, CG12, CG21, and CG22 generate carrier waves CW11, CW12, CW21, and CW22 of the same frequency. In general, a periodic voltage waveform such as a triangular wave or a sawtooth wave is used for the carrier waves CW11, CW12, CW21, and CW22. A phase difference is set between the carrier waves CW11, CW12, CW21, and CW22 by individually setting the initial phase. Specifically, the carrier waves CW11 and CW12 are in opposite phase to each other (phase difference of 180 [deg]), and a phase difference of 180 [deg] is also provided between the carrier waves CW21 and CW22.

[0037] As a result, in the first bridge 10, a phase difference of 180 [deg] is provided between the switching operation of leg 11 and the switching operation of leg 12. Similarly, in the second bridge 20, a phase difference of 180 [deg] is provided between the switching operation of leg 21 and the switching operation of leg 22. This allows the first bridge 10 and the second bridge 20 to operate as a DAB circuit.

[0038] Furthermore, a phase difference φ [deg] is provided between the carrier waves CW11 and CW12 corresponding to the first bridge 10 and the carrier waves CW21 and CW22 corresponding to the second bridge 20. Moreover, the peak-to-peak of each of the carrier waves CW11, CW12, CW21, and CW22 corresponds to the duty ratios DL and DH of 0 to 1.0.

[0039] In the gate signal generating unit 60, the phases of the carrier waves CW21 and CW22 corresponding to the second bridge 20 are adjusted using the second control command value REF2 from the control calculation unit 50 as the phase difference φ [deg].

[0040] As is well known, in the DAB circuit, the direction of power transmission between the first bridge 10 and the second bridge 20 as the DAB circuit is controlled by the direction (lead / lag) of the phase difference of the switching operation between the first bridge 10 and the second bridge 20. Furthermore, the transmitted power is controlled by the switching frequency and the amount of the phase difference. Specifically, it is known that, under the same switching frequency, the larger the amount of the phase difference, the larger the transmitted power (absolute value), and, under the same phase difference, the lower the switching frequency (i.e., the longer the switching period), the larger the transmitted power (absolute value).

[0041] When the DC voltage V2 is lower than the voltage command value VREF2, ΔV2>0 and REF2>0, so that the phase difference φ>0, i.e., the phase of the switching operation of the second bridge 20 lags behind the phase of the switching operation of the first bridge 10. As a result, the DC voltage V2 can increase toward the voltage command value VREF2 by transmitting power from the first bridge 10 to the second bridge 20. At this time, the larger the absolute value of ΔV2 is, the larger the amount of phase delay due to the phase difference φ is set, and the larger the power transmitted from the first bridge 10 to the second bridge 20 becomes.

[0042] The comparator CMP11 generates a gate signal S11L according to a voltage comparison between the first control command value REF1 (low-voltage side duty ratio DL) and the carrier wave CW11. Specifically, in a period in which REF1>CW11, the gate signal S11L is set to a high level (hereinafter, "H level"), and in a period in which REF1≦CW11, the gate signal S11L is set to a low level (hereinafter, "L level").

[0043] By similar voltage comparison, the comparator CMP12 generates a gate signal S12L according to a voltage comparison between the low-voltage side duty ratio DL and the carrier wave CW12. Also, the comparator CMP21 generates a gate signal S21L according to a voltage comparison between the low-voltage side duty ratio DL and the carrier wave CW21, and the comparator CMP22 generates a gate signal S22L according to a voltage comparison between the low-voltage side duty ratio DL and the carrier wave CW22.

[0044] As a result, each of the gate signals S11L, S12L, S21L, and S22L is generated to have an on-period length according to the first control command value REF1 for controlling the DC voltage V1 to the voltage command value VREF1. Furthermore, a phase difference φ is set between the gate signals S11L and S12L of the first bridge 10 and the gate signals S21L and S22 of the second bridge 20 according to the second control command value REF2 for controlling the DC voltage V2 to the voltage command value VREF2.

[0045] On the other hand, the high-voltage side gate signals S11H, S12H, S21H, and S22H during the discharge operation are generated with the third control command value REF3 calculated by the duty ratio calculation unit 70 as the high-voltage side duty ratio DH. 12H, Q21H, Q22H It is defined as the time ratio of the ON period to the switching period.

[0046] During a discharge operation, the duty ratio calculation unit 70 calculates a high-voltage side duty ratio DH using the low-voltage side duty ratio DL (first control command value REF1), the DC voltage V1, and the battery voltage VBAT so that the reactor currents IL1, IL2 are in a discontinuous current mode. Here, a method of calculating the high-voltage side duty ratio DH will be described by calculating the reactor current IL that includes the reactor currents IL1, IL2, assuming that the reactors L1, L2 have the same inductance value L.

[0047] As described above, in the first bridge 10, during the on-period of the low-voltage side switching elements Q11L, Q12L in accordance with the low-voltage side duty ratio DL, the battery voltage VBAT is applied across the reactors L1, L2. During the on-period of the low-voltage side switching elements, energy is stored in the reactors.

[0048] Therefore, the reactor current IL (IL>0) during this period of the discharge operation rises at a slope of (VBAT / L). For this reason, at the timing when the low-voltage side switching elements Q11L, Q12L are turned off according to the duty ratio DL, the reactor current IL reaches its maximum value (maximum current ILmax) within that switching period. If IL=0 at the timing when the switching elements Q11L, Q12L start to be turned on, this maximum current ILmax is expressed by the following equation (1) using the switching period length Ts (the reciprocal of the switching frequency fs corresponding to the frequency of the carrier wave, i.e., Ts=1 / fs).

[0049] ILmax = (VBAT / L) × DL × Ts … (1) Next, in the first bridge 10, the high-voltage side switching element Q11L, Q12L, which is provided after the low-voltage side switching elements Q11L and Q12L are turned off, H ,Q12 H During the on-period, a negative voltage (VBAT-V1) is applied across the reactors L1 and L2. Therefore, the reactor current IL during the on-period decreases at a slope of (VBAT-V1) / L, with IL=ILmax as the initial value. During this period, the energy stored in the reactors L1 and L2 during the on-period of the low-voltage side switching elements Q11L and Q12L and the energy from the storage battery 1 are supplied to the power line PL1, thereby achieving a boost function where V1>VBAT.

[0050] The high-voltage side switching elements Q11H, Q12H are controlled so that they are turned on until the reactor current IL falls to zero after the low-voltage side switching elements Q11L, Q12L are turned off, in other words, they are turned off just when IL = 0. By turning off the high-voltage side switching elements Q11H, Q12H at this timing, the path through which a negative current flows through the reactors L1, L2 can be cut off, thereby realizing a discontinuous current mode. Furthermore, by providing an on period for the high-voltage side switching elements Q11H, Q12H until the reactor current IL falls to zero, it is possible to ensure a power transmission path from the capacitor C1 (power line PL1) to the transformer 30 even during the period when the reactor currents IL1, IL2 decrease.

[0051] In order to set the on-period of the high-voltage side switching elements Q11H, Q12H in this manner, it is necessary to set the reactor current IL0 at the end of the on-period of the high-voltage side switching elements Q11H, Q12H to 0. First, IL0 is expressed by the following equation (2) using the high-voltage side duty ratio DH.

[0052] IL0=ILmax+DH×Ts×(VBAT-V1) / L…(2) From equations (1) and (2), it can be seen that the duty ratio DH at which IL0=0 is given by the following equation (3).

[0053] DH = DL × VBAT / (V1 - VBAT) … (3) The duty ratio calculation unit 70 calculates the third control command value REF3, that is, the duty ratio DH on the high voltage side, according to the above equation (3).

[0054] The logic unit LG11 receives a gate signal S11L of the low-voltage side switching element Q11L generated by the comparator CMP11 and a third control command value REF3 (REF 3=DH), the logic unit LG11 generates the gate signal S11H of the high-voltage side switching element Q11H. Specifically, the logic unit LG11 generates the gate signal S11H such that an H level period having a duration (DH×Ts) according to the duty ratio DH is provided, starting from the timing when the gate signal S11L transitions from H level to L level.

[0055] Using the same logic, the logic unit LG12 generates a gate signal S12H according to the gate signal S12L from the comparator CMP12 and the third control command value REF3 (REF3=DH). Also, the logic unit LG21 generates a gate signal S21H according to the gate signal S21L from the comparator CMP21 and the third control command value REF3 (REF3=DH), and the logic unit LG22 generates a gate signal S22H according to the gate signal S22L from the comparator CMP22 and the third control command value REF3 (REF3=DH).

[0056] By setting the high-voltage side duty ratio DH in this manner, even if the low-voltage side duty ratio DL is set to a small value by controlling the DC voltage V1 during discharge operation, the high-voltage side switching element can be turned off when the reactor current IL = 0. In other words, by turning off the high-voltage side switching element before the reactor current IL becomes a negative value, it becomes possible to control the reactor current IL (IL1, IL2) in a discontinuous current mode in which the period during which the reactor current IL = 0 continues.

[0057] That is, during discharge operation, the low-voltage side gate signals S11L, S12L, S21L, and S22L are PWM controlled in accordance with the first control command value REF1, while the high-voltage side gate signals S11H, S12H, S21H, and S22H are switching controlled to achieve both the introduction of discontinuous current mode and power transmission.

[0058] FIG. 3 shows a simulation waveform diagram illustrating an example of the operation of the power conversion device 100 during the discharging operation using the gate signals generated by the method shown in FIG.

[0059] 3, during the discharging operation, the power P12 transmitted from the first bridge 10 to the second bridge 20 is a positive value (P12>0). In addition, the DC voltage V1 of the capacitor C1 is controlled to be constant according to the voltage command value VREF1.

[0060] In the first bridge 10 and the second bridge 20, the H level period lengths of the low-voltage side gate signals S11L, S12L, S21L, and S22L are set according to the duty ratio DL (first control command value REF1) for controlling the DC voltage V1. Furthermore, the second control command value REF2 is reflected in the phase difference φ [deg] of the initial phases in the carrier wave generators CG11, CG12, CG21, and CG22 described in FIG.

[0061] During a discharging operation, since P12>0, and therefore REF2>0 (φ>0), the phase of the gate signals S21L, S22L of the second bridge 20 lags behind the phase of the gate signals S11L, S12L of the first bridge .

[0062] The high-voltage side gate signals S11H, S12H, S21H, and S22H are set to H level for a period length according to the high-voltage side duty ratio DH after the low-voltage side gate signals S11L, S12L, S21L, and S22L transition from H level to L level. As also shown in Fig. 3, in each of the legs 11, 12, 21, and 22, a so-called dead time is generally provided in which both gate signals are set to L level between the time when the low-voltage side gate signal changes to L level and the time when the high-voltage side gate signal changes to H level.

[0063] FIG. 3 further shows waveform diagrams of output end voltages VTrp, VTrs of the first bridge 10 and the second bridge 20, transformer currents ITrp and ITrs flowing through the primary winding 30p and secondary winding of the transformer 30, respectively, and reactor currents IL1, IL2 when the first bridge 10 and the second bridge 20 operate in accordance with the gate signals S11H, S11L, S12H, S12L, S21H, S21L, S22H, S22L.

[0064] The output voltages VTrp and VTrs change sequentially according to the switching pattern, which is a combination of H / L of the gate signals S11H, S11L, S12H, S12L, S21H, S21L, S22H, and S22L. In addition, in the simulation waveforms in Fig. 3, the transformer 30 is an ideal transformer (with infinite excitation inductance) and the turns ratio is 1:1, so ITrp is approximately equal to ITrs.

[0065] The reactor current IL1 rises (increases in absolute value) during the H level period of the gate signal S11L, and drops (decreases in absolute value) during the H level period of the gate signal S11H. By setting the duty ratio DH according to equation (3), IL1=0 at the time when the gate signal S11H changes from H level to L level, and a current discontinuous period can be provided in the reactor current IL1 thereafter.

[0066] Similarly, reactor current IL2 rises (increases in absolute value) during the H level period of gate signal S12L, and falls (decreases in absolute value) during the H level period of gate signal S12H. By setting duty ratio DH according to equation (3), IL2=0 when gate signal S12H changes from H level to L level, and a current discontinuous period can be provided in reactor current IL2 thereafter.

[0067] As a result, in the power conversion device according to the present embodiment, during the discharge operation, the high-voltage side switching elements Q11H and Q 12H, Q21H, Q22H In this way, it becomes possible to provide an on-period without stopping (fixing to off) the reactor currents IL1 and IL2, and to control the reactor currents IL1 and IL2 in a discontinuous current mode.

[0068] Furthermore, by providing an on-period of the high-voltage side switching elements Q11H, Q12H, transformer currents ITrs, ITrp are generated during the on-period of the diagonal switching elements Q12L, Q11L (low-voltage side). This allows power to be transferred from the first bridge 10 to the second bridge 20, in other words, from the capacitor C1 to the load 2 (capacitor C2) via the transformer 30.

[0069] In contrast, when a discontinuous current mode is established by stopping (fixing off) the high-voltage side switching elements Q11H, Q12H, power cannot be transmitted from the first bridge 10 to the second bridge 20 during the period when the absolute values ​​of the reactor currents IL1, IL2 are decreasing.

[0070] In this way, the power conversion device according to the present embodiment enables switching control of the first bridge 10 and the second bridge 20 that can simultaneously introduce a discontinuous current mode for suppressing conduction loss during low output and ensure power transmission operation.

[0071] (Control and circuit operation during charging) Next, control and operational waveform examples during the charging operation of the power conversion device 100 will be described with reference to FIGS.

[0072] 4, during charging operation, the control calculation unit 50 generates the first control command value REF1 in a mode in which a multiplier 54x that multiplies the output value of the controller 54 by "-1" is further provided. That is, the first control command value REF1 during charging operation is calculated so as to be limited to REF1=0 when the voltage command value VREF1 is higher than the DC voltage V1, as opposed to during discharging operation.

[0073] On the other hand, the voltage deviation ΔV2 is calculated by ΔV2=VREF2-V2, as in the discharging operation. That is, also in the charging operation, the first control command value REF1 is a command value for controlling the DC voltage V1 to the voltage command value VREF1, and the second control command value REF2 is a command value for controlling the DC voltage V2 to the voltage command value VREF2.

[0074] When the DC voltage V2 is higher than the voltage command value VREF2, ΔV2<0 and REF2<0, so that the phase difference φ<0, i.e., the phase of the switching operation of the second bridge 20 leads the phase of the switching operation of the first bridge 10. As a result, the DC voltage V2 can be reduced toward the voltage command value VREF2 by transmitting power from the second bridge 20 to the first bridge 10. At this time, the larger the absolute value of ΔV2 is, the larger the amount of phase lead due to the phase difference φ is set, and the more power is transmitted from the second bridge 20 to the first bridge 10.

[0075] During charging operation, in the gate signal generating unit 60, high-voltage side gate signals S11H, S12H, S21H, and S22H are generated with the first control command value REF1 from the control calculation unit 50 as the high-voltage side duty ratio DH. Note that the second control command value REF2 is given to the carrier wave generators CG21 and CG22 as the phase difference φ [deg] of the carrier waves CW21 and CW22 corresponding to the second bridge 20 with respect to the carrier waves CW11 and CW12 corresponding to the first bridge 10, as in the discharging operation.

[0076] Therefore, the comparators CMP11, CMP12, CMP21, and CMP22 generate high-voltage side gate signals S11H, S12H, S21H, and S22H by performing a voltage comparison between the first control command value REF1 (high-voltage side duty ratio DH) and each of the carrier waves CW11, CW12, CW21, and CW22 similar to those in FIG. 2.

[0077] The low-voltage side gate signals S11L, S12L, S21L, and S22L during charging operation are generated using the third control command value REF3 calculated by the duty ratio calculation unit 70 as the low-voltage side duty ratio DL.

[0078] During charging operation where the reactor current IL<0, the above equations (1) and (2) are transformed into the following equations (4) and (5) using the minimum current ILmin (ILmin<0), which is the maximum value for negative current.

[0079] ILmin=(VBAT-V1)×DH×Ts…(4) IL0=ILmin+(VBAT / L)×DL×Ts…(5) From equations (4) and (5), it can be seen that the duty ratio DL at which IL0=0 is given by the following equation (6).

[0080] DL = DH × (V1 - VBAT) / VBAT … (6) Therefore, during the discharging operation, the duty ratio calculation unit 70 calculates the third control command value REF3, that is, the duty ratio DL on the low voltage side, according to the above equation (6).

[0081] Then, the logic units LG11, LG12, LG21, and LG22 generate low-voltage side gate signals S11L, S12L, S21L, and S22L based on the high-voltage side gate signals S11H, S12H, S21H, and S22H and the third control command value REF3 (low-voltage side duty ratio DL).

[0082] Specifically, the low-voltage side gate signals S11L, S12L, S21L, and S22L are generated so that an H-level period of time (DL x Ts) according to the duty ratio DL is provided, starting from the transition timing from H level to L level of each of the high-voltage side gate signals S11H, S12H, S21H, and S22H.

[0083] In this way, during charging operation, in contrast to discharging operation, the high-voltage side gate signals S11H, S12H, S21H, and S22H are PWM controlled in accordance with the first control command value REF1, while the low-voltage side gate signals S11L, S12L, S21L, and S22L are switching controlled to achieve both the introduction of discontinuous current mode and power transmission.

[0084] FIG. 5 shows a simulation waveform diagram illustrating an example of an operation during a charging operation of the power conversion device 100 using the gate signals generated according to FIG.

[0085] 5, during charging, the power P12 transmitted from the first bridge 10 to the second bridge 20 is a negative value (P12<0). The DC voltage V1 of the capacitor C1 is controlled to a constant value in accordance with the voltage command value VREF1.

[0086] In the first bridge 10 and the second bridge 20, the H-level period lengths of the high-voltage side gate signals S11H, S12H, S21H, and S22H are set according to the duty ratio DH (first control command value REF1) for controlling the DC voltage V1.

[0087] During charging operation, since P12<0, REF2<0 (φ<0), the second control command value REF2 is reflected in the phase difference φ [deg] of the initial phases in the carrier wave generators CG11, CG12, CG21, and CG22 in FIG. 4, and as a result, the phases of the gate signals S21H, S22H of the second bridge 20 lead the phases of the gate signals S11H, S12H of the first bridge 10.

[0088] The low-voltage side gate signals S11L, S12L, S21L, and S22L are set to H level for a period according to the low-voltage side duty ratio DL after the high-voltage side gate signals S11H, S12H, S21H, and S22H transition from H level to L level. chargingDuring operation, in each of the legs 11, 12, 21, and 22, a dead time is generally provided between the time when the high-voltage side gate signal changes to L level and the time when the low-voltage side gate signal changes to H level.

[0089] FIG. 5 also shows waveform diagrams of the output end voltages VTrp, VTrs of the first bridge 10 and the second bridge 20, similar to those in FIG. 3 (during discharging operation), the transformer currents ITrp and ITrs flowing through the primary winding 30p and the secondary winding of the transformer 30, respectively, and the reactor currents IL1, IL2.

[0090] The reactor current IL1 decreases (increases in absolute value) during the H level period of the gate signal S11H, and increases (decreases in absolute value) during the H level period of the gate signal S11L. By setting the duty ratio DL according to equation (6), IL1=0 at the time when the gate signal S11L changes from H level to L level, and a current discontinuous period can be provided in the reactor current IL1 thereafter.

[0091] Similarly, reactor current IL2 decreases (increases in absolute value) during the H level period of gate signal S12H, and increases (decreases in absolute value) during the H level period of gate signal S12L. By setting duty ratio DL according to equation (6), IL2=0 when gate signal S12L changes from H level to L level, and a current discontinuous period can be provided in reactor current IL2 thereafter.

[0092] As a result, in the power conversion device of this embodiment, during charging operation, it is possible to control the reactor currents IL1 and IL2 in discontinuous current mode by providing an on-period without stopping (fixing to off) the low-voltage side switching elements Q11L, Q12L, Q21L, and Q22L.

[0093] Furthermore, by providing an on-period of the low-voltage side switching elements Q11L, Q12L, transformer currents ITrs, ITrp are generated during the on-period of the diagonal switching elements Q12H, Q12H (low-voltage side). This allows power to be transferred from the second bridge 20 to the first bridge 10, in other words, from the load 2 (capacitor C2) to the storage battery 1 via the transformer 30.

[0094] In contrast, when the discontinuous current mode is established by stopping the low-voltage side switching elements Q11L, Q12L, power cannot be transmitted from the first bridge 10 to the second bridge 20 during the period when the reactor currents IL1, IL2 decrease.

[0095] As described above, according to the control operation of the power conversion device of the first embodiment, in both the discharging operation and the charging operation, the power conversion performance can be improved by introducing the discontinuous current mode for suppressing the conduction loss at low output and the switching control of the first bridge 10 and the second bridge 20 for achieving both ensuring the power transmission operation.

[0096] The gate signal generating unit 60 executes the control operation of FIG. 2 during the discharging operation in accordance with the first control command value REF1 and the second control command value REF2 from the control calculation unit 50. charging During operation, the gate signal generating unit 60 is configured to have a control operation switching function for executing the control operation of Fig. 4. For example, the control operation during discharging operation (ΔV2 ≧ 0) and charging operation (ΔV2 < 0) can be switched according to the polarity (positive / negative) of the voltage deviation ΔV2. Alternatively, the gate signal generating unit 60 can be configured to perform control during discharging operation when V1 > V2, while performing control during charging operation when V1 ≦ V2, according to a comparison between the DC voltages V1 and V2.

[0097] A variation of the first embodiment. In the first embodiment, a control example has been described in which the second control command value REF2 by the control calculation unit 50 is reflected in the phase difference φ [deg] of the carrier wave, but it is also possible to reflect the second control command value REF2 in the frequency of the carrier wave.

[0098] FIG. 6 is a block diagram illustrating a control operation during a discharging operation of the power conversion device according to the modification of the first embodiment.

[0099] 6, in the modification of the first embodiment, in the control calculation unit 50, the second control command value REF2 for controlling the DC voltage V2 to the voltage command value VREF2 is calculated as the frequency change amount df of the carrier waves CW11, CW12, CW21, and CW22. The frequency change amount df is obtained as a positive value (df>0) during a discharging operation, that is, when ΔV2>0. On the other hand, the first control command value REF1 is calculated in the same manner as in the first embodiment (FIG. 2).

[0100] 2, the gate signal generating unit 60 further includes a carrier frequency setting unit 65. The carrier frequency setting unit 65 calculates a carrier frequency fcw by subtracting a frequency change amount df from a predetermined reference frequency fa.

[0101] In the modification of the first embodiment, the carrier wave generators CG11, CG12, CG21, and CG22 each generate a carrier wave CW11, CW12, CW21, and CW22 having a carrier frequency fcw set by the carrier frequency setting unit 65. That is, the periods (1 / fcw) of the carrier waves CW11, CW12, CW21, and CW22 are variable in accordance with the second control command value REF2.

[0102] In contrast, in the modification of the first embodiment, the phase difference between the carrier waves CW11, CW12, CW21, and CW22 is fixed. Specifically, the phase of the carrier waves CW21 and CW22 corresponding to the second bridge 20 is fixed to lag behind the phase of the carrier waves CW11 and CW12 corresponding to the first bridge 10 by a reference phase φa. That is, in the modification of the first embodiment, the phase difference φ [deg] between the first bridge 10 and the second bridge 20 is fixed to a constant value that does not depend on the second control command value REF2 (φ=+φa during a discharging operation).

[0103] The comparators CMP11, CMP12, CMP21, and CMP22 operate in the same manner as in FIG. 2, and generate low-voltage side gate signals S11L, S12L, S21L, and S22L by comparing the voltage of the first control command value REF1 (low-voltage side duty ratio DL) with each of the carrier waves CW11, CW12, CW21, and CW22, whose frequencies are variable as described above.

[0104] Furthermore, similar to FIG. 2, the duty ratio calculation unit 70 calculates the high-voltage side duty ratio DH according to the above-mentioned equation (3), and the logic units LG11, LG12, LG21, and LG22 generate high-voltage side gate signals S11H, S12H, S21H, and S22H so as to achieve both the introduction of discontinuous current mode and power transmission.

[0105] In the case of a discharge operation, i.e., when the voltage deviation ΔV2>0 (VREF2>V2), the greater the absolute value of the voltage deviation ΔV2, the greater the amount of decrease in the carrier frequency fcw, and therefore the greater the switching period length Ts. Since the power transmitted from the first bridge 10 to the second bridge 20 increases in proportion to the switching period length Ts (Ts=1 / fcw), it is possible to execute a control operation during a discharge operation similar to that described in FIG. 2 by controlling the second control command value REF2 for controlling the DC voltage V2 to be reflected in the carrier frequency fcw.

[0106] FIG. 7 is a block diagram illustrating a control operation during a charging operation of a power conversion device according to a modification of the first embodiment.

[0107] 7, in the modification of the first embodiment, the control calculation unit 50 in the charging operation calculates the second control command value REF2 for controlling the DC voltage V2 to the voltage command value VREF2 as the frequency change amount df of the carrier waves CW11, CW12, CW21, and CW22, similarly to the discharging operation (FIG. 6). On the other hand, the first control command value REF1 in the charging operation is calculated similarly to the first embodiment (FIG. 4).

[0108] In the gate signal generating unit 60, during charging operation, the carrier frequency setting unit 65 calculates the carrier frequency fcw by adding the frequency change amount df (second control command value REF2) obtained from the voltage deviation ΔV2 (ΔV2<0) which is set to a negative value to the reference frequency fa.

[0109] Similarly to the discharging operation (FIG. 6), each of the carrier wave generators CG11, CG12, CG21, and CG22 changes the frequency of the carrier waves CW11, CW12, CW21, and CW22 according to the carrier frequency fcw set by the carrier frequency setting unit 65. On the other hand, unlike the discharging operation, the charging operation fixes the phase of the carrier waves CW21 and CW22 corresponding to the second bridge 20 to lead by a reference phase φa with respect to the phase of the carrier waves CW11 and CW12 corresponding to the first bridge 10 (φ=-φa during the charging operation).

[0110] In this manner, in the modification of the first embodiment, the phase difference φ [deg] between the first bridge 10 and the second bridge 20 is switched between the lagging direction and the leading direction between the discharging operation and the charging operation, and the switching period length Ts in the first bridge 10 and the second bridge 20 is changed in accordance with the change in the switching period length Ts reflecting the second control command value REF2.

[0111] Comparators CMP11, CMP12, CMP21, and CMP22 operate in the same manner as in Figure 4, and generate high-voltage side gate signals S11H, S12H, S21H, and S22H by comparing the voltages of the first control command value REF1 (high-voltage side duty ratio DH) with each of the carrier waves CW11, CW12, CW21, and CW22 generated as described above.

[0112] Furthermore, similar to FIG. 4, the duty ratio calculation unit 70 calculates the low-voltage side duty ratio DL according to the above-mentioned equation (6), and the logic units LG11, LG12, LG21, and LG22 generate low-voltage side gate signals S11L, S12L, S21L, and S22L so as to achieve both the introduction of discontinuous current mode and power transmission.

[0113] In the charging operation, i.e., in the case of voltage deviation ΔV2<0 (V2>VREF2), the larger the absolute value of the voltage deviation ΔV2, the larger the amount of decrease in carrier frequency fcw becomes, and therefore the switching period length Ts becomes larger. Since the power transmitted from the second bridge 20 to the first bridge 10 becomes larger in proportion to the switching period length Ts, it is possible to execute the control operation during the charging operation similar to that described in FIG. 4 by controlling the second control command value REF2 for controlling the DC voltage V2 to be reflected in the carrier frequency fcw.

[0114] In this way, according to the control operation of the power conversion device related to the modified example of embodiment 1, even if the second control command value REF2 for controlling the DC voltage V2 is reflected in the frequency of the carrier wave, i.e., the switching period length, it is possible to provide control of the power conversion device similar to that of embodiment 1.

[0115] Embodiment 2 FIG. 8 is a block diagram illustrating a control operation during a discharging operation of the power conversion device according to the second embodiment.

[0116] Comparing Fig. 8 with Fig. 2 showing the control operation during the discharge operation according to the first embodiment, the second embodiment differs in that the gate signal generating unit 60 further includes an additional pulse applying unit 75. Other configurations of the control calculation unit 50 and the gate signal generating unit 60 are similar to those in Fig. 2.

[0117] Gate signals S21L, S21H, and S2 2 The gate signals S11H, S11L, S12H, and S12L of the first bridge 10 generated by the control calculation unit 50 and the gate signal generation unit 60 in Fig. 2 are input to the additional pulse application unit 75 as reference gate signals S11H*, S11L*, S12H*, and S12L*.

[0118] The additional pulse applying unit 75 generates gate signals S11H, S11L, S12H, and S12L for the gate signal of the first bridge 10 in such a manner that an on-pulse is added to the above-mentioned gate signals S11H*, S11L*, S12H*, and S12L* for alternately turning on and off the switching elements on the high-voltage side and the low-voltage side during the current discontinuous period (IL1=0, IL2=0) of the reactor currents IL1 and IL2.

[0119] FIG. 9 shows a simulation waveform diagram illustrating an example of the operation of the power conversion device 100 during the discharging operation using the gate signals generated according to FIG.

[0120] As shown in FIG. 9, the gate signals S21H, S21L, S22H, and S22L of the second bridge 20 have the same waveforms as those in FIG. 3 without any additional pulses being added.

[0121] In response to this, additional pulses 76, 77 are added to the gate signals S11H, S11L, S12H, and S12L of the first bridge 10. Specifically, the additional pulse 76 is added to the gate signals S11L and S11H corresponding to leg 11 during a period in which the reactor current IL1=0 (current discontinuous period) until the start of the next switching period. As a result, after the timing of the transition of the gate signal S11H in Fig. 3 (corresponding to the gate signal S11H* in Fig. 8) from the H level to the L level, the gate signals S11L and S11H are alternately set to the H level and the L level.

[0122] Similarly, the additional pulse applying unit 75 applies an additional pulse 77 to the gate signals S12L and S12H corresponding to leg 12 during a period (current discontinuous period) in which the reactor current IL2 is 0 until the start of the next switching period. As a result, after the timing of the transition of the gate signal S12H in Fig. 3 (corresponding to the gate signal S12H* in Fig. 8) from the H level to the L level, the gate signals S12L and S12H are alternately set to the H level and the L level.

[0123] By providing such additional pulses 76 and 77, a generation period of the transformer currents ITrp and ITrs can be added to the area surrounded by the dotted line in FIG. 9, compared to the operation waveform example in FIG. 3. As a result, under the same transmission power as in FIG. 3, the amplitude of the transformer currents ITrp and ITrs generated at the same timing as in FIG. 3 is suppressed. This makes it possible to suppress the maximum current of the transformer 30 and each switching element constituting the first bridge 10 and the second bridge 20. This makes it possible to reduce the power loss in each switching element and the transformer 30, thereby improving the power conversion efficiency and increasing the transmission power under the same circuit configuration (element specifications).

[0124] Here, the pulse width of each of the additional pulses 76 and 77 can be determined in advance within a range that does not increase the conduction loss in the reactors L1 and L2. Alternatively, the pulse width can be determined so that the length of time during which each switching element on the power transmission side can be turned on and off by zero voltage switching is secured.

[0125] 8, the calculated high-voltage side duty ratio DH, low-voltage side duty ratio DL, switching period length Ts, and pulse width PW are input to additional pulse applying unit 75. The duration of the current discontinuous period of reactor currents IL1 and IL2 is calculated by Ts×(1-DL-DH) using the duty ratio. Therefore, the number of pulses included in additional pulses 76 and 77 shown in FIG. 9 can be set by using the pulse width PW set from the power loss perspective as described above and the duration of the calculated zero current period.

[0126] In this way, the additional pulse applying unit 75 performs a logical operation to apply on-pulses of a given pulse width PW as additional pulses 76, 77 in accordance with the number of pulses that can be added within the period in which the reactor currents IL1, IL2 = 0, which is calculated by the above-mentioned calculation, and generates the gate signals S11H, S11L, S12H, and S12L of the first bridge 10.

[0127] The additional pulse applying unit 75 can also be applied to the control during charging operation. That is, the additional pulse applying unit 75 can be further arranged in the configuration of FIG. 4. In this case, the gate signals S11H, S11L, S12H, and S12L of the first bridge 10 generated by the control calculation unit 50 and the gate signal generating unit 60 of FIG. 4 can be input to the additional pulse applying unit 75 as the reference gate signals S11H*, S11L*, S12H*, and S12L*. As a result, the additional pulses 76 and 77 similar to those of FIG. 9 can be applied to the gate signals S11H, S11L, S12H, and S12L of the first bridge 10.

[0128] Also in the configurations of FIGS. 6 and 7, by adding an additional pulse applying section 75, it is possible to combine the modified example of the first embodiment with the second embodiment.

[0129] In this way, according to the control operation of the power conversion device according to the second embodiment, the application of the additional pulses 76 and 77 can provide an additional power transmission period between the first bridge 10 and the second bridge 20. This makes it possible to suppress the maximum current when transmitting the same power, thereby improving the power conversion efficiency by reducing the power loss, and increasing the power that can actually be transmitted, thereby further improving the power conversion performance.

[0130] Embodiment 3 In the third embodiment, a modified example of the main circuit configuration of the power conversion device will be described.

[0131] FIG. 10 is a circuit diagram illustrating a circuit configuration of a power conversion device 101 according to a first example of the third embodiment.

[0132] 10, the power conversion device 101 is different from the power conversion device 100 shown in Fig. 1 in the connection position of the capacitor C1. Specifically, the capacitor C1 is connected between the power line PL1 and the node Ni, not between the power lines PL1 and NL1 (Fig. 1). Therefore, in the power conversion device 101, the capacitor C1 and the storage battery 1 are connected in series between the power lines PL1 and NL1.

[0133] Therefore, in the power conversion device 101, the voltage command value VREF1 is obtained by subtracting the battery voltage VBAT from the command value of the DC voltage between the power lines PL1 and NL1 (corresponding to VREF1 in the first and second embodiments). Therefore, also in the third embodiment, the voltage deviation ΔV1 for calculating the first control command value REF1 can be calculated by ΔV1=VREF1-V1. That is, also in the third embodiment, the voltage command value VREF1 is a command value for controlling the DC voltage between the power lines PL1 and NL1. Note that the voltage command value VREF2 is the same as in the first and second embodiments. In the third embodiment, the sum of the DC voltage V1 and the battery voltage VBAT corresponds to the "first DC voltage". On the other hand, as in the first and second embodiments, the DC voltage V2 corresponds to the "second DC voltage".

[0134] In the power conversion device 101, the voltage applied across the reactors L1, L2 during the on-period of the switching elements Q11H and Q21H on the high-voltage side of the first bridge 10 is different from that in the power conversion device 100. This causes the slope of the reactor currents IL1, IL2 during the period to change, so that the duty ratio calculation unit 70 needs to change the calculation equation for the high-voltage side duty ratio DH during discharging operation and the calculation equation for the low-voltage side duty ratio DL during charging operation in order to appropriately introduce the discontinuous current mode.

[0135] Specifically, while the high-voltage side switching elements Q11H and Q21H are on, the voltage applied across reactors L1, L2 is (VBAT-V1) in FIG. 1, but is -V1 in FIG.

[0136] Therefore, during the discharging operation, the duty ratio DH on the high voltage side can be calculated by the following equation (7) in which (V1-VBAT) is replaced with V1 in the above equation (3).

[0137] DH = DL × VBAT / V1 … (7) Similarly, during charging operation, the duty ratio DL on the low voltage side can be calculated by the following equation (8) in which (V1-VBAT) is replaced with V1 in the above equation (6).

[0138] DL = DH × V1 / VBAT … (8) For the power conversion apparatus 101, in the above-mentioned first embodiment and its modification, and second embodiment, the calculation formula for the voltage deviation ΔV1 is changed, and the calculation formula used in the duty ratio calculation unit 70 is changed to the above-mentioned formulas (7) and (8), so that it is possible to apply a control operation similar to that for the power conversion apparatus 100. As a result, the effects of the above-mentioned first embodiment and its modification, and second embodiment can be obtained for the power conversion apparatus 101 according to the first example of the third embodiment.

[0139] FIG. 11 is a circuit diagram illustrating a circuit configuration of a power conversion device 102 according to a second example of the third embodiment.

[0140] 11, in the power conversion device 102, each of the first bridge 10 and the second bridge 20 in the power conversion device 100 is configured as a three-phase bridge. As a result, instead of the transformer 30 in FIG. 1, a transformer 31 (three-phase transformer) having three-phase primary windings 31p, 32p, and 33p and secondary windings 31s, 32s, and 33s is connected between the first bridge 10 and the second bridge. Each of the primary winding 31p and the secondary winding 31s, the primary winding 32p and the secondary winding 32s, and the primary winding 33p and the secondary winding 33s is magnetically coupled.

[0141] The first bridge 10 further includes a leg 13 in addition to the legs 11 and 12 similar to those in Fig. 1. The leg 13 is composed of a high-voltage side switching element Q13H and a low-voltage side switching element Q13L connected in series between the power lines PL1 and NL1 via a node N13. The switching elements Q13H and Q13L are on / off controlled according to gate signals S13H and S13L, respectively. A reactor L3 is connected between the node N13 of the leg 13 and a node Ni (the positive electrode of the storage battery 1).

[0142] Similarly, the second bridge 20 further includes a leg 23 in addition to the legs 21 and 22 similar to those in Fig. 1. Leg 23 is composed of a high-voltage side switching element Q23H and a low-voltage side switching element Q23L connected in series between power lines PL1 and NL1 via node N23. The switching elements Q23H and Q23L are on / off controlled according to gate signals S23H and S23L, respectively.

[0143] The three-phase transformer 31 is YY-connected, that is, both the primary side and the secondary side are star-connected. Specifically, one ends of the primary windings 31p, 32p, and 33p are interconnected, and the other ends of the primary windings 31p, 32p, and 33p are connected to nodes N11, N12, and N13, respectively. Similarly, one ends of the secondary windings 31s, 32s, and 33s are interconnected, and the other ends of the primary windings 31p, 32p, and 33p are connected to nodes N11, N12, and N13, respectively. Note that the first bridge 10 and the second bridge 20 may be connected to the transformer 31 in the form of Y-Δ connection, ΔY connection, or ΔΔ connection.

[0144] The reactors LT1 to LT3 are connected in series to the primary windings 31p to 33p, respectively. The reactors LT1 to LT3 can also be formed by leakage inductances of the magnetic coupling between the primary winding 31p and the secondary winding 31s, between the primary winding 32p and the secondary winding 32s, and between the primary winding 33p and the secondary winding 33s.

[0145] Fig. 12 is a block diagram for explaining an example of a control operation of the power conversion device 102 shown in Fig. 11. In Fig. 12, a block diagram for a control operation during a discharging operation is shown.

[0146] 12, in the power conversion device 102, the control calculation unit 50 is configured similarly to that in Fig. 2, that is, similarly to the control of the power conversion device 100. Meanwhile, in order to generate gate signals for the switching elements of the three legs, a gate signal generation unit 61 is provided instead of the gate signal generation unit 60 in Fig. 2.

[0147] The gate signal generating unit 61 includes carrier wave generators CG11, CG12, CG21, CG22, comparators CMP11, CMP12, CMP21, CMP22, a duty ratio calculation unit 70, and logic units LG11, LG12, LG21, LG22 similar to the gate signal generating unit 60, and further includes carrier wave generators CG13, CG23, comparators CMP13, CMP23, and logic units LG13, LG23 corresponding to legs 13, 23.

[0148] While the gate signal generating unit 60 switches the two legs of each of the first bridge 10 and the second bridge 20 with a phase difference of 180 degrees, the gate signal generating unit 61 switches the three legs of each of the first bridge 10 and the second bridge 20 with a phase difference of 120 degrees. This allows the first bridge 10 and the second bridge 20 to operate as a DAB circuit.

[0149] Therefore, in the carrier wave generators CG11, CG12, and CG13 corresponding to the first bridge 10, the initial phases are set to 0 [deg], 120 [deg], and 240 [deg], respectively, so that a phase difference of 120 [deg] occurs in the carrier waves CW11 to CW13. The initial phases of the carrier wave generators CG21, CG22, and CG23 corresponding to the second bridge 20 are set to have a phase difference φ [deg] with respect to the initial phases of the carrier wave generators CG11, CG12, and CG13 corresponding to the first bridge 10. As a result, during the discharge operation, the switching operations of the legs 21 to 23 have a phase difference φ [deg] in the lagging direction with respect to the switching operations of the legs 11 to 13, so that power is transmitted from the first bridge 10 to the second bridge 20.

[0150] The comparators CMP11-CMP13 generate gate signals S11L-S13L by comparing the voltages of the low-voltage side duty ratio DL (first control command value REF1) and the carrier waves CW11-CW13 (similar to FIG. 2). Similarly, the comparators CMP21-CMP23 generate gate signals S21L-S23L by comparing the voltages of the low-voltage side duty ratio DL (first control command value REF1) and the carrier waves CW21-CW23 (similar to FIG. 2).

[0151] The logic units LG11 to LG13 generate gate signals S11H to S13H based on the gate signals S11L to S13L and a high-voltage side duty ratio DH calculated by a duty ratio calculation unit 70 similar to that of FIG. 2. As described in the first embodiment, the gate signals S21H to S23H are generated such that an H-level period is provided for a period length according to the high-voltage side duty ratio DH after the gate signals S11L to S13L transition from H level to L level. Similarly, the logic units LG21 to LG23 generate gate signals S21H to S23H based on the gate signals S21L to S23L and a high-voltage side duty ratio DH calculated by a duty ratio calculation unit 70 similar to that of FIG. 2.

[0152] By controlling the on / off of each switching element of the first bridge 10 and the second bridge 20 in accordance with the gate signals S11L-S13L, S11H-S13H, S21L-S23L, and S21H-S23H generated in this manner, the power conversion device 102 shown in FIG. 11 also realizes switching control of the first bridge 10 and the second bridge 20 that can simultaneously introduce a discontinuous current mode and ensure power transmission operation, similar to the discharging operation of the power conversion device 100 described in the first embodiment.

[0153] For the power conversion device 102, in Figures 4 (charging operation), 7 (discharging operation with variable carrier frequency), and 8 (charging operation with variable carrier frequency), similar control can be achieved by modifying the arrangement to include carrier wave generators CG11-CG13, CG21-CG23, comparators CMP11-CMP13, CMP21-CMP23, and logic units LG11-LG13, LG21-LG23 for three legs, as in Figure 12.

[0154] Also, in the power conversion device 102 in the configuration of Fig. 11, it is possible to configure the capacitor C1 to be connected between the power line PL1 and the node Ni (the positive electrode of the storage battery 1) as in Fig. 9. In this case, as explained in Fig. 9, it is necessary to change the calculation of the voltage deviation ΔV1 and to change the calculation formula of the duty ratio in the duty ratio calculation unit 70 for introducing the discontinuous current mode.

[0155] In this manner, similarly to the power conversion devices 100 and 101, the power conversion device 102 in which the first bridge 10 and the second bridge 20 each have three legs can also improve the power conversion performance during both discharging and charging operations by introducing a discontinuous current mode for suppressing conduction loss during low output and by introducing switching control of the first bridge 10 and the second bridge 20 for achieving both ensuring power transmission operation.

[0156] The number of legs in each of the first bridge 10 and the second bridge 20 can be any number (N: N≧2). In this case, a phase difference of (360 / N) [deg] is provided in the switching operation between the N legs of the first bridge 10 and the second bridge 20. Increasing N makes it possible to reduce the passing current of the reactor and the switching element, but increases the circuit size.

[0157] It should be noted that the same circuit operation can be achieved even if the turns ratio of the primary winding and secondary winding in the transformers 30 and 31 is not 1:1. In particular, by making the number of turns of the primary winding greater than the number of turns of the secondary winding to a turns ratio of n:1 (n>1), the transformers 30 and 31 can be configured so that a voltage n times that of the secondary winding appears in the primary winding. This makes it possible to utilize the boost function on the primary side to perform a discharge operation even when VBAT>V2, thereby relaxing the conditions under which power can be transmitted.

[0158] The functions of each circuit element and each block shown in the block diagrams explaining the control operation, including Fig. 2, can be realized by at least one of software processing and hardware processing. That is, the control calculation unit 50 and the gate signal generation unit 60 can be configured as a computer-based configuration in which a CPU (Central Processing Unit) (not shown) executes a program stored in advance in a memory, or can be configured by an FPGA (Field Programmable Gate Array), an ASIC (Application Specific Integrated Circuit), an electronic circuit (analog circuit), or the like, or can be configured by a combination of these.

[0159] It should be noted that in the power conversion device according to the present embodiment, it is not necessary to constantly apply the switching control in which the discontinuous current mode described above is introduced, regardless of the load condition, etc. In other words, as long as there is a mode in which the switching control in which the discontinuous current mode described in the first to third embodiments is introduced is performed depending on the load condition (for example, when the transmitted power is small), switching control in which the discontinuous current mode does not exist may be performed under other load conditions (for example, when the transmitted power is large).

[0160] Although various exemplary embodiments and examples are described in this disclosure, the various features, aspects, and functions described in one or more embodiments are not limited to application to a particular embodiment, but may be applied to the embodiments alone or in various combinations.

[0161] Therefore, countless modifications not illustrated are assumed within the scope of the technology of the present disclosure, including, for example, modifying, adding, or omitting at least one component, and further, extracting at least one component and combining it with a component of another embodiment.

[0162] The embodiments disclosed herein should be considered to be illustrative and not restrictive in all respects. The scope of the present disclosure is defined by the claims, not the above description, and is intended to include all modifications within the meaning and scope of the claims. [Explanation of symbols]

[0163] 1 storage battery, 2 load, 10 first bridge, 11-13, 21-23 legs, N11-N13, N21-N23, Ni node, 20 second bridge, 30, 31 transformer, 30p, 31p, 32p, 33p primary winding, 30s, 31s, 32s, 33s secondary winding, 50 control calculation unit, 52, 56 deviation calculation unit, 54, 58 controller, 60, 61 gate signal generation unit, 65 carrier frequency setting unit, 70 duty ratio calculation unit, 75 additional pulse application unit, 76, 77 additional pulse, 100-102 power conversion device, C1, C2 capacitor, CG11-CG13, CG21-CG23 carrier wave generator, CMP11-CMP13, CMP21-CMP23 Comparator, CW11~CW13, CW21~CW23 Carrier wave, DH, DL Duty ratio, ITrp, ITrs Transformer current, L1~L3, LT, LT1~LT3 Reactor, LG11~LG13, LG21~LG23 Logic section, NL1, NL2, PL1, PL2 Power line, PW Pulse width, Q11H, Q11L, Q12H, Q12L, Q13H, Q13L, Q21L, Q21H, Q22H, Q22L, Q23H, Q23L Switching element, REF1 First control command value, REF2 Second control command value, REF3 Third control command value, S11H, S11L, S12H, S12L, S13H, S13L, S21H, S21L, S22H, S22L, S23L, S23H gate signal, Ts switching cycle length, V1, V2 DC voltage, VBAT battery voltage, VREF1, VREF2 voltage command value, df frequency change amount, fa reference frequency, fcw carrier frequency.

Claims

1. A power conversion device that performs DC voltage conversion, a first bridge having a plurality of first legs; a second bridge having a plurality of second legs; a transformer connected between the first bridge and the second bridge, Each of the plurality of first legs comprises: a high-voltage side switching element and a low-voltage side switching element connected in series between a first power line on a high-voltage side and a second power line on a low-voltage side via an intermediate node; Each of the plurality of second legs comprises: a high-voltage side switching element and a low-voltage side switching element connected in series between a third power line on a high-voltage side and a fourth power line on a low-voltage side via an intermediate node; The transformer is a primary winding connected to the intermediate nodes of the first legs; a secondary winding connected to the intermediate nodes of the second legs and magnetically coupled to the primary winding; The power conversion device is a plurality of reactors respectively connected between a rechargeable DC power source and the intermediate nodes of the first legs; a control circuit that controls on / off of each of the switching elements in each of the plurality of first legs and second legs, The control circuit includes: in the first bridge, in accordance with a first control command value for controlling a first DC voltage between the first and second power lines, an on-period of one of the high-voltage side and low-voltage side switching elements of each of the first legs is set to increase an absolute value of a reactor current flowing through each of the plurality of reactors, and after an on-period of the one switching element ends, an on-period of the other of the high-voltage side and low-voltage side switching elements is set until the absolute value of the reactor current returns to zero; and a second control command value for controlling a second DC voltage between the third and fourth power lines in the second bridge is reflected, and on / off of each switching cycle of each of the switching elements in each of the second legs is controlled so that on-period lengths of the switching elements on the high-voltage side and the switching elements on the low-voltage side are equivalent between the plurality of first legs and the plurality of second legs.

2. 2. The power conversion device according to claim 1, wherein, when the power conversion device operates to increase the second DC voltage, in each of the first legs, the one switching element is the low-voltage side switching element and the other switching element is the high-voltage side switching element.

3. 2. The power conversion device according to claim 1, wherein, when the power conversion device operates to reduce the second DC voltage, in each of the first legs, the one switching element is a switching element on the high-voltage side, and the other switching element is a switching element on the low-voltage side.

4. The power conversion device according to any one of claims 1 to 3, wherein the control circuit controls on / off of each switching period of each of the switching elements of each of the first legs so that, in each switching period in the first bridge, after an on period of the one switching element and an on period of the other switching element end, the one switching element and the other switching element are alternately turned on and off.

5. The power conversion device according to any one of claims 1 to 4, wherein the control circuit controls on / off of each switching period of each of the switching elements in each of the second legs so as to fix a time length of the switching period of each of the switching elements in the plurality of first legs and the plurality of second legs, and to change a phase difference of a switching operation of the plurality of second legs relative to the switching operation of the plurality of first legs in accordance with the second control command value.

6. The power conversion device according to any one of claims 1 to 4, wherein the control circuit controls on / off of each switching period of each of the switching elements of each of the second legs so as to change a time length of the switching period of each of the switching elements of the plurality of first legs and the plurality of second legs in accordance with the second control command value, after fixing a phase difference between the switching operations of the plurality of second legs and the switching operations of the plurality of first legs.

7. The first legs and the second legs are arranged in N pieces (N: an integer of 2 or more), In the first bridge, a phase difference of 360 / N [deg] is provided between the switching operations of the N first legs, The power conversion device according to any one of claims 1 to 6, wherein in the second bridge, a phase difference of 360 / N [deg] is provided between the switching operations of the N second legs.

8. The power conversion device according to any one of claims 1 to 7, wherein the plurality of reactors are configured to share magnetic components.

9. The power conversion device according to any one of claims 1 to 8, wherein the number of turns of the primary winding of the transformer is greater than the number of turns of the secondary winding.

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