Receiver Circuit

Abstract

An example of a receiver circuit is disclosed. The receiver circuit includes an input for receiving a radio frequency signal, an output for providing an output signal of the receiver circuit, a receive path connected between the input and output of the receiver circuit, and a feedback path whose input is connected to the output of the receiver circuit. The receiver circuit also includes a converter, where the output signal of the feedback path is combined with the received radio frequency signal in the receive path to form a first signal. The receive path includes a first mixer for downconverting the first signal by a local oscillator frequency and an amplifier for amplifying the downconverted first signal. The feedback path includes a second mixer for upconverting the signal in the feedback path by the local oscillator frequency. The combination of the output signal of the feedback path with the received radio frequency signal is achieved by the converter.

Description

[Technical field]

[0001] This relates to receiver circuits, in particular to receiver circuits for receiving radio frequency radio signals, and to communication devices including the receiver circuits. [Background technology]

[0002] In wireless communications, the frequency spectrum below 6 GHz is becoming increasingly full. The number of frequency bands is large and increasing, and they are used frequently. This raises issues regarding receiver linearity, which must provide high performance in the presence of strong interference. The need for high receiver linearity is further increased by the need to relax filter specifications in devices such as cellular phones to keep the cost of multi-band solutions with an ever-increasing number of frequency bands under control. While providing high linearity, the receiver should of course still meet the requirements for low noise and power consumption.

[0003] There are different techniques to obtain high linearity in a receiver. The key concept is to present a matched impedance only to a narrow band of frequencies where it is desired to receive signals, and a low impedance to other frequencies. The low impedance, ideally zero, would then reflect signals at these other frequencies, so that they do not enter the receiver. The effect of the low impedance is therefore to prevent interfering signals from inducing a voltage at the receiver input.

[0004] FIG. 1 is a block schematic diagram illustrating the general principles for one receiver architecture.

[0005] The receiver 10 shown in Figure 1 is a quadrature receiver and therefore has separate paths 12, 14 for the in-phase (I) and quadrature (Q) signals, respectively. Only the in-phase path (I path) 12 is shown in detail. The structure of the quadrature path (Q path) 14 is the same as that of the in-phase path 12.

[0006] A signal received at a radio frequency input (RF in) 16 of the receiver is applied to a mixer 18 which downconverts the signal to baseband by a local oscillator signal applied to the mixer 18 .

[0007] The output of mixer 18 is applied to amplifier 20 and then to low pass filter 22. The output of low pass filter 22 is the output of in-phase path 12 of receiver 10. The output of in-phase path 12 of receiver 10 and the output of quadrature path 14 of receiver 10 together form the quadrature baseband output of the receiver.

[0008] Therefore, the receiver 10 shown in FIG. 1 is a "mixer first" architecture receiver in which the received signal is passed directly to the mixer 18 without the need for a radio frequency amplifier. This means that the possible RF downconversion frequency span is completely determined by the mixer and local oscillator signals. This architecture can have higher linearity than an architecture that includes a radio frequency low noise amplifier to amplify the received signal, but this architecture will generally not have high performance in terms of noise. Using a bidirectional mixer, such as a passive mixer, the input match depends on the frequency upconverted baseband impedance.

[0009] Figure 2 is a block schematic diagram showing an evolution of the receiver architecture shown in Figure 1. In this architecture, frequency translating positive feedback is used to separate the input matching from the baseband impedance and provide flexibility in controlling both the phase and magnitude of the input impedance. An example of this is disclosed in the article "A positive feedback passive mixer-first receiver front-end", A. Nejdel, M. Abdulaziz, M. Tormanen, H. Sjoland, IEEE RFIC Symposium, 2015, pp. 79-82.

[0010] The receiver 30 shown in Figure 2 is a quadrature receiver and therefore has separate paths 32, 34 for the in-phase (I) and quadrature (Q) signals, respectively. Only the in-phase path (I path) 32 is shown in detail. The structure of the quadrature path (Q path) 34 is the same as that of the in-phase path 32.

[0011] A signal received at a radio frequency input (RF in) 36 of the receiver is applied to a mixer 38 in a forward receive path 40 and the signal is downconverted to baseband by a local oscillator signal applied to mixer 38 .

[0012] The output of the mixer 38 is applied to a transimpedance amplifier 42 and then to a lowpass filter 44. The output of the lowpass filter 44 is the output of the forward receive path 40 of the in-phase path 32 of the receiver 30. The output of the receiver's in-phase path 32 and the output of the quadrature path 34 of the receiver 30 together form the quadrature baseband output of the receiver.

[0013] The output of the forward receive path 40 is also applied to a feedback path 46, which includes a series connected adjustable resistor 48 and a second mixer 50. The adjustable resistor 48 serves to adjust the magnitude of the signal in the feedback path 46.

[0014] The second mixer 50 serves to upconvert the baseband output signal of the receive path 40 to the frequency of the received signal, for example using the same local oscillator signal applied to the mixer 38, described above.

[0015] The upconverted feedback signal generated by the second mixer 50 is then added to the input received signal in the current domain. This is positive feedback, since the feedback signal is added in phase with the input signal.

[0016] The passive mixer 38, followed by a transimpedance amplifier 42, presents a wideband low input impedance. The mixer-first architecture, where the mixer is connected directly to the receiver input, means that a low input impedance appears at the input.

[0017] The bandwidth of the feedback loop is limited primarily by the combination of the transimpedance amplifier 42 and low pass filter 44 (and corresponding circuitry in the Q path), which perform some channel selection filtering and have poles to reject interference. Positive shunt feedback (which is narrowband compared to the carrier frequency) boosts the input impedance in the receive channel around the local oscillator frequencies of the two mixers 38, 50.

[0018] By adjusting the amplitude of the feedback signal it is possible to frequency selectively boost the otherwise low input impedance to match the antenna port impedance across the channel bandwidth.

[0019] The addition of feedback will have a negligible effect on the noise figure since the effective shunt feedback resistance can be designed to be large.

[0020] Although this architecture has good performance, the use of positive feedback generally does not provide as high linearity as negative feedback, since negative feedback has the effect of improving linearity, whereas positive feedback does the opposite, and positive feedback architectures are inherently sensitive to instability. Summary of the Invention

[0021] A first aspect of the disclosure provides a receiver circuit comprising an input for receiving a radio frequency signal, an output for providing an output signal of the receiver circuit, a receive path connected between the input and the output of the receiver circuit, and a feedback path, the input of the feedback path connected to the output of the receiver circuit. The receiver circuit also comprises a converter, and the output signal of the feedback path is combined with a received radio frequency signal in the receive path to form a first signal. The receive path comprises a first mixer for down-converting the first signal by a local oscillator frequency and an amplifier for amplifying the down-converted first signal. The feedback path comprises a second mixer for up-converting a signal in the feedback path by the local oscillator frequency. The combination of the output signal of the feedback path with the received radio frequency signal is achieved by the converter.

[0022] Another aspect of the present disclosure provides a communication device comprising a receiver circuit according to the above aspect.

[0023] Examples of the present disclosure may have the advantage that the receiver circuit achieves excellent in-band and out-of-band linearity using a relatively simple structure.

[0024] For a better understanding of the present invention and to show how it may be carried into effect, reference will now be made, by way of example only, to the accompanying drawings in which: [Brief description of the drawings]

[0025] [Figure 1] FIG. 2 is a block schematic diagram showing a first receiver architecture; [Diagram 2] FIG. 11 is a block schematic diagram showing a second receiver architecture; [Diagram 3] FIG. 2 is a block schematic diagram illustrating a receiver architecture according to the present disclosure. [Figure 4] FIG. 4 is a schematic diagram showing a receiver in more detail according to the receiver architecture of FIG. 3; [Diagram 5] FIG. 5 is a schematic diagram showing a portion of the receiver of FIG. 4 in more detail. [Figure 6] FIG. 5 is a block schematic diagram of a communication device incorporating the receiver of FIG. 3 or FIG. 4. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0026] Embodiments will now be described with reference to the accompanying drawings, in which: It will be appreciated that these embodiments are provided by way of example only, and that variations and modifications may be made within the scope of the invention as defined by the claims.

[0027] As explained above, there are drawbacks to the use of positive feedback, such as increased nonlinearity and an inherent susceptibility to instability.

[0028] Thus, Figure 3 shows a receiver architecture in which negative feedback is used. As described in more detail below, the feedback path upconverts the attenuated baseband signal to an RF frequency. The converter effectively combines this signal with (e.g., subtracts it from) the RF input signal in the voltage domain. By adjusting the feedback signal level (and phase), it is possible to match the input to the port impedance, and the addition of feedback will have a negligible effect on the noise figure.

[0029] The receiver 60 shown in Figure 3 is a quadrature receiver and therefore has separate paths 62, 64 for the in-phase (I) and quadrature (Q) signals, respectively. Only the in-phase path (I path) 62 is shown in detail. The structure of the quadrature path (Q path) 64 is the same as that of the in-phase path 62.

[0030] 3 shows a pair of wires for each signal, and thus illustrates the use of differential signaling, although it will be appreciated that architectures using single-ended signals are also possible.

[0031] The signal received at the receiver's radio frequency input (RF in) 66 is applied to a mixer 68 in a forward receive path 70, where the signal is downconverted to baseband by a local oscillator signal applied to the mixer 68. In other embodiments, the signal may instead be downconverted to an intermediate frequency. In some instances of these embodiments, the separate I and Q lowpass filters 74 may be replaced with complex bandpass filters.

[0032] The mixer 68 in the forward receive path 70 of the in-phase path (I path) 62 and the mixer in the forward receive path of the quadrature path (Q path) 64 are passive mixers that operate with quadrature signals, i.e., the same local oscillator frequency is applied to the mixer 68 in the forward receive path 70 of the in-phase path (I path) 62 and the mixer in the forward receive path of the quadrature path (Q path) 64, but a phase shift is imposed between the local oscillator signals of the two mixers.

[0033] The output of the mixer 68 is applied to a transimpedance amplifier 72 and then to a lowpass filter 74. The output of the lowpass filter 74 is the output of the forward receive path 70 of the in-phase path 62 of the receiver 60. The output of the receiver's in-phase path 62 and the output of the quadrature path 64 of the receiver 60 may together form the quadrature baseband signal voltage output of the receiver 60 as an in-phase (I) voltage and a quadrature (Q) voltage. In some examples, the amplifier 72 and the filter 74 may be combined as a single block or component.

[0034] The output of the forward receive path 70 is also applied to a feedback path 76, which includes a voltage divider 78 and a second mixer 80. The voltage divider 78 is a passive voltage divider that may appear before and / or after the mixer 80 in the feedback path.

[0035] The voltage divider 78 serves to adjust the magnitude of the signal in the feedback path 76 .

[0036] The second mixer 80 receives the same local oscillator frequency signal as the mixer 68 in the forward receive path, and thus serves to upconvert the baseband output signal of the receive path 70 to the frequency of the received signal.

[0037] The upconverted feedback signal produced by the second mixer 80 is then combined with (e.g., subtracted from) the input received signal. Because the feedback signal is subtracted from the input signal, in some examples, this is negative feedback.

[0038] The synthesis is performed by converter 82 .

[0039] In particular, the input lines 84, 86 of the receiver 60 include respective windings 88, 90. A further winding consisting of two sections 92, 94 is connected across the line at the output of the feedback path 76. The configuration is such that there is a mutual inductance between the winding 88 on one input line and a first section 92 of the feedback path winding, and a mutual inductance between the winding 90 on the other input line and a second section 94 of the feedback path winding. The configuration is further such that the signal on the feedback path 76 is combined with (e.g., subtracted from) the received signal.

[0040] In some examples, the quadrature path 64 may be connected to the transformer 82 in parallel with the in-phase path 62. If a passive mixer 80 is used, in some examples, the mixer's local oscillator signals may not overlap between the in-phase and quadrature paths. In such examples, the feedback signals on both paths may simply be connected together. With non-overlapping local oscillator signals, when one mixer is active the other is at high impedance and vice versa.

[0041] Also connected across the line at the output of feedback path 76 is a controllable capacitance 96 .

[0042] Therefore, receiver 60 uses narrowband frequency translation negative feedback to provide a precise and controllable input impedance at the frequency of the signal to be received. At other frequencies the input impedance is low and therefore disturbances are reflected by the mismatch.

[0043] The receiver is a mixer-first structure with a quadrature passive mixer followed by a transimpedance amplifier 72. The low input impedance of the transimpedance amplifier 72 causes the passive mixer 68 to operate in current mode and present a low input impedance. At the output of the low pass filter 74, the baseband signal information is available as in-phase (I) and quadrature (Q) voltages.

[0044] These voltages are divided by a passive voltage divider 78 and fed to a quadrature mixer 80 which generates a high frequency feedback signal. This feedback signal is mixed with (e.g., subtracted from) the receiver input using a transformer 82, which then precisely sets the receiver input impedance within the bandwidth of the feedback loop. The bandwidth is set by the pole of the transimpedance amplifier, which is controlled by the low pass filter 74 or, as described in more detail below, by the RC shunt feedback impedance of the transimpedance amplifier. The transimpedance amplifier 72 may then, in some examples, also include the functionality of the low pass filter 74.

[0045] The receiver input impedance is set by the ratio of the feedback voltage signal from the converter to the input current of the mixer. The non-zero mixer input impedance adds to this ratio in the total receiver input impedance, which in a well-designed circuit is much lower than the ratio. The magnitude of the ratio is primarily controlled by the transimpedance amplifier 72 (e.g., the feedback resistors associated with the transimpedance amplifier 72), which determines the forward gain, and the passive voltage divider 78 in the feedback path. In some examples, the input impedance may be programmable or dynamically adjusted. For example, the feedback resistors in the passive voltage divider 78 (in the in-phase and / or quadrature paths) may include digitally programmable resistors, such as a bank of resistors controlled using MOS transistor switches. In some examples, the impedance due to feedback (equal or approximately equal to the ratio described above) may be greater than or much greater than the input impedance of the forward path. Otherwise, the performance improvement achievable by using feedback may be limited in some cases. The input impedance of mixer 68 (when connected to amplifier 72) may in some instances be less than 10-20% of the total input impedance, for example at least 80-90% of the input impedance may be due to feedback in a well designed circuit.

[0046] The phase of the ratio may also be controlled by applying a phase difference between the local oscillator signal applied to the forward path mixer 68 and the local oscillator signal applied to the feedback mixer 80. This may be done in any one or more of several suitable ways. For example, a variable delay line may be applied to the LO signal. Alternatively, the contribution for the I path may be added to or subtracted from the Q path, or vice versa, i.e., baseband coordinate rotation. This may be achieved, for example, by adding controllable resistors to a resistive voltage divider that also has a set of crossed and non-crossed resistors from I to Q and vice versa (to handle both positive and negative rotations). These may be added, for example, to the existing resistors 148a and 148b in FIG. 4, which may be described more fully below.

[0047] The use of negative feedback means that the linearity of the circuit is improved compared to positive feedback, especially the intermodulation performance.

[0048] FIG. 4 illustrates in greater detail an implementation of receiver circuit 60 of FIG. 3 in one exemplary embodiment.

[0049] As mentioned above, the receiver 60 shown in FIG. 3 is a quadrature receiver and thus has separate paths 62, 64 for the in-phase (I) and quadrature (Q) signals, respectively.

[0050] In the case of a quadrature receiver having two paths, the received signal is applied to two paths, each of which includes a mixer in the forward path and a mixer in the feedback path, and there is a phase difference between the local oscillator signal applied to the mixer in the in-phase path and the local oscillator signal applied to the mixer in the quadrature path.

[0051] More generally, the receiver can be any polyphase receiver with multiple paths.

[0052] In FIG. 4, only one of the multiple paths is shown to avoid overcomplicating the diagram.

[0053] In particular, Figure 4 shows a receiver 120 having a pair of input terminals 122, 124 at the radio frequency input (RF in) for receiving a differential signal, although it will be appreciated that receivers operating with single-ended signals are also possible.

[0054] A signal received at a radio frequency input (RF in) of the receiver is applied to a passive mixer 126 in a forward receive path 128 .

[0055] The mixer 126 in this illustrated example is a double-balanced passive mixer that includes CMOS transistors 130a, 130b, 130c, 130d to which positive and negative local oscillator signals (LO+ and LO-) are applied. In one embodiment, the CMOS transistors 130a, 130b, 130c, 130d each have a channel length of 20 nm with a total channel width of 240 μm, although this is just one particular example and any suitable values ​​may be used in other examples.

[0056] The mixer 126 serves to frequency downconvert the received signal to baseband, and the baseband output of the mixer 126 is applied to a transimpedance amplifier 132 .

[0057] A pair of bypass capacitors 134a, 134b are provided across the input terminals of the amplifier 132 to shunt out-of-band signal current to signal ground to reduce intermodulation distortion in the transimpedance amplifier. In one embodiment, the capacitors 134a, 134b each have a capacitance value of 32 pF, although this is just one particular example and any suitable value may be used in other examples.

[0058] The structure of the transimpedance amplifier 132 will be described in more detail below, but suffice it to say at this stage that the transimpedance amplifier 132 converts the current output of the mixer 126 to a baseband voltage output BBout on output terminals 136, 138 of the receiver circuit.

[0059] Low pass filtering of the baseband voltage output is accomplished by a pair of feedback resistors 140a, 140b, each in parallel with a respective feedback capacitor 142a, 142b. In one embodiment, the feedback resistors 140a, 140b have a resistance value of 4kΩ and the feedback capacitors 142a, 142b have a capacitance value of 6pF, although this is just one particular example and any suitable values ​​may be used in other examples.

[0060] The feedback resistors 140a, 140b in the transimpedance amplifier 132 may be selected to produce a total gain from RF input to output of, for example, about 30 dB. The feedback capacitors 142a, 142b may be selected to give a receive bandwidth of, for example, about 10 MHz. The in-band voltage gain of the circuit may be determined as follows. For example, the in-band voltage gain may be proportional to the values ​​of the feedback resistors 140a and 140b. Thus, if an additional gain of, for example, 6 dB is desired, the resistance should therefore be doubled. The bandwidth is inversely proportional to the RC product of the feedback resistor 140 and the feedback capacitor 142. When increasing the gain by increasing R, to maintain the bandwidth, C may be reduced by the same factor. On the other hand, to change the bandwidth and not change the gain, the capacitance of the capacitor 142 may be changed without changing the resistance of the resistor 140. For example, the bandwidth can be halved by doubling the capacitance of feedback capacitors 142a and 142b and keeping the resistors unchanged.

[0061] As previously explained, the baseband voltage outputs BBout on the output terminals 136, 138 of the receiver circuitry and the outputs of the corresponding circuits may together form a composite baseband signal voltage as the output of the receiver.

[0062] The output of the forward receive path 128 is also applied to a feedback path 144, which includes a passive voltage divider 146. The voltage divider 146 includes a respective resistor 148a, 148b in each signal line with an associated resistor 150a, 150b connected to signal ground.

[0063] In one embodiment, the sum of the resistance of each pair of resistors, i.e., resistors 148a and 150a plus resistors 148b, 150b, is 100 Ω. This sum is split so that resistors 148a, 148b in each signal line have a resistance of 96.25 Ω and resistors 150a, 150b connected to ground have a resistance of 3.75 Ω, although this is just one particular example and any suitable values ​​may be used in other examples.

[0064] Thus, the voltage divider 146 acts to adjust the magnitude of the signal in the feedback path 144 .

[0065] The signal in the feedback path is then applied to a second mixer 152, which receives the same local oscillator frequency signal as the mixer 126 in the forward receive path, and thus serves to upconvert the baseband output signal of the receive path 144 to the frequency of the received signal.

[0066] The second mixer 152 in this illustrated example is a double-balanced passive mixer that includes CMOS transistors 154a, 154b, 154c, 154d to which the positive and negative local oscillator signals (LO+ and LO-) are applied. In one embodiment, the CMOS transistors 154a, 154b, 154c, 154d each have a channel length of 20 nm with a total channel width of 60 μm.

[0067] As mentioned above, the voltage divider 146 may appear before and / or after the second mixer 152 in the feedback path.

[0068] A capacitance 156 is connected between the two signal lines of the feedback path, which in one embodiment consists of two segments 158a, 158b, each having a capacitance value of 14 pF, although this is just one particular example and any suitable value may be used in other examples.

[0069] The resulting signal is then combined with (eg, subtracted from) the input signal by converter 160 .

[0070] The converter 160 comprises respective windings 162, 164 in input lines 166, 168 of the receiver 120. A further winding consisting of two parts 170, 172 is connected between the two signal lines at the output of the feedback path 144. The connection point of the two winding parts 170, 172 may be connected to the connection point of the two capacitor segments 158a, 158b.

[0071] The winding configuration is such that there is a mutual inductance between winding 162 on one input line and a first portion 170 of the feedback path winding, and a mutual inductance between winding 164 on the other input line and a second portion 172 of the feedback path winding. The configuration is further such that the signal on feedback path 144 is combined with (e.g., subtracted from) the received signal before being input to mixer 126.

[0072] In one embodiment, each of the windings 162, 164 in the input lines 166, 168 of the receiver 120 and each of the winding portions 170, 172 in the feedback path 144 have an inductance of 300 pH and a parasitic series resistance of 0.19 Ω, although this is just one particular example and any suitable values ​​may be used in other examples.

[0073] In some examples, there is a mutual coupling coefficient of 0.8 between winding 162 and winding portion 170 and between winding 164 and winding portion 172 .

[0074] Therefore, receiver 120 uses narrowband frequency translating negative feedback to provide a precise and controllable input impedance at the frequencies at which it is desired to receive the signal. At other frequencies the input impedance is low and therefore disturbances are reflected by the mismatch.

[0075] With the input impedance tuned to a desired value, the frequency may in some instances track the local oscillator frequency. However, in some instances the range is limited by the tuned transducer. Thus, if a wider frequency range of operation is desired, some (coarse) tuning may need to be applied to capacitor 158.

[0076] The receiver input impedance is set by the ratio of the feedback voltage signal from the converter to the input current of the mixer. The non-zero mixer input impedance adds to this ratio in the total receiver input impedance, which in a well designed circuit is much lower than the ratio. The magnitude of the ratio is primarily controlled by the feedback resistors 140a and 140b in the transimpedance amplifier and the passive voltage divider 146 in the feedback path, which determines the forward gain.

[0077] By adjusting the feedback, the input impedance of the receiver 120 can be optimized for matching and at the same time the input impedance of the mixer 126 can be low, which can allow for high linearity.

[0078] Thus, in summary, the receiver 120 shown in Figure 4 has a converter 160 at its input that combines a feedback voltage with the input signal, which comes from a frequency up-conversion mixer 152 with a baseband frequency input. The feedback is negative, i.e., it acts to reduce the receiver's output signal level. The forward path of the receiver has a low input impedance and performs frequency down-conversion to baseband frequencies, which is performed by a passive mixer 126 followed by a transimpedance amplifier 132.

[0079] The transimpedance amplifier 132 shown in FIG. 4 can be implemented in many ways.

[0080] FIG. 5 shows an illustrative example configuration of a suitable transimpedance amplifier 132, which in this case is a simple class-A fully differential two stage operational amplifier with a common-mode control loop.

[0081] The common mode control 190 regulates the DC common mode voltage of the output stages 192, 194 to half the supply voltage, for example at 400 mV for an 800 mV supply. The input DC voltage is shifted lower, to about 250 mV, by two current sources 196, 198 connected to the inputs In+, In- of the amplifier 132. The current sources can be implemented using NMOS transistors. The current sources 196, 198 are chosen such that their DC current will provide a 150 mV DC drop across the feedback resistors 140a, 140b seen in FIG. 4. That is, the value of the current provided by each of the current sources 196, 198 is 150 mV divided by the resistance of each feedback resistor 140a, 140b. This helps bias the PMOS input pair 200 so that there is enough headroom for the PMOS current mirror 202 that controls the tail current.

[0082] In one embodiment where all transistors have a unit width of 10 μm, the transistors of the input pair 200 have a channel length of 200 nm and a multiplicator of 750, the transistors of the current mirror 202 have a channel length of 200 nm, a multiplicator for the output transistor of 400, and 20 input transistors, which scales the current up by 20, the reference current for the current mirror 202 provided by the current source 216 is 700 μA, the transistors 204, 206 of the output stages 192, 194 each have a channel length of 20 nm and a multiplicator of 20, and the values ​​of the Miller capacitance 208, 210 and Miller resistance 212, 214 are 230 fF and 10 Ω, respectively.

[0083] Thus, the architecture shown in FIG. 3 and described in more detail with reference to FIGS. 4 and 5 has a high degree of linearity, both in-band and out-of-band, due to the negative feedback that linearizes the circuit.

[0084] The effect of the transimpedance low-pass filtering caused by the feedback resistors 140a, 140b and feedback capacitors 142a, 142b around the transimpedance amplifier 132 is that the feedback to the receiver input drops and therefore the input impedance becomes less ohmic as the received RF input signal moves away from the local oscillator frequency. At even higher frequency offsets, the bypass capacitors 134a, 134b at the input of the amplifier 132 will help to attenuate the blockers by shunting their current to signal ground, preventing them from causing intermodulation in the amplifier. The bypass capacitors 134a, 134b contribute to further reducing the input impedance and increasing the mismatch, if necessary.

[0085] The architecture shown in FIG. 3 and described in more detail with reference to FIGS. 4 and 5 also has a programmable input impedance and may, for example, use only a single converter without an additional inductor.

[0086] FIG. 6 is a block schematic diagram showing a communications device incorporating the receiver of FIG. 3 or FIG.

[0087] 6 illustrates a communications device 230, which may be, for example, a mobile phone, a laptop or tablet computer, a wireless sensor, etc., commonly referred to as a user equipment device. Additionally, the communications device may be a node of a communications network, for example, a node of a radio access network, such as a base station, a relay, or a remote radio unit. It will be appreciated that the communications device 230 will have other features and functions in addition to those described and illustrated herein, but which are not necessary for an understanding of the present disclosure.

[0088] FIG. 6 shows a communications device 230 having an antenna 232 suitable for transmitting and receiving wireless signals.

[0089] 6 shows a device having transmit circuitry 234 for generating signals suitable for transmission, it will be appreciated that the present disclosure also relates to devices that are only suitable for receiving signals.

[0090] FIG. 6 shows a receiver circuit 236, which may be a receiver circuit as shown in and described with reference to FIG. 3 or FIG.

[0091] In this example, the receiver circuitry 236 provides its output signal to a processor 238, which demodulates and decodes the signal and performs any necessary signal processing tasks. The processor 238 is in communication with a memory 240.

[0092] FIG. 6 also shows a local oscillator signal generator 242 for providing a suitable local oscillator signal to the receiver circuit 236 for use therein, as described with reference to FIGS.

[0093] It should be noted that the above-described embodiments are illustrative rather than limiting of the present invention, and that those skilled in the art can design many alternative embodiments without departing from the scope of the appended claims. The word "comprising" does not exclude the presence of elements or steps other than those listed in a claim, and "a" or "an" does not exclude a plurality, and a single feature or other unit may fulfill the functions of several units recited in a claim. Any reference signs in the claims should not be interpreted as limiting their scope.

Claims

1. A receiver circuit (60, 120), comprising: an input (66, 84, 86, 122, 124) for receiving a radio frequency signal; an output (136, 138) for providing an output signal of the receiver circuit; a receive path (70, 128) connected between the input and the output of the receiver circuit; a feedback path (76, 144) having an input connected to the output of the receiver circuit; A converter (82, 160) Equipped with an output signal of the feedback path is subtracted from the received radio frequency signal in the receive path to form a first signal; The receiving path is a first mixer (68, 126) for down-converting the first signal by a local oscillator frequency; an amplifier (72, 132) for amplifying the downconverted first signal; Equipped with The feedback path is a second mixer (80, 152) for up-converting the signal in the feedback path by the local oscillator frequency; Equipped with the subtraction of the output signal of the feedback path from the received radio frequency signal is accomplished by the converter; the converter (82, 160) comprises a first winding connected to the input (66, 84, 86, 122, 124) for receiving the radio frequency signal and a second winding connected to the feedback path (76, 144), there being mutual inductance between the first winding and the second winding such that the output signal of the feedback path is subtracted from the received radio frequency signal; A receiver circuit (60, 120).

2. 2. The receiver circuit of claim 1, wherein the input (66, 84, 86, 122, 124), the output (136, 138), the receive path (70, 128) and the feedback path (76, 144) comprise respective first and second signal lines such that the receiver circuit is configured to handle differential signals.

3. The first winding of the converter (82, 160) comprises a first winding portion in the first signal line of the input (66, 84, 86, 122, 124) and a second winding portion in the second signal line of the input; the second winding of the transformer comprises a first segment and a second segment connected between the first signal line and the second signal line of the feedback path (76, 144); there is a mutual inductance between the first winding portion of the first winding and the first segment of the second winding; 3. The receiver circuit of claim 2, wherein there is a mutual inductance between the second winding portion of the first winding and the second segment of the second winding.

4. 4. The receiver circuit of claim 1, wherein the first mixer and the second mixer are passive mixers.

5. 5. The receiver circuit of claim 1, wherein the amplifier (72, 132) is a transimpedance amplifier.

6. 6. The receiver circuit of claim 1, wherein the feedback path (76, 144) comprises a voltage divider (78, 146).

7. 7. The receiver circuit of claim 6, wherein the voltage divider (78, 146) is a passive voltage divider.

8. 8. The receiver circuit of claim 6 or 7, wherein the voltage divider (78, 146) is located before the second mixer (80, 152) in the feedback path.

9. A receiver circuit as described in any one of claims 1 to 8, wherein the receiver circuit is a multi-phase receiver.

10. a plurality of receive paths (70, 128), a plurality of feedback paths (76, 144), and at least one converter (82, 160) for subtraction of the output signal of each of the feedback paths from the received radio frequency signal in the receive paths, 10. The receiver circuit of claim 9, wherein local oscillator signals provided to each first mixer (68, 126) in each said receive path (70, 128) have a phase difference therebetween.

11. Receiver circuit according to claim 9 or 10, wherein the receiver circuit is at least a three-phase receiver.

12. 11. The receiver circuit of claim 10, wherein the receiver circuit is a polyphase receiver, the polyphase receiver being a quadrature receiver comprising respective receive paths (70, 128) and feedback paths (76, 144) for in-phase and quadrature signals.

13. A receiver circuit as described in any one of claims 1 to 12, wherein the effect of subtracting the output signal of the feedback path from the received radio frequency signal is to change the input impedance of the receiver circuit.

14. 14. The receiver circuit of claim 1, comprising a filter (74, 140, 142) in at least one of the receive path (70, 128) and the feedback path (76, 144) for attenuating out-of-band signals.

15. A communication device comprising a receiver circuit (60, 120) according to any one of claims 1 to 14.

16. The communication device of claim 15 , wherein the communication device is a user equipment device.

17. The communication device of claim 15 , wherein the communication device is a network node of a wireless communication network.

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