AC motor drive control device and drive control method

Abstract

To provide a drive control device for an AC motor for suppressing torque vibration caused by an angle error due to an eccentricity of a position sensor and influence of signal noise in control with a position sensor of the AC motor.SOLUTION: A drive control device of an AC motor driven by a power converter using vector control includes: a phase difference calculation unit for calculating a phase difference between rotational position information on the AC motor and a control phase used when performing vector control to output a first phase difference; a filter unit for outputting a second phase difference attenuated by filtering processing of a specific frequency component from the first phase difference; and a speed and phase calculation unit for calculating rotation speed and a control phase of the AC motor based on the second phase difference.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a drive control device and a drive control method for an AC motor, and is applicable to electric railway cars, electric automobiles, industrial inverters, wind power generation systems, diesel generator systems, and the like. [Background technology]

[0002] Due to the increasing demand for energy saving and reducing the environmental impact in recent years, drive control devices for AC motors are becoming increasingly popular in a wide range of applications, such as home appliances, industrial equipment, and infrastructure. One method of driving and controlling an AC motor is to use an angle sensor that detects the rotor position of the AC motor. Examples of such angle sensors include resolvers and magnetic sensors. In particular, resolvers are angle sensors that use electromagnetic induction to convert mechanical angles into electrical signals, and have become widely used as angle detection sensors for machine tools and robots. Recently, resolvers have also been increasingly used as angle detection sensors for automotive permanent magnet synchronous motors, due to their excellent environmental resistance.

[0003] However, when an AC motor is controlled using a resolver, if the output voltage signal from the resolver has a DC offset error due to eccentricity, an amplitude error due to gain imbalance, a transmission delay, or a manufacturing error in the output winding, an angle error occurs due to these, and it is known that a vibration component of one or two times the drive frequency of the AC motor occurs in the drive control device of the AC motor. Therefore, the faster the AC motor rotates, the more vibration phenomena of torque and speed proportional to the drive frequency occur, and there is a risk of damage to the devices constituting the drive control device of the AC motor. Therefore, it is necessary to suppress the vibration phenomenon due to the error in the angle signal caused by the resolver sensor noise, etc.

[0004] Generally, control methods for suppressing speed and torque vibrations caused by angle errors of angle sensors such as resolvers include a method of inserting a filter that reduces vibration components in the speed detection value of an AC motor calculated based on the output signal of the angle sensor, and a method of inserting a filter that reduces vibration components in a control loop of speed feedback or current feedback of the AC motor.

[0005] For example, Patent Document 1 discloses a technology for reducing the influence of errors in an angle sensor by applying a notch filter process to the electrical angular frequency of an AC motor obtained by calculation based on the angle detected by an angle sensor, for a frequency to be cut off. [Prior art documents] [Patent documents]

[0006] [Patent Document 1] JP 2015-136218 A Summary of the Invention [Problem to be solved by the invention]

[0007] The technology disclosed in Patent Document 1 calculates the electrical angular frequency of an AC motor from the change over time in the angle detected by an angle sensor, and applies notch filter processing to cut off frequency components proportional to the calculated electrical angular frequency value, thereby reducing torque vibrations and speed vibrations in a drive control device for an AC motor caused by angle errors in the angle sensor.

[0008] In addition, by making the cutoff frequency of the notch filter variable depending on the calculation result of the electrical angle frequency, it is possible to reduce the vibration component of the speed calculation value due to the angle error even in a transient state where the rotation speed command and the mechanical angle rotation speed do not match.

[0009] However, since it is not possible to remove the angle error that occurs in the electrical angle of the AC motor based on the angle detected by the angle sensor, the torque control and speed control of the AC motor will vibrate.

[0010] Therefore, in view of the above problems, an object of the present invention is to provide a drive control device and a drive control method for an AC motor that suppresses torque vibration of the AC motor caused by angle error due to the influence of sensor noise of an angle sensor. [Means for solving the problem]

[0011] In order to solve the above problems, one representative drive control device for an AC motor of the present invention is a drive control device for an AC motor driven by a power converter using vector control, and includes a phase difference calculation unit that calculates the phase difference between rotational position information of the AC motor and a control phase used when performing vector control and outputs a first phase difference, a filter unit that outputs a second phase difference obtained by attenuating specific frequency components from the first phase difference through filtering, and a speed and phase calculation unit that calculates the rotational speed and control phase of the AC motor based on the second phase difference. Effect of the Invention

[0012] According to the present invention, by reducing the effect of angle error due to sensor noise of an angle sensor on vector control, it is possible to suppress torque vibration of an AC motor and realize a drive control device for an AC motor with high control stability. Problems, configurations and effects other than those described above will become apparent from the description of the following embodiments. [Brief description of the drawings]

[0013] [Figure 1] 1 is a block diagram illustrating an example of the configuration of a drive control device for an AC motor according to a first embodiment of the present invention. [Diagram 2] FIG. 2 is a diagram showing the definition of a coordinate system and symbols used in the control of an AC motor according to the present invention. [Diagram 3] FIG. 2 is a block diagram illustrating an example of the configuration of a phase / velocity calculation unit according to the first embodiment. [Figure 4] 4 is a block diagram illustrating an example of the configuration of a phase difference calculation unit according to the first embodiment. FIG. [Diagram 5]2 is a block diagram illustrating a configuration example of a proportional-integral control unit according to the first embodiment. FIG. [Figure 6] FIG. 11 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a second embodiment of the present invention. [Figure 7] FIG. 11 is a block diagram illustrating an example of the configuration of a phase / velocity calculation unit according to a second embodiment. [Figure 8] FIG. 11 is a block diagram illustrating an example of the configuration of a phase difference calculation unit according to a second embodiment. [Figure 9] FIG. 11 is a block diagram illustrating a configuration example of a phase difference FF compensation unit according to a second embodiment. [Figure 10] FIG. 11 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a third embodiment of the present invention. [Figure 11] FIG. 11 is a block diagram illustrating an example of the configuration of a phase / velocity calculation unit according to a third embodiment. [Figure 12] FIG. 11 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a fourth embodiment of the present invention. [Figure 13] FIG. 13 is a block diagram illustrating an example of the configuration of a phase / velocity calculation unit according to a fourth embodiment. [Figure 14] FIG. 11 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a fifth embodiment of the present invention. [Figure 15] FIG. 13 is a block diagram illustrating a configuration example of a phase / velocity calculation unit according to a fifth embodiment. [Figure 16] FIG. 11 is a block diagram showing a schematic configuration of a part of a railway vehicle equipped with a drive control device for an AC motor using any one of the first to fifth embodiments, as a sixth embodiment of the present invention. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0014] Hereinafter, with reference to the drawings, embodiments 1 to 6 will be described in detail as modes for carrying out the present invention. In principle, the same reference numerals are used for the same elements in all the drawings. Furthermore, duplicated descriptions of parts having the same functions will be omitted. The embodiments and modified examples described below may be combined in part or in whole as long as they are not inconsistent. EXAMPLES

[0015] Fig. 1 is a block diagram showing an example of the configuration of a drive control device for an AC motor according to a first embodiment of the present invention. Fig. 1 includes an AC motor 103 to be controlled, a power converter 102 that drives the AC motor 103, a controller 101 that controls the power converter 102, a command generator 105 that generates a torque command Tm* for the AC motor 103, a phase current detection unit 121 that detects a current flowing through the AC motor 103, and a rotational position detection unit 124 that detects a rotor position of the AC motor 103.

[0016] The AC motor 103 is a motor controlled by AC power output from the power converter 102. In each embodiment of the present invention, the invention will be described taking a permanent magnet synchronous motor as an example of one type of AC motor, but the invention is not limited to this motor and can be applied to all other AC motors. In addition, although an AC motor is taken as an example in each embodiment of the present invention, the same effects as in the case of an AC motor can be obtained even when an AC generator is controlled.

[0017] Power converter 102 includes input terminals 123a and 123b for supplying DC power to power converter 102, a main circuit section 132 composed of six switching elements Sup to Swn, a gate driver 133 for directly driving main circuit section 132, a DC resistor 134 and a smoothing capacitor 131 that are attached for overcurrent protection of power converter 102. Power converter 102 also converts DC power supplied from input terminals 123a and 123b into AC power based on a gate command signal generated by controller 101, and supplies the converted AC power to AC motor 103.

[0018] The phase current detection unit 121 detects AC currents iu and iw flowing from the power converter 102 to the permanent magnet synchronous machine 103. This phase current detection unit 121 is realized by a current sensor using a Hall element, for example. Although the phase current detection unit 121 shown in Fig. 1 is configured to detect AC current by two-phase detection, it may also be configured to detect three phases. Also, instead of using a phase current sensor, an AC current value estimated from a current value flowing through a DC resistor 134 attached for overcurrent protection of the power converter 102 may be used.

[0019] The rotational position detection unit 124 outputs an angle signal φ corresponding to the rotor position (rotation angle) of the AC motor 103. The rotational position detection unit 124 is realized by, for example, a resolver, an encoder, or a magnetic sensor.

[0020] The phase / speed calculation unit 110 calculates, from the angle signal φ corresponding to the detected rotor position, the electrical angular frequency ωre of the rotational speed of the AC motor 103 and the control phase θdc used in the dq coordinate transformation of the current detection value for vector control and the UVW coordinate transformation of the voltage command value.

[0021] The command generator 105 is a controller located at a higher level than the controller 101, and generates a torque command Tm* to the AC motor.

[0022] The controller 101 controls the torque generated by the AC motor 103 based on a torque command Tm* from a command generator 105. As this controller, for example, a current controller is used to control the current flowing through the AC motor 103, or a speed controller or a position controller is used to control the rotation speed or position. In the first embodiment, the controller operates as a torque controller intended to control the torque, but a speed controller or a position controller may be used as a higher-level controller.

[0023] The controller 101 includes a phase / speed calculation unit 110, a current command calculation unit 111, a vector control unit 112, a current detection unit 113, a dq coordinate conversion unit 114, a polar coordinate conversion unit 115, a UVW coordinate conversion unit 116, and a PWM signal controller 117.

[0024] In addition, the controller 101 generates a gate command signal for driving the switching elements of the power converter 102 from the calculation results of a current control system and a phase control system based on an angle signal φ corresponding to the rotor position (rotation angle) of the AC motor 103 from the rotational position detection unit 124, AC current detection values ​​Iu and Iw which are detection values ​​of the AC currents iu and iw flowing through the AC motor 103, and a torque command Tm* from the command generator 105, and supplies the gate command signal to a gate driver 133 of the power converter 102.

[0025] 2 is a diagram showing a coordinate system and definitions of symbols used in the control of AC motor 103 according to the present invention. Here, as described above, a permanent magnet synchronous motor will be taken as an example of AC motor 103.

[0026] 2, the a-axis coordinate system defined by the a-axis and the b-axis is a stator coordinate system that represents the phase of the stator winding of the permanent magnet synchronous motor 103a. The a-axis is generally taken as the reference for the u-phase winding phase of the permanent magnet synchronous motor 103a.

[0027] The dq-axis coordinate system defined by the d-axis and q-axis is a rotating coordinate system that represents the magnetic pole position of the rotor of the permanent magnet synchronous motor 103a, and rotates in synchronization with the rotor magnetic pole position of the permanent magnet synchronous motor 103a. In the case of a permanent magnet synchronous motor, the d-axis is also called the magnetic pole axis because the north pole direction of the magnetic pole of the permanent magnet attached to the rotor is generally taken as the reference.

[0028] The dc-qc axis coordinate system defined by the dc axis and qc axis is the control phase for vector control, that is, the coordinate system controlled by the controller 101 as the d axis and q axis directions, and is also called the control axis.

[0029] The z-axis is a rotating coordinate system that indicates the phase of the angle signal φ output by the rotational position detection unit 124 in accordance with the rotor position (rotation angle) of the permanent magnet synchronous motor 103a. The coordinate axes combined in each coordinate system are all orthogonal to each other.

[0030] In each of the above coordinate systems, the phases of the d-axis, dc-axis, and z-axis relative to the a-axis are expressed as θd, θdc, and φ, respectively, as shown in Figure 2. The deviation of the dc-axis from the d-axis is expressed as Δθerr, and the deviation of the dc-axis from the z-axis is expressed as Δθc.

[0031] Here, when vector control is performed using the angle signal φ corresponding to the rotor position (rotation angle) of the permanent magnet synchronous motor 103a output by the rotational position detection unit 124, if the d-axis and z-axis are aligned, the dq-axis coordinate system and the dc-qc-axis coordinate system will be aligned, and the deviation Δθerr of the dc-axis with respect to the d-axis will be zero.

[0032] However, if there is an angle error due to sensor noise or the like in the angle signal φ corresponding to the rotor position (rotation angle) of the permanent magnet synchronous motor 103a output by the rotational position detection unit 124, a deviation Δθerr of the dc axis from the d axis will occur.

[0033] The configuration of the controller 101 shown in FIG. 1 will be described in detail below. The phase / speed calculation unit 110 calculates and outputs the electrical angular frequency ωre of the rotational speed and the control phase θdc of the dc-qc axis coordinate system for performing vector control, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124.

[0034] The current command calculation unit 111 calculates and outputs current command values ​​Idc* and Iqc* on the dc-qc axis coordinate system based on the electrical angular frequency ωre of the rotation speed of the permanent magnet synchronous motor 103a output from the phase / speed calculation unit 110, the voltage detection value Ecf of the smoothing capacitor 131, and the torque command Tm* output from the command generator 105. The current command values ​​Idc* and Iqc* can be calculated, for example, by using optimal d-axis and q-axis current command values ​​for the torque command Tm*, the voltage detection value Ecf, and the inverter frequency ω1, as a reference table, a function formula, an approximation formula, a design formula, or a theoretical formula, which are previously obtained by testing or analysis.

[0035] The vector control unit 112 performs current control so that the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system output by the dq coordinate conversion unit 114 coincide with the current command values ​​Idc* and Iqc* on the dc-qc axis coordinate system output by the current command calculation unit 111. Based on the result of this current control and the electrical angular frequency ωre of the rotation speed of the permanent magnet synchronous motor 103a, voltage commands Vdc* and Vqc* on the dc-qc axes, which are the rotating coordinate system, are calculated and output.

[0036] In the vector control unit 112, the voltage commands Vdc* and Vqc* on the dc-qc axes are calculated, for example, by the following equation (1) using the current command values ​​Idc* and Iqc* on the dc-qc axis coordinate system, the electrical angular frequency ωre of the permanent magnet synchronous motor 103a, and the motor constants R1, Ld, Lq and Ke (back electromotive force constant).

number

[0037] The current detection unit 113 calculates three-phase current detection values ​​Iu, Iv, and Iw from the AC currents iu and iw flowing through the permanent magnet synchronous motor 103 a detected by the phase current detection unit 121 , and outputs them to the dq coordinate conversion unit 114 .

[0038] The dq coordinate conversion unit 114 converts the three-phase current detection values ​​Iu, Iv, and Iw output by the current detection unit 113 into current detection values ​​Idc and Iqc on the dc-qc axis coordinate system using the control phase θdc calculated by the phase / speed calculation unit 110 on the basis of the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output by the rotational position detection unit 124, and outputs the converted values.

[0039] A polar coordinate conversion unit 115 converts the voltage commands Vdc* and Vqc* on the dc- and qc-axes output by the vector control unit 112 into a voltage amplitude command V1* and a voltage phase command δ and outputs them.

[0040] The UVW coordinate conversion unit 116 converts the voltage amplitude command V1* and voltage phase command δ output by the polar coordinate conversion unit 115 and the control phase θdc output by the phase / speed calculation unit 110 into three-phase AC voltage commands Vu*, Vv*, and Vw* and outputs them to a PWM signal controller 117.

[0041] The PWM controller 117 generates a triangular wave carrier based on an arbitrary carrier frequency fc and a voltage detection value Ecf of the smoothing capacitor 131, and performs a magnitude comparison between the triangular wave carrier and a modulated wave based on three-phase AC voltage commands Vu*, Vv*, and Vw* to perform pulse width modulation. The switching elements of the power converter 102 are controlled on / off by a gate command signal generated as a result of the pulse width modulation.

[0042] Next, the phase / velocity calculation unit 110, which is a characteristic part of the first embodiment, will be described in detail. FIG. 3 is a block diagram illustrating an example of the configuration of the phase / velocity calculation unit 110 according to the first embodiment. The phase / velocity calculation section 110 shown in FIG. 3 includes a phase difference calculation section 201, a proportional / integral control section 202, a control phase calculation section 203, and a filter processing section 204.

[0043] The phase / speed calculation unit 110 calculates and outputs the electrical angular frequency ωre and the control phase θdc in the dc-qc axis coordinate system of the permanent magnet synchronous motor 103a for vector control, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124.

[0044] FIG. 4 is a block diagram illustrating an example of the configuration of the phase difference calculation unit 201 according to the first embodiment. The phase difference calculation section 201 shown in FIG. 4 includes a ±π processing section 301, a ±π processing section 302, and an adder / subtractor 801.

[0045] The phase difference calculation unit 201 calculates and outputs a phase difference Δθc' between the angle signal φ and the control phase θdc, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124 and the control phase θdc output from the control phase calculation unit 203. Note that in each embodiment of the present invention, the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124 has angle information that changes from -π to +π (-180 degrees to +180 degrees).

[0046] The ±π processing unit 301 converts the control phase θdc output from the control phase calculation unit 203 into a magnitude between -π and +π, and the ±π processing unit 302 converts the phase difference Δθc obtained by subtracting the converted control phase θdc from the angle signal φ into a magnitude between -π and +π. Δθc' obtained by converting the phase difference Δθc between the angle signal φ and the control phase θdc into ±π is calculated, for example, by the following equation (2).

number

[0047] The phase difference Δθc' between the angle signal φ and the control phase θdc output from the phase difference calculation unit 201 is obtained by converting the phase difference Δθc between the angle signal φ and the control phase θdc into a magnitude of -π to +π (-180 degrees to +180 degrees).

[0048] FIG. 5 is a block diagram illustrating an example of the configuration of the proportional-integral control unit 202 according to the first embodiment. The proportional / integral control unit 202 shown in FIG. 5 includes a proportional unit 401 , a proportional unit 402 , an integrator 403 , and an adder 802 .

[0049] The proportional-integral control unit 202 calculates and outputs the electrical angular frequency ωre of the permanent magnet synchronous motor 103a based on the phase difference Δθc′ between the angle signal φ and the control phase θdc.

[0050] Here, the electrical angular frequency ωre of the permanent magnet synchronous motor 103a is obtained by adding the proportional calculation result and the proportional integral calculation result in order to control the phase difference Δθc' between the angle signal φ and the control phase θdc to 0. The electrical angular frequency ωre of the permanent magnet synchronous motor 103a used in vector control is calculated, for example, by the following equation (3).

number

[0051] Here, the proportional gain Kp-pll of the proportional unit 401 and the integral gain Ki-pll of the proportional unit 402 are set, for example, by the following equation (4).

number

[0052] The proportional gain Kp-pll and integral gain Ki-pll of the proportional / integral control unit 202 may be constants or may be calculated as a function of a current command value, a current detection value, a speed, or a vector calculation period.

[0053] 3 calculates and outputs a control phase θdc of a dc-qc axis coordinate system used in vector control, based on the electrical angular frequency ωre of the permanent magnet synchronous motor 103a output by the proportional-integral control unit 202. Here, the control phase θdc is obtained by integrating the electrical angular frequency ωre.

[0054] The filter processing unit 204 shown in FIG. 3 calculates and outputs a filtered phase difference ΔθcN′ based on the phase difference Δθc′ output by the phase difference calculation unit 201 and the electrical angular frequency ωre output by the proportional / integral control unit 202.

[0055] The phase difference ΔθcN' filtered by the filter processing unit 204 is the phase difference Δθc' between the angle signal φ and the control phase θdc that has been filtered in order to reduce the electrical angular frequency ωre of the permanent magnet synchronous motor 103a used in vector control and the frequency components and angle errors proportional to the electrical angular frequency ωre of the permanent magnet synchronous motor 103a included in the control phase θd.

[0056] Here, the filter processing unit 204 may have a notch filter function that reduces only a specific frequency component and passes other frequency components. The notch filter processing is based on a transfer function Gn(s) of the notch filter in the following formula (5), for example.

number

[0057] In the first embodiment, the angle signal φ output from the rotational position detection unit 124 contains an angle error resulting from the DC offset error and amplitude error of the angle sensor, and in order to suppress the torque vibration of the drive control device of the AC motor caused by this angle error, a notch filter is used whose notch frequency ωn is the electrical angular frequency ωre of the permanent magnet synchronous motor 103a.

[0058] In the first embodiment, the electrical angular frequency ωre of the permanent magnet synchronous motor 103a outputted by the proportional-integral control unit 202 is used as the notch frequency ωn of the notch filter process, but the present invention is not limited to this. When it is desired to suppress mechanical resonance vibration of a drive control device including an AC motor, this mechanical resonance frequency may be set as the notch frequency ωn.

[0059] As described above, the filter processing unit 204 in the first embodiment is described as a notch filter processing, but a first-order lag filter processing may be used. For example, the phase difference Δθc′ output by the phase difference calculation unit 201 may be subjected to a first-order lag filter processing to be output to the proportional-integral control unit 202 as a phase difference ΔθcN′. Here, by setting the time constant of the first-order lag filter to a value that is at least twice the reciprocal of the electrical angular frequency ωre output by the proportional / integral control unit 202, the vibration component proportional to the electrical angular frequency ωre can be reduced from the phase difference Δθc'.

[0060] Next, the principle of suppressing torque vibration in a drive control device for an AC motor by using the phase / speed calculation unit 110 of the first embodiment will be described. When an angle error proportional to the drive frequency component of the AC motor is included in the detected angle signal of the angle sensor, the detected angle signal is an AC quantity proportional to the rotation speed of the AC motor, so that it is not possible to directly apply filtering to the detected angle signal to reduce the drive frequency component of the AC motor.

[0061] Therefore, in the first embodiment, the detected angle signal of the angle sensor is not directly used for the vector control of the AC motor, but a control phase θdc for vector control is prepared as a second detected angle synchronized with the detected angle signal of the angle sensor. The control phase θdc is compared with the angle signal φ and feedback-controlled to follow the change in the angle signal due to the acceleration / deceleration of the AC motor.

[0062] Furthermore, the angle error of the angle signal φ due to the DC offset error and amplitude error of the angle sensor appears as a frequency component proportional to the drive frequency component of the AC motor in the phase difference Δθc between the angle signal φ and the control phase θdc, so the vibration component due to the angle error can be reduced by filtering this phase difference Δθc.

[0063] In addition, by inserting a filter process into the phase difference Δθc in the phase feedback control loop and feeding back the calculation result of the electrical angular frequency ωre of the permanent magnet synchronous motor 103a to the notch frequency setting value of the filter process, the effect of reducing the vibration frequency component of the angle error is improved.

[0064] In this way, the calculation results of the electrical angular frequency ωre and the control phase θdc of the permanent magnet synchronous motor 103a output by the phase / speed calculation unit 110 are those in which the vibration components of the angle error resulting from the DC offset error and amplitude error of the angle sensor are reduced, so that vibrations in the vector control system using them are suppressed, and the occurrence of torque vibrations and speed vibrations can be suppressed.

[0065] As described above, according to the first embodiment, a phase feedback control system is constructed in which the difference between the detected angle of the angle sensor and the control phase of the vector control is passed through a filter that attenuates vibration components, and the electrical angle frequency and control phase of the rotation speed of the AC motor are calculated based on the filter output value. This makes it possible to suppress torque vibration of the AC motor caused by angle errors in the detected value of the angle sensor, and improve the control stability of the drive control device for the AC motor. EXAMPLES

[0066] In the second embodiment, in the calculation by the phase / speed calculation unit 110 of the electrical angular frequency ωre of the rotational speed of the AC motor and the control phase θdc of the dc-qc axis coordinate system for vector control, a feedforward (FF) compensation process using a first-order lag filter is added to the phase difference Δθc' between the angle signal φ corresponding to the rotor position of the AC motor 103 output from the rotational position detection unit 124 and the control phase θdc output from the control phase calculation unit 203.

[0067] This makes it possible to compensate for the phase difference between the angle signal φ and the control phase θdc that occurs when the rotational speed of the AC motor 103 accelerates or decelerates, and realizes a drive control device for an AC motor that can improve control tracking and torque accuracy compared to Example 1.

[0068] Fig. 6 is a block diagram showing a configuration example of a drive control device for an AC motor according to a second embodiment of the present invention. Only differences in configuration will be described below in comparison with the first embodiment shown in Fig. 1. In the second embodiment, a phase / speed calculation unit 110b is used instead of the phase / speed calculation unit 110 of the first embodiment.

[0069] FIG. 7 is a block diagram illustrating an example of the configuration of a phase / velocity calculation unit 110b according to the second embodiment. 7 includes a phase difference calculator 201b, a proportional / integral controller 202, a control phase calculator 203, and a filter processor 204. That is, in the configuration of the phase / velocity calculator 110 of the first embodiment shown in FIG. 3, the phase difference calculator 201 is replaced with the phase difference calculator 201b.

[0070] The phase / speed calculation unit 110b calculates and outputs the electrical angular frequency ωre and the control phase θdc in the dc-qc axis coordinate system of the permanent magnet synchronous motor 103a for vector control, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124.

[0071] Here, the configuration of the phase difference calculation unit 201b shown in FIG. 7 will be described in detail. FIG. 8 is a block diagram illustrating an example of the configuration of a phase difference calculation unit 201b according to the second embodiment. The phase difference calculation section 201b shown in FIG. 8 includes a ±π processing section 301, a ±π processing section 302, an adder / subtractor 801, and a phase difference FF compensation section 303b.

[0072] The phase difference calculation unit 201b calculates and outputs the phase difference Δθc-ff' between the angle signal φ and the control phase θdc based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124 and the control phase θdc output from the control phase calculation unit 203.

[0073] Next, the phase difference FF compensation unit 303b, which is a characteristic feature of the second embodiment, will be described in detail. FIG. 9 is a block diagram illustrating an example of the configuration of the phase difference FF compensation unit 303b according to the second embodiment. The phase difference FF compensation unit 303b shown in FIG. 9 includes a first-order lag filter unit 501b, a proportional unit 502b, and an adder 802.

[0074] The phase difference FF compensation unit 303b calculates and outputs a phase difference Δθc-ff' that compensates for the phase difference between the angle signal φ and the control phase θdc due to acceleration or deceleration of the rotational speed of the permanent magnet synchronous motor 103a, based on the phase difference Δθc' output from the ±π processing unit 302.

[0075] In the phase difference FF compensation unit 303b, the phase difference Δθc-ff', which is an input signal to the filter processing unit 204 shown in Fig. 7, is obtained by subjecting the phase difference Δθc-ff' between the angle signal φ and the control phase θdc output from the ±π processing unit 302 shown in Fig. 8 to first-order lag filter processing in the first-order lag filter unit 501, followed by proportional calculation in the proportional unit 502b, and adding the result in the adder 802. For example, the calculation is performed using the following equation (6).

number

[0076] Here, the first-order lag filter time constant Tpll-ff of the first-order lag filter unit 501b and the proportional gain Kpll-ff of the proportional converter 502b are set, for example, by the following equation (7).

number

[0077] Also, as a means for compensating for the phase difference between the angle signal φ and the control phase θdc that occurs when the rotation speed of the AC motor 103 accelerates or decelerates, it may be realized by adding a double integral calculation to the proportional calculation and integral calculation in the proportional / integral control unit 202, instead of the phase difference FF compensation unit 303b. For example, it is assumed that the calculation is performed according to the following equation (8).

number

[0078] Therefore, in the second embodiment, the phase difference between the angle signal φ and the control phase θdc that occurs when the rotation speed of the AC motor accelerates or decelerates can be compensated for by adding a phase FF compensation value using a first-order lag filter to the phase difference Δθc' between the angle signal φ of the rotational position detection unit 124 and the control phase θdc for performing vector control. This makes it possible to improve the control tracking ability and torque accuracy of the drive control device for the AC motor compared to the first embodiment. EXAMPLES

[0079] In the third embodiment, in the calculation by the phase / speed calculation unit 110 of the electrical angular frequency ωre of the rotational speed of the AC motor and the control phase θdc of the dc-qc axis coordinate system for performing vector control, the phase difference Δθc' between the angle signal φ output from the rotational position detection unit 124 and the control phase θdc used in vector control and the phase difference ΔθN' obtained by filtering this phase difference Δθc' to reduce frequency components proportional to the electrical angular frequency ωre and angle errors are switched according to the electrical angular frequency ωre of the rotational speed of the AC motor.

[0080] This makes it possible to avoid the phenomenon in which control becomes unstable due to the filter processing frequency based on the electrical angular frequency ωre and the control response angular frequency of phase feedback control becoming closer to each other, and to realize a drive control device for an AC motor that can improve control stability compared to the first embodiment.

[0081] Fig. 10 is a block diagram showing a configuration example of a drive control device for an AC motor according to a third embodiment of the present invention. Only different parts of the configuration will be described in comparison with the first embodiment shown in Fig. 1. In the third embodiment, a phase / speed calculation unit 110c is used instead of the phase / speed calculation unit 110 of the first embodiment.

[0082] FIG. 11 is a block diagram illustrating a configuration example of a phase / velocity calculation unit 110c according to the third embodiment. 11 includes a phase difference calculation section 201, a proportional / integral control section 202, a control phase calculation section 203, a filter processing section 204, and a phase difference switching section 601c. That is, the phase difference switching section 601c is added to the configuration of the phase difference calculation section 110 of the first embodiment shown in FIG.

[0083] The phase / speed calculation unit 110c calculates and outputs the electrical angular frequency ωre and the control phase θdc in the dc-qc axis coordinate system of the permanent magnet synchronous motor 103a for vector control, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124.

[0084] Next, the phase difference switching unit 601c, which is a characteristic feature of the third embodiment, will be described in detail. The phase difference switching unit 601c switches between the phase difference Δθc' output from the phase difference calculation unit 201 and the phase difference ΔθcN' obtained by reducing the frequency components proportional to the electrical angle frequency ωre from the phase difference Δθc' by the filter processing unit 204 in accordance with the electrical angle frequency ωre, and outputs the switched phase difference to the proportional / integral control unit 202.

[0085] For example, when the filter processing unit 204 performs notch filter processing with the electrical angle frequency ωre as the notch frequency ωn, if the electrical angle frequency ωre is close to the control response angular frequency ωpll* of the proportional / integral control unit 202, the phase difference switching unit 601c outputs the phase difference Δθc' output from the phase difference calculation unit 201, and if the electrical angle frequency ωre is away from the control response angular frequency ωpll* of the proportional / integral control unit 202, the phase difference switching unit 601c switches to output the filtered phase difference ΔθcN' output from the filter processing unit 204. This makes it possible to avoid instability due to control interference.

[0086] More specifically, the condition of the switching speed of the phase difference switching unit 601c may be set, for example, so that the filtered phase difference ΔθcN′ output from the filter processing unit 204 is output in a speed range in which the electrical angular frequency ωre is three or more times greater than the control response angular frequency ωpll* of the proportional / integral control unit 202.

[0087] Furthermore, in a drive control device for an AC motor, when the vibration frequency at which vibration becomes a problem is close to the control response angular frequency of phase feedback control, the setting value of the control response angular frequency can be shifted from the problematic vibration frequency to obtain the effect of suppressing vibration.

[0088] Therefore, in the third embodiment, the phase difference Δθc' output from the phase difference calculation unit 201 and the phase difference ΔθcN' obtained by filtering the phase difference Δθc' are switched in accordance with the electrical angular frequency ωre and output to the proportional / integral control unit 202 for phase feedback control, thereby preventing control from becoming unstable due to the filter processing frequency based on the electrical angular frequency ωre being close to the control response angular frequency of the phase feedback control. This makes it possible to improve the control stability of the drive control device for an AC motor compared to the first embodiment. EXAMPLES

[0089] In the fourth embodiment, in calculation by the phase / speed calculation unit 110 of the electrical angular frequency ωre of the rotational speed of the AC motor and the control phase θdc of the dc-qc axis coordinate system for vector control, a filter frequency setting value for filtering the phase difference Δθc' between the angle signal φ output from the rotational position detection unit 124 and the control phase θdc used in vector control is determined based on the vibration components extracted from the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system.

[0090] As a result, it becomes possible to suppress the vibration components generated in the drive control device for an AC motor, and a drive control device for an AC motor capable of suppressing vibration more effectively than the first embodiment can be realized.

[0091] Fig. 12 is a block diagram showing a configuration example of a drive control device for an AC motor according to a fourth embodiment of the present invention. Only different parts of the configuration will be described in comparison with the first embodiment shown in Fig. 1. In the fourth embodiment, a phase / speed calculation unit 110d is used instead of the phase / speed calculation unit 110 of the first embodiment.

[0092] FIG. 13 is a block diagram illustrating a configuration example of a phase / velocity calculation unit 110d according to the fourth embodiment. 13 includes a phase difference calculation section 201, a proportional / integral control section 202, a control phase calculation section 203, a filter processing section 204, and a vibration frequency extraction section 701d. That is, the vibration frequency extraction section 701d is added to the configuration of the phase difference calculation section 110 of the first embodiment shown in FIG.

[0093] The phase / speed calculation unit 110d calculates and outputs the electrical angular frequency ωre of the permanent magnet synchronous motor 103a for vector control and the control phase θdc of the dc-qc coordinate system, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124 and the current detection values ​​Idc and Iqc on the dc-qc coordinate system output from the dq coordinate conversion unit 114.

[0094] Next, the vibration frequency extraction unit 701d, which is a characteristic feature of the fourth embodiment, will be described in detail. The vibration frequency extraction unit 701d extracts an vibration component from at least one of the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system output from the dq coordinate transformation unit 114, and determines and outputs the setting value of the notch filter frequency ωn of the filter processing unit 204 so that the vibration component is attenuated.

[0095] For example, vibration components may be extracted from the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system, and the vibration frequency of the vibration component with the largest amplitude among the vibration components may be set as the setting value of the notch filter frequency ωn of the filter processing unit 204. In this case, the setting value of the notch filter frequency ωn transitions so that the amplitude of the vibration component extracted from at least one of the current detection values ​​Idc and Iqc becomes smaller. In this case, if there are multiple vibration frequencies with the largest amplitude, a condition for determining the setting value of the notch filter frequency ωn may be added to the transition condition of the setting value of the notch filter frequency ωn by providing hysteresis for the difference in amplitude between the vibration frequencies, or by squaring the amplitudes of the vibration frequencies and minimizing the sum of the squared values.

[0096] In addition, the vibration frequency extraction unit 701d may be configured to correct the setting value of the notch filter frequency ωn of the filter processing unit 204 based on the electrical angular frequency ωre output by the proportional / integral control unit 202 so that the vibration components extracted from the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system are attenuated thereto.

[0097] Therefore, in the fourth embodiment, a filter frequency for filtering the phase difference Δθc' between the angle signal φ of the rotational position detector 124 and the control phase θdc for performing vector control is determined based on a vibration component extracted from at least one of the detected current values ​​Idc and Iqc on the dc-qc axis coordinate system. This makes it possible to realize vibration suppression control according to the vibration component generated in the drive controller for an AC motor, thereby improving the vibration suppression effect of the drive controller for an AC motor compared to the first embodiment. EXAMPLES

[0098] In the fifth embodiment, in the calculation by the phase / speed calculation unit 110 of the electrical angular frequency ωre of the rotational speed of the AC motor and the control phase θdc of the dc-qc axis coordinate system for vector control, the notch frequency ωn of the filter processing unit 204, which filters the phase difference Δθc' between the angle signal φ output from the rotational position detection unit 124 and the control phase θdc used in vector control, the damping gain Qmin which is the gain value at the notch frequency ωn, and the damping ratio ζdn of the filter pole which determines the width of the notch are made variable according to the operating state of the drive control device for the AC motor.

[0099] This makes it possible to avoid control instability caused by a high gain in the control loop, and to realize a drive control device for an AC motor that can improve control stability and vibration suppression effects compared to the first embodiment.

[0100] Fig. 14 is a block diagram showing a configuration example of a drive control device for an AC motor according to a fifth embodiment of the present invention. When comparing the fifth embodiment with the first embodiment shown in Fig. 1, only differences in the configuration are shown. In the fifth embodiment, a phase / speed calculation unit 110e is used instead of the phase / speed calculation unit 110 of the first embodiment.

[0101] FIG. 15 is a block diagram illustrating a configuration example of a phase / velocity calculation unit 110e according to the fifth embodiment. 15 includes a phase difference calculator 201, a proportional / integral controller 202, a control phase calculator 203, a filter processor 204, and a filter setting parameter calculator 901e. That is, the filter setting parameter calculator 901e is added to the configuration of the phase / velocity calculator 110 of the first embodiment shown in FIG.

[0102] The phase / speed calculation unit 110e calculates and outputs the electrical angular frequency ωre of the permanent magnet synchronous motor 103a for vector control and the control phase θdc of the dc-qc coordinate system, based on the angle signal φ corresponding to the rotor position of the permanent magnet synchronous motor 103a output from the rotational position detection unit 124, the current detection values ​​Idc and Iqc on the dc-qc coordinate system output from the dq coordinate conversion unit 114, and the torque command Tm* output from the command generator 105.

[0103] Next, the filter setting parameter calculation unit 901e, which is a characteristic part of the fifth embodiment, will be described in detail. The filter setting parameter calculation unit 901e derives the transfer characteristic from the voltage input to the current output of the permanent magnet synchronous motor 103a from, for example, the motor constant of the permanent magnet synchronous motor 103a, the electrical angular frequency ωre of the permanent magnet synchronous motor 103a, and the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system, and calculates the attenuation gain Qmin of the notch filter based on the resonance gain of the transfer characteristic. Note that, instead of the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system, the torque command Tm* output from the command generator 105 may be used.

[0104] By making the damping gain Qmin of the notch filter variable according to the magnitude of the resonance gain of the AC motor to be controlled, the ωre vibration component of the angle error caused by the DC offset error and amplitude error of the angle sensor can be suppressed to an arbitrary magnitude regardless of the operating state.

[0105] Furthermore, in the high speed range, the resonant gain of the permanent magnet synchronous motor 103a does not change, but the frequency band narrows, so by increasing or decreasing the damping ratio ζdn of the notch filter in accordance with the electrical angular frequency ωre, the effect of suppressing the ωre vibration component of the angle error resulting from the DC offset error and amplitude error of the angle sensor is improved.

[0106] In addition, the filter setting parameter calculation unit 901e may simply vary the attenuation gain Qmin of the notch filter based on at least one of the electrical angular frequency ωre of the permanent magnet synchronous motor 103a, the current detection values ​​Idc and Iqc on the dc-qc axis coordinate system, and the torque command Tm* output from the command generator 105.

[0107] Furthermore, the filter setting parameter calculation unit 901e calculates the damping gain Qmin of the notch filter based on, for example, the calculation period of the controller 101 including the filter processing unit 204 and the vector control unit 112 and the electrical angular frequency ωre of the permanent magnet synchronous motor 103a. In particular, when the effect of the calculation delay of the filter processing unit 204 and the vector control unit 112 becomes large in the high-speed range, if the damping gain Qmin of the notch filter is kept increased, the control system will become unstable, and therefore it is necessary to lower the damping gain Qmin.

[0108] In addition, as a modified example of the third embodiment, the filter setting parameter calculation unit 901e makes the attenuation gain Qmin of the notch filter variable in accordance with the electrical angular frequency ωre of the permanent magnet synchronous motor 103a and the control response angular frequency ωpll* of the proportional-integral control unit 202, thereby making it possible to avoid instability due to control interference.

[0109] For example, when the electrical angle frequency ωre is close to the control response angular frequency ωpll* of the proportional / integral control unit 202, the attenuation gain Qmin of the notch filter is set to zero, and when the electrical angle frequency ωre is away from the control response angular frequency ωpll* of the proportional / integral control unit 202, the attenuation gain Qmin is changed to any value other than zero.

[0110] Therefore, in the fifth embodiment, the notch frequency ωn, the damping gain Qmin, and the damping ratio ζdn, which are setting parameters of the notch filter of the filter processor 204, are changed according to the operating state of the AC motor. This makes it possible to optimize the effect of reducing the ωre vibration component by the notch filter and to avoid control instability due to high gain of the control loop, thereby realizing a drive control device for an AC motor that can improve control stability and vibration suppression effect more than those of the first embodiment. EXAMPLES

[0111] A sixth embodiment of the present invention is an AC motor drive control device using any one of the first to fifth embodiments, which is applied to a railway vehicle.

[0112] FIG. 16 is a diagram showing a schematic configuration of a part of a railway vehicle equipped with a drive control device for an AC motor using any one of the first to fifth embodiments, as a sixth embodiment of the present invention.

[0113] The railway vehicle shown in FIG. 16 has a bogie on which AC motors 103a and 103b are mounted, and a bogie on which AC motors 103c and 103d are mounted, and is equipped with an AC motor drive control device including a controller 101, a power converter 102, a command generator 105, and a phase current detection unit 121.

[0114] In the railway vehicle according to the sixth embodiment, power supplied from an overhead line via a current collector is converted into AC power by a power converter 102 in accordance with a torque command value Tm* generated by a command generator 105 based on an operation command input by a driver via a master controller, and the AC power is supplied to the AC motor 103 to drive the AC motor 103. The AC motor 103 is connected to the axle of the railway vehicle, and the running of the railway vehicle is controlled by the AC motor 103.

[0115] In the railway vehicle according to the sixth embodiment, by applying any one of the first to fifth embodiments as a drive control device for an AC motor, it is possible to improve the control stability and the control response.

[0116] Although the embodiments of the present invention have been described above, the present invention can be similarly applied to other AC motors. For example, the present invention can be similarly applied to permanent magnet synchronous motors, wound-rotor synchronous motors, and synchronous machines using reluctance torque. The present invention can also be similarly applied to generator control. Furthermore, the present invention is not limited to the above-described embodiments, and various modifications can be made without departing from the spirit of the present invention. [Explanation of symbols]

[0117] 101 controller, 102 power converter, 103 AC motor, 105 command generator, 110 phase / speed calculation unit, 111 current command calculation unit, 112 vector control unit, 113 current detection unit, 114 dq coordinate conversion unit, 115 polar coordinate conversion unit, 116 UVW coordinate conversion unit, 117 PWM signal controller, 121 phase current detection unit, 123 input terminal, 124 rotation position detection unit, 131 smoothing capacitor, 132 main circuit section, 133 gate driver, 134 DC resistor, 201 phase difference calculation unit, 202 proportional / integral control unit, 203 control phase calculation unit, 204 filter processing section, 301, 302 ±π processing section, 303b phase difference FF compensation section, 401, 402, 502b proportional unit, 403 integrator, 501b first-order lag filter unit, 601c Phase difference switching unit, 701d Vibration frequency extraction unit, 801 Addition / subtraction unit, 802 Adder, 901e Filter setting parameter calculation section

Claims

1. A drive control device for an AC motor driven by a power converter using vector control, a phase difference calculation unit that calculates a phase difference between rotational position information of the AC motor and a control phase used when performing the vector control, and outputs a first phase difference; a filter unit that outputs a second phase difference obtained by attenuating a specific frequency component from the first phase difference through filtering; a speed and phase calculation unit that calculates a rotation speed and the control phase of the AC motor based on the second phase difference; A drive control device for an AC motor comprising:

2. 2. The drive control device for an AC motor according to claim 1, The filter unit performs notch filtering and attenuates the specific frequency component based on a notch frequency, an attenuation gain at the notch frequency, and an attenuation ratio related to a notch width. A drive control device for an AC motor comprising:

3. 3. The drive control device for an AC motor according to claim 2, The filter unit sets the notch frequency based on a rotation speed of the AC motor. A drive control device for an AC motor comprising:

4. 3. The drive control device for an AC motor according to claim 2, The filter unit sets the notch frequency based on a mechanical resonance frequency of the drive control device including the AC motor. A drive control device for an AC motor comprising:

5. 2. The drive control device for an AC motor according to claim 1, The filter unit performs a first-order lag filter process. A drive control device for an AC motor comprising:

6. 6. The drive control device for an AC motor according to claim 5, The filter unit changes a time constant of the first-order lag filtering process based on a rotation speed of the AC motor. A drive control device for an AC motor comprising:

7. 7. A drive control device for an AC motor according to claim 1, The phase difference calculation unit calculates a phase difference between the rotational position information and the control phase, and outputs the phase difference after feedforward compensation as the first phase difference. A drive control device for an AC motor comprising:

8. 7. A drive control device for an AC motor according to claim 1, a phase difference switching unit that switches the first phase difference or the second phase difference in response to a rotation speed of the AC motor, The speed and phase calculation unit calculates a rotation speed and a control phase of the AC motor based on the first phase difference or the second phase difference. A drive control device for an AC motor comprising:

9. The drive control device for an AC motor according to any one of claims 2 to 4, The filter unit sets the notch frequency based on a frequency of an oscillation component of at least one of two axial currents flowing through the AC motor. A drive control device for an AC motor comprising:

10. The drive control device for an AC motor according to any one of claims 2 to 4, The filter unit sets the attenuation gain based on at least one of a rotation speed of the AC motor, a torque command of the AC motor, a detected current of the AC motor, and a calculation period of the drive control device. A drive control device for an AC motor comprising:

11. A railway vehicle equipped with a drive control device for an AC motor according to any one of claims 1 to 10.

12. A drive control method for an AC motor that performs vector control on an AC motor driven by a power conversion device, comprising: calculating a phase difference between rotational position information of the AC motor and a control phase used in the vector control, and outputting a first phase difference; outputting a second phase difference obtained by filtering and attenuating a specific frequency component from the first phase difference; Calculating the rotation speed of the AC motor and the control phase based on the second phase difference. A drive control method for an AC motor comprising:

13. A drive control method for an AC motor according to claim 12, comprising the steps of: The filtering process is a notch filtering process, and the specific frequency component is attenuated based on a notch frequency, an attenuation gain at the notch frequency, and an attenuation ratio related to a notch width. A drive control method for an AC motor comprising:

14. A drive control method for an AC motor according to claim 12, comprising the steps of: The filtering process is a first-order lag filtering process, and a time constant of the first-order lag filtering process is changed based on a rotation speed of the AC motor. A drive control method for an AC motor comprising:

15. A drive control method for an AC motor according to any one of claims 12 to 14, comprising the steps of: The first phase difference is calculated by calculating the phase difference between the rotational position information and the control phase, and is further subjected to feedforward compensation and output. A drive control method for an AC motor comprising:

16. A drive control method for an AC motor according to any one of claims 12 to 14, comprising the steps of: switching the first phase difference or the second phase difference in response to a rotation speed of the AC motor; Calculating a rotation speed of the AC motor and the control phase based on the first phase difference or the second phase difference. A drive control method for an AC motor comprising:

Images