Control device for AC rotating machine and electric power steering device

Abstract

This control device for an AC rotating machine comprises: an inverter that applies voltage to the AC rotating machine; a current detector that detects rotating machine current flowing through the AC rotating machine; a first axis voltage command value calculation unit that calculates a voltage command value of the first axis of the rotating two axes of the AC rotating machine; and a second axis voltage command value calculation unit that calculates a voltage command value of the second axis of the rotating two axes. The first axis voltage command value calculation unit performs an integration calculation using a first deviation and a second deviation, said first deviation being the deviation between the current command value of the first axis and the detected current value of the first axis of the rotating machine current, said second deviation being the deviation between the current command value of the second axis and the detected current value of the second axis of the rotating machine current, and calculates the voltage command value of the first axis on the basis of the result of the integration calculation.

Description

[Technical field]

[0001] The present disclosure relates to a control device for an AC rotating machine and an electric power steering device. [Background technology]

[0002] Patent Document 1 discloses a control device that compensates for interference terms in two rotating axes (e.g., d-axis and q-axis) in vector control of an AC rotating machine. This control device includes a current PI controller and a decoupling error corrector. The current PI controller has two integrators corresponding to the d-axis and q-axis, respectively, to perform proportional-integral control. Similarly, the decoupling error corrector also has two integrators corresponding to the d-axis and q-axis, respectively. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] JP 2010-119245 A Summary of the Invention [Problem to be solved by the invention]

[0004] In such a control device, a phenomenon in which the accuracy of control is reduced (so-called overshoot, etc.) may occur due to windup in the integrator. As a method for suppressing the windup of the integrator, it is effective to perform anti-windup, such as limiting the integral term to an appropriate value. Here, in the configuration of Patent Document 1, for example, when calculating a voltage command value of the d-axis, the calculation result of the integrator in the current PI controller and the calculation result of the integrator in the non-interference error corrector are used. In this way, when the calculation results of two integrators are used for one control target value (for example, a voltage command value of the d-axis), it is difficult to perform anti-windup appropriately. As a result, there is a problem that the control becomes unstable.

[0005] The present disclosure has been made in consideration of the above circumstances, and has an object to provide a control device for an AC rotating machine and an electric power steering device that are capable of performing more stable control. [Means for solving the problem]

[0006] In order to solve the above problems, a control device for an AC rotating machine according to one embodiment of the present disclosure includes an inverter that applies a voltage to the AC rotating machine, a current detector that detects a rotating machine current flowing through the AC rotating machine, a first axis voltage command value calculator that calculates a voltage command value for a first axis among two rotating shafts of the AC rotating machine, and a second axis voltage command value calculator that calculates a voltage command value for a second axis among the two rotating shafts, wherein the first axis voltage command value calculator performs an integration operation using a first deviation that is a deviation between a current command value on the first axis and a detected current value of the rotating machine current on the first axis, and a second deviation that is a deviation between a current command value on the second axis and a detected current value of the rotating machine current on the second axis, and calculates the voltage command value for the first axis based on a result of the integration operation.

[0007] An electric power steering device according to one aspect of the present disclosure includes the above-mentioned AC rotating machine control device, the AC rotating machine, and a driving force transmission mechanism that transmits driving force of the AC rotating machine to a steering system. Effect of the Invention

[0008] According to the present disclosure, it is possible to provide a control device for an AC rotating machine and an electric power steering device capable of performing more stable control. [Brief description of the drawings]

[0009] [Figure 1] 1 is a block diagram showing a schematic configuration of a control device for a rotating machine according to a first embodiment. [Diagram 2] 3 is a diagram for explaining a generation principle of a switching signal in the first embodiment. FIG. [Diagram 3] 2 is a block diagram showing the configuration of a d-axis voltage command value calculator in FIG. 1. [Figure 4] 2 is a block diagram showing a configuration of a q-axis voltage command value calculator in FIG. 1. [Diagram 5] 2 is a flowchart showing an example of a process performed in the corrected voltage generator of FIG. 1. [Figure 6] 5 is a diagram showing an example of waveforms of a voltage command value and a corrected voltage command value in the first embodiment. FIG. [Figure 7] 4 is a Bode plot showing the transfer characteristics from a q-axis current command value to a q-axis detected current value when the rotation angular velocity is high in the control device according to the first embodiment. FIG. [Figure 8] FIG. 11 is a block diagram showing a schematic configuration of a control device for a rotating machine according to a second embodiment. [Figure 9] FIG. 11 is a block diagram showing a configuration of a d-axis voltage command value calculator according to the second embodiment. [Figure 10] FIG. 11 is a block diagram showing a configuration of a q-axis voltage command value calculator according to the second embodiment. [Figure 11] FIG. 11 is a diagram showing the relationship between a proportional gain, a limiting gain, and a voltage amplitude in the second embodiment. [Figure 12] 11 is a graph showing an example of a waveform of a voltage across a shunt resistor in the second embodiment. [Figure 13] FIG. 11 is a diagram showing a case where some corrected voltage command values ​​are greater than the upper limit value in the second embodiment. [Figure 14] FIG. 11 is a block diagram showing a schematic configuration of an electric power steering device according to a third embodiment. [Figure 15] 11 is a graph showing the relationship between the steering torque, the q-axis current command value, and the vehicle speed in the third embodiment. [Figure 16] FIG. 11 is a diagram showing the relationship between vehicle speed and proportional gain in the third embodiment. [Figure 17] FIG. 11 is a block diagram showing a schematic configuration of an electric power steering device according to a fourth embodiment. [Figure 18] FIG. 13 is a diagram showing the relationship between the amount of change in steering torque and a limit gain in the fourth embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0010] Hereinafter, a rotating machine control device and an electric power steering device according to the present disclosure will be described in detail with reference to the drawings.

[0011] Embodiment 1 Fig. 1 is a block diagram showing a schematic configuration of a control device for a rotating machine according to embodiment 1. As shown in Fig. 1, the control device 1 includes a rotor position detector 11, an inverter 12, and a controller 13. The control device 1 controls the rotating machine 10 based on a torque command T_ref as a control command input from outside the control device 1.

[0012] The rotating machine 10 is a three-phase AC rotating machine having three-phase windings U, V, and W. The rotating machine 10 is an AC rotating machine that can be controlled based on two rotating shafts. In this specification, the "two rotating shafts" refer to two axes that rotate in synchronization with the rotor of the rotating machine 10 and are perpendicular to each other in a cross section. The "cross section" is a cross section perpendicular to the central axis of the rotor. For example, the two rotating shafts may be dq axes. The d axis is an axis that connects the central axis of the rotor and the magnetic poles. The q axis is an axis that is perpendicular to both the d axis and the central axis. The two rotating shafts may be γ-δ axes. The γ axis is an axis that is shifted in the rotational direction from the d axis. The δ axis is an axis that is perpendicular to both the γ axis and the central axis. One of the two rotating shafts is called the first axis, and the other is called the second axis. For example, when the d axis is called the first axis, the q axis is called the second axis. The q axis may be the first axis and the d axis may be the second axis. Similarly, when the γ axis is called the first axis, the δ axis is called the second axis.

[0013] In the following, a case will be described in which the rotating machine 10 is a permanent magnet synchronous rotating machine and the two rotating shafts are dq axes. However, the rotating machine 10 may be, for example, a wound field synchronous rotating machine, an induction rotating machine, a synchronous reluctance motor, etc. Also, the d axis and the q axis in the following disclosure may be replaced with the δ axis and the γ axis.

[0014] The rotor position detector 11 includes a resolver, an encoder, an MR (magnetic resistance) sensor, etc., and detects the rotor position θ using these. The rotor position θ is the position in the rotation direction of the rotor of the rotating machine 10. In this embodiment, the rotor position θ of the rotating machine 10 is detected using the rotor position detector 11. However, it is also possible to employ a configuration in which the rotor position θ of the rotating machine 10 is estimated without using the rotor position detector 11. In other words, in the present disclosure, the control device 1 does not need to include the rotor position detector 11.

[0015] The inverter 12 is a power converter that applies a voltage to the rotating machine 10. Specifically, under the control of the controller 13, the inverter 12 converts DC power supplied from the DC power supply BT into AC power and supplies the converted AC power to the rotating machine 10. The DC power supply BT includes devices for supplying DC power, such as a DC-DC converter, a diode rectifier, and a PWM rectifier, in addition to a battery. In this specification, the output voltage (DC bus voltage) of the DC power supply BT is represented as Vdc.

[0016] The inverter 12 includes upper arm switching elements Sup, Svp, Swp, lower arm switching elements Sun, Svn, Swn, and shunt resistors Ru, Rv, Rw. The upper arm switching elements Sup, Svp, Swp are connected to the positive electrode of a DC power supply BT. The lower arm switching elements Sun, Svn, Swn are connected to the upper arm switching elements Sup, Svp, Swp, respectively, and are connected to the negative electrode of the DC power supply BT via the shunt resistors Ru, Rv, Rw, respectively.

[0017] Here, a U-phase series circuit is formed by the upper arm switching element Sup, the lower arm switching element Sun, and the shunt resistor Ru. In this U-phase series circuit, the connection point between the upper arm switching element Sup and the lower arm switching element Sun is connected to the winding U of the rotating machine 10.

[0018] Furthermore, a V-phase series circuit is formed by the upper arm switching element Svp, the lower arm switching element Svn, and the shunt resistor Rv. In this V-phase series circuit, a connection point between the upper arm switching element Svp and the lower arm switching element Svn is connected to the winding V of the rotating machine 10.

[0019] Further, the upper arm switching element Swp, the lower arm switching element Swn, and the shunt resistor Rw form a W-phase series circuit. In this W-phase series circuit, a connection point between the upper arm switching element Swp and the lower arm switching element Swn is connected to the winding W of the rotating machine 10.

[0020] The upper arm switching elements Sup, Svp, Swp and the lower arm switching elements Sun, Svn, Swn may be semiconductor switches such as IGBTs (Insulated Gate Bipolar Transistors), bipolar transistors, and MOSFETs (Metal-Oxide-Semiconductor Field Effect Transistors).

[0021] The upper arm switching elements Sup, Svp, Swp are respectively input with switching signals Gup, Gvp, Gwp output from the controller 13. The lower arm switching elements Sun, Svn, Swn are respectively input with switching signals Gun, Gvn, Gwn output from the controller 13. The upper arm switching elements Sup, Svp, Swp and the lower arm switching elements Sun, Svn, Swn are turned on or off by the switching signals Gup, Gvp, Gwp, Gun, Gvn, Gwn output from the controller 13. In this specification and drawings, the switching signals Gup, Gvp, Gwp, Gun, Gvn, Gwn may be collectively referred to as "switching signals Gup-Gwn".

[0022] For example, when the switching signal Gup is an "on command (=1)", the upper arm switching element Sup is in the on state, and when the switching signal Gup is an "off command (=0)", the upper arm switching element Sup is in the off state. The same is true for the other switching elements (upper arm switching elements Svp, Swp and lower arm switching elements Sun, Svn, Swn). In this way, the inverter 12 generates AC power to be supplied to the rotating machine 10 from the DC power supplied by the DC power supply BT.

[0023] The shunt resistors Ru, Rv, and Rw are resistive elements for detecting current. The shunt resistor Ru outputs to the controller 13 a voltage VRu (=-Ru×iu) proportional to a current (rotating machine current) iu flowing through the winding U of the rotating machine 10. The shunt resistor Rv outputs to the controller 13 a voltage VRv (=-Rv×iv) proportional to a current (rotating machine current) iv flowing through the winding V of the rotating machine 10. The shunt resistor Rw outputs to the controller 13 a voltage VRw (=-Rw×iw) proportional to a current (rotating machine current) iw flowing through the winding W of the rotating machine 10. In this specification and drawings, the voltages VRu, VRv, and VRw may be collectively referred to as "voltages VRu to VRw."

[0024] Here, the voltages VRu, VRv, VRw are values ​​obtained by multiplying the rotating machine currents iu, iv, iw by the resistance values ​​of the shunt resistors Ru, Rv, Rw, and are proportional to the currents iu, iv, iw. Therefore, the voltages VRu, VRv, VRw can be said to be values ​​of detected currents (detected values ​​of the rotating machine currents). The inverter 12 may be integrated with the rotating machine 10. The integrated inverter 12 and rotating machine 10 are called a power pack.

[0025] The controller 13 uses the torque command T_ref, the voltages VRu, VRv, VRw, and the rotor position θ as input values, and generates switching signals Gup to Gwn for driving the inverter 12 based on these. The controller 13 is a PWM controller realized by a discrete time calculator such as a microcomputer or a DSP (Digital Signal Processor). The controller 13 includes a current command value calculator 21, a current detector 22, a coordinate converter 23 (detection coordinate converter), a voltage command value calculator 24, a coordinate converter 25 (control coordinate converter), a corrected voltage generator 26, a PWM signal generator 27, and a speed calculator 28.

[0026] The speed calculator 28 performs a differential operation or a difference operation on the rotor position θ to obtain the rotational angular velocity ω of the rotating machine 10. The speed calculator 28 inputs the obtained rotational angular velocity ω to the current command value calculator 21 as well as to the d-axis voltage command value calculator 24d and the q-axis voltage command value calculator 24q of the voltage command value calculator 24. The current command value calculator 21 calculates current command values ​​id_ref, iq_ref based on the torque command T_ref, the DC bus voltage Vdc, and the rotational angular velocity ω. The current command values ​​id_ref, iq_ref are command values ​​(target values) of the currents supplied to the rotating machine 10. id_ref is also called a "field-weakening current command value," and iq_ref is also called a "torque current command value." As a calculation method executed by the current command value calculator 21, known maximum torque per amp (MTPA) control, maximum torque per voltage (MTPV) control, and flux-weakening control may be appropriately combined for each operation range (range of speed-torque characteristics). The rotational angular velocity ω may be converted into the rotor speed, and each control may be performed using the speed.

[0027] Next, the PWM signal generator 27 will be described. The PWM signal generator 27 outputs switching signals Gup to Gwn that are PWM (Pulse Width Modulation) modulated based on the corrected voltage command values ​​vu', vv', and vw' output from the corrected voltage generator 26. The corrected voltage command values ​​vu', vv', and vw' will be described later.

[0028] 2 is a diagram for explaining the principle of generating switching signals in the first embodiment. The PWM signal generator 27 generates switching signals Gup to Gwn by comparing the corrected voltage command values ​​vu', vv', and vw' with a carrier triangular wave (carrier wave) C having a cycle Tc (frequency fc). The corrected voltage command values ​​vu', vv', and vw' correspond to the U phase, V phase, and W phase, respectively.

[0029] Specifically, if the corrected voltage command value vu' is greater than the carrier triangular wave C, the PWM signal generator 27 turns on the switching signal Gup ("1") and turns off the switching signal Gun ("0"). Conversely, if the corrected voltage command value vu' is smaller than the carrier triangular wave C, the PWM signal generator 27 turns off the switching signal Gup ("0") and turns on the switching signal Gun ("1").

[0030] Furthermore, if the corrected voltage command value vv' is greater than the carrier triangular wave C, the PWM signal generator 27 turns the switching signal Gvp on ("1") and turns the switching signal Gvn off ("0"). Conversely, if the corrected voltage command value vv' is smaller than the carrier triangular wave C, the PWM signal generator 27 turns the switching signal Gvp off ("0") and turns the switching signal Gvn on ("1").

[0031] Furthermore, if the corrected voltage command value vw' is greater than the carrier triangular wave C, the PWM signal generator 27 turns on the switching signal Gwp ("1") and turns off the switching signal Gwn ("0"). Conversely, if the corrected voltage command value vw' is smaller than the carrier triangular wave C, the PWM signal generator 27 turns off the switching signal Gwp ("0") and turns on the switching signal Gwn ("1").

[0032] Incidentally, a short circuit prevention time (dead time) may be provided for the switching signals Gup to Gwn so that the upper arm switching elements Sup, Svp, Swp and the lower arm switching elements Sun, Svn, Swn of the inverter 12 are not turned on simultaneously.

[0033] The switching signals Gup to Gwn include a pattern in which all of the lower arm switching elements Sun, Svn, and Swn are turned on during one electrical angle period of the rotating machine 10. Specifically, a pattern in which all of the switching signals Gun, Gvn, and Gwn are turned on (1) is included, as in section D in FIG.

[0034] Here, the PWM-modulated voltage applied from the inverter 12 to the rotating machine 10 contains components of the corrected voltage command values ​​vu', vv', vw' as well as components that are integer multiples of the period Tc of the carrier triangular wave C. As a result, a current having components that are integer multiples of the period Tc flows through the rotating machine 10, and depending on the value of the period Tc, the rotating machine 10 may produce abnormal noise.

[0035] In order to prevent such abnormal noise, for example, when the rotating machine 10 is used as a motor that assists steering of an electric power steering, the period Tc of the carrier triangular wave C may be set to 60 [μs] or less. By setting Tc=60 [μs], the frequency fc (=1 / Tc) of the abnormal noise becomes 16.6 kHz, which is unlikely to be a noise that is unpleasant to humans. More preferably, the period Tc of the carrier triangular wave C may be set to about 50 [μs]. By setting Tc=50 [μs], the frequency fc (=1 / Tc) of the abnormal noise becomes about 20 kHz, which is almost inaudible to humans. The frequency band that humans can hear is about 20 Hz to 20 kHz. In the following, the explanation will be given assuming Tc=50 [μs].

[0036] Next, the current detector 22 shown in Fig. 1 will be described. The current detector 22 outputs pre-conversion detection currents ius, ivs, iws using voltages VRu, VRv, VRw across the shunt resistors Ru, Rv, Rw and switching signals Gup-Gwn output from the PWM signal generator 27. Specifically, the current detector 22 acquires voltages VRu, VRv, VRw across the shunt resistors Ru, Rv, Rw at timing "X" shown in Fig. 2. This timing "X" is the timing when the carrier triangular wave C reaches a maximum value (DC bus voltage Vdc).

[0037] 2, at timing "X", the switching signals Gun, Gvn, Gwn input to the lower-arm switching elements Sun, Svn, Swn are all on ("1"), so the current detector 22 obtains the values ​​of pre-conversion detection currents ius, ivs, iws by dividing the voltages VRu, VRv, VRw across the shunt resistors Ru, Rv, Rw by -Ru, -Rv, -Rw, respectively.

[0038] The coordinate converter 23 performs coordinate conversion based on the pre-conversion detected currents ius, ivs, iws detected by the current detector 22 and the rotor position θ detected by the rotor position detector 11. As a result, the coordinate converter 23 calculates the detected current values ​​id, iq on the two rotation axes (d-axis, q-axis). The coordinate converter 23 also inputs the calculation results (detected current values ​​id, iq after coordinate conversion) to the voltage command value calculation unit 24.

[0039] The voltage command value calculation unit 24 calculates voltage command values ​​vd, vq on the two rotation axes (d-axis, q-axis) based on the current command values ​​id_ref, iq_ref calculated by the current command value calculator 21, the detected current values ​​id, iq, the rotational angular velocity ω, and the DC bus voltage Vdc. The voltage command value calculation unit 24 will be described in detail below.

[0040] The voltage command value calculator 24 includes a first deviation calculator 24a, a second deviation calculator 24b, a d-axis voltage command value calculator 24d, and a q-axis voltage command value calculator 24q. The first deviation calculator 24a calculates a d-axis current deviation ed, which is the deviation between the d-axis current command value id_ref and the d-axis detected current value id. The second deviation calculator 24b calculates a q-axis current deviation eq, which is the deviation between the q-axis current command value iq_ref and the q-axis detected current value iq. The value of the d-axis current deviation ed calculated by the first deviation calculator 24a is input to the d-axis voltage command value calculator 24d and the q-axis voltage command value calculator 24q. The value of the q-axis current deviation eq calculated by the second deviation calculator 24b is input to the d-axis voltage command value calculator 24d and the q-axis voltage command value calculator 24q.

[0041] The d-axis voltage command value calculator 24d calculates a d-axis voltage command value vd using the d-axis current deviation ed, the q-axis current deviation eq, the DC bus voltage Vdc, and the rotational angular velocity ω. The q-axis voltage command value calculator 24q calculates a q-axis voltage command value vq using the q-axis current deviation eq, the d-axis current deviation ed, the DC bus voltage Vdc, and the rotational angular velocity ω.

[0042] In this specification, the deviation between the current command value and the detected current value on the first axis of the two rotating axes may be referred to as the first deviation. Similarly, the deviation between the current command value and the detected current value on the second axis of the two rotating axes may be referred to as the second deviation. For example, when the d axis is the first axis, the d axis current deviation ed, which is the deviation between the current command value id_ref on the d axis and the d axis detected current value id, is the "first deviation" and the q axis current deviation eq is the "second deviation". Similarly, when the q axis is the first axis, the q axis current deviation eq is the "first deviation" and the d axis current deviation ed is the "second deviation".

[0043] FIG. 3 shows the d-axis voltage command value calculator 24d in detail, and FIG. 4 shows the q-axis voltage command value calculator 24q in detail. The d-axis voltage command value calculator 24d will be described below with reference to Fig. 3. The d-axis voltage command value calculator 24d includes a d-axis proportional amplifier 101d, a limiter 102d, a d-axis integral amplifier 103d, a high-pass filter (HPF) 104d, a response angular frequency amplifier 105d, an inductance amplifier 106d, a multiplier 107d, a subtractor 108d, an integrator 109d, a limiter 110d, and an adder 111d.

[0044] The d-axis proportional amplifier 101d calculates the d-axis proportional output Vdp by multiplying the d-axis current deviation ed by Kpd. That is, Vdp = ed × Kpd. "Kpd" is the d-axis proportional gain that is multiplied to obtain a preferable response of the rotational electrical machine current actually flowing with respect to the current command value id_ref. For example, Kpd = ωcc × Ld. Here, ωcc is the response angular frequency (more specifically, the reciprocal of the time constant of the feedback control system) for adjusting the frequency response of the rotational electrical machine current with respect to the current command value to be in a preferable range, and Ld is the d-axis inductance of the rotational electrical machine 10. However, the value of Kpd is not limited to ωcc × Ld and may be appropriately adjusted by actually measuring the responsiveness of the rotational electrical machine current actually flowing with respect to the current command value id_ref, etc. The d-axis proportional output Vdp calculated by the d-axis proportional amplifier 101d is input to the limiter 102d.

[0045] The limiter 102d compares the d-axis proportional output Vdp with the upper limit value (Vlimit) and the lower limit value (-Vlimit), and outputs the limited d-axis proportional output Vdp’ based on the comparison result. However, Vlimit = Kmax·Vdc / 2 0.5 is set. Kmax is the maximum voltage utilization rate of the inverter 12 and is appropriately set according to the desired output. For example, Kmax may be 1.

[0046] The specific operations performed by the limiter 102d are as follows. (A) When Vdp < -Vlimit, Vdp’ = -Vlimit (B) When -Vlimit ≤ Vdp ≤ Vlimit, Vdp’ = Vdp (C) When Vlimit < Vdp, Vdp’ = Vlimit

[0047] That is, when the d-axis proportional output Vdp exceeds an upper limit value (Vlimit) or falls below a lower limit value (-Vlimit), the limiter 102d limits the value of Vdp and outputs it as the limited d-axis proportional output Vdp'. When the d-axis proportional output Vdp is equal to or greater than the lower limit value and equal to or less than the upper limit value, the limiter 102d outputs the value of Vdp as is as the limited d-axis proportional output Vdp'. The value of the limited d-axis proportional output Vdp' output by the limiter 102d is input to the limiter 110d and the adder 111d.

[0048] The d-axis integral amplifier 103d calculates its own axis integral input Aid by multiplying the d-axis current deviation ed by Kid. That is, Aid=ed×Kid. "Kid" is an integral gain by which the steady-state value of the d-axis current deviation ed is made zero. For example, Kid=ωcc×R, where R is the winding resistance value of the rotating machine 10. However, the value of Kid is not limited to ωcc×R, and may be adjusted as appropriate based on actual measurement results, etc. The own axis integral input Aid calculated by the d-axis integral amplifier 103d is input to a subtractor 108d.

[0049] The high-pass filter 104d reduces the low-frequency components of the q-axis current deviation eq and outputs the result to a response angular frequency amplifier 105d. The response angular frequency amplifier 105d multiplies the q-axis current deviation eq, the low-frequency components of which have been reduced by the high-pass filter 104d, by ωcc, and outputs the multiplied value to the multiplier 107d.

[0050] The inductance amplifier 106d multiplies the rotational angular velocity ω by Lq and outputs the multiplied value to the multiplier 107d. Here, “Lq” is the value of the q-axis inductance of the rotating machine 10. The multiplier 107d obtains the other axis integral input Bid by multiplying the output of the response angular frequency amplifier 105d by the output of the inductance amplifier 106d. The multiplier 107d outputs the other axis integral input Bid to the subtractor 108d.

[0051] The subtractor 108d obtains the d-axis integrated input Cid by subtracting the self-axis integrated input Aid from the other-axis integrated input Bid. The subtractor 108d outputs the d-axis integrated input Cid to the integrator 109d. The integrator 109d performs an integration operation on the d-axis integrated input Cid and outputs the result to the limiter 110d as the d-axis integrated output Vdi.

[0052] The limiter 110d compares the d-axis integrated output Vdi with the upper limit value (Vlimit - Vdp’) and the lower limit value (-Vlimit - Vdp’), and outputs the limited d-axis integrated output Vdi’ based on the comparison result. The specific operations performed by the limiter 110d are as follows. (D) When Vdi < -Vlimit - Vdp’, Vdi’ = -Vlimit - Vdp’ (E) When -Vlimit - Vdp’ ≤ Vdi ≤ Vlimit - Vdp’, Vdi’ = Vdi (F) When Vlimit - Vdp’ < Vdi, Vdi’ = Vlimit - Vdp’

[0053] That is, when the d-axis integrated output Vdi exceeds the upper limit value (Vlimit - Vdp’) or is less than the lower limit value (-Vlimit - Vdp’), the limiter 110d limits the value of Vdi and outputs it as the limited d-axis integrated output Vdi’. Also, when the d-axis integrated output Vdi is greater than or equal to the lower limit value and less than or equal to the upper limit value, the limiter 110d outputs the value of Vdi as the limited d-axis integrated output Vdi’ as it is. The value of the limited d-axis integrated output Vdi’ output by the limiter 110d is input to the adder 111d.

[0054] Here, the reason why the limiter 110d performs the limit determination using the limited d-axis proportional output Vdp’ is to prevent the excessive accumulation of the limited d-axis integrated output Vdi’ and obtain the anti-windup effect. The adder 111d adds the limited d-axis proportional output Vdp’ and the limited d-axis integrated output Vdi’ to obtain the d-axis voltage command value vd. The obtained voltage command value vd is input to the coordinate converter 25 as shown in FIG. 1.

[0055] Next, the q-axis voltage command value calculator 24q will be described with reference to FIG. 4. The q-axis voltage command value calculator 24q includes a q-axis proportional amplifier 101q, a limiter 102q, a q-axis integral amplifier 103q, a high-pass filter (HPF) 104q, a response angular frequency amplifier 105q, an inductance amplifier 106q, a multiplier 107q, an adder 108q, an integrator 109q, a limiter 110q, and an adder 111q.

[0056] The q-axis proportional amplifier 101q calculates a q-axis proportional output Vqp by multiplying the q-axis current deviation eq by Kpq. That is, Vqp = eq × Kpq. "Kpq" is the q-axis proportional gain that is multiplied to obtain a preferable response of the rotational machine current actually flowing with respect to the current command value iq_ref. For example, Kpq = ωcc × Lq. Here, as described above, ωcc is the response angular frequency and Lq is the q-axis inductance. However, the value of Kpq is not limited to ωcc × Lq, and may be appropriately adjusted by actually measuring the responsiveness of the rotational machine current actually flowing with respect to the current command value iq_ref or the like. The q-axis proportional output Vqp calculated by the q-axis proportional amplifier 101q is input to the limiter 102q.

[0057] The limiter 102q compares the q-axis proportional output Vqp with an upper limit value (Vlimit) and a lower limit value (-Vlimit), and outputs a limited q-axis proportional output Vqp' based on the comparison result. The specific calculation performed by the limiter 102q is as follows. (G) When Vqp < -Vlimit, Vqp' = -Vlimit (H) When -Vlimit ≤ Vqp ≤ Vlimit, Vqp' = Vqp (I) When Vlimit < Vqp, Vqp' = Vlimit

[0058] That is, when the q-axis proportional output Vqp exceeds the upper limit (Vlimit) or falls below the lower limit (-Vlimit), the limiter 102q limits the value of Vqp and outputs it as the limited q-axis proportional output Vqp'. When the q-axis proportional output Vqp is equal to or greater than the lower limit and equal to or less than the upper limit, the limiter 102q outputs the value of Vqp as is as the limited q-axis proportional output Vqp'. The value of the limited q-axis proportional output Vqp' output by the limiter 102q is input to the limiter 110q and the adder 111q.

[0059] The q-axis integral amplifier 103q calculates its own axis integral input Aiq by multiplying the q-axis current deviation eq by Kiq. That is, Aiq=eq×Kiq. Here, Kiq is an integral gain by which the steady-state value of the q-axis current deviation eq is multiplied to zero. For example, Kiq=ωcc×R. However, the value of Kiq is not limited to ωcc×R, and may be adjusted appropriately based on actual measurement results, etc. The own axis integral input Aiq calculated by the q-axis integral amplifier 103q is input to an adder 108q.

[0060] The high-pass filter 104q reduces the low-frequency components of the d-axis current deviation ed and outputs the result to a response angular frequency amplifier 105q. The response angular frequency amplifier 105q multiplies the d-axis current deviation ed, the low-frequency components of which have been reduced by the high-pass filter 104q, by ωcc, and outputs the result to the multiplier 107q.

[0061] The inductance amplifier 106q multiplies the rotational angular velocity ω by Ld and outputs the result to the multiplier 107d. The multiplier 107q multiplies the output of the response angular frequency amplifier 105d by the output of the inductance amplifier 106d to obtain the other axis integral input Biq. The multiplier 107q outputs the other axis integral input Biq to the adder 108q.

[0062] The adder 108q calculates the q-axis integral input Ciq by adding the self-axis integral input Aiq and the other-axis integral input Biq. The adder 108q outputs the q-axis integral input Ciq to the integrator 109q. The integrator 109q performs an integration operation on the q-axis integration input Ciq and outputs the result as the q-axis integration output Vqi to the limiter 110q.

[0063] The limiter 110q compares the q-axis integration output Vqi with the upper limit value (Vlimit - Vqp’) and the lower limit value (-Vlimit - Vqp’), and outputs the limited q-axis integration output Vqi’ based on the comparison result. The specific operations performed by the limiter 110q are as follows. (J) When Vqi < -Vlimit - Vqp’, Vqi’ = -Vlimit - Vqp’ (K) When -Vlimit - Vqp’ ≤ Vdi ≤ Vlimit - Vqp’, Vqi’ = Vqi (L) When Vlimit - Vqp’ < Vqi, Vqi’ = Vlimit - Vqp’

[0064] That is, when the q-axis integration output Vqi exceeds the upper limit value (Vlimit - Vqp’) or is less than the lower limit value (-Vlimit - Vqp’), the limiter 110q limits the value of Vqi and outputs it as the limited q-axis integration output Vqi’. Also, when the q-axis integration output Vqi is greater than or equal to the lower limit value and less than or equal to the upper limit value, the limiter 110q outputs the value of Vqi as the limited q-axis integration output Vqi’ as it is. The value of the limited q-axis integration output Vqi’ output by the limiter 110q is input to the adder 111q.

[0065] Here, the reason why the limiter 110q performs the limit determination using the limited q-axis proportional output Vqp’ is to prevent the excessive accumulation of the limited q-axis integration output Vqi’ and obtain the anti-windup effect. The adder 111q adds the limited q-axis proportional output Vqp’ and the limited q-axis integration output Vqi’ to obtain the voltage command value vq for the q-axis. The obtained voltage command value vq is input to the coordinate converter 25.

[0066] 1, voltage command values ​​vd, vq on two rotation axes (d-axis, q-axis) and a rotor position θ are input to the coordinate converter 25. The coordinate converter 25 performs coordinate conversion of the voltage command values ​​vd, vq based on the rotor position θ, and calculates voltage command values ​​vu, vv, vw on the three-phase coordinate system.

[0067] The modified voltage generator 26 generates modified voltage command values ​​vu', vv', vw' based on the voltage command values ​​vu, vv, vw output from the coordinate converter 25 and the offset voltage voffset. Fig. 5 is a flowchart showing the processing performed by the modified voltage generator 26 in the first embodiment.

[0068] 5 is started, first, step S11 is executed. In step S11, the corrected voltage generator 26 selects the smallest value from among the voltage command values ​​vu, vv, and vw, and sets the selected value as Vmin. Next, in step S12, the correction voltage generator 26 sets the value of the offset voltage voffset to 0.5 Vdc-Vmin. Next, in step S13, the corrected voltage generator 26 obtains corrected voltage command values ​​vu', vv', and vw' by subtracting the offset voltage voffset from the voltage command values ​​vu, vv, and vw, respectively.

[0069] FIG. 6 is a diagram showing an example of waveforms of the voltage command values ​​vu, vv, vw and the corrected voltage command values ​​vu′, vv′, vw′ in the first embodiment. In FIG. 6, the waveform graphs of the voltage command values ​​vu, vv, vw are shown on the upper side, and the waveform graphs of the corrected voltage command values ​​vu′, vv′, vw′ are shown on the lower side. However, in each graph shown in FIG. 6, the DC bus voltage Vdc=10V. Here, the voltage range that the inverter 12 can output is 0 (minimum value of the carrier triangular wave C) to Vdc (maximum value of the carrier triangular wave C). Therefore, when the DC bus voltage Vdc=10V, the voltage range that the inverter 12 can output is 0V to 10V as shown in FIG. 6.

[0070] By performing the process in steps S11 to S13 shown in Fig. 5, the minimum value of the corrected voltage command values ​​vu', vv', vw' always coincides with the inverter output lower limit value, as shown in the lower graph of Fig. 6. This means that the corrected voltage command values ​​vu', vv', vw' are offset downward as equally as possible for all three phases without causing voltage saturation, and the on-times of the lower-arm switching elements Sun, Svn, Swn are made as long as possible. This is advantageous from the viewpoint of current detection in an inverter having shunt resistors Ru, Rv, Rw in series with the lower-arm switching elements Sun, Svn, Swn as in this embodiment.

[0071] The calculation method of the offset voltage voffset is not limited to the above-mentioned method. For example, in step S11 of FIG. 5, in addition to the process of selecting the smallest one of the voltage command values ​​vu, vv, and vw and setting it to Vmin, the process of selecting the largest one of the voltage command values ​​vu, vv, and vw and setting it to Vmax may be performed. Then, in step S12, the process of setting voffset to "(Vmax+Vmin) / 2" may be performed. In this way, the method of setting voffset=(Vmax+Vmin) / 2 is called third harmonic addition or HIP modulation. Also, so-called two-phase modulation may be adopted in which the upper arm switching element corresponding to the phase with the largest voltage command value among the three phases is always in the on state.

[0072] The modified voltage command values ​​vu', vv', vw' generated by the modified voltage generator 26 through the above calculations are input to the PWM signal generator 27. Then, the PWM signal generator 27 performs PWM modulation on the modified voltage command values ​​vu', vv', vw', and outputs the result to the inverter 12 as switching signals Gup to Gwn.

[0073] Fig. 7 is a Bode plot showing the transfer characteristics from the q-axis current command value iq_ref to the q-axis detected current value iq when the rotational angular velocity ω is high in the control device 1 according to the first embodiment. In Fig. 7, the upper part shows a gain diagram, and the lower part shows a phase diagram. In both the gain diagram and the phase diagram, "the present disclosure" shows the transfer characteristics according to the first embodiment, and "other axis integral input (Bid, Biq) = 0" is the characteristic when 0 (zero) is forcibly input as the values ​​of Bid and Biq (see Figs. 3 and 4).

[0074] As shown in the gain diagram of Fig. 7, "other axis integral input (Bid, Biq) = 0" has a drop in gain near 100 Hz. In contrast, in "the present disclosure," there is no drop near 100 Hz, and an ideal first-order lag transfer characteristic is obtained. In other words, according to the control device 1 according to the present disclosure, it is possible to obtain favorable control results even when the rotational angular velocity ω is high.

[0075] In addition, the number of integrators included in the d-axis voltage command value calculator 24d and the q-axis voltage command value calculator 24q in the present disclosure is one each. Therefore, as compared with the case of using two integrators per axis as in Patent Document 1, it is possible to easily perform anti-windup processing. As a result, it is possible to provide a stable control device 1 in which overshooting due to excessive accumulation of integral values ​​is unlikely to occur even when there is a sudden change in the current command values ​​id_ref and iq_ref. In addition, as compared with the case of using two integrators per axis, according to the present disclosure, it is also possible to suppress the amount of calculation of the controller 13.

[0076] As described above, the control device 1 according to the first embodiment includes the inverter 12 that applies a voltage to the AC rotating machine 10, the current detector 22 that detects the rotating machine current flowing through the rotating machine 10, the d-axis voltage command value calculator 24d (first axis voltage command value calculator) that calculates a voltage command value vd of the d-axis (first axis) of the two rotating axes of the rotating machine 10, and the q-axis voltage command value calculator 24q (second axis voltage command value calculator) that calculates a voltage command value vq of the q-axis (second axis) of the two rotating axes. The d-axis voltage command value calculator 24d performs an integral calculation using a d-axis current deviation ed (first deviation) which is the deviation between a current command value id_ref on the d-axis and a detected current value id on the d-axis, and a q-axis current deviation eq (second deviation) which is the deviation between a current command value iq_ref on the q-axis and a q-axis detected current value iq on the q-axis, and calculates a voltage command value vd of the d-axis based on the result of the integral calculation. Further, the q-axis voltage command value calculator 24q performs an integral calculation using the q-axis current deviation eq and the d-axis current deviation ed, and calculates a q-axis voltage command value vq based on the result of the integral calculation.

[0077] This configuration makes it possible to reduce the number of integrators required to calculate the voltage command values ​​vd and vq, and as a result, makes it easier to perform anti-windup. This makes it possible to provide a control device 1 that is less likely to cause overshoots and is capable of more stable control.

[0078] Moreover, the d-axis voltage command value calculator 24d performs the integration calculation on the q-axis current deviation eq by using a value obtained by multiplying the rotational angular velocity ω of the AC rotating machine, the q-axis inductance Lq, and the response angular frequency ωcc for adjusting the frequency response of the rotating machine current to the current command value. This makes it possible to obtain a desired frequency response independent of the rotational angular velocity ω of the rotating machine. Similarly, the q-axis voltage command value calculator 24q performs the integration calculation on the d-axis current deviation ed by using a value obtained by multiplying the rotational angular velocity ω of the AC rotating machine, the inductance Ld on the d-axis, and the response angular frequency ωcc for adjusting the frequency response of the rotating machine current to the current command value.

[0079] Moreover, the d-axis voltage command value calculator 24d performs the integral calculation using the result of reducing the low-frequency components of the q-axis current deviation eq by the high-pass filter 104d. In this way, by reducing the low-frequency components of the current deviation eq of the axis (i.e., the q-axis) that is not the control target of the d-axis voltage command value calculator 24d, it is possible to steadily set the current deviation ed of the axis (i.e., the d-axis) that is the control target to 0 (zero). Similarly, the q-axis voltage command value calculator 24q performs the integral calculation using the result of reducing the low-frequency components of the d-axis current deviation ed by the high-pass filter 104q.

[0080] Embodiment 2 Next, a second embodiment of the present disclosure will be described, but the basic configuration is the same as that of the first embodiment. Therefore, the description of the parts that overlap with the first embodiment will be omitted, and the description will focus on the differences. Fig. 8 is a block diagram showing a schematic configuration of a control device 2 for a rotating machine according to the second embodiment. As shown in Fig. 8, the control device 2 differs from the control device 1 according to the first embodiment in that it includes an amplitude calculator 29. Figs. 9 and 10 are block diagrams showing configurations of a d-axis voltage command value calculator 24d and a q-axis voltage command value calculator 24q according to the second embodiment, respectively. The configurations of the d-axis voltage command value calculator 24d and the q-axis voltage command value calculator 24q also differ between the first embodiment and the second embodiment.

[0081] 8, the d-axis voltage command value vd, the q-axis voltage command value vq, and the DC bus voltage Vdc are input to the amplitude calculator 29. The amplitude calculator 29 calculates a voltage amplitude m based on the following formula (2-1). The voltage amplitude m is a numerical value obtained by standardizing the magnitudes of the d-axis voltage command value vd and the q-axis voltage command value vq using the DC bus voltage Vdc of the inverter 12. m = {(vd 2 +vq 2 ) / (Vdc / 2)} 0.5 …(2-1)

[0082] As shown in FIG. 9, the d-axis voltage command value calculator 24d includes an amplifier 112d. The amplifier 112d multiplies the output from the multiplier 107d by K and inputs the result to a subtractor 108d. "K" is a limiting gain that is multiplied when integrating and inputting the current deviation eq in the q-axis that is not the control target as viewed from the d-axis voltage command value calculator 24d. A voltage amplitude m is input to the amplifier 112d, and the value of the limiting gain K is determined based on the value of the voltage amplitude m. In the d-axis voltage command value calculator 24d according to this embodiment, the voltage amplitude m is also input to the d-axis proportional amplifier 101d. This embodiment differs from the first embodiment in that the d-axis proportional gain Kpd and the limiting gain K are changed according to the voltage amplitude m.

[0083] As shown in FIG. 10, the q-axis voltage command value calculator 24q of the second embodiment includes an amplifier 112q. The amplifier 112q multiplies the output from the multiplier 107q by K and inputs the result to the adder 108q. "K" is a limiting gain by which the current deviation ed in the d-axis, which is not the control target, is multiplied when it is integrated and input from the viewpoint of the q-axis voltage command value calculator 24q. A voltage amplitude m is input to the amplifier 112q, and the value of the limiting gain K is determined based on the value of the voltage amplitude m. In the q-axis voltage command value calculator 24q according to the present embodiment, the voltage amplitude m is also input to the q-axis proportional amplifier 101q. This embodiment differs from the first embodiment in that the q-axis proportional gain Kpq and the limiting gain K are changed according to the voltage amplitude m.

[0084] Fig. 11 shows the relationship between the voltage amplitude m and the proportional gains Kpd, Kpq, and the limiting gain K. The vertical axis of the upper graph in Fig. 11 is the proportional gains Kpd, Kpq, and the vertical axis of the lower graph is the limiting gain K. In both the upper and lower graphs, the horizontal axis is the voltage amplitude m. In Fig. 11, m1 is the first voltage threshold, and m2 is the second voltage threshold. The proportional gains Kpd and Kpq are switched between a first proportional value Kp_H and a second proportional value Kp_L. The second proportional value Kp_L is smaller than the first proportional value Kp_H. The limited gain K is switched between a first limited value K_H and a second limited value K_L. The second limited value K_L is smaller than the first limited value K_H.

[0085] 11, when the voltage amplitude m is higher than the first voltage threshold m1, the values ​​of the proportional gains Kpd and Kpq are set to the second proportional value Kp_L, and when the voltage amplitude m is lower than the first voltage threshold m1, the proportional gains Kpd and Kpq are set to the first proportional value Kp_H. When the voltage amplitude m is higher than the second voltage threshold m2, the limiting gain K is set to the first limiting value K_H. When the voltage amplitude m is lower than the second voltage threshold m2, the limiting gain K is set to the second limiting value K_L. Here, the first voltage threshold m1 is higher than the second voltage threshold m2. Therefore, when the values ​​of the proportional gains Kpd and Kpq are Kp_L, the value of the limiting gain K is K_H.

[0086] In this way, only when the limiting gain K is a high value K_H (i.e., the decoupling control between the d and q axes is sufficiently effective), the proportional gains Kpd and Kpq are reduced, so that the stability of the control can be ensured even when the proportional gains Kpd and Kpq are low. When the limiting gain K is a low value K_L, the proportional gains Kpd and Kpq are set to a high value Kp_H that can ensure the stability of the control even when the decoupling control between the d and q axes is not sufficient. The values ​​of Kp_H and Kp_L are specifically set so that the response from the current command values ​​id_ref and iq_ref to the rotating machine currents iu, iv, and iw is good within the ranges of 300 Hz to 1000 Hz and 100 Hz to 300 Hz, respectively. For example, the value of K_H is set to 0.7 or more, and the value of K_L is set to 0.3 or less.

[0087] Next, a preferred setting of the first voltage threshold m1 will be described. Gxn shown in FIG. 12 shows an example of the waveform of one of the switching signals Gun, Gvn, and Gwn in the lower arm switching elements Sun, Svn, and Swn. Also, VRx shows an example of the waveform of one of the voltages VRu, VRv, and VRw across the shunt resistors Ru, Rv, and Rw. In the example shown in FIG. 12, after the signal Gxn changes from 0 to 1, ringing occurs in the voltage VRx across the shunt resistor for several μs. Ringing is a phenomenon in which the voltage across the shunt resistor fluctuates for a certain period of time when switching is performed in the inverter 12. If the current detector 22 obtains the pre-conversion detection currents ius, ivs, and iws based on the voltages VRu, VRv, and VRw across the shunt resistor that include this ringing, an error will be included in the detection result. If the pre-conversion detection currents ius, ivs, and iws include an error, the detection current values ​​id and iq after coordinate conversion will also include an error.

[0088] In order to accurately obtain the pre-conversion detection currents ius, ivs, and iws, it is preferable that the on-time of the corresponding lower-arm switching elements Sun, Svn, and Swn is longer than the time threshold value Tmin set according to the convergence time of the ringing. In order to make the on-time of the lower-arm switching elements Sun, Svn, and Swn longer than the time threshold value Tmin, it is preferable that the voltage input to the PWM signal generator 27 is equal to or less than Vdc×(Tc-Tmin) / Tc shown by the dashed line in FIG. 2.

[0089] In FIG. 2, the corrected voltage commands vu', vv', vw' (i.e., the voltages input to the PWM signal generator 27) are all equal to or less than Vdc×(Tc-Tmin) / Tc. Therefore, it is possible to eliminate errors due to ringing and accurately obtain the pre-conversion detection currents ius, ivs, iws based on the voltages VRu, VRv, VRw across the shunt resistors. This detection method for determining the pre-conversion detection currents ius, ivs, iws based on the voltages VRu, VRv, VRw across the shunt resistors corresponding to the three phases is called "three-phase detection."

[0090] However, when the voltage amplitude m increases due to an increase in the rotation speed of the rotating machine 10, some of the corrected voltage commands vu', vv', vw' may take large values ​​close to the maximum value Vdc of the carrier triangular wave C. In the example of FIG. 13, the value of the corrected voltage command vu' is larger than the value of Vdc×(Tc-Tmin) / Tc (hereinafter also referred to as the "upper limit value"). In this case, the difference between the time when Gun switches from 0 to 1 and the time at timing X becomes small. Therefore, the voltage VRu across the u-phase shunt resistor acquired at timing X includes the effect of ringing.

[0091] Therefore, the pre-conversion detection current for the phase in which the on-time of the switching signals Gun, Gvn, and Gwn of the lower-arm switching elements is shorter than the time threshold Tmin may be generated from the other two phases. A detection method in which the pre-conversion detection current of one of the three phases is obtained based on the pre-conversion detection currents of the remaining two phases in this manner is called "two-phase detection." For example, when the on-time of the switching signal Gun is shorter than the time threshold Tmin, the pre-conversion detection current ius of the U phase may be calculated by ius = -ivs -iws. Similarly, when the on-time of the switching signal Gvn is shorter than the time threshold Tmin, the calculation may be made as ivs = -ius -iws, and when the on-time of the switching signal Gwn is shorter than the time threshold Tmin, the calculation may be made as iws = -ius -ivs.

[0092] The relationship between the number of phases used for current detection and the voltage amplitude m is as follows. That is, when the voltage amplitude m is low and the instantaneous values ​​of the three-phase corrected voltage commands vu', vv', and vw' are all equal to or lower than the upper limit value (Vdc×(Tc-Tmin) / Tc), "three-phase detection" is used. Alternatively, when the voltage amplitude m is high and the instantaneous value of any of the three-phase corrected voltage commands vu', vv', and vw' is equal to or higher than the upper limit value (Vdc×(Tc-Tmin) / Tc), "two-phase detection" is used. Here, when "three-phase detection" is compared with "two-phase detection," "three-phase detection" has better accuracy. Therefore, in order to improve the control accuracy of the rotating machine 10, it is preferable to use "three-phase detection" as much as possible. However, from the viewpoint of increasing the output of the rotating machine 10, there are cases where it is preferable to increase the voltage amplitude m and use "two-phase detection."

[0093] Therefore, in this embodiment, the value of the first voltage threshold m1 is set to "(Tc-Tmin) / Tc". It can be said that "(Tc-Tmin) / Tc" is a numerical value obtained by dividing the above-mentioned upper limit value "Vdc×(Tc-Tmin) / Tc" by Vdc and normalizing it. By setting the first voltage threshold m1 in this manner, the following effects can be obtained. When the voltage amplitude m is larger than the first voltage threshold m1, "two-phase detection" is used, and the proportional gains Kpd and Kpq are set to a low value Kp_L at that time. This reduces the influence of a decrease in the accuracy of current detection, and prevents an increase in vibration and noise generated by the rotating machine 10. At the same time, the limit gain K is set to a high value K_H, so that non-interference control is effective and control stability is maintained. Also, when the voltage amplitude m is smaller than the first voltage threshold m1, "three-phase detection" is used, and the proportional gains Kpd and Kpq are set to a large value Kp_H at that time (see FIG. 11). Since the current detection accuracy is good in three-phase detection, the dependency on the decoupling control is reduced, and the vibration and noise generated in the rotating machine 10 due to the noise contained in the rotation angular velocity ω can be reduced.

[0094] The second voltage threshold m2 may be set to a value lower than the first voltage threshold m1. However, if noise contained in the detected value of the rotational angular velocity ω does not pose a problem, it is not necessarily necessary to switch the value of the limiting gain K at the second voltage threshold m2. Specifically, if the voltage amplitude m is equal to or greater than the first voltage threshold m1, the values ​​of the proportional gains Kpd, Kpq may be set to Kp_L, and if not, the values ​​of the proportional gains Kpd, Kpq may be set to Kp_H, and the value of the limiting gain K may be set to a constant (K_H) regardless of the voltage amplitude m.

[0095] In addition, including the AC rotating machine 10 according to the present embodiment, the voltage applied to the rotating machine and the rotation speed (or rotation angular velocity ω) of the rotating machine are generally in a proportional relationship. Therefore, the proportional gains Kpd, Kpq and the limiting gain K may be switched using a threshold related to the rotation speed (or rotation angular velocity ω) instead of the threshold related to the voltage amplitude m described above. For example, the rotation speed when the voltage amplitude m matches the first voltage threshold m1 may be set as the first rotation speed threshold n1, and the rotation speed when the voltage amplitude m matches the second voltage threshold m2 may be set as the second rotation speed threshold n2. In this case, the switching of the proportional gains Kpd, Kpq or the limiting gain K based on the comparison of the voltage amplitude m with the thresholds m1 and m2 described above can be replaced with switching based on the comparison of the rotation speed with the thresholds n1 and n2. Similarly, the rotation angular velocity ω when the voltage amplitude m matches the first voltage threshold m1 may be set as the first rotation angular velocity threshold ω1, and the rotation angular velocity ω when the voltage amplitude m matches the second voltage threshold m2 may be set as the second rotation angular velocity threshold ω2. In this case, the switching of the proportional gains Kpd, Kpq or the limiting gain K based on the comparison of the voltage amplitude m with the thresholds m1, m2 described above can be replaced with switching based on a comparison of the rotational angular velocity ω with ω1, ω2.

[0096] Alternatively, switching according to the voltage amplitude m and switching according to the rotation speed (or rotation angular velocity ω) may be used in combination. As a specific example, the proportional gains Kpd, Kpq may be switched based on a comparison between the voltage amplitude m and a first voltage threshold m1, and the limit gain K may be switched based on a comparison between the rotation speed (or rotation angular velocity ω) and a second rotation speed threshold n2 (or a second rotation angular velocity threshold ω2).

[0097] In the above explanation, the values ​​of the proportional gains Kpd, Kpq are switched based on the comparison between the voltage amplitude m and the first voltage threshold m1, or the comparison between the rotation speed and the first rotation speed threshold n1, or the comparison between the rotation angular velocity ω and the first rotation angular velocity threshold ω1. However, for example, when the rotating machine current becomes excessive and exceeds the threshold, the values ​​of the proportional gains Kpd, Kpq may be constant at Kp_H regardless of the voltage amplitude m or the rotation speed. When the rotating machine current becomes excessive, the state in which the rotating machine current becomes excessive can be shortened by setting the proportional gains Kpd, Kpq to large values. This makes it possible to prevent failure of the control device for the rotating machine 10.

[0098] In addition, when a failure occurs in part of the control device of the rotating machine 10 and operation continues using the remaining non-faulty parts, the proportional gains Kpd and Kpq may be set to always large values ​​(Kp_H) to increase the safety of the system. Furthermore, when the values ​​of the proportional gains Kpd and Kpq are switched between Kp_L and Kp_H, the switching may be performed gradually with a slope so as to prevent discontinuity in the control. Similarly, when the value of the limit gain K is switched between K_L and K_H, the switching may be performed gradually with a slope.

[0099] As described above, in the second embodiment, the first axis voltage command value calculator (e.g., the d-axis voltage command value calculator 24d) calculates the voltage command value (e.g., Vd) of the first axis by using a value obtained by multiplying the first deviation (e.g., the d-axis current deviation ed) by the proportional gain (e.g., Kpd). The proportional gain is switched between a first proportional value Kp_H and a second proportional value Kp_L smaller than the first proportional value Kp_H. When the voltage amplitude m calculated based on the voltage command value is equal to or larger than the first voltage threshold value m1, or when the rotation speed of the AC rotating machine 10 is equal to or larger than the first rotation speed threshold value n1, or when the rotation angular velocity ω of the AC rotating machine 10 is equal to or larger than the first rotation angular velocity threshold value ω1, the proportional gain is set to the second proportional value Kp_L. According to this configuration, when the rotation angular velocity ω is large, the proportional gain is set to a small value (Kp_L), thereby making it possible to reduce high-frequency noise components included in the first deviation, and to provide a control device 2 for the AC rotating machine 10 with low noise.

[0100] In the second embodiment, the first axis voltage command value calculator performs an integral calculation using a value obtained by multiplying the second deviation by the limit gain K, and the limit gain K is switched between a first limit value K_H and a second limit value K_L smaller than the first limit value K_H. When the voltage amplitude m is equal to or smaller than a second voltage threshold m2 smaller than a first voltage threshold m1, or when the rotation speed of the AC rotating machine 10 is equal to or smaller than a second rotation speed threshold n2 smaller than a first rotation speed threshold n1, or when the rotation angular velocity ω of the AC rotating machine 10 is equal to or smaller than a second rotation angular velocity threshold ω2 smaller than a first rotation angular velocity threshold ω1, the limit gain K is set to the second limit value K_L. According to this configuration, when the rotation angular velocity ω is small, the limit gain is set to a small value (K_L), thereby limiting the input of the second deviation to the integral calculation. Therefore, the high frequency noise components included in the second deviation can be reduced, and a low noise control device 2 for the AC rotating machine 10 can be provided.

[0101] The inverter 12 also includes three sets of upper arm switching elements Sup, Svp, Swp, lower arm switching elements Sun, Svn, Swn, and shunt resistors Ru, Rv, Rw, which correspond to the three phases (U, V, W). When the voltage amplitude m is smaller than the first voltage threshold m1, the current detector 22 detects the rotating machine currents iu, iv, iw in the three phases based on the voltages VRu, VRv, VRw across the corresponding shunt resistors (i.e., uses "three-phase detection"). This configuration makes it possible to suppress errors due to the influence of ringing from being included in the detection results of the rotating machine currents iu, iv, iw. Therefore, the AC rotating machine 10 can be controlled with high accuracy.

[0102] Furthermore, when the rotating machine current exceeds a threshold value (i.e., is excessive), the proportional gains Kpd and Kpq may be set to the first proportional value Kp_H regardless of the voltage amplitude m, etc. In this case, the time during which the rotating machine current remains excessive can be shortened. Therefore, a breakdown in the control device 2 can be prevented.

[0103] Furthermore, when a failure occurs at at least one location in the control device 2, the proportional gains Kpd, Kpq may be set to the first proportional value Kp_H regardless of the voltage amplitude m, etc. In this case, it is possible to improve the stability of the system when a failure occurs and suppress the occurrence of a secondary failure.

[0104] Embodiment 3 Next, a description will be given of a third embodiment. In this embodiment, a case will be described in which the technique described in the first or second embodiment is applied to the control of a rotating machine included in an electric power steering device. As shown in Fig. 14, an electric power steering device 100 according to this embodiment includes a control device 3, a steering wheel 101, a rotating machine 10, and a torque detector 103. The electric power steering device 100 is mounted on a vehicle. The steering wheel 101 is operated by a driver. Front wheels 102 of the vehicle are driven by operating the steering wheel 101. The basic configuration of the control device 3 is similar to that of the control device 1 in the first embodiment, so a detailed description will be omitted and differences will be mainly described.

[0105] The torque detector 103 detects the steering torque Ts of the steering wheel 101 by the driver and outputs the detection result to the control device 3. The driving force of the rotating machine 10 is transmitted to a steering system 100s of the vehicle via a driving force transmission mechanism 104. The steering system 100s includes the steering wheel 101, front wheels 102, etc. The electric power steering device 100 uses the driving force generated by the rotating machine 10 as an assist torque to assist the driver in steering the vehicle.

[0106] The current command value calculator 21 in the control device 3 differs from that in the first embodiment in the calculation of the torque current command value (q-axis current command value iq_ref), and this point will be described below. The steering torque Ts and the vehicle running speed S are input to the current command value calculator 21, and the q-axis current command value iq_ref is calculated based on these inputs.

[0107] 15 is a graph showing a set value of the q-axis current command value iq_ref according to the steering torque Ts and the vehicle speed S. As shown in this graph, the greater the steering torque Ts, the greater the value of the current command value iq_ref, and the greater the gradient of the change. Also, the greater the vehicle speed S, the smaller the value of the current command value iq_ref. The value of the current command value iq_ref may be determined by further taking into account a damping torque calculated based on the rotational angular velocity ω, etc.

[0108] 15, the gradient of the q-axis current command value iq_ref with respect to the steering torque Ts increases as the vehicle travel speed S decreases. As a result, when detection noise is included in the steering torque Ts, the effect on the q-axis current command value iq_ref increases, leading to the generation of vibrations and noise from the rotating machine 10, or an unpleasant feel to the steering wheel 101.

[0109] Therefore, in the voltage command value calculator 24 in the third embodiment, the proportional gain Kpd in the d-axis voltage command value calculator 24d and the proportional gain Kpq in the q-axis voltage command value calculator 24q are varied according to the vehicle running speed S, as shown in Fig. 16. "Kp_H1" is referred to as the first speed reference value, and "Kp_L1" is referred to as the second speed reference value. The second speed reference value Kp_L1 is smaller than the first speed reference value Kp_H1.

[0110] When the vehicle speed S is equal to or lower than s1, the values ​​of the proportional gains Kpd and Kpq are set to Kp_L1. In this way, by setting the values ​​of the proportional gains Kpd and Kpq low, it is possible to suppress the q-axis current command value iq_ref from reacting sensitively to the pulsation of the steering torque Ts. Also, when the vehicle speed S is equal to or higher than s2, the proportional gains Kpd and Kpq are set to a high value Kp_H1 because the q-axis current command value iq_ref is less affected by the pulsation of the steering torque Ts. In this way, by switching the proportional gains Kpd and Kpq according to the vehicle speed S, it is possible to provide an electric power steering device that suppresses the generation of vibrations and abnormal sounds of the rotating machine 10 caused by noise contained in the steering torque detection value and the discomfort felt by the driver from the steering wheel 101.

[0111] As shown in Fig. 16, when the vehicle speed S is within the range of s1 to s2, the values ​​of the proportional gains Kpd and Kpq are changed continuously within the range of Kp_H1 to Kp_L1. This makes it possible to suppress discomfort felt by the driver due to a sudden change in the set values ​​of the proportional gains Kpd and Kpq. The proportional gains Kpd and Kpq may be switched in a stepwise manner between Kp_H1 and Kp_L1.

[0112] As described above, the electric power steering device 100 according to this embodiment includes the control device 3, the AC rotating machine 10, and the driving force transmission mechanism 104 that transmits the driving force of the AC rotating machine 10 to the steering system 100s of the vehicle. With this configuration, it is possible to provide an electric power steering device 100 that achieves both quietness and steering stability.

[0113] In the present embodiment, the proportional gains Kpd, Kpq are switched between a first speed reference value Kp_H1 and a second speed reference value Kp_L1 smaller than the first speed reference value Kp_H1. When the vehicle travel speed S is smaller than a threshold value s2, the voltage command value calculation unit 24 sets the proportional gains Kpd, Kpq to values ​​lower than the first speed reference value Kp_H1. This configuration can improve quietness in a situation where the vehicle speed is low and steering noise is likely to resonate.

[0114] Fourth embodiment Next, a fourth embodiment will be described. As shown in Fig. 17, an electric power steering device 100 according to this embodiment includes a control device 3 similar to that of the third embodiment. This embodiment differs from the technology described in the third embodiment in that the detection value input to the voltage command value calculation unit 24 is changed from the vehicle travel speed S to the steering torque Ts, and that some of the calculation contents in the voltage command value calculation unit 24 are different. Other points are the same as those of the third embodiment, and therefore description thereof will be omitted.

[0115] 16 described in the third embodiment, whether the vehicle travel speed S is low or high, the smaller the steering torque Ts is, the smaller the gradient of the q-axis current command value iq_ref with respect to the steering torque Ts is. In other words, the smaller the steering torque Ts is, the more leeway there is in steering stability, and therefore there is room to reduce the limit gain K described in the second embodiment. Therefore, in the voltage command value calculation unit 24 in the fourth embodiment, the limit gain K is varied as shown in FIG.

[0116] In FIG. 18, "ΔTs" is the amount of change in the steering torque Ts per unit time, "K_H1" is the first torque reference value, and "K_L1" is the second torque reference value. The second torque reference value K_L1 is smaller than the first torque reference value K_H1. The limited gain K varies within the range of K_L1 to K_H1 depending on the magnitude of the amount of change ΔTs. Specifically, when ΔTs is equal to or smaller than the first torque threshold value ΔTs1, the value of the limited gain K is set to K_L1. When ΔTs is equal to or larger than the second torque threshold value ΔTs2, the value of the limited gain K is set to K_H1. When ΔTs is within the range of ΔTs1 to ΔTs2, the value of the limited gain K varies continuously within the range of K_L1 to K_H1. This makes it possible to reduce vibration and noise of the rotating machine 10 in a region where the fluctuation of the steering torque Ts is small.

[0117] In FIG. 18, the horizontal axis is ΔTs. However, the same effect can be obtained by using the gradient (Δiq_ref / ΔTs) of the graph in FIG. 15 as the horizontal axis. The gradient (Δiq_ref / ΔTs) is referred to as the "current torque gradient" in this specification. In other words, the current torque gradient is the ratio of the change amount (Δiq_ref) of the q-axis current command value iq_ref to the change amount (ΔTs) of the steering torque Ts. When the current torque gradient is smaller than a threshold value, the proportional gains Kpd and Kpq may be set to a small value Kp_L1. Also, when at least one of the current command values ​​iq_ref and id_ref is smaller than a threshold value or when the steering torque Ts is smaller than a threshold value, the proportional gains Kpd and Kpq may be set to a small value (for example, Kp_L1).

[0118] When the steering torque Ts or the current command values ​​iq_ref, id_ref are small, the steering noise becomes noticeable. Therefore, in the fourth embodiment, a configuration is proposed in which the proportional gains Kpd, Kpq are set to a value lower than the first speed reference value Kp_H1 when the current command values ​​iq_ref, id_ref are equal to or lower than a threshold value, when the gradient (current torque gradient Δiq_ref / ΔTs) in the graph showing the relationship between the steering torque Ts and the torque current command value iq_ref is smaller than a threshold value, or when the change amount ΔTs per unit time of the steering torque Ts is smaller than a first torque threshold value ΔTs1. This makes it possible to weaken the non-interference control and ensure quietness in a situation where the steering noise is noticeable.

[0119] In the third and fourth embodiments, instead of inputting the rotational angular velocity ω to the inductance amplifiers 106d, 106q in accordance with the non-interference control, the result of the rotational angular velocity ω passing through a low-pass filter may be input. The cutoff frequency of the low-pass filter is preferably higher than the upper limit (e.g., 5 Hz) of the steering frequency of the electric power steering device 100. In this case, there is an effect of reducing abnormal noise and vibration of the rotating machine 10 caused by noise components contained in the rotational angular velocity ω, which are generated by introducing the non-interference control. Therefore, a quiet electric power steering device 100 can be provided.

[0120] Although the first to fourth embodiments have been described above, the present disclosure is not limited to the above-mentioned embodiments, and can be freely modified without departing from the spirit of the present disclosure. Also, the first to fourth embodiments described above can be appropriately combined.

[0121] Each of the components of the rotating machine control devices 1-4 and the electric power steering device 100 described above has an internal computer system. A program for implementing the functions of each of the components of the rotating machine control devices 1-4 and the electric power steering device 100 described above may be recorded in a computer-readable recording medium, and the program recorded in the recording medium may be read into a computer system and executed to perform processing in each of the components of the rotating machine control devices 1-4 and the electric power steering device 100 described above. Here, "reading the program recorded in the recording medium into a computer system and executing it" includes installing the program into the computer system. The "computer system" referred to here includes hardware such as an OS and peripheral devices.

[0122] Furthermore, a "computer system" may include multiple computer devices connected via a network including the Internet or communication lines such as a WAN, LAN, or leased line. Furthermore, a "computer-readable recording medium" refers to portable media such as flexible disks, optical magnetic disks, ROMs, and CD-ROMs, and storage devices such as hard disks built into a computer system. In this way, the recording medium storing the program may be a non-transitory recording medium such as a CD-ROM.

[0123] The recording medium also includes a recording medium installed inside or outside the storage medium and accessible from a distribution server to distribute the program. The program may be divided into a plurality of parts, downloaded at different times, and then integrated by the respective components of the rotary machine control devices 1-4 and the electric power steering device 100, and each of the divided programs may be distributed by a different distribution server. Furthermore, the "computer-readable recording medium" includes a storage medium that holds a program for a certain period of time, such as a volatile memory (RAM) in a computer system that becomes a server or a client when a program is transmitted via a network. The program may be a storage medium for implementing part of the above-mentioned functions. Furthermore, the program may be a so-called difference file (difference program) that can realize the above-mentioned functions in combination with a program already recorded in the computer system. [Explanation of symbols]

[0124] Reference Signs List 1 to 4: Control device 10: AC rotating machine 12: Inverter 22: Current detector 24d, 24q: (first axis voltage command value calculator, second axis voltage command value calculator) 100: Electric power steering device 104: Driving force transmission mechanism id, iq: Detected current value id_ref, iq_ref: Current command value iq_ref: Torque current command value iu, iv, iw: Rotating machine current K: Limiting gain K_H: First limiting value K_L: Second limiting value Kp_H: First proportional value Kp_H1: First speed reference value Kp_L: Second proportional value Kp_L1: Second speed reference value Kpd, Kpq: Proportional gain Ld...d-axis inductance m...voltage amplitude m1...first voltage threshold m2...second voltage threshold n1...first rotation speed threshold n2...second rotation speed threshold Ru,Rv,Rw...shunt resistance S...traveling speed 100s...vehicle steering system Sun,Svn,Swn...lower arm switching elements Sup,Svp,Swp...upper arm switching elements Ts...steering torque vd,vq...(voltage command value for the first axis, voltage command value for the second axis) VRu,VRv,VRw...voltage at both ends vu,vv,vw...voltage command value ΔTs...change in steering torque ω...rotational angular velocity ω1...first rotational angular velocity threshold ω2...second rotational angular velocity threshold ωcc...response angular frequency

Claims

1. A control device for an AC rotating machine, comprising: an inverter that applies a voltage to the AC rotating machine; a current detector for detecting a rotating machine current flowing through the AC rotating machine; a first axis voltage command value calculator that calculates a voltage command value of a first axis among two rotating shafts of the AC rotary machine; a second axis voltage command value calculator that calculates a voltage command value of a second axis of the two rotational axes, the first-axis voltage command value calculator performs an integration operation using a first deviation, which is a deviation between a current command value on the first axis and a detected current value of the rotating machine current on the first axis, and a second deviation, which is a deviation between a current command value on the second axis and a detected current value of the rotating machine current on the second axis, and calculates a voltage command value for the first axis based on a result of the integration operation; the first axis voltage command value calculator executes the integral calculation on the second deviation by using a value obtained by multiplying a rotational angular velocity of the AC rotating machine, an inductance in the second axis, and a response angular frequency for adjusting a frequency response of the rotating machine current with respect to the current command value.

2. 2. The control device for an AC rotating machine according to claim 1, wherein the first axis voltage command value calculator executes the integral calculation using a result obtained by reducing a low-frequency component of the second deviation.

3. the first axis voltage command value calculator calculates the voltage command value of the first axis by using a value obtained by multiplying the first deviation by a proportional gain; the proportional gain is switched between a first proportional value and a second proportional value smaller than the first proportional value; 3. The control device for an AC rotating machine according to claim 1, wherein the proportional gain is set to the second proportional value when a voltage amplitude calculated based on the voltage command value is equal to or greater than a first voltage threshold value, when a rotation speed of the AC rotating machine is equal to or greater than a first rotation speed threshold value, or when a rotation angular velocity of the AC rotating machine is equal to or greater than a first rotation angular velocity threshold value.

4. the first axis voltage command value calculator executes the integral calculation using a value obtained by multiplying the second deviation by a limit gain; the limited gain is switched between a first limit value and a second limit value smaller than the first limit value; 4. The control device for an AC rotating machine according to claim 3, wherein the limit gain is set to the second limit value when the voltage amplitude is equal to or less than a second voltage threshold value which is smaller than the first voltage threshold value, or when the rotation speed of the AC rotating machine is equal to or less than a second rotation speed threshold value which is smaller than the first rotation speed threshold value, or when the rotation angular velocity of the AC rotating machine is equal to or less than a second rotation angular velocity threshold value which is smaller than the first rotation angular velocity threshold value.

5. the inverter includes three sets of upper arm switching elements, lower arm switching elements, and shunt resistors corresponding to three phases, respectively; 5. The control device for an AC rotating machine according to claim 3, wherein the current detector detects the rotating machine currents in the three phases based on voltages across the corresponding shunt resistors when the voltage amplitude is smaller than the first voltage threshold value.

6. 6. The control device for an AC rotating machine according to claim 3, wherein the first axis voltage command value calculator sets the proportional gain to the first proportional value when the rotating machine current exceeds a threshold value.

7. 7. The control device for an AC rotating machine according to claim 3, wherein the first axis voltage command value calculator sets the proportional gain to the first proportional value when a fault occurs at at least one point in the control device for an AC rotating machine.

8. A control device for an AC rotating machine according to any one of claims 1 to 7; The AC rotating machine; a driving force transmission mechanism that transmits the driving force of the AC rotating machine to a steering system of a vehicle.

9. the first axis voltage command value calculator calculates the voltage command value of the first axis by using a value obtained by multiplying the first deviation by a proportional gain; the proportional gain is switched between a first speed reference value and a second speed reference value that is less than the first speed reference value; 9. The electric power steering device according to claim 8, wherein the first axis voltage command value calculator sets the proportional gain to a value smaller than the first speed reference value when a traveling speed of the vehicle is smaller than a threshold value.

10. 10. The electric power steering device according to claim 9, wherein the first axis voltage command value calculator sets the proportional gain to a value smaller than the first speed reference value when the current command value is equal to or smaller than a threshold value, when the steering torque is equal to or smaller than a threshold value, when a gradient in a graph showing a relationship between the steering torque and the torque current command value is smaller than a threshold value, or when an amount of change in the steering torque per unit time is smaller than a threshold value.

11. 11. The electric power steering device according to claim 8, wherein the first axis voltage command value calculator executes the integral calculation using a result of passing the rotational angular velocity of the AC rotating machine through a low-pass filter having a cut-off frequency higher than an upper limit value of a steering frequency of the electric power steering device.

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