Power Conversion Equipment

Abstract

To provide a power conversion device capable of avoiding torque shock in a case different phase error estimation technique is respectively used for a different frequency range.SOLUTION: A power conversion device includes a power converter for outputting a signal for varying an output frequency, an output voltage, and an output current of a motor to the motor, and a control part for controlling the power converter, the control part calculates a first electric power from the output voltage and the output current, calculates a second electric power from the output current, an electric circuit parameter, and a frequency estimation value, calculates a first phase error estimation value in a first frequency range and calculates a second phase error estimation value in a second frequency range different from the first frequency region, so that the first electric power follows the second electric power, and controls the frequency estimation value so that the first phase error estimation value or the second phase error estimation value follows a command value for a phase error estimation value.SELECTED DRAWING: Figure 1

Description

[Technical field]

[0001] The present invention relates to a power conversion device. [Background technology]

[0002] As a stable and highly accurate control method in the low-speed range of position sensorless control, there is a technology described in Patent Document 1 in which reactive power is calculated and the frequency of the magnet motor is estimated based on a voltage command value to a power converter, a current detection value, electric circuit parameters of the magnet motor, and a frequency estimate value. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] JP 2006-197712 A Summary of the Invention [Problem to be solved by the invention]

[0004] The technology described in Patent Document 1 calculates two types of reactive power (Q and Qhat) and calculates an inverter frequency estimate value so as to make the deviation ΔQ zero. Since the frequency estimate value can be made less sensitive to temperature changes in the winding resistance value of the magnet motor, highly accurate control characteristics can be achieved.

[0005] Incidentally, there are cases where different technologies are used for speed control when the frequency is high (medium-high speed) and when the frequency is low (low speed). In such cases, for example, in the low speed range, the technology of Patent Document 1 is used to calculate two types of reactive power, and the first motor frequency is estimated so that the deviation ΔQ is zero. On the other hand, for the medium-high speed range, for example, a different technology is used to estimate the phase error (control phase and magnet motor phase) using extended induced voltage, and the second motor frequency is estimated using PI control so as to follow zero. When the first motor frequency and the second motor frequency are switched between the low speed range and the medium-high speed range, if there is a difference between the two frequencies, a torque shock (torque fluctuation) due to current change may occur.

[0006] In addition, by reducing the sensitivity of the winding resistance of the magnet motor to temperature changes, it is possible to achieve stable and highly accurate control characteristics without the need to adjust electrical circuit parameters such as resistance values.

[0007] An object of the present invention is to provide a power conversion device that can prevent torque shock when the frequency of a motor changes, does not require adjustment of electrical circuit parameters, and has stable and highly accurate control characteristics. [Means for solving the problem]

[0008] The present invention includes a power converter that outputs signals to the motor to vary the output frequency, output voltage, and output current of the motor; A control unit for controlling the power converter, The control unit is calculating a first power from the output voltage and the output current, calculating a second power from the output current, an electrical circuit parameter, and a frequency estimate, and calculating the first phase error estimate in a first frequency domain such that the first power tracks the second power; computing a second phase error estimate in a second frequency domain different from the first frequency domain; The power conversion device controls a frequency estimate so that the first phase error estimate or the second phase error estimate follows a command value of the phase error estimate. Effect of the Invention

[0009] According to the present invention, it is possible to prevent torque shock when the motor frequency changes, and to realize stable and highly accurate control characteristics without the need to adjust electrical circuit parameters. [Brief description of the drawings]

[0010] [Figure 1] FIG. 1 is a system configuration diagram of a power conversion device and the like in a first embodiment. [Diagram 2] FIG. 4 is a diagram showing the configuration of a phase error estimation calculation unit using an extended induced voltage in the high-speed range. [Diagram 3] FIG. 4 is a configuration diagram of a phase error estimation calculation unit in the low speed range in the first embodiment. [Figure 4] FIG. 3 is a configuration diagram of a frequency and phase estimation calculation unit in the first embodiment. [Diagram 5] FIG. 13 is a diagram showing control characteristics when the extended induced voltage method for the medium to high speed range is used in the low speed range. [Figure 6] FIG. 4 is a diagram showing control characteristics in the first embodiment. [Figure 7] FIG. 4 is a diagram showing control characteristics when switching between a low speed range and a medium to high speed range in the first embodiment. [Figure 8] FIG. 1 is a configuration diagram for confirming the feasibility of the present invention. [Figure 9] FIG. 11 is a system configuration diagram of a power conversion device and the like in a second embodiment. [Figure 10] FIG. 11 is a configuration diagram of a phase error estimation calculation unit for a low speed range in the second embodiment. [Figure 11] FIG. 11 is a system configuration diagram of a power conversion device and the like in a third embodiment. [Figure 12] FIG. 11 is a configuration diagram of a phase error estimation calculation unit for medium to high speed ranges in the third embodiment. [Figure 13] FIG. 13 is a system configuration diagram of a power conversion device and the like in a fourth embodiment. [Figure 14] FIG. 13 is a configuration diagram of a phase error estimation calculation unit in the medium to high speed range in the fourth embodiment. [Figure 15] FIG. 13 is a system configuration diagram of a power conversion device and the like in a fifth embodiment. [Figure 16] FIG. 13 is a system configuration diagram of a power conversion device and the like in a sixth embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0011] The present embodiment will be described in detail below with reference to the drawings. Note that the same reference numerals are used to designate common components in the various drawings. Furthermore, the embodiments described below are not limited to the examples shown in the drawings. EXAMPLES

[0012] FIG. 1 is a diagram showing the configuration of a system including a power conversion device and a magnet motor according to a first embodiment.

[0013] The power conversion device of this embodiment achieves stable and highly accurate control characteristics in position sensorless control that omits encoders and other devices that detect the magnet phase of the magnet motor, even in the low speed range from a stop to about 10% of the base frequency.

[0014] The magnet motor 1 outputs a motor torque that is a combination of a torque component due to the magnetic flux of the permanent magnet and a torque component due to the inductance of the armature winding.

[0015] The power converter 2 includes a semiconductor element as a switching element. The power converter 2 outputs a three-phase AC voltage command value v u * , v v * , v w * and the three-phase AC voltage command value v u * , v v * , v w * The magnet motor 1 is driven based on the output of the power converter 2, and the output voltage value, output frequency value, and output current of the magnet motor 1 are variably controlled. IGBTs (Insulated Gate Bipolar Transistors) may be used as switching elements.

[0016] The DC power supply 3 supplies a DC voltage and a DC current to the power converter 2 .

[0017] The current detector 4 detects the three-phase AC current i u , i v , i w The detected value of i uc , i vc , i wc The current detector 4 detects AC currents in two of the three phases of the magnet motor 1, for example, the u and w phases, and the v phase AC current is output under the AC condition (i u +i v+i w = 0), i v =-(i u +i w ) may also be calculated.

[0018] In this embodiment, the current detector 4 is provided inside the power conversion device, but may be provided outside the power conversion device.

[0019] The control unit includes a coordinate conversion unit 5, a speed control calculation unit 6, a vector control calculation unit 7, a medium to high speed range phase error estimation calculation unit 8, a low speed range phase error estimation calculation unit 9, a frequency and phase estimation calculation unit 10, and a coordinate conversion unit 11, which are described below. The control unit controls the output of the power converter 2 so that the output voltage value, output frequency value, and output current of the magnet motor 1 are variably controlled.

[0020] The control unit is composed of a semiconductor integrated circuit (arithmetic control means) such as a microcomputer (microcomputer) or a DSP (Digital Signal Processor). Any or all of the control unit can be composed of hardware such as an ASIC (Application Specific Integrated Circuit) or an FPGA (Field Programmable Gate Array). The CPU (Central Processing Unit) of the control unit reads out a program stored in a recording device such as a memory, and executes the processing of each unit such as the coordinate conversion unit 5 described above.

[0021] Next, each component of the control unit will be described.

[0022] The coordinate conversion unit 5 converts the three-phase AC current i u , i v , i w Detection value i uc , i vc , i wc and the phase estimate θ dc to the d-axis and q-axis current detection values ​​i dc , i qc Output.

[0023] The speed control calculation unit 6 calculates a frequency command value ω r * and the frequency estimate ω dc Based on the torque command value τ * The current command value i of the q axis is calculated by dividing it by the torque coefficient. q * The frequency command value ω is output. r * is used to judge whether the speed is in the medium or high speed range or the low speed range, and the frequency estimate ω dc corresponds to the motor speed estimate (motor rotation speed estimate).

[0024] The vector control calculation unit 7 calculates the d-axis and q-axis current command values ​​i d * , i q * , current detection value i dc , i qc , the frequency estimate ω dc and the d-axis and q-axis voltage command values ​​v dc ** , v qc ** Output.

[0025] The phase error estimation calculation unit 8 for the medium to high speed range is c axis and q c Axis voltage command value v dc ** , v qc ** , the frequency estimate ω dc , current detection value i dc , i qc Using the electric circuit parameters of the magnet motor 1, the phase θ dc and the phase θ of the magnet of magnet motor 1 d The estimated value Δθ of the phase error Δθ, which is the deviation from c _ H Output.

[0026] The phase error estimation calculation unit 9 for the low speed region is c axis and q c Axis voltage command value v dc ** , vqc ** , the frequency estimate ω dc , current detection value i dc , i qc and the electric circuit parameters of the magnet motor 1 are used to determine the phase θ dc and the phase θ of the magnet of magnet motor 1 d The estimated value Δθ of the phase error Δθ, which is the deviation from c _ L Calculate the following.

[0027] The frequency and phase estimation calculation unit 10 calculates an estimated value Δθ of the phase error in the low speed range. c _ L Or the estimated phase error in the medium to high speed range Δθ c _ H Based on dc and the phase estimate θ dc Output.

[0028] The coordinate conversion unit 11 is d c axis and q c Axis voltage command value v dc * , v qc ** and the phase estimate θ dc to the three-phase AC voltage command value v u * , v v * , v w * Output.

[0029] First, the basic operation of the sensorless vector control method when the phase error estimation calculation unit 9 for the low speed range is used will be described.

[0030] The speed control calculation unit 6 calculates a frequency command value ω r * The frequency estimate ω dc In order to follow the torque command τ according to (Equation 1) using proportional and integral control, * and the q-axis current command value i q * Calculate the following.

[0031]

number

[0032] First, the vector control calculation unit 7 calculates the set value R of the winding resistance, which is an electric circuit parameter of the permanent magnet motor 1. * , d-axis inductance setting value L d * , the set value of the q-axis inductance L q * , the value of the induced voltage coefficient K e * , d c axis and q c Axis current command value i d * , i q * and the frequency estimate ω dc Using this, according to (Equation 2), c axis and q c Axis voltage reference value v dc * , v qc * Output.

[0033]

number

[0034] Secondly, the vector control calculation unit 7 calculates d c axis and q c Axis current command value i d * , i q * The current detection value of each component i dc , i qc By proportional and integral control, d is controlled according to (Equation 3). caxis and q c Axis voltage correction value Δv dc , Δv qc Calculate the following.

[0035]

number

[0036] Furthermore, the vector control calculation unit 7 calculates d c axis and q c Axis voltage command value v dc ** , v qc ** Calculate the following.

[0037]

number

[0038] FIG. 2 shows a block diagram of the medium-to-high-speed phase error estimation calculation unit 8. The medium-to-high-speed phase error estimation calculation unit 8 is c axis and q c Axis voltage command value v dc ** , v qc ** , current detection value i dc , i qc and the electric circuit parameters of magnet motor 1 (R * , L q * ), the calculation unit 81 for the estimated value of the phase error in the medium to high speed range by the extended induced voltage method calculates the estimated value Δθ of the phase error in the medium to high speed range according to (Equation 5). c_H Calculate the following.

[0039]

number

[0040] Here, a description will be given of the low speed range phase error estimation calculation unit 9. A block diagram of the low speed range phase error estimation calculation unit 9 is shown in FIG.

[0041] The low-speed phase error estimation calculation unit 9 estimates d c Axis voltage command value v dc ** and q c Axis voltage command value v qc ** and d as the output current of magnet motor 1 c Axis current detection value i dc , q c Axis current detection value i qc Using the above, the first reactive power Q c is calculated according to (Equation 6).

[0042]

number

[0043] The second reactive power calculation unit 92 calculates d c Axis current detection value i dc , and q c Axis current detection value i qc , the frequency estimate ω dc , the electric circuit parameters of magnet motor 1 (L d * , L q * , K e * ) to calculate the second reactive power Q c ^ is calculated according to (Equation 7).

[0044]

number

[0045] The subtractor 93 includes the first reactive power Q c and the second reactive power Q c ^ is input, and its deviation, ΔQ c Calculate the reactive power deviation ΔQ c is input to a PI control calculation unit 95 (hereinafter referred to as a PI control unit) so as to follow "0", which is a command value 94 of the deviation of reactive power. The PI control unit 95 calculates an estimated value Δθ of the phase error Δθ in the low speed range according to (Equation 8) by P (proportional) + I (integral) control calculation. c_L Calculate the following.

[0046]

number

[0047] The frequency and phase estimation calculation unit 10 will now be described.

[0048] The switching unit 101 receives an estimated value Δθ of the phase error in the low speed range. c_L and the estimated phase error in the mid-high range, Δθ c_H , and the frequency command value ω r * The switching unit 101 receives the frequency command value ω r * Depending on the size of Δθ c = Δθ c_L , Δθ for medium to high speed range c = Δθ c_H , the estimated phase error Δθ c The output is as follows:

[0049] The subtractor 103 subtracts the estimated phase error Δθ c is the phase error command value Δθ c * The estimated phase error Δθ c and Δθ c *The deviation between the frequency and the frequency is input to the PI control unit 104. The PI control unit 104 calculates the frequency estimate value ω dc In addition, an I control calculation unit (I control unit) 105 calculates a phase estimate value θ based on the output of the PI control unit 104 according to (Equation 10). dc Calculate the following.

[0050]

number

[0051]

number

[0052] Next, the principle by which the present invention provides stable and highly accurate control characteristics will be described.

[0053] FIG. 5 shows a phase error estimation calculation unit 9 for the low-speed range according to the present invention (ΔQ c_H FIG. 1 is a diagram showing the control characteristics when a frequency command value ω r * is set to 2% of the base frequency. c axis and q c Axis voltage command value v dc ** , v dc ** The set value R of the resistance R included in the calculation formula for the phase error in the medium to high speed range shown in (Formula 5) is * When there is no error in (a)R * / R=1 (reference).

[0054] Resistance value R set value R * When there is an error in (b) R * The simulation results for (a) and (b) are shown in Figure 5 when / R=0.5.

[0055] In Fig. 5, the upper part shows the load torque TL , the middle row is the frequency command ω r * and the motor frequency ω r The lower part shows the phase error Δθ. A ramp-like load torque is applied from point A in the figure, and is changed to 100% at point B. From point B to the right, the load torque is maintained.

[0056] (a)R * When / R=1 (standard) is set, the phase error Δθ is zero in the steady state, and the motor frequency ω r is the frequency command ω r * (b) R * When R is set to 0.5, the phase error Δθ becomes negative and the motor frequency ω r is stagnating near zero, and magnet motor 1 is out of step.

[0057] In this embodiment, d c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc Using the above, the first reactive power Q c is calculated using (Equation 6). c axis and q c Axis current detection value i dc , i qc , the frequency estimate ω dc , the setting value of the electric circuit parameters of the magnet motor 1 (L d * , L q * , K e * ) to calculate the second reactive power Q c ^ is calculated using (Equation 7).

[0058] The result of the calculation, Q c ^ and Q c In order to track the deviation of the phase error in the low-speed range, the estimated value Δθ c _L is automatically adjusted using (Equation 8) to estimate the value Δθ c _ L By using this in the frequency and phase estimation calculation unit 10, the sensitivity to the resistance value can be reduced, thereby improving the control characteristics.

[0059] The control characteristics in the low speed range according to this embodiment are shown in FIG. * When R is set to 0.5, the low-speed phase error estimation calculation unit 9 and the frequency and phase estimation calculation unit 10 are operated, and a load torque similar to that shown in FIG. 4 is applied. The estimated value Δθ of the phase error is calculated by the reactive power, which is insensitive to the resistance value R. c _ L Therefore, even if R* / R is set to 0.5, the actual phase error Δθ can be suppressed to zero.

[0060] Furthermore, in this embodiment, the frequency command value ω r * The frequency command value ω r * The estimated phase error values ​​for the low speed range and the medium to high speed range are switched at a magnitude of 10%.

[0061] ω r * In the low-speed range where ω is less than 10% of the base frequency, it is calculated using (Equation 8). r * The mid-to-high speed range where the frequency is 10% or more of the base frequency is calculated from (Formula 5). The switching characteristics from the low speed range to the mid-to-high speed range, or from the mid-to-high speed range to the low speed range in this embodiment, are shown in FIG.

[0062] As shown in the middle of Fig. 7, in the C region, the speed is switched from the low speed range to the medium-high speed range, and in the D region, the speed is switched from the medium-high speed range to the low speed range. Looking at the phase error Δθ in the lower part, the magnitude changes slightly depending on the timing of the switch, but the motor frequency ω r There is no shock in the torque and there is no shock in the torque, and it is clear that the effect of this embodiment is obvious.

[0063] In this embodiment, ω r* is 10% of the base frequency, and the estimated phase error value is switched between the low-speed range and the medium-speed range. r * There is no problem if the value is switched between zero and 10% or less of the base frequency.

[0064] In addition, the estimated phase error in the low-speed region, Δθ c _ L The taper gain, which changes between "1" and "0", is G _L , the estimated phase error in the medium to high speed range Δθ c _ H The taper gain G changes between "0" and "1". _H The average phase error estimate is multiplied by Δθ c It is also possible to use the following.

[0065] Here, a verification method when this embodiment is adopted will be described with reference to Fig. 8. A voltage detector 21 and a current detector 22 are attached to a power conversion device 20 that drives a magnet motor 1, and an encoder 23 is attached to the shaft of the magnet motor 1.

[0066] The vector voltage and current component calculation unit 24 receives the three-phase AC voltage detection value (v uc , v vc , v wc ), three-phase AC current detection value (i uc , i vc , i wc ) and the encoder output position θ are input, and the vector voltage component v dc , v qc , vector current component i dc , i qc and the detected value ω obtained by differentiating the position θ dc Calculate the following.

[0067] In the observation unit 25 of each waveform, the phase error Δθ_ is calculated using (Equation 11). cal Calculate the following.

[0068]

number

[0069] The parameters to be set in the controller of power converter 2 (R * ,L d * ,L q * ,K e * ) to change the size of Δθ_ in (Equation 11). cal If it coincides with the actual phase error Δθ, it becomes clear that the present invention is employed.

[0070] According to this embodiment, in position sensorless control that omits an encoder or the like that detects the magnet phase of the magnet motor, a power conversion device can be realized that achieves stable and highly accurate control characteristics even in the low speed range from stopped to about 10% of the base frequency, without adjusting the electrical circuit parameters and control gains of the magnet motor that are set in the control unit (controller).

[0071] Furthermore, in the low-speed range, instead of estimating the motor frequency as in Patent Document 1, according to this embodiment, the phase error is estimated from the deviation of two types of reactive power in the same way as in the medium-high speed range. With such a configuration, in the low-speed range, as in the medium-high speed range, the motor frequency (motor rotation speed) is estimated so that the estimated value of the phase error follows its command value, making it possible to prevent torque shock. EXAMPLES

[0072] FIG. 9 is a system configuration diagram including a power conversion device and a magnet motor according to the second embodiment.

[0073] In the first embodiment, the phase error estimation calculation unit 9 for the low speed range is d c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc to the first reactive power Q cIn this embodiment, the first reactive power calculation unit 9a1 calculates the amplitude value V1 of the voltage command for one phase of the three-phase AC as the output voltage of the magnet motor 1. * , the amplitude value i1 of the current detection, and the phase difference θ between the voltage command value and the current detection value vi The first reactive power Q c Calculate the following.

[0074] The phase error estimation calculation unit 8 for the medium to high speed range from the magnet motor 1, the frequency and phase estimation calculation unit 10, and the coordinate conversion unit 11 in Fig. 9 are the same as those in Fig. 1. Description of the same contents as those in the first embodiment will be omitted.

[0075] 10 shows an estimation calculation unit 9a for a phase error in a low speed range according to this embodiment. The functional block in FIG. 10 corresponds to the phase error estimation calculation unit 9 in FIG.

[0076] 3. Furthermore, the second reactive power calculation unit 9a2, subtraction unit 9a3, reactive power deviation command value 9a4, and PI control unit 9a5 in FIG. 10 are the same as the second reactive power calculation unit 92, subtraction unit 93, reactive power deviation command value 94, and PI control unit 95 in FIG.

[0077] In the figure, the first reactive power calculation unit 9a1 calculates the amplitude value V1 of the three-phase AC voltage command. * (Formula 12), the amplitude value i1 of the current detection value is (Formula 13), and the phase θ vi is calculated from (Formula 14). Then, using (Formula 15), the voltage amplitude value V1 for one phase of the three-phase AC as the output voltage of the magnet motor 1 is calculated. * , the current amplitude value i1, and the phase difference θ between the voltage command value and the current detection value vi Using the sinusoidal signal of reactive power Q c Calculate the following.

[0078]

number

[0079]

number

[0080]

number

[0081]

number

[0082] According to this embodiment, the first reactive power Q c Since there are few parameters to calculate, the amount of calculation can be reduced.

[0083] Even if this embodiment, which is an AC quantity, is used, it is possible to realize highly accurate control characteristics, similar to the first embodiment. EXAMPLES

[0084] 11 is a system configuration diagram having a power conversion device and a magnet motor in the third embodiment. In the first embodiment, the phase error estimation calculation unit 8 for the medium to high speed range calculates an estimated value Δθ of the phase error in the medium to high speed range according to (Equation 5). c_H Calculate the following.

[0085] In the third embodiment, the phase error estimation calculation unit 8a for the medium to high speed range is c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc to the first active power P c Calculate the following.

[0086] 12 is a diagram showing the configuration of an estimation calculation unit 8a for a phase error in the medium to high speed range according to the third embodiment. FIG. 12 corresponds to the medium to high speed range phase error estimation calculation unit 8 in FIG.

[0087] In Fig. 11, the magnet motor 1 to the vector control calculation unit 7, and the low-speed range phase error estimation calculation unit 9 to the frequency and phase estimation calculation unit 10 are the same as those in Fig. 1. Descriptions of the same contents as in the first or second embodiment will be omitted.

[0088] In the phase error estimation calculation unit 8a for the medium to high speed range in FIG. 12, the first active power calculation unit 8a1 calculates d c axis and q c Axis voltage command value v dc ** , v qc ** And, d c axis and q c Axis current detection value i dc , i qc The first active power P c is calculated according to (Equation 16).

[0089]

number

[0090] The second active power calculation unit 8a2 calculates d c axis and q c Axis current detection value i dc , i qc , the frequency estimate ω dc , the electric circuit parameters of magnet motor 1 (R * , L d * , L q * , K e * ) to obtain the second active power P c ^ is calculated according to (Equation 17).

[0091]

number

[0092] The subtractor 8a3 receives the first active power P c and the second active power P c ^is input, and the deviation is ΔP c Calculate the deviation of active power ΔP c The active power deviation ΔP c and the command value of the deviation of the active power are input to the PI control unit 8a5.

[0093] The PI control unit 8a5 calculates an estimated value Δθ of the phase error Δθ in the medium to high speed range according to (Equation 18) by P (proportional) + I (integral) control. c_H Calculate the following.

[0094]

number

[0095] According to this embodiment, compared to the estimation calculation of the phase error in the medium to high speed range by the extended induced voltage method of the first embodiment, the estimation calculation of the phase error in the medium to high speed range can be performed without being affected by the estimation error of the inductance because the set value of the inductance is not used. Also, the same effects as those of the first embodiment are obtained. EXAMPLES

[0096] FIG. 13 is a diagram showing the configuration of a system including a power conversion device and a magnet motor according to the fourth embodiment.

[0097] In Example 3, d c axis and q c Axis voltage command value v dc ** , v qc ** and the current detection value i dc , i qc to the first active power P c was calculated.

[0098] In this embodiment, the amplitude value V1 of the three-phase AC voltage command * and the amplitude value i1 and phase θ of the current detection viThe first active power calculation unit 9b1 calculates the first active power P c The description of the same contents as in the first to third embodiments will be omitted.

[0099] Fig. 14 is a diagram showing a phase error estimation calculation unit 8b in the medium to high speed range of the embodiment 3. Fig. 14 corresponds to the medium to high speed range phase error estimation calculation unit 8 in Fig. 2. The magnet motor 1 to the vector control calculation unit 7, and the low speed range phase error estimation calculation unit 9 to the coordinate conversion unit 11 in Fig. 13 are the same as those in Fig. 1.

[0100] The second active power calculation unit 8b2, subtraction unit 8b3, active power deviation command value 8b4, and PI control unit 8b5 in Figure 14 are the same as the second active power calculation unit 8a2, subtraction unit 8a3, active power deviation command value 8a4, and PI control unit 8a5 in Figure 12.

[0101] In FIG. 14, the first active power calculation unit 9b1 calculates the amplitude value V1 of the three-phase AC voltage command. * is calculated using the above-mentioned (Formula 12), the amplitude value i1 of the current detection value is calculated using the above-mentioned (Formula 13), and the phase difference θ between the voltage command value and the current detection value is calculated using the above-mentioned (Formula 14). vi is calculated from the above-mentioned (Formula 14). The first active power calculation unit 9b1 then calculates the voltage amplitude value V1 for one phase of the three-phase AC by using (Formula 19). * and the current amplitude value i1, and the phase difference θ between the voltage command value and the current detection value vi The first active power P c Calculate the following.

[0102]

number

[0103] According to this embodiment, similarly to the third embodiment, compared to the first embodiment, it is possible to perform an estimation calculation of the phase error in the medium to high speed range without being affected by the estimation error of the inductance.

[0104] Similarly to the first embodiment, this embodiment can also realize highly accurate control characteristics. EXAMPLES

[0105] FIG. 15 is a system configuration diagram including a power conversion device, a magnet motor, and an IOT (Internet of Things) controller according to the fifth embodiment.

[0106] In the first to fourth embodiments, the electrical circuit parameters of the magnet motor 1 are set in a controller (such as a microcomputer) of a power converter, but in the present embodiment, the state quantity of the control is fed back to a higher-level IOT controller, and the machine-learned electrical circuit parameters are reset in the controller of the power converter.

[0107] The magnet motor 1 to the coordinate conversion unit 11 in Fig. 15 are the same as the functional blocks in Fig. 1. The IOT controller 12 is an IOT controller that executes machine learning. Descriptions of the same contents as in the first to fourth embodiments will be omitted.

[0108] In this embodiment, the voltage command value v dc ** , v qc ** and the current detection value i dc , i qc , the estimated phase error Δθ c The electric circuit parameters (R * , L d * , L q * , K e * ) is reset in the controller of the power converter 2, which is the control unit.

[0109] According to this embodiment, the electrical circuit parameters can be reconfigured from the upper IOT controller 12 by machine learning of the driving situation.

[0110] In this embodiment as well, similarly to the first embodiment, more stable and highly accurate control characteristics can be realized. EXAMPLES

[0111] FIG. 16 is a system configuration diagram having a power conversion device, a magnet motor, and devices such as a personal computer, a tablet, and a smartphone in the sixth embodiment.

[0112] This embodiment is applied to a magnet motor drive system. In the figure, the components of the magnet motor 1 and the coordinate conversion units 5 to 11 are the same as those in Fig. 1. Explanations of the same contents as in the first to fifth embodiments will be omitted.

[0113] The magnet motor 1, which is a component of Fig. 16, is driven by a power conversion device 20. In the power conversion device 20, the coordinate conversion unit 5, speed control calculation unit 6, vector control calculation unit 7, medium-to-high speed range phase error estimation calculation unit 8, low speed range phase error estimation calculation unit 9, frequency and phase estimation calculation unit 10, and coordinate conversion unit 11 of Fig. 1 are implemented as software 20a. In addition, the power converter 2, DC power supply 3, and current detector 4 of Fig. 1 are implemented in the power conversion device 20 as hardware.

[0114] In addition, the digital operator 20b, a personal computer 28, a tablet 29, a smartphone 30, or other higher-level device can control the "low-speed range / medium-high-speed range switching frequency 26, ω chg ", "ω c " can be set and changed.

[0115] The frequency and phase estimation calculation unit 10 shown in FIG. 4 is input with a low speed range / medium / high speed range switching frequency 26, ω chg is input from a higher-level device such as a personal computer 28, a tablet 29, or a smartphone 30. The frequency and phase estimation calculation unit 10 then calculates the frequency command value ω r * and ω, which is the low speed / medium / high speed switching frequency 26 chg The configuration is such that a comparison unit is provided for comparing the above.

[0116] Frequency command value ω r * is the low speed / medium / high speed switching frequency ω chg If the output is less than Δθ 26, the switching unit 101 sets the output to the low speed range. c = Δθ c_L Also, the frequency command value ω r * is the low speed / medium / high speed switching frequency ω chg If it is higher than 26, the switching unit 101 sets Δθ c = Δθ c_H is the estimated phase error Δθ c Switch the output to

[0117] The control response of phase error in the low-speed region is ω c is input to the low-speed phase error estimation calculation unit 9 in FIG. 3 from a host device such as a personal computer 28, a tablet 29, or a smartphone 30. With such a configuration, Kp pll (Proportional gain of PLL control), Ki pll (Integral gain of PLL control) is the control response of phase error in the low speed range ω c The control can be based on the above.

[0118] In addition, the low speed / medium / high speed switching frequency is 26, ω chg ", "ω c " may be configured on a fieldbus such as a programmable logic controller, a local area network connecting to a computer, or an IOT controller.

[0119] Furthermore, the configuration of the power conversion device is not limited to that of the first embodiment, and the configurations of the second to fifth embodiments may be used.

[0120] If this embodiment is applied to a magnet motor drive system, highly accurate control characteristics can be achieved in position sensorless vector control.

[0121] According to the sixth embodiment, the set values ​​such as the low speed range / medium / high speed range switching frequency 26 can be changed from outside the power conversion device. Also in this embodiment, as in the first embodiment, more stable and highly accurate control characteristics can be realized.

[0122] In the first to fifth embodiments, the first reactive power Q c (Equation 6) and the second reactive power Q c ^ In (Equation 7), the current detection value i dc , i qc was used, but the current command value i d * , i q * Also, the first active power P c (Equation 16) and the second reactive power P c ^ In (Equation 17), the current detection value i dc , i qc was used, but the current command value i d * , i q * may also be used.

[0123] Furthermore, in the first to fifth embodiments, the current command value i d * , i q * and the current detection value i dc , i qc to voltage correction value Δv dc , Δv qc The voltage correction value is added to the voltage reference value of the vector control (Equation 4). d * , i q * and the current detection value i dc , i qc The intermediate current command value i shown in (Equation 20) is used for vector control calculation. d ** , i q ** and calculate the frequency estimate ω dcMoreover, a vector control calculation shown in (Formula 21) may be performed using the electric circuit parameters of the magnet motor 1.

[0124]

number

[0125]

number

[0126] Or the current command value i d * , i q * and the current detection value i dc , i qc d used for vector control calculation c Voltage correction value Δv of the proportional calculation component of the axis d_p * , d c Voltage correction value Δv of the integral calculation component of the axis d_i * , q c Voltage correction value Δv of the proportional calculation component of the axis q_p * , q c Voltage correction value Δv of the integral calculation component of the axis q_i * is created using (Equation 22), and the frequency value estimate ω dc Moreover, a vector control calculation shown in (Formula 23) using the electric circuit parameters of the magnet motor 1 may be performed.

[0127]

number

[0128]

number

[0129] Also d c Axis current command value i d * and q c Axis current detection value i qc First-order lag signal i qctd , the frequency estimate ω dc Then, a vector control calculation shown in (Formula 24) may be performed using the electric circuit parameters of the magnet motor 1.

[0130]

number

[0131] In the first to sixth embodiments, the switching elements constituting the power converter 2 may be Si (silicon) semiconductor elements or wide band gap semiconductor elements such as SiC (silicon carbide) and GaN (gallium nitride). [Explanation of symbols]

[0132] 1...Magnet motor, 2...Power converter, 3...DC power supply, 4...Current detector, 5...Coordinate conversion section, 6...Speed ​​control calculation section, 7...Vector control calculation section, 8...Phase error estimation calculation section for medium to high speed range, 9...Phase error estimation calculation section for low speed range, 10...Frequency and phase estimation calculation section, 11...Coordinate conversion section, 20...Power conversion device, 20a...Software section of power conversion device, 20b...Digital operator of power conversion device, 21...Voltage detector, 22...Current detector, 23...Encoder, 24...Vector current component calculation section, 25...Observation section of waveforms of each section, 26...Predetermined low speed range / medium to high speed range switching frequency, 27...Control response of predetermined low speed range, 28...Personal computer, 29...Tablet, 30...Smartphone, i d * …d-axis current command value, i q * …q-axis current command value, ω dc …frequency estimate, ω r …frequency of the magnet motor, v dc * , v dc ** , v dc ** , v dc *** , v dc **** , v dc ***** …d-axis voltage command value, v qc * , v qc ** , v qc *** , v qc **** , v qc ***** …q-axis voltage command value, Q C …First reactive power, Q C ^ …Second reactive power, P C …First active power, P C ^ …Second active power, Δθ c_L …Estimated phase error in the low-speed range, Δθ c_H …Estimated phase error in the medium to high speed range, Δθ c …Estimated phase error

Claims

1. a power converter that outputs signals to the motor to vary the output frequency, output voltage, and output current of the motor; A control unit for controlling the power converter, The control unit is a first power is calculated from the output voltage and the output current, a second power is calculated from the output current, an electric circuit parameter, and a frequency estimate, and a first phase error estimate is calculated as a deviation between a control phase and a phase of a magnet of a magnet motor in a low speed range so that the first power follows the second power, and a second phase error estimate is calculated as a deviation between the control phase and a phase of a magnet of a magnet motor in a medium to high speed range; a deviation between the first phase error estimated value or the second phase error estimated value and a phase error command value is input to a PI control unit so that the first phase error estimated value or the second phase error estimated value follows the phase error command value.

2. 2. The power conversion device according to claim 1, A power conversion device, wherein the first power and the second power are first reactive power and second reactive power, respectively.

3. 2. The power conversion device according to claim 1, The control unit is a power conversion device that calculates the first phase error estimate value by proportional control and integral control so as to make the deviation between the first power and the second power zero;

4. The power conversion device according to claim 3, The control unit is If the rotation speed of the motor is in a low speed range, a first reactive power and a second reactive power are calculated; performing proportional control and integral control so as to make the deviation between the first reactive power and the second reactive power zero, and calculating the first phase error estimate value; a power conversion device that, when the rotation speed of the motor is in a mid-high range, calculates a first active power and a second active power, and performs proportional control and integral control so as to make a deviation between the first active power and the second active power zero, thereby calculating the second phase error estimate value.

5. The power conversion device according to claim 3, The control unit is If the rotation speed of the motor is in a low speed range, a first reactive power and a second reactive power are calculated; calculating the first phase error estimate value by performing proportional control and integral control so as to make the deviation between the first reactive power and the second reactive power zero; If the rotation speed of the motor is in a medium to high range, the power conversion device calculates the second phase error estimate value using an extended induced voltage method.

6. 2. The power conversion device according to claim 1, The control unit is the output voltage, the output current, and the first phase error estimate or the second phase error estimate, The data is fed back to the higher-level IOT controller for analysis, A power converter that modifies electrical circuit parameters of the motor.

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