Audio power amplifier and distortion reduction method wherein

Abstract

The audio power amplifier distortion reduction method is based on the common negative feedback introduction mode, at which the resistors-based R5, R6, Rnfb linear adder (12) sums input signal voltage Uin and output signal voltage Uout, as well as on the absence of the voltage amplifier stages. The signal is current-amplified only. First stages (10) convert the output voltage of the adder Uin + Ufnb into the output currents. Resistor Rcm (11) sets the current for two current mirrors at once, stabilizing the emitter currents of the transistors of first stages (10). Current amplifiers are supplied from the ground. Frequency dependent chains Rf, Cf (20, 21) and Rca, Cfc (40, 41) form the frequency dependence of amplifiers (30, 31) current gain. Current mirrors (50, 51) carry out the additional signal current amplification. Circuit (13) sets the bias voltage of the output stage transistors, and the output stage converts the voltage into the amplifier output current. Transistor blocks (10, 11, 50, 51) are supplied via power buses +U and –U over RC-filters (60, 61).

Description

Audio power amplifier and distortion reduction method wherein

[0001] This invention is a high-quality transistor audio power amplifier all stages of which are DC coupled and a large-amount negative feedback is applied in order to obtain ultra-low nonlinear (harmonic and intermodulation) signal distortion and noise reduction.

[0002] Transistor audio power amplifiers are usually built using 2-, 3-, 4- or 5-stage architectures as, for example, described in Douglas Self (2013) “Audio Power Amplifier Design”. There is a wide variety of audio power amplifier circuits, and almost all of them have audio signal voltage amplifier stages. Nowadays, a double differential stage with current sources in emitter circuits as an input signal voltage amplifier appears to be the best solution for the input stage of a transistor audio power amplifier. Moreover, such a circuit design allows to develop a fully symmetrical amplifier based on the complementary transistors. The patent FR2700082 ‘Low noise integrated circuit audio amplifier e.g. for hi=fi units’ is an example of such a solution. Even if a large-amount negative feedback is applied in those amplifiers, the level of total harmonic distortion (further, THD) less than 0.001% is achieved only for some values of the operating range frequencies and output powers. Moreover, the signal-to-noise ratio (further, SNR) is rarely above 120 dB at maximum power of the output harmonic signal.

[0003] Current bipolar transistor amplifier has a higher linearity of the transfer characteristic (current gain), compared to voltage amplifier. This is the basis of the audio power amplifier distortion reduction method. The absence of voltage amplifier stages in the audio signal path reduces the overall harmonic and intermodulation distortion factor of the amplifier. The input signal of an amplifier constructed in accordance with this method passes through the amplifier stages as follows: the voltages of the input and output signals pass through the linear resistive adder and the voltage-to-current converters; the signal current passes through the intermediate current amplifiers and the precision current mirrors, which perform a slight current amplification and a parametric current-voltage to signal voltage conversion at their outputs junction point; the output stage on powerful MOSFETs performs a final voltage-current conversion into the output current of the entire amplifier. The linear resistive adder determines the voltage gain of the entire amplifier, which is usually greater than 1. By virtue of Ohm's law, the signal voltage at the junction point of the current mirror outputs is a function of the output currents of the mirrors and their dynamic resistances. Thus, the voltage amplification is parametric and is carried out by current mirrors.

[0004] In the audio power amplifier, a signal from the adder (12) is fed to the input of the first stages (10). The based on resistors R5, R6 and Rnfb adder (12) sums the voltage of input signal “In” with the voltage of output signal “Out”, herewith R5=R6, and the voltage gain of the power amplifier is determined by the resistance ratio of Rnfb to R6, which is equal to Rnfb / R6=20. Low-value enough capacitor Cnfb prevents the amplifier excitation at high frequencies. Based on NPN transistors Q1, Q3 and PNP transistors Q2, Q4, the first common collector – common base (further, CC-CB) stages are symmetrical with respect to ground and appear to be the voltage to current converters (10). Emitters currents of the first double CC-CB stage (10) are stabilized by a circuit based on precision current mirrors R1, R3, Q5, Q9, Q6, Q10 and R2, R4, Q8, Q12, Q7, Q11 with one current-setting resistor Rcm (11). Transistors in the CC-CB stages (10) and in the precision current mirrors (11) are low power and low-noise, such as BC546B-BC556. Output currents of transistors Q3 and Q4 (10) flow in the direction of positive +U and negative -U power buses, amplified by the emitter followers (30, 31). A distinctive feature of the emitter followers is that the sum of their supply and input currents is equal to the output currents of the stages (30, 31). This is achieved by joint connection of base resistor R9 (20) and emitter resistor R11 (40) to the input of the current mirror, collector Q18 (50). Similarly, the junction point of base resistor R10 (21) and emitter resistor R12 (41) is connected to the input of the current mirror, collector Q18 (51). Since the emitter followers (30, 31) power supply appears to be the ground, the supply voltage ripples are not included in the signal they amplify. Frequency dependent chains R7, C1 (20) and R8, C4 (21) weaken the amplification of emitter followers (30) and (31) at the frequencies significantly higher than the operating frequency range of the power amplifier. Capacitors C2 (40) and C3 (41), which are connected in parallel to emitter resistors R11 (40) and R12 (41) respectively, raise the current gain of emitter followers (30) and (31) within the operating frequency range of the power amplifier. The output currents of emitter followers (30) and (31) are the input currents of precision current mirrors R13, R18, Q17, Q21, Q18, Q22 (50) and R17, R23, Q20, Q24, Q19, Q23 (51), respectively. The current mirrors (50, 51) transmission factor is slightly more than 2. Transistors in precision current mirrors (50, 51) should be middle-power, high-voltage, such as MJE340-MJE350. The outputs of precision current mirrors (50) and (51) are interconnected via the based on resistors R14, R15, R16 and transistors Q15, Q16 (13) circuit. Circuit (13) creates a bias voltage between the gates of high-power MOSFETs M1, M2 and M3, M4 and determines the quiescent current of output stage (14). Limiting resistors R19, R20 and R21, R22 are included in the gate chain of powerful MOSFETs M1, M2 and M3, M4. Current mirrors (11, 50, 51) and the collectors of input transistors Q1 and Q2 (10) are powered by buses +U and -U via the frequency filters based on resistors R24, R25, R26, capacitors C7, C8, C9 (60) and on resistors R27, R28, R29, capacitors C10, C11, C12 (61), respectively. Frequency filters (60, 61) are placed on a frequency of 50 Hz or 60 Hz, depending on the ripple frequency of the amplifier supply voltage. According to the simulation in the Micro-Cap 12 software, the maximum amplifier current gain from the outputs of the CC-CB stages (10) to the power amplifier output reaches approximately 196 dB for both amplification paths (upper, through the current amplifier (30), and lower, through the current amplifier (31)). Moreover, it practically does not depend on supply voltages +U and -U. The greater linearity of current amplifiers compared to voltage amplifiers reduces the level of harmonic and intermodulation distortion of the amplifier to a very low values, and the level of the supply voltage ripple penetration into the output signal is small. Amplification of the input signal voltage occurs in the amplifier indirectly. Neglecting the voltage drop across the bias circuit assembled on transistors Q15 and Q16 and resistors R14, R15, R16 (13), we can assume that the voltage of the amplified signal at the junction point of the outputs of current mirrors (50, 51) is a function of the output currents of these mirrors. The output voltage maximum value at this point is equal to the power bus +U voltage subtracting the voltage drop across resistors R24, R25, R26 (60) and R18, transistors Q21 and Q22 (50). The output voltage minimum value at the same point is equal to the power bus -U voltage subtracting the voltage drop across resistors R27, R28, R29 (61) and R23, transistors Q23 and Q24 (51).

[0005] Amplifier power supply is from AC mains (15). AC mains feeds voltage to the primary winding of step-down transformer X1. From the secondary center-tapped winding connected to the ground a reduced voltage is applied to the terminals of the bridge rectifier based on powerful Schottky diodes D1-D4, to the outputs of which simple filters, based on capacitors C5 and C6, are connected. Thus, we obtain positive +U and negative -U supply voltages.

[0006] The audio power amplifierwith a supply voltage of ±43.5 V in an operating temperature (2-42 °C) and a frequency (20-20000 Hz) ranges, with a load impedance of 4 Ohm, with a sinusoidal signal output power of 144 W, has the following calculated characteristics: a low output impedance ~0.00015 Ω, THD level ~0.0002 %, total level of intermodulation distortion according to DIN tests ~0.00035 %, to SMPTE tests ~0.00035 %, to CCIF (19 kHz, 20 kHz) tests ~0.0013 %, SNR >126 dB. The amplifier has an effectively amplified frequency band of 5-100000 Hz.

[0007] The graphs of the amplifier main characteristics are shown in—.

[0008] Circuit diagram of audio power amplifier.

[0009] Output voltage amplitude on frequency of audio power amplifier input signal dependence at temperature of 22 °C, frequency variation is less than 0.001 dB in operating range.

[0010] Output voltage phase on frequency of audio power amplifier input signal dependence at temperature of 22 °C, frequency variation is less than 0.4 degrees in operating range.

[0011] THD on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 2 °С, (%). THD=0.000172% for Pout=144 W, F=20 kHz.

[0012] THD on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 2 °С, (dB). THD=-115.3 dB for Pout=144 W, F=20 kHz.

[0013] THD on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 22 °С, (%). THD=0.000186% for Pout=144 W, F=20 kHz.

[0014] THD on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 22 °С, (dB). THD=-114.6 dB for Pout=144 W, F=20 kHz.

[0015] THD on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 42 °С, (%). THD=0.000204% for Pout=144 W, F=20 kHz.

[0016] THD on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 42 °С, (dB). THD=-113.8 dB for Pout=144 W, F=20 kHz.

[0017] Total harmonic distortion plus noise (THDN) on audio amplifier output power dependence for frequencies 20 Hz, 50 Hz, 100 Hz, 200 Hz, 500 Hz, 1 kHz, 2 kHz, 5 kHz, 10 kHz, 20 kHz at temperature of 22 °С, (dB).

[0018] Dependence of intermodulation distortion (IM2 and IM3) on audio amplifier output power in range from 0.1 mW to level determined by maximum output signal in CCIF test (F1=19 kHz, F2=20 kHz) at temperature of 22 °С, (dB).

[0019] Dependence of intermodulation distortion (IM2 and IM3) on audio amplifier output power in range from 0.1 mW to level determined by maximum output signal in DIN test (F1=250 Hz, F2=8 kHz) at temperature of 22 °С, (dB).

[0020] Dependence of intermodulation distortion (IM2 and IM3) on audio amplifier output power in range from 0.1 mW to level determined by maximum output signal in SMPTE test (F1=60 Hz, F2=7 kHz) at temperature of 22 °С, (dB).

[0021] Signal to noise ratio (SNR) for output power from 0.1 mW to maximum level at temperature of 22 °С, (dB).

[0022] Block diagram of audio power amplifier.

[0023] The invention is an electronic circuit of an audio power amplifier, which can obviously be used in conjunction with a corresponding high-quality audio signal source and electro-audio converters (passive acoustic speakers) for listening to high-quality audio by the end user. The best way of carrying out the electronic circuit of the audio power amplifier is to use low-power transistor microassemblies similar to discrete transistors BC546B-BC556B (10, 11, 13, 30, 31) and medium-power transistor microassemblies similar to discrete transistors MJE340-MJE350 (50, 51), in order to ensure the most complete symmetry of the circuit.

Claims

An audio power amplifier distortion reduction method is characterized by replacing the amplifier stages which contain involved in the formation of the transfer function non-linear elements with the stages ensuring its linearity, wherein:a signal is current amplified only, since the bipolar transistor current amplifier has a greater linearity of the transfer characteristic (current gain is a constant value);a summation of the input and output signal voltages is carried out by a linear resistive adder, which eliminates the very first source of distortion in the amplifier;current amplifiers (emitter followers) are powered by the ground, (i.e. the source without ripples of the rectified mains voltage), which results in less signal distortion in the frequency range close to and lower than the double frequency of the amplifier mains supply source (50 or 60 Hz).An audio power amplifier, comprising:a resistive adder, which is based on a first, a second and a third resistors, wherein:a first terminal of said first resistor is connected to an input terminal of the amplifier, anda first terminal of said second resistor is connected to an output terminal of the amplifier, anda first terminal of said third resistor is connected to the ground, andinterconnected second terminals of first, second and third resistors form the adder output, which is connected to inputs of voltage-to-current converters;based on complementary transistor pairs a first and a second common collector - common emitter (further, CC-CB) stages of the voltage-to-current converters, wherein:interconnected bases of a first NPN transistor of the first CC-CB stage and of a first PNP transistor of the second CC-CB stage are the voltage-to-current converters input, andcollectors of the first NPN transistor of the first CC-CB stage and of the first PNP transistor of the second CC-CB stage are connected to positive and negative power buses, respectively, andinterconnected emitters of NPN transistors of the first CC-CB stage and interconnected emitters of PNP transistors of the second CC-CB stage are connected to the outputs of the first and the second precision current mirror, respectively, providing the necessary operating current of the voltage-to-current converters andcollectors of the second NPN and PNP transistors of the first CC-CB stage are the current outputs of the voltage-to-current converters;a first and a second current mirrors powered by the negative and the positive power buses, respectively, wherein:a first and a second current mirror inputs are connected to each other via the resistor, which sets the current mirrors current and, hence, the operating current of the voltage-to-current converters;current outputs of the voltage-to-current converters, which are connected to the first terminals of the first and the second bias resistors;second terminals of the bias resistors, which are connected to the outputs of the first and the second current amplifiers (emitter followers), respectively;frequency correction chains of the first resistor and the first capacitor and the second resistor and the second capacitor in series, connected in parallel to the first and the second bias resistors, respectively;connected in parallel to the first bias resistor a chain of the base-emitter junction of PNP transistor of the first current amplifier and the first emitter resistor connected in series with it;connected in parallel to the second bias resistor a chain of the base-emitter junction of NPN transistor of the second current amplifier and the second emitter resistor connected in series with it;a first and a second local feedback blocking capacitors connected in parallel to the first and the second emitter resistors, respectively;a first and a second current amplifiers (emitter followers), the outputs of which are connected to the inputs of the third and the fourth precision current mirrors, respectively, and their collectors are connected to the ground;a third and a fourth precision current mirrors powered by positive and negative power buses, respectively, and being a dynamic load for each other, since their outputs are connected to each other via the bias circuit of N-Channel and P-Channel MOSFETs of high-power output stage;a high-power output stage, based on high-power N-Channel and P-Channel MOSFETs, which drains are connected to positive and negative terminals of the power source respectively, and all of their sources are connected to the output terminal of the amplifier.The circuit according to claim 2, wherein: the input voltage-current converters are carried out within another circuits, comprising one input connected to the resistive adder output and two antipodal current outputs connected to the output of the first and the second current amplifiers (emitter followers).The circuit according to claim 2, wherein: the amplification of signal currents instead of the first and the second current amplifiers is carried out by the other current amplifier circuits which are double-pole with respect to the signal passage.The circuit according to claim 2, wherein: the output signals of the voltage-to-current converters are amplified by several current amplifiers connected in series.The circuit according to claim 2, wherein: the first and the second current mirrors and the current setting resistor are replaced by other circuits of the stable current sources based on bipolar and field-effect transistors.The circuit according to claim 2, wherein: the third and the fourth current mirrors are replaced by other current mirror circuits.

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