Power conversion device

Abstract

In a DAB converter, a control circuit (13) performs control so that, when transmitting power from a first DC part (E1, Ca) to a second DC part (E2, Cb) by boosting the voltage, a second bridge circuit (12) includes a first period in which both ends of a secondary winding (n2) of an isolation transformer (TR1) are short-circuited within the second bridge circuit (12), and a second period in which the secondary winding (n2) of the isolation transformer (TR1) and the second DC part (E2, Cb) are conducted. The control circuit (13) generates a first pattern in which the first period shifts to the second period by turn-off of a seventh switching element (Q7), a second pattern in which the first period shifts to the second period by turn-off of a fifth switching element (Q5), a third pattern in which the first period shifts to the second period by turn-off of a sixth switching element (Q6), and a fourth pattern in which the first period shifts to the second period by turn-off of an eighth switching element (Q8).

Description

Power Conversion Device

[0001] The present disclosure relates to a power conversion device that converts DC power into DC power of another voltage.

[0002] Power conditioners used in solar power generation systems and vehicle-to-home (V2H) systems require highly efficient power conversion. V2H systems can charge and discharge power between a storage battery installed in an electric vehicle (e.g., an EV or PHEV) and a commercial power grid or a home load. For example, power generated by a home solar power generation system can be charged to the storage battery of an electric vehicle. Furthermore, the storage battery installed in an electric vehicle can be used for peak load shifting or backup purposes. DC / DC converters used in V2H systems must be highly efficient, isolated, and have a wide voltage range. This is because the voltage of storage batteries installed in electric vehicles varies greatly depending on the vehicle model. One DC / DC converter that meets these requirements is the dual active bridge (DAB) converter.

[0003] In a typical DAB converter, losses increase due to hard switching when the potential difference between input and output is large, and losses increase due to the flow of reactive current unrelated to power transmission. To address this issue, for example, a DAB converter has been proposed that uses a phase-shift method to boost or buck the voltage while turning off all PWM (Pulse Width Modulation) signals input to one leg of a bridge circuit in which four switching elements are bridge-connected (see, for example, Patent Document 1).

[0004] WO 16 / 125373

[0005] In the bridge circuit on the secondary side of the proposed DAB converter, losses are uneven between the switching elements that are all off and the switching elements used for phase shift control. This results in uneven heat generation, which leads to an increase in the size of components (e.g., heat sinks) used to cool the switching elements. This also leads to greater variations in the life span of the switching elements.

[0006] The present disclosure has been made in view of these circumstances, and its purpose is to provide a technique for equalizing losses in switching elements included in a DAB converter.

[0007] In order to solve the above problems, a power conversion device according to one aspect of the present disclosure includes: a first bridge circuit having a first leg in which a first switching element and a second switching element are connected in series, and a second leg in which a third switching element and a fourth switching element are connected in series, the first leg and the second leg being connected in parallel to a first DC section; a second bridge circuit having a third leg in which a fifth switching element and a sixth switching element are connected in series, and a fourth leg in which a seventh switching element and an eighth switching element are connected in series, the third leg and the fourth leg being connected in parallel to a second DC section; an isolation transformer connected between the first bridge circuit and the second bridge circuit; and a control circuit that controls the first switching element and the eighth switching element. When power is boosted from the first DC section to the second DC section and transmitted, the control circuit controls the second bridge circuit to include a first period in which both ends of the secondary winding of the isolation transformer are short-circuited within the second bridge circuit and a second period in which the secondary winding of the isolation transformer and the second DC section are conductive, and generates a first pattern in which the seventh switching element conducting in the forward direction is turned off and a transition to the second period occurs from the first period, a second pattern in which the fifth switching element conducting in the forward direction is turned off and a transition to the second period occurs, a third pattern in which the sixth switching element conducting in the forward direction is turned off and a transition to the second period occurs, and a fourth pattern in which the eighth switching element conducting in the forward direction is turned off and a transition to the second period occurs.

[0008] According to the present disclosure, it is possible to equalize the losses of the switching elements included in the DAB converter.

[0009] 1A to 1C are diagrams illustrating the configuration of a power conversion device according to an embodiment. FIGS. 2A to 2C are diagrams illustrating the operating state of a power conversion device according to Comparative Example 1. FIGS. 3A to 3C are diagrams illustrating the current flow during step-down operation of a power conversion device according to Comparative Example 2. FIGS. 3A to 3C are diagrams illustrating the switching pattern of the first switching element to the eighth switching element during step-down operation of a power conversion device according to Comparative Example 2. FIGS. 5A to 5C are diagrams illustrating the current flow during step-up operation of a power conversion device according to Comparative Example 2. FIGS. 5A to 5C are diagrams illustrating the switching pattern of the first switching element to the eighth switching element during step-up operation of a power conversion device according to Comparative Example 2. FIGS. 8A to 8D are diagrams illustrating a first example of switching patterns and current flows during step-down operation of a power conversion device according to an example (part 1). FIGS. 9A to 9D are diagrams illustrating a first example of switching patterns and current flows during step-down operation of a power conversion device according to an example (part 2). 10(a)-10(d) are diagrams illustrating a second example of a switching pattern and a current flow during a step-down operation according to an embodiment of the power conversion device (part 1). 11(a)-11(d) are diagrams illustrating a second example of a switching pattern and a current flow during a step-down operation according to an embodiment of the power conversion device (part 2). 12(a)-12(d) are diagrams illustrating a third example of a switching pattern and a current flow during a step-down operation according to an embodiment of the power conversion device (part 1). 13(a)-13(d) are diagrams illustrating a third example of a switching pattern and a current flow during a step-down operation according to an embodiment of the power conversion device (part 2). This figure shows an example of a switching pattern and a transition of a transformer current during reverse transmission of the first switching element to the eighth switching element during a step-down operation according to an embodiment of the power conversion device. This figure shows an example of a switching pattern and a transition of a transformer current during a step-up operation according to an embodiment of the power conversion device.FIGS. 16(a)-(d) are diagrams illustrating a first example of switching patterns and current flows during boost operation in the embodiment of the power conversion device (part 1). FIGS. 17(a)-(d) are diagrams illustrating a first example of switching patterns and current flows during boost operation in the embodiment of the power conversion device (part 2). FIGS. 18(a)-(d) are diagrams illustrating a second example of switching patterns and current flows during boost operation in the embodiment of the power conversion device (part 1). FIGS. 19(a)-(d) are diagrams illustrating a second example of switching patterns and current flows during boost operation in the embodiment of the power conversion device (part 2). Specific example 1 shows a periodic generation of switching patterns for the first switching element to the eighth switching element during boost operation in the embodiment of the power conversion device. Specific example 2 shows an irregular generation of switching patterns for the first switching element to the eighth switching element during boost operation in the embodiment of the power conversion device. FIGS. 22(a)-(d) are diagrams illustrating a third example of switching patterns and current flows during boost operation in the embodiment of the power conversion device (part 1). 23(a)-(d) are diagrams illustrating a third example of switching patterns and current flows during boost operation in the embodiment of the power conversion device (part 2). This diagram illustrates a switching pattern during reverse transmission of the first switching element and the eighth switching element during boost operation in the embodiment of the power conversion device, and an example of transitions in the transformer current. This diagram illustrates a switching pattern during reverse transmission of the first switching element and the eighth switching element during boost / buck operation in the embodiment of the power conversion device, and an example of transitions in the transformer current. FIG. 26(a) is a diagram schematically illustrating the relationship between the duty ratio and the transmission power when a boost / buck mode is not provided. FIG. 26(b) is a diagram schematically illustrating the relationship between the duty ratio and the transmission power when a boost / buck mode is provided. This diagram illustrates an example configuration of a control circuit. FIGS. 28(a)-(b) are graphs illustrating examples of variables and constants in a theoretical formula for calculating a synchronous rectification period. FIGS. 29(a)-(d) are diagrams illustrating yet another example of switching patterns and current flows during buck operation in the embodiment of the power conversion device (part 1).30(a) to 30(d) are diagrams for explaining still another example of switching patterns and current flows in the step-down operation according to the embodiment of the power conversion device (part 2).

[0010] 1 is a diagram illustrating the configuration of a power conversion device 1 according to an embodiment. The power conversion device 1 is an isolated bidirectional DC / DC converter (DAB converter) that converts DC power supplied from a first DC power source E1 and transmits it to a second DC power source E2. The power conversion device 1 also converts DC power supplied from the second DC power source E2 and transmits it to the first DC power source E1. The power conversion device 1 can transmit power by stepping down or stepping up the voltage.

[0011] The first DC power source E1 may be, for example, a storage battery or electric double layer capacitor mounted on the EV, or a stationary storage battery or electric double layer capacitor. The second DC power source E2 may be, for example, a DC bus connected to a commercial power system via an inverter. Other storage batteries, solar cells, fuel cells, etc. may be connected to the DC bus via other DC / DC converters.

[0012] The power conversion device 1 includes a primary side capacitor Ca, a first bridge circuit 11, a first inductance L1, an isolation transformer TR1, a second inductance L2, a second bridge circuit 12, a secondary side capacitor Cb, and a control circuit 13.

[0013] A primary-side capacitor Ca is connected in parallel with the first DC power supply E1. A secondary-side capacitor Cb is connected in parallel with the second DC power supply E2. For example, electrolytic capacitors are used for the primary-side capacitor Ca and the secondary-side capacitor Cb. In this specification, the first DC power supply E1 and the primary-side capacitor Ca are collectively referred to as a first DC unit, and the second DC power supply E2 and the secondary-side capacitor Cb are collectively referred to as a second DC unit.

[0014] The first bridge circuit 11 is a full-bridge circuit configured by connecting a first leg in series with a first switching element Q1 and a second switching element Q2, and a second leg in series with a third switching element Q3 and a fourth switching element Q4, connected in parallel. The first bridge circuit 11 is connected in parallel with the first DC section, and the midpoint of the first leg and the midpoint of the second leg are connected to both ends of the primary winding n1 of the isolation transformer TR1, respectively. The first bridge circuit 11 can convert the primary-side DC voltage supplied from the first DC section into an AC voltage and output it to the primary winding n1 of the isolation transformer TR1. The first bridge circuit 11 can also convert the AC voltage supplied from the primary winding n1 of the isolation transformer TR1 into a DC voltage and output it to the first DC section.

[0015] The second bridge circuit 12 is a full-bridge circuit configured by connecting a third leg in series, in which the fifth switching element Q5 and the sixth switching element Q6 are connected, and a fourth leg in series, in which the seventh switching element Q7 and the eighth switching element Q8 are connected, in parallel. The second bridge circuit 12 is connected in parallel with the second DC section, and the midpoint of the third leg and the midpoint of the fourth leg are connected to both ends of the secondary winding n2 of the isolation transformer TR1, respectively. The second bridge circuit 12 can convert the secondary-side DC voltage supplied from the second DC section into an AC voltage and output it to the secondary winding n2 of the isolation transformer TR1. The second bridge circuit 12 can also convert the AC voltage supplied from the secondary winding n2 of the isolation transformer TR1 into a DC voltage and output it to the second DC section.

[0016] A first diode D1 to an eighth diode D8 are formed in anti-parallel or connected to the first switching element Q1 to the eighth switching element Q8, respectively. In addition, a first capacitor C1 to an eighth capacitor C8 are formed in parallel or connected to the first switching element Q1 to the eighth switching element Q8, respectively.

[0017] The first switching element Q1 to the eighth switching element Q8 can be, for example, a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) or an IGBT (Insulated Gate Bipolar Transistor). When MOSFETs are used for the first switching element Q1 to the eighth switching element Q8, parasitic diodes formed between the drain and source of the first switching element Q1 to the eighth switching element Q8 are used as the first diode D1 to the eighth diode D8, respectively, or external diode elements are connected as the first diode D1 to the eighth diode D8, respectively. Furthermore, parasitic capacitances formed between the drain and source of the first switching element Q1 to the eighth switching element Q8 are used as the first capacitance C1 to the eighth capacitance C8, respectively, or external capacitors are connected between the drain and source of the first switching element Q1 to the eighth switching element Q8, respectively.

[0018] When IGBTs are used for the first switching element Q1 to the eighth switching element Q8, external diodes are connected between the collectors and emitters of the first switching element Q1 to the eighth switching element Q8 as the first diode D1 to the eighth diode D8, respectively. Also, external capacitors are connected between the collectors and emitters of the first switching element Q1 to the eighth switching element Q8 as the first capacitance C1 to the eighth capacitance C8, respectively, or parasitic capacitances formed between the collectors and emitters of the first switching element Q1 to the eighth switching element Q8 are used as the first capacitance C1 to the eighth capacitance C8, respectively.

[0019] MOSFETs do not generate tail current compared to IGBTs, which reduces switching loss during turn-off and reduces heat generation, allowing for the miniaturization of coolers (e.g., heat sinks). Furthermore, MOSFETs can be driven at higher frequencies than IGBTs, allowing for the miniaturization of passive components (e.g., transformers and capacitors). However, MOSFETs typically have a higher on-resistance than IGBTs.

[0020] In recent years, switching elements using wide-gap semiconductors (for example, silicon carbide (SiC), gallium nitride (GaN), gallium oxide (Ga2O3), and diamond (C)) have become popular as switching elements with high withstand voltage and low loss. In this embodiment, it is assumed that SiC-MOSFETs are used for the first switching element Q1 to the eighth switching element Q8. SiC-MOSFETs have a higher withstand voltage than Si-MOSFETs, and therefore can be made smaller in size and have a lower on-resistance per unit area.

[0021] The isolation transformer TR1 is connected between the AC terminals of the first bridge circuit 11 and the AC terminals of the second bridge circuit 12. The isolation transformer TR1 converts the output voltage of the first bridge circuit 11 connected to the primary winding n1 in accordance with the turns ratio between the primary winding n1 and the secondary winding n2, and outputs the converted voltage to the second bridge circuit 12 connected to the secondary winding n2. The isolation transformer TR1 also converts the output voltage of the second bridge circuit 12 connected to the secondary winding n2 in accordance with the turns ratio between the secondary winding n2 and the primary winding n1, and outputs the converted voltage to the first bridge circuit 11 connected to the primary winding n1.

[0022] The first inductance L1 is connected or formed in series between the AC terminal of the first bridge circuit 11 and the primary winding n1 of the isolation transformer TR1. The second inductance L2 is connected or formed in series between the AC terminal of the second bridge circuit 12 and the secondary winding n2 of the isolation transformer TR1. In the example shown in Fig. 1 , the first inductance L1 is formed by a reactor element connected between the midpoint of the first leg of the first bridge circuit 11 and the primary winding n1 of the isolation transformer TR1. The second inductance L2 is formed by a reactor element connected between the midpoint of the third leg of the second bridge circuit 12 and the secondary winding n2 of the isolation transformer TR1.

[0023] The first inductance L1 may be formed by the leakage inductance of the primary winding n1 formed between the midpoint of the first leg of the first bridge circuit 11 and the primary winding n1 of the isolation transformer TR1. The second inductance L2 may be formed by the leakage inductance of the secondary winding n2 formed between the midpoint of the third leg of the second bridge circuit 12 and the secondary winding n2 of the isolation transformer TR1. Either the first inductance L1 or the second inductance L2 may be omitted.

[0024] The control circuit 13 executes the following control as basic control. When transmitting power from the first DC unit to the second DC unit (when discharging from the first DC power supply E1), the control circuit 13 controls the first switching element Q1 to the eighth switching element Q8 so that the current value detected by the secondary-side current sensor 23 maintains the current command value. Note that the voltage value detected by the first voltage sensor 21 on the primary side, the secondary-side voltage value detected by the second voltage sensor 22 on the secondary side, and the current value detected by a primary-side current sensor (not shown) may be controlled as target values.

[0025] Furthermore, when transmitting power from the second DC unit to the first DC unit (when charging the first DC power supply E1), the control circuit 13 controls the first switching element Q1 to the eighth switching element Q8 so that the current value detected by a primary-side current sensor (not shown) maintains the current command value. Note that the voltage value detected by the first voltage sensor 21 on the primary side, the secondary-side voltage value detected by the second voltage sensor 22 on the secondary side, and the current value detected by the secondary-side current sensor 23 may be controlled as target values.

[0026] As described above, the DAB converter has a symmetrical configuration between the primary side and the secondary side, and is capable of transmitting power in both directions. The operation of the power conversion device 1 will now be described.

[0027] 2(a) to 2(c) are diagrams illustrating operating states according to Comparative Example 1 of the power conversion device 1. In the first state shown in Fig. 2(a), the control circuit 13 controls the first switching element Q1, the fourth switching element Q4, the sixth switching element Q6, and the seventh switching element Q7 to be in the on state, and the second switching element Q2, the third switching element Q3, the fifth switching element Q5, and the eighth switching element Q8 to be in the off state. In this state, power is charged from the first DC power source E1 to the first inductance L1, and power is charged from the second DC power source E2 to the second inductance L2.

[0028] 2(b), the control circuit 13 controls the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 to be on, and the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 to be off. In this state, the power of the first DC power supply E1, the power stored in the first inductance L1, and the power stored in the second inductance L2 are transmitted to the second DC power supply E2.

[0029] In a third state (not shown), the control circuit 13 controls the second switching element Q2, the third switching element Q3, the fifth switching element Q5, and the eighth switching element Q8 to be on, and the first switching element Q1, the fourth switching element Q4, the sixth switching element Q6, and the seventh switching element Q7 to be off. In this state, power is charged from the first DC power supply E1 to the first inductance L1, and power is charged from the second DC power supply E2 to the second inductance L2.

[0030] In a fourth state (not shown), the control circuit 13 controls the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 to be on, and the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 to be off. In this state, the power of the first DC power supply E1, the power stored in the first inductance L1, and the power stored in the second inductance L2 are transmitted to the second DC power supply E2.

[0031] In the control according to Comparative Example 1, in the first state (see FIG. 2(a)) and the third state (not shown), the power of the second DC power supply E2 is charged to the second inductance L2. In the subsequent second state (see FIG. 2(b)) and the fourth state (not shown), the power stored in the second inductance L2 is discharged to the second DC power supply E2. That is, a reactive current unrelated to power transmission flows on the secondary side. This flow of reactive current causes unnecessary loss.

[0032] 2(c) shows the current flow when the voltage of the first DC power supply E1 drops significantly relative to the voltage of the second DC power supply E2 in the second state shown in FIG. 2(b). When the voltage of the second DC power supply E2 becomes higher than the voltage of the first DC power supply E1, the direction of the current reverses, and current flows back from the second DC power supply E2 to the first DC power supply E1. In this state, when the first switching element Q1 and the fourth switching element Q4 are turned off and the second switching element Q2 and the third switching element Q3 are turned on to transition to the next state, the second switching element Q2 and the third switching element Q3 enter hard switching mode, and the first diode D1 of the first switching element Q1 and the fourth diode D4 of the fourth switching element Q4 enter recovery mode, resulting in increased loss.

[0033] (Comparative Example 2) Figures 3(a) and 3(b) are diagrams illustrating the flow of current during a step-down operation according to Comparative Example 2 of the power conversion device 1. Figure 4 is a diagram illustrating the switching patterns of the first switching element Q1 to the eighth switching element Q8 during a step-down operation according to Comparative Example 2 of the power conversion device 1. Figures 5(a) and 5(b) are diagrams illustrating the flow of current during a step-up operation according to Comparative Example 2 of the power conversion device 1. Figure 6 is a diagram illustrating the switching pattern of the first switching element Q1 to the eighth switching element Q8 during a step-up operation according to Comparative Example 2 of the power conversion device 1.

[0034] A phase shift method is adopted in Comparative Example 2. The duty ratio of the first switching element Q1 to the sixth switching element Q6 is fixed at 50%, and the seventh switching element Q7 and the eighth switching element Q8 are maintained in an all-off state.

[0035] Fig. 3(a) shows a state in which power is transmitted from the first DC power supply E1 to the second DC power supply E2 (hereinafter referred to as the transmission state). Fig. 3(b) shows a state in which the first DC power supply E1 and the isolation transformer TR1 are disconnected and power is transmitted from the first inductance L1 and the second inductance L2 to the second DC power supply E2 (hereinafter referred to as the commutation state). In the step-down operation, the voltage or current of the transmitted power is controlled by the ratio between the transmission state and the commutation state. The higher the ratio of the commutation state, the lower the voltage or current of the transmitted power is controlled.

[0036] Specifically, as shown in FIG. 4 , the phase of the first leg (the first switching element Q1 and the second switching element Q2) is fixed, and the phase of the second leg (the third switching element Q3 and the fourth switching element Q4) is variable. By controlling the phase of the second leg, the phase difference between the first and second legs is controlled. The third leg (the fifth switching element Q5 and the sixth switching element Q6) is controlled in synchronization with the second leg. When increasing power transmitted from the primary side to the secondary side, the control circuit 13 controls the phase difference to be smaller (shifting the phase of the second leg to the left). When decreasing power transmitted from the primary side to the secondary side, the control circuit 13 controls the phase difference to be larger (shifting the phase of the second leg to the right).

[0037] Fig. 5(a) shows a state in which the isolation transformer TR1 and the second DC power supply E2 are disconnected and power is stored in the first inductance L1 and the second inductance L2 from the first DC power supply E1 (hereinafter referred to as the storage state). Fig. 5(b) shows a transmission state in which power is transmitted from the first DC power supply E1, the first inductance L1, and the second inductance L2 to the second DC power supply E2. In the boost operation, the voltage or current of the transmitted power is controlled by the ratio between the transmission state and the storage state. The higher the ratio of the storage state, the higher the voltage or current of the transmitted power is controlled.

[0038] 6 , the phases of the first and second legs are fixed, the phase of the third leg is variable, and the phase difference between the first and second legs and the third leg is controlled by controlling the phase of the third leg. When increasing the power transmitted from the primary side to the secondary side, the control circuit 13 controls the phase difference to increase (shifting the phase of the third leg to the right), and when decreasing the power transmitted from the primary side to the secondary side, the control circuit 13 controls the phase difference to decrease (shifting the phase of the third leg to the left).

[0039] In Comparative Example 2, the seventh switching element Q7 and the eighth switching element Q8 are maintained in an all-off state, so that current does not flow back from the second DC power supply E2 to the second inductance L2, the first inductance L1, and the first DC power supply E1. That is, no reactive current is generated due to current flowing back from the second DC power supply E2 as shown in FIG. 2A of Comparative Example 1. Furthermore, even if the voltage of the second DC power supply E2 becomes higher than the voltage of the first DC power supply E1 during voltage step-down operation, current does not flow back from the second DC power supply E2 to the first DC power supply E1. Therefore, the next on / off switching of the first switching element Q1 to the fourth switching element Q4 on the primary side can be prevented from becoming hard switching.

[0040] 4 , in the step-down operation of Comparative Example 2, the first switching element Q1 and the second switching element Q2 included in the first leg are turned off in the transmission state, and the third switching element Q3 and the fourth switching element Q4 included in the second leg are turned off in the commutation state. Because the turn-off occurs in the transmission state when a larger current is flowing, the turn-off loss of the first switching element Q1 and the second switching element Q2 included in the first leg is larger than the turn-off loss of the third switching element Q3 and the fourth switching element Q4 included in the second leg. As shown in FIG. 4 , the turn-off of the first switching element Q1 and the second switching element Q2 is a turn-off that changes the absolute value of the current from an increase to a decrease, so the switching loss is large.

[0041] As shown in Fig. 6, in the boost operation of Comparative Example 2, the fourth leg is completely turned off. Therefore, the switching loss of the fifth switching element Q5 and the sixth switching element Q6 included in the third leg is larger than the switching loss of the seventh switching element Q7 and the eighth switching element Q8 included in the fourth leg. As shown in Fig. 6, the turn-off of the fifth switching element Q5 and the sixth switching element Q6 causes the absolute value of the current to change from increasing to decreasing, so the switching loss is large.

[0042] (Example (Step-Down)) FIG. 7 is a diagram showing an example of the switching patterns of the first switching element Q1 to the eighth switching element Q8 and the transition of the transformer current IL during step-down operation according to the example of the power conversion device 1. The transformer current IL is a current flowing through the isolation transformer TR1. FIGS. 8(a) to 8(d) are diagrams for explaining a first example of the switching patterns and current flows during step-down operation according to the example of the power conversion device 1 (part 1). FIGS. 9(a) to 9(d) are diagrams for explaining a first example of the switching patterns and current flows during step-down operation according to the example of the power conversion device 1 (part 2). FIGS. 10(a) to 10(d) are diagrams for explaining a second example of the switching patterns and current flows during step-down operation according to the example of the power conversion device 1 (part 1). FIGS. 11(a) to 11(d) are diagrams for explaining a second example of the switching patterns and current flows during step-down operation according to the example of the power conversion device 1 (part 2).

[0043] This embodiment employs a combination of PWM and phase-shift control. When transferring power by stepping down the voltage from the first DC section to the second DC section, the control circuit 13 controls the power transfer so that it includes a first period T1 (step-down) and a second period T2 (step-down). The first period T1 (step-down) is a period during which the first bridge circuit 11 conducts power between the first DC section and the primary winding n1 of the isolation transformer TR1, and the second bridge circuit 12 conducts power between the secondary winding n2 of the isolation transformer TR1 and the second DC section. The second period T2 (step-down) is a period during which both ends of the primary winding n1 of the isolation transformer TR1 are short-circuited within the first bridge circuit 11, and the second bridge circuit 12 conducts power between the secondary winding n2 of the isolation transformer TR1 and the second DC section.

[0044] The control circuit 13 generates the first period T1 (step-down) and the second period T2 (step-down) using a first pattern (step-down)-fourth pattern (step-down). A first example of the switching patterns and current flow during step-down operation shown in FIGS. 8(a)-(d) and 9(a)-(d) is an example in which the switching elements in the second bridge circuit 12 that should be controlled to a rectifying state are controlled to an on state to perform synchronous rectification. A second example of the switching patterns and current flow during step-down operation shown in FIGS. 10(a)-(d) and 11(a)-(d) is an example in which the switching elements in the second bridge circuit 12 that should be controlled to a rectifying state are controlled to an off state to perform diode rectification.

[0045] In the first period T1-1 (step-down) of the first pattern (step-down), the first switching element Q1 and the fourth switching element Q4 are in the ON state, the fifth switching element Q5 and the eighth switching element Q8 are in the rectifying state, and the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the OFF state (see Figures 8(a) and 10(a)).

[0046] In the second period T2-1 (step-down) of the first pattern (step-down), the first switching element Q1 is in the ON state, the third switching element Q3, the fifth switching element Q5, and the eighth switching element Q8 are in the rectifying state, and the second switching element Q2, the fourth switching element Q4, the sixth switching element Q6, and the seventh switching element Q7 are in the OFF state (see Figures 8(b) and 10(b)).

[0047] In the first pattern (step-down), the control circuit 13 turns off the fourth switching element Q4, which is conducting in the forward direction, during the first period T1-1 (step-down), to transition to the second period T2-1 (step-down). Here, "conducting in the forward direction" refers to a state in which a current flows from the drain to the source or from the collector to the emitter of the switching element.

[0048] In the first period T1-2 (step-down) of the second pattern (step-down), the second switching element Q2 and the third switching element Q3 are in the ON state, the sixth switching element Q6 and the seventh switching element Q7 are in the rectifying state, and the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the OFF state (see Figures 8(c) and 10(c)).

[0049] In the second period T2-2 (step-down) of the second pattern (step-down), the third switching element Q3 is in the ON state, the first switching element Q1, the sixth switching element Q6, and the seventh switching element Q7 are in the rectifying state, and the second switching element Q2, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the OFF state (see Figures 8(d) and 10(d)).

[0050] In the second pattern (step-down), the control circuit 13 turns off the second switching element Q2 that is forward conductive during the first period T1-2 (step-down), and transitions to the second period T2-2 (step-down).

[0051] In the first period T1-3 (step-down) of the third pattern (step-down), the first switching element Q1 and the fourth switching element Q4 are in the ON state, the fifth switching element Q5 and the eighth switching element Q8 are in the rectifying state, and the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the OFF state (see Figures 9(a) and 11(a)).

[0052] In the second period T2-3 (step-down) of the third pattern (step-down), the fourth switching element Q4 is in the ON state, the second switching element Q2, the fifth switching element Q5, and the eighth switching element Q8 are in the rectifying state, and the first switching element Q1, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the OFF state (see Figures 9(b) and 11(b)).

[0053] In the third pattern (step-down), the control circuit 13 turns off the first switching element Q1 that is forward conductive during the first period T1-3 (step-down), and transitions to the second period T2-3 (step-down).

[0054] In the first period T1-4 (step-down) of the fourth pattern (step-down), the second switching element Q2 and the third switching element Q3 are in the ON state, the sixth switching element Q6 and the seventh switching element Q7 are in the rectifying state, and the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the OFF state (see Figures 9(c) and 11(c)).

[0055] In the second period T2-4 (step-down) in the fourth pattern (step-down), the second switching element Q2 is in the ON state, the fourth switching element Q4, the sixth switching element Q6, and the seventh switching element Q7 are in the rectifying state, and the first switching element Q1, the third switching element Q3, the fifth switching element Q5, and the eighth switching element Q8 are in the OFF state (see Figures 9(d) and 11(d)).

[0056] In the fourth pattern (step-down), the control circuit 13 turns off the third switching element Q3 that is forward conductive during the first period T1-4 (step-down), and transitions to the second period T2-4 (step-down).

[0057] The control circuit 13 supplies different drive signals to the first switching element Q1 to the fourth switching element Q4 included in the first bridge circuit 11. Specifically, the control circuit 13 fixes the duty ratio of the drive signals supplied to two diagonal switching elements (e.g., the first switching element Q1 and the fourth switching element Q4) of one of the first switching element Q1 to the fourth switching element Q4 to 50%, regardless of the amount of power transmission, excluding dead time. The control circuit 13 changes the first phase difference φ1 between the two drive signals supplied to the two diagonal switching elements, respectively, within a range of 0 to 180° depending on the amount of power transmission, excluding dead time. The first phase difference φ1 is 0° when the turn-on timings of the two drive signals supplied to the two diagonal switching elements are simultaneous.

[0058] The control circuit 13 changes the duty ratio of the drive signal supplied to the two switching elements (e.g., the second switching element Q2 and the third switching element Q3) on the other diagonal of the first switching element Q1 to the fourth switching element Q4, within a range of 0 to 100%, excluding dead time, depending on the amount of power transmission. Specifically, the control circuit 13 reduces the duty ratio of the drive signal supplied to one of the two switching elements on the other diagonal from 50%, excluding dead time, by an amount corresponding to the first phase difference φ1. At the same time, the control circuit 13 increases the duty ratio of the drive signal supplied to the other of the two switching elements on the other diagonal from 50%, excluding dead time, by an amount corresponding to the first phase difference φ1.

[0059] In this way, in the first bridge circuit 11, the duty ratio and the phase shift amount are controlled using the same control variable according to the amount of power transmission. As a result, as shown in FIG. 7, the turn-off that reverses the increase in the absolute value of the current can be distributed between the first switching element Q1 and the fourth switching element Q4, and loss and heat generation can be distributed between the first switching element Q1 and the fourth switching element Q4. In Comparative Example 2 shown in FIG. 4 above, loss and heat generation are concentrated between the first switching element Q1 and the second switching element Q2.

[0060] The control circuit 13 synchronizes the turn-on timing of the drive signals supplied to two diagonal switching elements, the fifth switching element Q5 and the eighth switching element Q8. The control circuit 13 fixes the duty ratio of the drive signal supplied to one (e.g., the fifth switching element Q5) of the two diagonal switching elements (e.g., the fifth switching element Q5 and the eighth switching element Q8) at 50% regardless of the amount of power transmission, excluding dead time. The control circuit 13 changes the duty ratio of the drive signal supplied to the other (e.g., the eighth switching element Q8) of the two diagonal switching elements within a range of 50% or less, depending on the amount of power transmission, excluding dead time. The control circuit 13 switches the settings of the duty ratios of the drive signals supplied to the two switching elements at the start of each switching period.

[0061] In this way, in the second bridge circuit 12, by fixing the phase and controlling the duty ratio according to the amount of power transmission, it is possible to distribute loss and heat generation to the fifth switching element Q5 to the eighth switching element Q8.

[0062] As shown in FIG. 7, the control circuit 13 generates the first pattern (step-down) and the fourth pattern (step-down) twice each in units of four switching cycles. In this case, the first pattern (step-down) and the fourth pattern (step-down) are generated the same number of times. This allows the switching loss generated on the primary side during step-down operation to be evenly allocated to the first switching element Q1 to the fourth switching element Q4, thereby dispersing heat generation.

[0063] The control circuit 13 may perform control so that the number of times that the first pattern (step-down) and the fourth pattern (step-down) occur differs. For example, the control circuit 13 may generate the first pattern (step-down) and the second pattern (step-down) four times each, and the third pattern (step-down) and the fourth pattern (step-down) twice each, over three switching cycles. Even in this case, it is possible to distribute losses to a certain extent, and the peak heat generation on the primary side can be suppressed compared to the control shown in Comparative Example 2.

[0064] 12(a)-(d) are diagrams (part 1) for explaining a third example of a switching pattern and a current flow in a step-down operation according to an embodiment of the power conversion device 1. FIG. 13(a)-(d) are diagrams (part 2) for explaining a third example of a switching pattern and a current flow in a step-down operation according to an embodiment of the power conversion device 1.

[0065] In the third example (step-down), the control circuit 13 controls the switching elements in the second bridge circuit 12 that should be in a rectifying state to the on state during the first period T1 (step-down), thereby performing synchronous rectification. The control circuit 13 controls the switching elements in the second bridge circuit 12 that should be in a rectifying state to the off state during the second period T2 (step-down), thereby performing diode rectification. Compared to the second example (step-down), the third example (step-down) allows for smaller losses in the first period T1 (step-down) because the conduction loss due to synchronous rectification is smaller than the forward loss Vf due to diode rectification.

[0066] The switching pattern shown in FIG. 7 represents a combination of the first example (step-down) and the second example (step-down). In principle, during the second period T2 (step-down), the control circuit 13 controls the two diagonal switching elements of the second bridge circuit 12 that should be in a rectifying state to the on state, thereby performing synchronous rectification. During the second period T2 (step-down), when the absolute value of the transformer current IL falls below a predetermined threshold Ith, the control circuit 13 turns off at least one of the two diagonal switching elements to switch to diode rectification. Note that both switching elements may be turned off, although this reduces efficiency.

[0067] The threshold value Ith is set to 0 A if the transformer current IL can be ideally measured and the delay in the control system and drive system is ideally zero. In practice, taking into consideration voltage measurement errors and delays in the drive signals supplied to the switching elements, the threshold value Ith is set to (0 ± margin) A. The margin value is set based on experiments and simulations by the designer so that the sign of the transformer current IL does not reverse within half a switching period Tc / 2 and the synchronous rectification period T2s of the second period T2 (step-down) is maximized.

[0068] If the absolute value of the transformer current IL does not decrease to the threshold value Ith within a predetermined time, the control circuit 13 skips turning off at least one of the two diagonal switching elements included in the second bridge circuit 12. The predetermined time is set to half a cycle Tc / 2 minus the dead time. That is, if the absolute value of the transformer current IL does not decrease to the threshold value Ith within the predetermined time, the two diagonal switching elements included in the second bridge circuit 12 enter a dead time while remaining in the on state. Entering the dead time causes the control circuit 13 to control all of the fifth switching element Q5 through the eighth switching element Q8 to be turned off. If the absolute value of the transformer current IL decreases slowly and there is no risk of power backflow, there is no need to switch from synchronous rectification to diode rectification. On the other hand, if there is a risk of power backflow, as shown in FIG. 7, the control circuit 13 switches from synchronous rectification to diode rectification. If power backflow occurs, a reactive current unrelated to power transmission flows, leading to increased loss. Switching to diode rectification prevents increased loss due to reactive current.

[0069] FIG. 14 is a diagram showing an example of the switching patterns of the first switching element Q1 through the eighth switching element Q8 during reverse transmission during step-down operation of the power conversion device 1 according to the embodiment, and a transition example of the transformer current IL. The switching patterns of the first switching element Q1 through the eighth switching element Q8 shown in FIG. 7 illustrate an example in which power is transmitted by stepping down from the first DC section to the second DC section. In this regard, it is also possible to transmit power by stepping down from the second DC section to the first DC section. In this case, as shown in FIG. 14, the control circuit 13 simply switches the drive signals supplied to the first switching element Q1 through the fourth switching element Q4 and the drive signals supplied to the fifth switching element Q5 through the eighth switching element Q8.

[0070] (Example (Boost)) FIG. 15 is a diagram showing switching patterns of the first switching element Q1 to the eighth switching element Q8 and an example of transition of the transformer current IL during boost operation according to the example of the power conversion device 1. FIGS. 16(a) to 16(d) are diagrams for explaining a first example of switching patterns and current flows during boost operation according to the example of the power conversion device 1 (part 1). FIGS. 17(a) to 17(d) are diagrams for explaining a first example of switching patterns and current flows during boost operation according to the example of the power conversion device 1 (part 2). FIGS. 18(a) to 18(d) are diagrams for explaining a second example of switching patterns and current flows during boost operation according to the example of the power conversion device 1 (part 1). FIGS. 19(a) to 19(d) are diagrams for explaining a second example of switching patterns and current flows during boost operation according to the example of the power conversion device 1 (part 2).

[0071] This embodiment employs a combination of PWM and phase-shift control. When transmitting power by boosting it from the first DC section to the second DC section, the control circuit 13 controls the power transmission to include a first period T1 (boost) and a second period T2 (boost). The first period T1 (boost) is a period during which the first bridge circuit 11 conducts power between the first DC section and the primary winding n1 of the isolation transformer TR1, and both ends of the secondary winding n2 of the isolation transformer TR1 are short-circuited within the second bridge circuit 12, resulting in a storage state. The second period T2 (boost) is a period during which the first bridge circuit 11 conducts power between the first DC section and the primary winding n1 of the isolation transformer TR1, and the second bridge circuit 12 conducts power between the second DC section and the secondary winding n2 of the isolation transformer TR1.

[0072] The control circuit 13 generates the first period T1 (boost) and the second period T2 (boost) using a first pattern to a fourth pattern (boost). A first example of the switching patterns and current flow during the boost operation shown in FIGS. 16(a)-(d) and 17(a)-(d) is an example in which the switching elements in the second bridge circuit 12 that should be controlled to the rectifying state are controlled to the on state, thereby performing synchronous rectification. A second example of the switching patterns and current flow during the boost operation shown in FIGS. 18(a)-(d) and 19(a)-(d) is an example in which the switching elements in the second bridge circuit 12 that should be controlled to the rectifying state are controlled to the off state, thereby performing diode rectification.

[0073] In the first period T1-1 (boost) of the first pattern (boost), the first switching element Q1, the fourth switching element Q4, and the seventh switching element Q7 are in the on state, the fifth switching element Q5 is in the rectifying state, and the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the eighth switching element Q8 are in the off state (see Figures 16(a) and 18(a)).

[0074] In the second period T2-1 (boost) of the first pattern (boost), the first switching element Q1 and the fourth switching element Q4 are in the on state, the fifth switching element Q5 and the eighth switching element Q8 are in the rectifying state, and the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the off state (see Figures 16(b) and 18(b)).

[0075] In the first pattern (boost), the control circuit 13 turns off the seventh switching element Q7, which is forward conductive, during the first period T1-1 (boost), to transition to the second period T2-1 (boost).

[0076] In the first period T1-2 (boost) of the second pattern (boost), the second switching element Q2, the third switching element Q3, and the fifth switching element Q5 are in the on state, the seventh switching element Q7 is in the rectifying state, and the first switching element Q1, the fourth switching element Q4, the sixth switching element Q6, and the eighth switching element Q8 are in the off state (see Figures 16(c) and 18(c)).

[0077] In the second period T2-2 (boost) of the second pattern (boost), the second switching element Q2 and the third switching element Q3 are in the on state, the sixth switching element Q6 and the seventh switching element Q7 are in the rectifying state, and the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the off state (see Figures 16(d) and 18(d)).

[0078] In the second pattern (boost), the control circuit 13 turns off the fifth switching element Q5, which is forward conductive, during the first period T1-2 (boost), to transition to the second period T2-2 (boost).

[0079] In the first period T1-3 (boost) of the third pattern (boost), the first switching element Q1, the fourth switching element Q4, and the sixth switching element Q6 are in the on state, the eighth switching element Q8 is in the rectifying state, and the second switching element Q2, the third switching element Q3, the fifth switching element Q5, and the seventh switching element Q7 are in the off state (see Figures 17(a) and 19(a)).

[0080] In the second period T2-3 (boost) of the third pattern (boost), the first switching element Q1 and the fourth switching element Q4 are in the on state, the fifth switching element Q5 and the eighth switching element Q8 are in the rectifying state, and the second switching element Q2, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the off state (see Figures 17(b) and 19(b)).

[0081] In the third pattern (boost), the control circuit 13 turns off the sixth switching element Q6 that is forward conductive during the first period T1-3 (boost), and transitions to the second period T2-3 (boost).

[0082] In the first period T1-4 (boost) of the fourth pattern (boost), the second switching element Q2, the third switching element Q3, and the eighth switching element Q8 are in the on state, the sixth switching element Q6 is in the rectifying state, and the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the seventh switching element Q7 are in the off state (see Figures 17(c) and 19(c)).

[0083] In the second period T2-4 (boost) of the fourth pattern (boost), the second switching element Q2 and the third switching element Q3 are in the on state, the sixth switching element Q6 and the seventh switching element Q7 are in the rectifying state, and the first switching element Q1, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the off state (see Figures 17(d) and 19(d)).

[0084] In the fourth pattern (boost), the control circuit 13 turns off the eighth switching element Q8, which is forward conductive, during the first period T1-4 (boost), to transition to the second period T2-4 (boost).

[0085] The control circuit 13 supplies the same drive signal with synchronized on / off timing to the first switching element Q1 and the fourth switching element Q4 on one diagonal corner included in the first bridge circuit 11, and supplies the same drive signal with synchronized on / off timing to the second switching element Q2 and the third switching element Q3 on the other diagonal corner.

[0086] The control circuit 13 fixes the duty ratio of the drive signals supplied to the first switching element Q1 and the fourth switching element Q4 to 50% regardless of the amount of power transmission except during dead time. The control circuit 13 fixes the phase difference between the phase of the drive signals supplied to the first switching element Q1 and the fourth switching element Q4 and the phase of the drive signals supplied to the second switching element Q2 and the third switching element Q3 to 180° regardless of the amount of power transmission.

[0087] In this way, in the first bridge circuit 12, by fixing the duty ratio and phase of the drive signal regardless of the amount of power transmission, it is possible to minimize loss and maximize power transmission efficiency.

[0088] The control circuit 13 supplies different drive signals to the fifth switching element Q5 to the eighth switching element Q8 included in the second bridge circuit 12. Specifically, the control circuit 13 changes the duty ratio of the drive signals supplied to two switching elements (e.g., the fifth switching element Q5 and the eighth switching element Q8) on the first diagonal of the fifth switching element Q5 to the eighth switching element Q8, within a range of 0 to 100%, excluding dead time, depending on the amount of power transmission.

[0089] The control circuit 13 changes the duty ratio of the drive signal supplied to two switching elements (e.g., the sixth switching element Q6 and the seventh switching element Q7) on the second diagonal of the fifth switching element Q5 to the eighth switching element Q8 within a range of 0 to 50% in accordance with the amount of power transmission, excluding dead time. The control circuit 13 controls a second phase difference φ2 between the turn-on timing of the drive signal supplied to the two switching elements on the second diagonal and the turn-on timing of the first switching element Q1 to the fourth switching element Q4 in accordance with the amount of power transmission, excluding dead time. More precisely, of the two switching elements on the second diagonal, the switching element with a duty ratio fixed at 50% undergoes a phase shift, while the switching element whose duty ratio varies within a range of 0 to 50% does not undergo a phase shift.

[0090] The control circuit 13 reduces the duty ratio of the drive signal supplied to one of the two switching elements on the first diagonal from 50% except during dead time by an amount corresponding to the second phase difference φ2, and increases the duty ratio of the drive signal supplied to the other of the two switching elements from 50% except during dead time by an amount corresponding to the second phase difference φ2. The control circuit 13 alternately switches between the two switching elements on the first diagonal whose duty ratio is reduced from 50% and the switching element whose duty ratio is increased from 50%. The control circuit 13 fixes the duty ratio of the drive signal supplied to one of the two switching elements on the second diagonal to 50% except during dead time, and reduces the duty ratio of the drive signal supplied to one of the two switching elements from 50% except during dead time by an amount corresponding to the second phase difference φ2. The control circuit 13 alternately switches between the two switching elements on the second diagonal whose duty ratio is fixed at 50% and the switching element whose duty ratio is reduced from 50%.

[0091] In this way, the second bridge circuit 12 simultaneously controls the duty ratio and the phase shift amount according to the amount of power transmission. As a result, as shown in FIG. 15, the turn-off that reverses the increase in the absolute value of the current can be distributed among the fifth switching element Q5 to the eighth switching element Q8, and loss and heat generation can be distributed among the fifth switching element Q5 to the eighth switching element Q8. In Comparative Example 2 shown in FIG. 6, loss and heat generation are concentrated among the fifth switching element Q5 to the sixth switching element Q6.

[0092] FIG. 20 illustrates a specific example 1 in which switching patterns of the first switching element Q1 through the eighth switching element Q8 are periodically generated during boost operation according to the embodiment of the power conversion device 1. In the switching patterns illustrated in FIG. 20 , the control circuit 13 generates each of the first pattern (boost) and the fourth pattern (boost) twice every four switching cycles. In the example illustrated in FIG. 20 , the patterns are generated in the following order: first pattern (boost) → second pattern (boost) → third pattern (boost) → second pattern (boost) → third pattern (boost) → fourth pattern (boost) → first pattern (boost) → fourth pattern (boost). In this specific example 1, the first pattern (boost) and the fourth pattern (boost) are generated the same number of times. This allows the switching loss generated on the secondary side during boost operation to be evenly distributed among the fifth switching element Q5 through the eighth switching element Q8, thereby dispersing heat generation.

[0093] FIG. 21 is a diagram illustrating a specific example 2 in which switching patterns of the first switching element Q1 through the eighth switching element Q8 are generated irregularly during boost operation according to the embodiment of the power conversion device 1. In the switching patterns illustrated in FIG. 21 , the control circuit 13 controls the switching patterns so that the number of occurrences of the first pattern (step-down) and the fourth pattern (step-down) differs. In the example illustrated in FIG. 21 , the first pattern (step-up) is generated twice, the second pattern (step-up) three times, the third pattern (step-up) twice, and the fourth pattern (step-up) once per four switching cycles. Even in the specific example 2, loss dispersion to a certain extent is possible, and the peak heat generation on the secondary side can be suppressed compared to the control illustrated in the comparative example 2.

[0094] 22(a)-(d) are diagrams (part 1) for explaining a third example of a switching pattern and a current flow in a boost operation according to an embodiment of the power conversion device 1. FIG. 23(a)-(d) are diagrams (part 2) for explaining a third example of a switching pattern and a current flow in a boost operation according to an embodiment of the power conversion device 1.

[0095] In the third example (boost), the control circuit 13 controls the switching elements in the second bridge circuit 12 that should be in a rectifying state to the on state during the first period T1 (boost), thereby performing synchronous rectification. The control circuit 13 controls the switching elements in the second bridge circuit 12 that should be in a rectifying state to the off state during the second period T2 (boost), thereby performing diode rectification. Because the conduction loss due to synchronous rectification is smaller than the forward loss Vf due to diode rectification, the loss during the first period T1 (boost) can be reduced.

[0096] The switching pattern shown in FIG. 15 represents a combination of the first example (boost) and the second example (boost). In principle, during the second period T2 (boost), the control circuit 13 controls the two diagonal switching elements of the second bridge circuit 12 that should be in a rectifying state to the on state, thereby performing synchronous rectification. During the second period T2 (boost), when the absolute value of the transformer current IL becomes equal to or less than the threshold value Ith, the control circuit 13 turns off at least one of the two diagonal switching elements to switch to diode rectification. Note that both switching elements may be turned off, although this reduces efficiency.

[0097] If the absolute value of the transformer current IL does not decrease to the threshold value Ith within a predetermined time, the control circuit 13 skips turning off at least one of the two diagonal switching elements included in the second bridge circuit 12. The predetermined time is set to half a cycle Tc / 2 minus the dead time. That is, if the absolute value of the transformer current IL does not decrease to the threshold value Ith within the predetermined time, the two diagonal switching elements included in the second bridge circuit 12 enter a dead time while remaining in the on state. Entering the dead time causes the control circuit 13 to control all of the fifth switching element Q5 through the eighth switching element Q8 to be turned off. If the absolute value of the transformer current IL decreases slowly and there is no risk of power backflow, there is no need to switch from synchronous rectification to diode rectification. On the other hand, if there is a risk of power backflow, as shown in FIG. 15, the control circuit 13 switches from synchronous rectification to diode rectification. If power backflow occurs, a reactive current unrelated to power transmission flows, leading to increased loss. Switching to diode rectification prevents increased loss due to reactive current.

[0098] FIG. 24 is a diagram showing an example of the switching patterns of the first switching element Q1 through the eighth switching element Q8 during reverse transmission during boost operation in the embodiment of the power conversion device 1, and a transition example of the transformer current IL. The switching pattern of the first switching element Q1 through the eighth switching element Q8 shown in FIG. 15 illustrates an example in which power is boosted from the first DC section to the second DC section and transmitted. However, it is also possible to boost power from the second DC section to the first DC section and transmit it. In this case, as shown in FIG. 24, the control circuit 13 simply switches the drive signals supplied to the first switching element Q1 through the fourth switching element Q4 and the drive signals supplied to the fifth switching element Q5 through the eighth switching element Q8.

[0099] (Example (Step-Up / Step-Down)) FIG. 25 is a diagram showing an example of the switching patterns of the first switching element Q1 to the eighth switching element Q8 and the transition of the transformer current IL during step-up / step-down operation according to an example of the power conversion device 1. The step-up / step-down operation is an operation inserted in the period between step-down operation and step-up operation, or between step-up operation and step-down operation. When power is transferred by step-up / step-down from the first DC unit to the second DC unit during the step-up / step-down switching period, the control circuit 13 controls the period to include a first period T1 (step-up), a second period T2 (step-up), and a third period T3. The third period T3 is a period during which both ends of the primary winding n1 of the isolation transformer TR1 are short-circuited within the first bridge circuit 11, and the second bridge circuit 12 enters a commutation state in which the secondary winding n2 of the isolation transformer TR1 and the second DC unit are conductive.

[0100] 25 , in the step-up / step-down operation, the control circuit 13 imposes a phase difference between the drive signal supplied to the first switching element Q1 and the drive signal supplied to the fourth switching element Q4, and also imposes a phase difference between the drive signal supplied to the second switching element Q2 and the drive signal supplied to the third switching element Q3. The period corresponding to this phase difference is the third period T3. In the step-up / step-down operation, the third period T3 (in which the first DC unit and the isolation transformer TR1 are in a cutoff state) is inserted before the end of the second period T2 (in which the first DC unit and the isolation transformer TR1 are in a conduction state), thereby suppressing the energy supplied from the first DC unit to the second DC unit.

[0101] FIG. 26( a) is a schematic diagram illustrating the relationship between the duty ratio and the transmission power when the buck-boost mode is not provided. FIG. 26( b) is a schematic diagram illustrating the relationship between the duty ratio and the transmission power when the buck-boost mode is provided. As shown in FIG. 26( a), when the buck-boost mode is not provided, one inflection point occurs at the time of switching between the buck mode and the boost mode, where the slope of the transmission power (which can be considered as the transmission current) changes. As shown in FIG. 26( b), when the buck-boost mode is provided, two inflection points occur at the time of switching between the buck mode and the buck-boost mode, and at the time of switching between the buck-boost mode and the boost mode. When the buck-boost mode is provided, the number of inflection points increases to two, but the change in the slope of each inflection point is more gradual than the slope of the inflection point when the buck-boost mode is not provided. As shown in FIG. 26( b), by inserting the buck-boost mode between the boost mode and the buck mode, the circuit gain can be smoothly changed and current oscillation can be suppressed.

[0102] FIG. 27 is a diagram showing an example configuration of the control circuit 13. The control circuit 13 includes detectors 131a-c, a subtraction unit 132, a controller 133, a synchronous rectification period calculation unit 135, and a PWM generation unit 136. The detector 131a includes an A / D converter. The A / D converter converts the analog output current value Io detected by the current sensor 23 into a digital output current value at a predetermined sampling rate. The detector 131b includes an A / D converter. The A / D converter converts the analog input voltage value V1 detected by the first voltage sensor 21 into a digital input voltage value at a predetermined sampling rate. The detector 131c includes an A / D converter. The A / D converter converts the analog output voltage value V2 detected by the second voltage sensor 22 into a digital input voltage value at a predetermined sampling rate. The A / D converter may be shared by the detectors 131a-c in a time-sharing manner.

[0103] The subtraction unit 132 calculates a deviation err between the current command value Iref, which should be the target value, and the output current value input from the detector 131. The controller 133 performs PI control of the deviation err using a predetermined control gain to calculate a control operation amount duty. The controller 133 outputs the calculated control operation amount duty to the synchronous rectification period calculation unit 135 and the PWM generation unit 136.

[0104] The synchronous rectification period calculation unit 135 calculates the synchronous rectification period in the second bridge circuit 12 based on the differential voltage (V1-V2) or (V2-V1) between the input voltage V1 detected by the first voltage sensor 21 and the output voltage V2 detected by the second voltage sensor 22.

[0105] 28(a) and 28(b) are graphs showing an example of variables and constants in a theoretical formula for calculating the synchronous rectification period. Hereinafter, with reference to FIG. 28(a), a method for calculating the transformer current IL in the step-down mode and the step-up mode and the synchronous rectification period T2s of the second period T2 will be described.

[0106] The synchronous rectification period calculation unit 135 calculates the transformer current IL for the first period T1 (step-down) using the following (Equation 1): V1-V2=L (IL / T1) IL=(V1-V2)T1 / L (Equation 1) where L is the inductance between the first bridge circuit 11 and the second bridge circuit 12 (in this embodiment, L=L1+L2). T1 is the duration of the first period T1 (step-down) based on feedback control, and in this embodiment, is determined by the control operation amount duty calculated by the controller 133.

[0107] The synchronous rectification period calculation unit 135 calculates the transformer current (Ith-IL) during the synchronous rectification period T2s of the second period T2 (step-down) using the following formula (2): 0-V2=L·((Ith-IL) / T2s) Ith-IL=-V2·T2s / L (formula 2)

[0108] The synchronous rectification period calculation unit 135 calculates the synchronous rectification period T2s of the second period T2 (step-down) using the following (Equation 3): T2s=((V1-V2)T1 / V2)-((L / V2)Ith) (Equation 3) Because the inductance L and the threshold value Ith are constants, the synchronous rectification period T2s of the second period T2 (step-down) can be calculated by detecting the input voltage V1 and the output voltage V2.

[0109] The synchronous rectification period calculation unit 135 calculates the transformer current IL of the first period T1 (voltage step-up) using the following (Equation 4): V1=L (IL / T1) IL=V1T1 / L (Equation 4) where L is the inductance between the first bridge circuit 11 and the second bridge circuit 12 (in this embodiment, L=L1+L2). T1 is the duration of the first period T1 (voltage step-up) based on feedback control, and in this embodiment, is determined by the control operation amount duty calculated by the controller 133.

[0110] The synchronous rectification period calculation unit 135 calculates the transformer current (Ith-IL) during the synchronous rectification period T2s of the second period T2 (boost) using the following equation (5): V1-V2=L·((Ith-IL) / T2s) Ith-IL=(V1-V2)·T2s / L (Equation 5)

[0111] The synchronous rectification period calculation unit 135 calculates the synchronous rectification period T2s of the second period T2 (boost) using the following (Equation 6): T2s=(V1·T1 / (V2−V1))−(L / (V2−V1)·Ith) (Equation 6) Because the inductance L and the threshold value Ith are constants, the synchronous rectification period T2s of the second period T2 (boost) can be calculated by detecting the input voltage V1 and the output voltage V2.

[0112] In the buck-boost mode, when the absolute value of the transformer current IL becomes equal to or less than the threshold value Ith during the second period T2 (boost), the synchronous rectification period T2s during the second period T2 (boost) can be calculated using the above equations 4 to 6. In the buck-boost mode, when the absolute value of the transformer current IL does not become equal to or less than the threshold value Ith during the second period T2 (boost) but becomes equal to or less than the threshold value Ith during the third period T3, a method for calculating the transformer current IL and the synchronous rectification period T3s during the third period T3 will be described with reference to FIG.

[0113] The synchronous rectification period calculation unit 135 calculates the transformer current IL1 for the first period T1 (voltage step-up) in the voltage step-up / buck mode using the following equation (7): V1=L (IL1 / T1) IL1=V1T1 / L (Equation 7) where L is the inductance between the first bridge circuit 11 and the second bridge circuit 12 (in this embodiment, L=L1+L2). T1 is the duration of the first period T1 (voltage step-up) based on feedback control, and in this embodiment, is determined by the control operation amount duty calculated by the controller 133.

[0114] In the step-up / step-down mode, when the absolute value of the transformer current IL during the second period T2 (step-up) does not become equal to or less than the threshold value Ith, the synchronous rectification period calculation unit 135 calculates the transformer current IL2 during the second period T2 (step-up) using the following equation (8): V1-V2=L ((IL2-IL1) / T2) IL2-IL1=(V1-V2) T2 / L (Equation 8)

[0115] In the step-up / step-down mode, when the absolute value of the transformer current IL during the third period T3 is equal to or less than the threshold value Ith, the synchronous rectification period calculation unit 135 calculates the transformer current (Ith-IL2) during the synchronous rectification period T3s of the third period T3 using the following equation (9): -V2=L·((Ith-IL2) / T3s) Ith-IL2=-V2·T2 / L (Equation 9)

[0116] The synchronous rectification period calculation unit 135 calculates the synchronous rectification period T3s of the third period T3 in the buck-boost mode when the absolute value of the transformer current IL is equal to or less than the threshold value Ith during the third period T3, using the following equation (10): T3s=((V1·T1+(V1−V2)·T2) / V2)−(L / V2·Ith) (Equation 10) Because the inductance L and the threshold value Ith are constants, the synchronous rectification period T3s of the third period T3 in the buck-boost mode can be calculated by detecting the input voltage V1 and the output voltage V2.

[0117] The PWM generating unit 136 generates a PWM signal for driving the first switching element Q1 to the eighth switching element Q8 based on the control operation amount duty calculated by the controller 133 and the synchronous rectification period T2s of the second period T2 or the synchronous rectification period T3s of the third period T3 calculated by the synchronous rectification period calculating unit 135.

[0118] As described above, this embodiment makes it possible to equalize the heat generation of multiple switching elements included in the DAB converter. In particular, in step-down mode, the loss and heat generation of the first switching element Q1 through the fourth switching element Q4 can be equalized, and in step-up mode, the loss and heat generation of the fifth switching element Q5 through the eighth switching element Q8 can be equalized. This reduces the peak loss and heat generation of the switching element that generates the largest loss, enabling the miniaturization and cost reduction of components (e.g., heat sinks) used to cool the switching elements. Furthermore, this reduces the variation in lifespan between switching elements, facilitating product management.

[0119] Furthermore, a highly efficient DAB converter can be realized by performing synchronous rectification in the second bridge circuit 12 during the second period T2 (step-down), the second period T2 (step-up), and the third period T3. In this case, if the transformer current IL drops to around 0 A, the second bridge circuit 12 can be switched from synchronous rectification to diode rectification to prevent a decrease in efficiency due to the generation of reactive current. This embodiment is particularly effective for DAB converters using SiC-MOSFETs, which have large parasitic diode losses.

[0120] The present disclosure has been described above based on the embodiments. The embodiments are merely examples, and it will be understood by those skilled in the art that various modifications are possible in the combination of the respective components and processing steps, and that such modifications are also within the scope of the present disclosure.

[0121] In the above embodiment, an example has been described in which the transformer current IL is calculated using the above (Equation 1) to (Equation 10) without installing a current sensor between the first bridge circuit 11 and the isolation transformer TR1, and between the isolation transformer TR1 and the second bridge circuit 12. In this regard, a current sensor may be installed between the first bridge circuit 11 and the isolation transformer TR1, or between the isolation transformer TR1 and the second bridge circuit 12, and the current measured by the current sensor may be defined as the transformer current IL, which is then compared with the threshold value Ith.

[0122] 29(a)-(d) are diagrams (part 1) illustrating another example of switching patterns and current flows in the step-down operation of the embodiment of the power conversion device 1. FIG. 30(a)-(d) are diagrams (part 2) illustrating another example of switching patterns and current flows in the step-down operation of the embodiment of the power conversion device 1.

[0123] In this example, during the second period T2-1 (step-down) in the first pattern (step-down), the first switching element Q1, the third switching element Q3, the fifth switching element Q5, and the eighth switching element Q8 are in the on state, and the second switching element Q2, the fourth switching element Q4, the sixth switching element Q6, and the seventh switching element Q7 are in the off state (see FIG. 29(b)).

[0124] In the second period T2-2 (step-down) of the second pattern (step-down), the first switching element Q1, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the on state, and the second switching element Q2, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the off state (see FIG. 29(d)).

[0125] In the second period T2-3 (step-down) of the third pattern (step-down), the second switching element Q2, the fourth switching element Q4, the fifth switching element Q5, and the eighth switching element Q8 are in the on state, and the first switching element Q1, the third switching element Q3, the sixth switching element Q6, and the seventh switching element Q7 are in the off state (see Figure 30 (b)).

[0126] In the second period T2-4 (step-down) in the fourth pattern (step-down), the second switching element Q2, the fourth switching element Q4, the sixth switching element Q6, and the seventh switching element Q7 are in the on state, and the first switching element Q1, the third switching element Q3, the fifth switching element Q5, and the eighth switching element Q8 are in the off state (see Figure 30 (d)).

[0127] In the second period T2, loss can be reduced to the maximum by performing synchronous rectification on both the primary side and the secondary side.

[0128] The embodiment may be specified by the following items.

[0129] [Item 1] A power supply comprising: a first bridge circuit (11) having a first leg in which a first switching element (Q1) and a second switching element (Q2) are connected in series, and a second leg in which a third switching element (Q3) and a fourth switching element (Q4) are connected in series, the first leg and the second leg being connected in parallel to a first DC unit (E1, Ca); a second bridge circuit (12) having a third leg in which a fifth switching element (Q5) and a sixth switching element (Q6) are connected in series, and a fourth leg in which a seventh switching element (Q7) and an eighth switching element (Q8) are connected in series, the third leg and the fourth leg being connected in parallel to a second DC unit (E2, Cb); an isolation transformer (TR1) connected between the first bridge circuit (11) and the second bridge circuit (12); and a control circuit (13) that controls the first switching element (Q1) to the eighth switching element (Q8), The control circuit (13) controls the second bridge circuit (12) to include a first period in which both ends of the secondary winding (n2) of the isolation transformer (TR1) are short-circuited within the second bridge circuit (12) and a second period in which the secondary winding (n2) of the isolation transformer (TR1) and the second DC unit (E2, Cb) are conductive, when power is transferred by boosting the voltage from the first DC unit (E1, Ca) to the second DC unit (E2, Cb), and controls the first period to include: a first pattern in which the seventh switching element (Q7) conducting in the forward direction is turned off and transition to the second period; a second pattern in which the fifth switching element (Q5) conducting in the forward direction is turned off and transition to the second period; and a third pattern in which the sixth switching element (Q6) conducting in the forward direction is turned off and transition to the second period. and generating a fourth pattern in which the eighth switching element (Q8) conducting in the forward direction is turned off to transition to the second period. This allows the switching loss occurring in the second bridge circuit (12) during boost operation to be leveled and allocated to the fifth switching element (Q5)-eighth switching element (Q8), thereby dispersing heat generation.[Item 2] The power conversion device (1) according to Item 1, wherein the control circuit (13) generates the first pattern and the fourth pattern twice each in units of four switching cycles. This allows the switching loss generated in the second bridge circuit (12) to be evenly allocated to the fifth switching element (Q5) through the eighth switching element (Q8). [Item 3] The power conversion device (1) according to Item 1, wherein the control circuit (13) performs control so that the number of times the first pattern and the fourth pattern are generated differs. This allows the switching loss generated in the second bridge circuit (12) to be evenly allocated to the fifth switching element (Q5) through the eighth switching element (Q8).[Item 4] In the first period in the first pattern, the first switching element (Q1), the fourth switching element (Q4), and the seventh switching element (Q7) are in an on state, and the fifth switching element (Q5) is in a rectifying state; in the second period in the first pattern, the first switching element (Q1) and the fourth switching element (Q4) are in an on state, and the fifth switching element (Q5) and the eighth switching element (Q8) are in a rectifying state; in the first period in the second pattern, the second switching element (Q2), the third switching element (Q3), and the fifth switching element (Q5) are in an on state, and the seventh switching element (Q7) is in a rectifying state; in the second period in the second pattern, the second switching element (Q2) and the third switching element (Q3) are in an on state, and the sixth switching element (Q6) and the seventh switching element (Q7) are in a rectifying state; Item 1. The power conversion device (1) according to item 1, wherein: during the first period in the third pattern, the first switching element (Q1), the fourth switching element (Q4), and the sixth switching element (Q6) are in an on state, and the eighth switching element (Q8) is in a rectifying state; during the second period in the third pattern, the first switching element (Q1) and the fourth switching element (Q4) are in an on state, and the fifth switching element (Q5) and the eighth switching element (Q8) are in a rectifying state; during the first period in the fourth pattern, the second switching element (Q2), the third switching element (Q3), and the eighth switching element (Q8) are in an on state, and the sixth switching element (Q6) is in a rectifying state; and during the second period in the fourth pattern, the second switching element (Q2) and the third switching element (Q3) are in an on state, and the sixth switching element (Q6) and the seventh switching element (Q7) are in a rectifying state. This allows the switching loss occurring in the second bridge circuit (12) to be leveled and allocated to the fifth switching element (Q5) to the eighth switching element (Q8).[Item 5] The control circuit (13) controls the power conversion device (1) according to Item 1, when transferring power from the first DC unit (E1, Ca) to the second DC unit (E2, Cb) during a switching period between voltage step-up and voltage step-down, to include the first period, the second period, and a third period in which both ends of the primary winding (n1) of the isolation transformer (TR1) are short-circuited within the first bridge circuit (11) and the second bridge circuit (12) conducts the secondary winding (n2) of the isolation transformer (TR1) to the second DC unit (E2, Cb). This allows for smooth change in circuit gain during the switching period between voltage step-up and voltage step-down, and suppresses current oscillation. [Item 6] The control circuit (13) supplies drive signals having a duty ratio fixed at 50% to the first switching element (Q1) and the fourth switching element (Q4), supplies drive signals having a duty ratio fixed at 50% to the second switching element (Q2) and the third switching element (Q3), and fixes a first phase difference between the phase of the drive signals supplied to the first switching element (Q1) and the fourth switching element (Q4) and the phase of the drive signals supplied to the second switching element (Q2) and the third switching element (Q3) to 180°. According to this, by fixing the duty ratio and phase of the first switching element (Q1)-fourth switching element (Q4), it is possible to minimize losses in the first bridge circuit (11) and maximize power transmission efficiency. [Item 7] The power conversion device (1) according to Item 4, wherein the control circuit (13) supplies different drive signals to the fifth switching element (Q5) through the eighth switching element (Q8), the duty ratios of which change according to the amount of transmitted power, and controls a second phase difference between the turn-on timing of the drive signals supplied to two switching elements on one diagonal of the fifth switching element (Q5) through the eighth switching element (Q8) and the turn-on timing of the first switching element (Q1) through the fourth switching element (Q4), according to the amount of transmitted power.According to this, by simultaneously controlling the duty ratio and phase difference of the fifth switching element (Q5) and the eighth switching element (Q8) in accordance with the amount of power transmission, it is possible to realize loss dispersion and heat dispersion in the second bridge circuit (12). [Item 8] The control circuit (13) fixes the duty ratio of the drive signal supplied to one of the two switching elements on one diagonal to 50%, and decreases the duty ratio of the drive signal supplied to one of the two switching elements from 50% by an amount corresponding to the second phase difference, and decreases the duty ratio of the drive signal supplied to one of the two switching elements on the other diagonal of the fifth switching element (Q5) and the eighth switching element (Q8) from 50% by an amount corresponding to the second phase difference, and increases the duty ratio of the drive signal supplied to the other of the two switching elements from 50% by an amount corresponding to the second phase difference. According to this, by simultaneously controlling the duty ratios and phase differences of the fifth switching element (Q5) and the eighth switching element (Q8) using the same control variable in accordance with the amount of power transmission, loss distribution and heat distribution in the second bridge circuit (12) can be achieved. [Item 9] The power conversion device (1) according to Item 4, wherein the control circuit (13) controls the switching elements included in the second bridge circuit (12) to be in a rectifying state to an on state during the first period. According to this, by increasing the synchronous rectification period, forward loss due to diodes, which is greater than conduction loss due to synchronous rectification, can be reduced, thereby reducing overall loss. [Item 10] The power conversion device (1) according to Item 4, wherein the control circuit (13) controls two diagonal switching elements included in the second bridge circuit (12) to be in a rectifying state to an on state during the second period, and turns off at least one of the two diagonal switching elements when the absolute value of the current flowing through the isolation transformer (TR1) becomes equal to or less than a predetermined threshold. This makes it possible to increase the synchronous rectification period while suppressing the reactive current in the second period, thereby reducing the overall loss.[Item 11] The power conversion device (1) according to Item 10, wherein the control circuit (13) calculates a period T2s during which two diagonal switching elements included in the second bridge circuit (12) are controlled to an on state during the second period by using a first derivation formula, wherein the first derivation formula is: T2s = (V1 T1 / (V2 - V1)) - (L / (V2 - V1) Ith), where V1 is the voltage of the first DC unit (E1, Ca), V2 is the voltage of the second DC unit (E2, Cb), T1 is the duration of the first period based on feedback control, L is the inductance between the first bridge circuit (11) and the second bridge circuit (12), and Ith is the predetermined threshold. This makes it possible to derive an optimal timing for switching from synchronous rectification to diode rectification during the second period. [Item 12] The power conversion device (1) according to Item 5, wherein the control circuit (13) controls two diagonal switching elements included in the second bridge circuit (12) that should be in a rectifying state to be in an on state during the third period, and turns off at least one of the two diagonal switching elements when the absolute value of the current flowing through the isolation transformer (TR1) becomes equal to or less than a predetermined threshold. This makes it possible to increase the synchronous rectification period while suppressing reactive current during the third period, thereby reducing overall loss. [Item 13] The power conversion device (1) according to Item 12, wherein the control circuit (13) calculates a period T3s during which two diagonal switching elements included in the second bridge circuit (12) are controlled to be in an on state during the third period by using a second derivation formula, wherein the second derivation formula is: T3s = ((V1 T1 + (V1 - V2) T2) / V2) - (L / V2 Ith), where V1 is the voltage of the first DC unit (E1, Ca), V2 is the voltage of the second DC unit (E2, Cb), T1 is the duration of the first period based on feedback control, T2 is the duration of the second period, L is the inductance between the first bridge circuit (11) and the second bridge circuit (12), and Ith is the predetermined threshold. This makes it possible to derive an optimal timing for switching from synchronous rectification to diode rectification during the third period.[Item 14] The power conversion device (1) according to Item 1, wherein, when boosting and transmitting power from the second DC unit (E2, Cb) to the first DC unit (E1, Ca), the control circuit (13) switches the drive signal supplied to the first switching element (Q1) through the fourth switching element (Q4) with the drive signal supplied to the fifth switching element (Q5) through the eighth switching element (Q8). This enables bidirectional operation.

[0130] The present invention can be used in a solar power generation system or a V2H system.

[0131] E1 First DC power supply, E2 Second DC power supply, 1 Power conversion device, 11 First bridge circuit, 12 Second bridge circuit, 13 Control circuit, 21 First voltage sensor, 22 Second voltage sensor, 23 Current sensor, 131 Detector, 132 Subtraction unit, 133 Controller, 135 Synchronous rectification period calculation unit, 136 PWM generation unit, Q1-Q8 Switching elements, D1-D8 Diodes, C1-C8 Capacitance, L1 First inductance, L2 Second inductance, TR1 Isolation transformer, n1 Primary winding, n2 Secondary winding, Ca Primary side capacitor, Cb Secondary side capacitor.

Claims

1. A first bridge circuit having a first leg in which a first switching element and a second switching element are connected in series, and a second leg in which a third switching element and a fourth switching element are connected in series, the first leg and the second leg being connected in parallel to a first DC section; a second bridge circuit having a third leg in which a fifth switching element and a sixth switching element are connected in series, and a fourth leg in which a seventh switching element and an eighth switching element are connected in series, the third leg and the fourth leg being connected in parallel to a second DC section; an isolation transformer connected between the first bridge circuit and the second bridge circuit; and a control circuit for controlling the first switching element to the eighth switching element, wherein when boosting power from the first DC section to the second DC section and transmitting the power, the control circuit controls the second bridge circuit to include a first period in which both ends of the secondary winding of the isolation transformer are short-circuited within the second bridge circuit, and a second period in which the secondary winding of the isolation transformer and the second DC section are conductive, and from the first period, a first pattern in which the seventh switching element conducting in the forward direction turns off and transitions to the second period, a second pattern in which the fifth switching element conducting in the forward direction turns off and transitions to the second period, a third pattern in which the sixth switching element conducting in the forward direction turns off and transitions to the second period, and a fourth pattern in which the eighth switching element conducting in the forward direction turns off and transitions to the second period are generated. A power conversion device.

2. The power conversion device according to claim 1, wherein the control circuit generates the first pattern to the fourth pattern twice each in units of four switching cycles.

3. The power conversion device according to claim 1, wherein the control circuit controls so that the number of occurrences of the first pattern to the fourth pattern is different.

4. In the first period of the first pattern, the first switching element, the fourth switching element, and the seventh switching element are in the on state, and the fifth switching element is in the rectifying state. In the second period of the first pattern, the first switching element and the fourth switching element are in the on state, and the fifth switching element and the eighth switching element are in the rectifying state. In the first period of the second pattern, the second switching element, the third switching element, and the fifth switching element are in the on state, and the seventh switching element is in the rectifying state. In the second period of the second pattern, the second switching element and the third switching element are in the on state, and the sixth switching element and the seventh switching element are in the rectifying state. In the first period of the third pattern, the first switching element, the fourth switching element, and the sixth switching element are in the on state, and the eighth switching element is in the rectifying state. In the second period of the third pattern, the first switching element and the fourth switching element are in the on state, and the fifth switching element and the eighth switching element are in the rectifying state. In the first period of the fourth pattern, the second switching element, the third switching element, and the eighth switching element are in the on state, and the sixth switching element is in the rectifying state. In the second period of the fourth pattern, the second switching element and the third switching element are in the on state, and the sixth switching element and the seventh switching element are in the rectifying state. The power conversion device according to claim 1.

5. The control circuit controls such that, when transmitting power from the first DC section to the second DC section during the switching period between step-up and step-down, it includes the first period, the second period, and a third period in which both ends of the primary winding of the isolation transformer are short-circuited within the first bridge circuit and the second bridge circuit conducts the secondary winding of the isolation transformer to the second DC section. The power conversion device according to claim 1.

6. The control circuit supplies drive signals with a fixed duty ratio of 50% to the first switching element and the fourth switching element, supplies drive signals with a fixed duty ratio of 50% to the second switching element and the third switching element, and fixes a first phase difference between the phase of the drive signal supplied to the first switching element and the fourth switching element and the phase of the drive signal supplied to the second switching element and the third switching element at 180°. The power conversion device according to claim 4.

7. The control circuit supplies different drive signals with duty ratios that change according to the power transmission amount to the fifth switching element - the eighth switching element, and controls a second phase difference between the turn-on timing of the drive signal supplied to one of the two diagonal switching elements of the fifth switching element - the eighth switching element and the turn-on timing of the first switching element - the fourth switching element according to the power transmission amount. The power conversion device according to claim 4.

8. The control circuit fixes the duty ratio of the drive signal supplied to one of the two diagonal switching elements at 50%, decreases the duty ratio of the drive signal supplied to one of the two switching elements by an amount corresponding to the second phase difference from 50%, decreases the duty ratio of the drive signal supplied to one of the two diagonal switching elements of the other diagonal of the fifth switching element - the eighth switching element by an amount corresponding to the second phase difference from 50%, and increases the duty ratio of the drive signal supplied to the other of the two switching elements by an amount corresponding to the second phase difference from 50%. The power conversion device according to claim 7.

9. The control circuit controls the switching element to be in a rectifying state included in the second bridge circuit to be in an on state during the first period. The power conversion device according to claim 4.

10. The control circuit controls two diagonal switching elements to be in a rectifying state included in the second bridge circuit to be in an on state during the second period, and turns off at least one of the two diagonal switching elements when the absolute value of the current flowing through the isolation transformer becomes equal to or less than a predetermined threshold value. The power conversion device according to claim 4.

11. In the second period, the control circuit calculates a period T2s for controlling two diagonal switching elements included in the second bridge circuit to be in an on state using a first derivation formula. The first derivation formula is: T2s = (V1·T1 / (V2 - V1)) - (L / (V2 - V1)·Ith), where V1 is the voltage of the first DC part, V2 is the voltage of the second DC part, T1 is the time of the first period based on feedback control, L is the inductance between the first bridge circuit and the second bridge circuit, and Ith is the predetermined threshold value. The power conversion device according to claim 10.

12. In the third period, the control circuit controls two diagonal switching elements to be in an on state, which should be in a rectifying state, included in the second bridge circuit. When the absolute value of the current flowing through the isolation transformer becomes equal to or less than a predetermined threshold value, at least one of the two diagonal switching elements is turned off. The power conversion device according to claim 5.

13. In the third period, the control circuit calculates a period T3s for controlling two diagonal switching elements included in the second bridge circuit to be in an on state using a second derivation formula. The second derivation formula is: T3s = ((V1·T1 + (V1 - V2)·T2) / V2) - (L / V2·Ith), where V1 is the voltage of the first DC part, V2 is the voltage of the second DC part, T1 is the time of the first period based on feedback control, T2 is the time of the second period, L is the inductance between the first bridge circuit and the second bridge circuit, and Ith is the predetermined threshold value. The power conversion device according to claim 12.

14. When boosting the voltage from the second DC part to the first DC part to transmit power, the control circuit switches the drive signals supplied to the first switching element - the fourth switching element and the drive signals supplied to the fifth switching element - the eighth switching element. The power conversion device according to claim 1.

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