Power conversion device

Abstract

PROBLEM TO BE SOLVED To provide a power conversion device capable of minimizing excitation of switching ripple in a GND common type dual inverter. SOLUTION The present invention comprises: a primary-side inverter connected to one end of a winding of an open-end structure; and a secondary-side inverter connected to the other end of the winding. The differential voltage between the primary-side inverter and the secondary-side inverter is applied to the motor. The primary-side inverter is connected to a DC power supply, the secondary-side inverter is connected to a capacitor, the negative-side power supply lines of the primary-side inverter and the secondary-side inverter are shared, and a control device 6 is provided for eliminating or suppressing the variation of the zero-phase voltage vz which is the potential difference between the zero-phase voltage vz1 of the primary-side inverter and the zero-phase voltage vz2 of the secondary-side inverter.

Description

Power Conversion Device

[0001] The present invention relates to a power conversion device that applies AC output to a motor having open-ended windings with both ends open, using two inverters, a primary inverter and a secondary inverter.

[0002] For example, when driving an electric compressor used in an air conditioning system for an electric vehicle such as an electric car or a hybrid car, a three-phase permanent magnet synchronous motor is used. A three-phase modulation motor control method using an inverter is d and q-axis current i q Vector control, which controls the above, is common.

[0003] Here, when the input voltage of a motor driven by an inverter drops, the number of rotations at which it can be driven decreases, or the torque that can be output at high rotations decreases, so the high-speed driving range tends to become narrower. In the above-mentioned electric compressor for vehicle use, which drives the compression mechanism with a motor housed in a housing, the required range of input voltage has traditionally not been wide, but in recent years, the requirements for the driving range in response to changes in input voltage have become stricter even for electric compressors.

[0004] However, in-vehicle electric compressors use the battery of the electric vehicle as a DC power source, which can cause a drop in input voltage. If the input voltage drops, the torque that can be output will decrease in the high rotation speed range.

[0005] Therefore, as a method for increasing the output voltage to a motor relative to the input voltage, various dual inverter power conversion devices have been proposed, in which a motor with a so-called open-end winding structure (a motor in which the neutral point of the motor is not connected but extended to the outside) having multiple stator windings with both ends open is sandwiched between two inverters, one on the primary side and one on the secondary side, and the differential voltage between the primary inverter and the secondary inverter is applied to the motor to drive it (see, for example, Patent Document 1 and Patent Document 2).

[0006] JP 2006-149153 A JP 2005-33984 A

[0007] In a dual inverter power conversion device that drives a motor with an open-end winding structure, there are two inverters, one on the primary side and one on the secondary side, so there is a degree of freedom in how the voltage command value is distributed to the two inverters.

[0008] Conventional dual inverter power conversion devices can be broadly divided into three types, as shown in Figures 19 to 21. Figure 19 shows a typical common power supply dual inverter power conversion device. In Figure 19, M is a motor equipped with the open-end structure windings described above, INV1 is a primary inverter, and INV2 is a secondary inverter, which sandwich the stator windings of motor M.

[0009] The common power supply dual inverter is the simplest method, in which the primary and secondary sides share a common power supply. Because there is only one power supply, the withstand voltage of all semiconductor elements and passive elements (capacitors, etc.) on the primary and secondary sides must be set according to the power supply voltage. In addition, because there is a path through which zero-phase current flows, measures must be taken to prevent zero-phase current, but both the primary and secondary sides can output active power P and reactive power Q.

[0010] In this method, since a common power supply is used for the primary and secondary sides, half of the voltage command value is output from each of the inverters INV1 and INV2.

[0011] Next, Figure 20 shows a power conversion device with an isolated power supply type dual inverter. In Figure 20, M is a motor with open-ended windings, INV1 is a primary-side inverter, and INV2 is a secondary-side inverter, which sandwich the stator winding of motor M, but in this system, two isolated power supplies are provided on the primary and secondary sides, respectively. Since two isolated power supplies must be provided, there is a concern that the system may become larger, but it has the advantages of not allowing zero-phase current to flow and that both the primary and secondary sides can output active power P and reactive power Q.

[0012] In this method, the primary and secondary sides are driven by separate power supplies (voltages), so the output is determined according to the voltage ratio. For example, if the voltage V of the primary side power supply is dc1is the secondary power supply voltage V dc2 When the voltage command value is 1 / 2 of the voltage command value, the primary side inverter INV1 outputs 1 / 3 of the voltage command value, and the secondary side inverter INV2 outputs 2 / 3 of the voltage command value.

[0013] Next, Fig. 21 shows a floating capacitor type dual inverter power conversion device. In Fig. 21, M is a motor with open-ended windings, INV1 is a primary-side inverter, and INV2 is a secondary-side inverter, which sandwich the stator windings of motor M. In this system, a floating capacitor C is placed in the secondary-side inverter, and the secondary-side voltage is controlled by charging and discharging the capacitor from the motor side.

[0014] The floating capacitor method has the advantage that the secondary side voltage can be controlled and no zero-phase current flows, but on the other hand, it is necessary to prepare an additional drive power supply to drive the switching elements of the secondary side inverter.

[0015] In addition, since the power supply of the secondary inverter is a capacitor, it can only supply reactive power Q. Therefore, the primary inverter INV1 outputs an effective voltage V of the voltage command value. real The secondary inverter INV2 outputs a reactive voltage V imag In addition, the secondary inverter INV2 outputs a reactive voltage V imag is also borne by the primary side inverter INV1.

[0016] In the floating capacitor type dual inverter power conversion device shown in FIG. 21, the primary side inverter INV1 can output active power P and reactive power Q, and the secondary side inverter INV2 can output only reactive power Q. Therefore, with control as shown in the block diagram of FIG. 22, the voltage command values ​​of the primary side inverter INV1 and the secondary side inverter INV2 are distributed according to the following formulas (I) and (II) to drive the motor M.

[0017]

[0018] In addition, in FIG. 22, in formula (I) and formula (II), v d ref is the d-axis voltage command value, vq ref is the q-axis voltage command value, i d is the d-axis current, i q is the q-axis current, θ Im is the motor current phase, V real is the effective voltage, V imag is the reactive voltage, v d1 ref is the d-axis voltage command value of the primary inverter INV1, v q1 ref is the q-axis voltage command value of the primary side inverter INV1, v d2 ref is the d-axis voltage command value of the secondary inverter INV2, v q2 ref is the q-axis voltage command value of the secondary side inverter INV2.

[0019] Furthermore, in a floating capacitor type dual inverter power conversion device, in addition to driving the motor M, it is necessary to control the DC link voltage of the secondary side inverter INV2, i.e., the secondary side voltage Vdc2, and this adds to the voltage vector components required to charge the capacitor C.

[0020] In this case, in the power conversion device of the floating capacitor type dual inverter, since no zero-phase current flows, the motor current I m and the voltage vector, the secondary voltage V dc2 That is, as shown in FIG. 23, the phase difference of the voltage vector between the primary inverter INV1 and the secondary inverter INV2 is controlled, and the secondary voltage V dc2 Control the charging and discharging modes of the battery.

[0021] Specifically, when the output voltage vector V2 of the secondary inverter INV2 is advanced by 90 degrees relative to the current phase and then brought closer to the current phase (+β), the charging mode is achieved, and when the phase is further advanced (-β), the discharging mode is achieved. In addition to this method, there are many other ways to create a phase difference and distribute vectors. This floating capacitor type dual inverter power conversion device has the advantage that it can drive the motor M while boosting the capacitor C provided in the secondary inverter INV2, thereby enabling the voltage that can be applied to the motor M to be even higher than in a common power supply type.

[0022] However, the conventional dual inverter power conversion devices described above all require a large number of components, making it difficult to reduce costs and size. In particular, in a floating capacitor type dual inverter power conversion device, the secondary side inverter INV2 has a floating potential, so it is necessary to create a gate drive power supply (insulated power supply) for the switching elements of the secondary side inverter INV2, which has a different reference potential, in addition to the gate drive power supply for the switching elements of the primary side inverter INV1. This makes it difficult to adopt this type of power conversion device in devices with limited board area, such as on-board electric compressors, in terms of component mounting area and cost.

[0023] Therefore, the applicant previously proposed a power conversion device as shown in FIG. 1. In this system, the positive-side power supply line of the common power supply type shown in FIG. 19 is separated between the primary-side inverter INV1 and the secondary-side inverter INV2, and only the negative-side power supply line is shared, with a capacitor disposed on the secondary side as in the floating capacitor type shown in FIG. 21. Hereinafter, this type of power conversion device will be referred to as a common-GND type dual inverter. In this system, the secondary-side voltage can be controlled in the same manner as the floating capacitor type dual inverter power conversion device described in FIGS. 22 and 23. Furthermore, since the secondary-side voltage can be stepped up or down, the operating range can be expanded by stepping up the secondary-side voltage.

[0024] Furthermore, in a common-ground dual-inverter power conversion device, it is possible to share the power supply for the drive circuit of the secondary-side inverter INV2, just like in a common-power-supply type.The primary-side inverter INV1 on the power supply side can be configured with high-voltage components, and the controllable secondary-side inverter INV2 can be configured with low-voltage components.The common-ground type, which can share the power supply for the drive circuit of the secondary-side inverter INV2 with the primary-side inverter INV1, has great advantages in applications where the power supply voltage fluctuates greatly.

[0025] However, such a common-ground dual inverter power conversion device has a problem of zero-phase current flow. In particular, when the secondary-side voltage is boosted and driven, the instantaneous potential difference due to PWM between the primary-side inverter INV1 and the secondary-side inverter INV2 tends to increase, resulting in zero-phase current ripple. Because this zero-phase impedance is smaller than the three-phase (UVW) impedance, instantaneous zero-phase voltage fluctuations caused by switching between the primary-side inverter INV1 and the secondary-side inverter INV2 excite a large current ripple, which can increase losses and conduction noise. Furthermore, the ripple increases the current amplitude, requiring a larger maximum rated current.

[0026] The present invention has been made to solve the above-mentioned conventional technical problems, and provides a power conversion device that can minimize the excitation of switching ripples in the above-mentioned common-GND dual inverter.

[0027] The power conversion device of the present invention includes a primary inverter connected to one end of an open-ended winding of a motor, and a secondary inverter connected to the other end of the winding, and applies a differential voltage between the primary inverter and the secondary inverter to the motor. The primary inverter is connected to a DC power supply, and the secondary inverter is connected to a capacitor. The negative power supply lines of the primary inverter and the secondary inverter are common, and the zero-phase voltage v of the primary inverter is z1 and the zero-phase voltage v of the secondary inverter z2 The zero-phase voltage V is the potential difference z The present invention is characterized by the inclusion of a control device that eliminates or suppresses fluctuations in the temperature.

[0028] A power conversion device according to a second aspect of the present invention is characterized in that the primary-side inverter and the secondary-side inverter are each composed of a plurality of switching elements, the positive-side input terminal of the primary-side inverter is connected to a positive power supply line of a DC power supply, the negative-side input terminal of the primary-side inverter is connected to a negative power supply line of the DC power supply, the output terminal of the primary-side inverter is connected to one end of a winding, the output terminal of the secondary-side inverter is connected to the other end of the winding, a capacitor is connected between the positive-side input terminal and the negative-side input terminal of the secondary-side inverter, the positive-side input terminal of the secondary-side inverter is not connected to the positive power supply line of the DC power supply, and the negative-side input terminal of the secondary-side inverter is connected to the negative power supply line of the DC power supply, and an AC output is generated from the DC power supply by switching each switching element using a control device.

[0029] The power conversion device of the present invention is characterized in that the control device is a secondary side voltage command value v dc2 ref Based on the zero-phase current command value i z ref and a secondary voltage control unit that generates a zero-phase current command value i z ref Based on the zero-phase voltage command value v z ref and a z-axis current control unit that generates a dq-axis voltage command value v dq ref and generating a primary-side output voltage vector command value and a secondary-side output voltage vector command value from the primary-side output voltage vector command value, and generating a primary-side output voltage command value v for switching the primary-side inverter from the primary-side output voltage vector command value. uvw1 ref and generates a secondary-side output voltage command value v for switching the secondary-side inverter from the secondary-side output voltage vector command value. uvw2 ref and an output voltage command generating unit that generates an output voltage command.

[0030] In the power conversion device of the invention of claim 4, the output voltage command generating unit generates a dq-axis voltage command value v dq ref is the effective voltage V real and reactive voltage V imag The effective voltage V realis the primary side output voltage vector command value, and the reactive voltage V imag is set as the secondary side output voltage vector command value.

[0031] The power conversion device of the present invention is characterized in that the control device is a primary side output voltage command value v uvw1 ref a primary-side modulator that generates a switching signal for the primary-side inverter from the secondary-side output voltage command value v uvw2 ref and a secondary side modulation unit that generates a switching signal for the secondary side inverter from the output of the inverter.

[0032] The power conversion device of the invention of claim 6 is characterized in that in the above invention, the secondary side modulation unit performs either pulse width modulation that outputs only odd voltage vectors during one control period, pulse width modulation that outputs only even voltage vectors during one control period, or pulse width modulation that switches between pulse width modulation that outputs only odd voltage vectors during one control period and pulse width modulation that outputs only even voltage vectors during one control period according to a voltage vector command value to generate a switching signal, thereby eliminating or suppressing fluctuations in the zero-phase sequence voltage.

[0033] The power conversion device of the invention of claim 7 is characterized in that in the above invention, the primary side modulation unit performs pulse width modulation to match the timing of the rise and fall of the phase voltage of a specific phase with the rise and fall of the phase voltage of another phase, thereby generating a switching signal, thereby eliminating or suppressing fluctuations in the zero-phase voltage.

[0034] The power conversion device of the invention of claim 8 is the power conversion device of the invention of claim 6 or claim 7, wherein the output voltage command generating unit generates a zero-phase voltage command value v z ref and the zero-phase voltage v of the secondary inverter z2 The sum of the above is added to the primary side output voltage vector command value.

[0035] The power conversion device of the invention of claim 9 is characterized in that, in the invention of claim 5, the primary side modulation unit and the secondary side modulation unit perform pulse width modulation to match the timing of the rise and fall of the phase voltage of a specific phase with the rise and fall of the phase voltage of another phase, thereby generating a switching signal and eliminating or suppressing fluctuations in the zero-phase voltage.

[0036] In the power conversion device of the present invention, the output voltage command generating unit calculates a zero-phase voltage command value v z ref The feature is that the above is distributed and added.

[0037] The power conversion device of the invention of claim 11 is characterized in that in the above invention, the output voltage command generation unit may have a case where the gain M1 to be distributed to the primary side output voltage vector command value is set to 1 and the gain M2 to be distributed to the secondary side output voltage vector command value is set to 0.

[0038] The power conversion device of the invention of claim 12 is characterized in that in the invention of claim 10, the output voltage command generation unit may have a case where the gain M1 to be distributed to the primary side output voltage vector command value is set to 0 and the gain M2 to be distributed to the secondary side output voltage vector command value is set to 1.

[0039] The power conversion device of the invention of claim 13 is characterized in that, in the invention of claim 5, the primary side modulation unit performs either pulse width modulation that outputs only odd voltage vectors during one control period, pulse width modulation that outputs only even voltage vectors during one control period, or pulse width modulation that switches between pulse width modulation that outputs only odd voltage vectors during one control period and pulse width modulation that outputs only even voltage vectors during one control period according to a voltage vector command value to generate a switching signal, thereby eliminating or suppressing fluctuations in the zero-phase sequence voltage.

[0040] The power conversion device of the invention of claim 14 is characterized in that in the above invention, the secondary side modulation unit performs pulse width modulation to match the timing of the rise and fall of the phase voltage of a specific phase with the rise and fall of the phase voltage of another phase, thereby generating a switching signal, thereby eliminating or suppressing fluctuations in the zero-phase voltage.

[0041] The power conversion device of the present invention according to claim 15 is the power conversion device according to claim 13 or 14, wherein the output voltage command generating unit generates a zero-phase voltage command value v z ref and the zero-phase voltage v of the primary inverter z1 The sum of the above is added to the secondary side output voltage vector command value.

[0042] According to the present invention, in a power conversion device that includes a primary side inverter connected to one end of an open-ended winding of a motor and a secondary side inverter connected to the other end of the winding, and that applies a differential voltage between the primary side inverter and the secondary side inverter to the motor, the primary side inverter is connected to a DC power supply, the secondary side inverter is connected to a capacitor, and the negative side power supply lines of the primary side inverter and the secondary side inverter are shared, thereby using the above-mentioned common-GND dual inverter, the capacitor is charged by the secondary side inverter from the DC power supply via the motor, and AC output is applied to the motor using the voltage charged in the capacitor by the secondary side inverter, making it possible to expand the drivable range in response to changes in input voltage.

[0043] Furthermore, in a power conversion device with a common GND type dual inverter, as in the invention of claim 2, the positive input terminal of the secondary inverter is not connected to the positive power line of the DC power supply, and the negative input terminal of the secondary inverter is connected to the negative power line of the DC power supply, so the secondary inverter does not have a floating potential, and the reference voltages of the primary inverter and the secondary inverter are aligned.

[0044] This eliminates the need to create a gate drive power supply (insulated power supply) for the switching elements of the secondary inverter separately from the gate drive power supply (insulated power supply) for the switching elements of the primary inverter, and the switching elements of the primary inverter and the switching elements of the secondary inverter can be switched using a common gate drive power supply, making it possible to expand the drivable range in response to changes in input voltage without using a separate boost converter, etc.

[0045] Furthermore, since there is no longer a need to create a gate drive power supply for each inverter, the increase in component mounting area can be suppressed, and miniaturization and cost reduction can be achieved. This is extremely effective for devices such as automotive electric compressors, where the input voltage varies greatly, there is a strong demand for cost reduction, and the board area is limited, requiring the power conversion device to be miniaturized.

[0046] In particular, in the present invention, the control device is configured to detect the zero-phase voltage v of the primary inverter. z1 and the zero-phase voltage v of the secondary inverter z2 The zero-phase voltage V is the potential difference z Since the fluctuations in the switching ripple are eliminated or suppressed, particularly when the secondary side voltage is driven in a boosted state, it is possible to minimize the excitation of switching ripple and reduce the current amplitude, preventing an increase in loss and conduction noise and making it unnecessary to increase the maximum rated current.

[0047] Further, as in the invention of claim 3, the control device controls the secondary side voltage command value v dc2 ref Based on the zero-phase current command value i z ref and a secondary voltage control unit that generates a zero-phase current command value i z ref Based on the zero-phase voltage command value v z ref and a z-axis current control unit that generates a dq-axis voltage command value v dq ref and generating a primary-side output voltage vector command value and a secondary-side output voltage vector command value from the primary-side output voltage vector command value, and generating a primary-side output voltage command value v for switching the primary-side inverter from the primary-side output voltage vector command value.uvw1 ref and generates a secondary-side output voltage command value v for switching the secondary-side inverter from the secondary-side output voltage vector command value. uvw2 ref If the configuration has an output voltage command generating unit that generates the above, it becomes possible to perform stable secondary side voltage control and motor drive control.

[0048] In this case, for example, as in the invention of claim 4, the output voltage command generating unit generates the dq axis voltage command value v dq ref is the effective voltage V real and reactive voltage V imag The effective voltage V real is the primary side output voltage vector command value, and the reactive voltage V imag is the secondary side output voltage vector command value.

[0049] Furthermore, as in the invention of claim 5, the control device uvw1 ref a primary-side modulator that generates a switching signal for the primary-side inverter from the secondary-side output voltage command value v uvw2 ref The secondary side modulator generates a switching signal for the secondary side inverter from the above.

[0050] Then, for example, as in the invention of claim 6, the secondary side modulation unit generates a switching signal by executing either pulse width modulation that outputs only odd voltage vectors during one control period, or pulse width modulation that outputs only even voltage vectors during one control period, or pulse width modulation that switches between pulse width modulation that outputs only odd voltage vectors during one control period and pulse width modulation that outputs only even voltage vectors during one control period according to the voltage vector command value, thereby eliminating or suppressing fluctuations in the zero-phase sequence voltage, and making it possible to effectively suppress the excitation of switching ripple.

[0051] In addition, as in the invention of claim 7, the primary side modulation unit generates a switching signal by performing pulse width modulation that matches the rising and falling timing of the phase voltage of a specific phase with the rising and falling timing of the phase voltage of another phase, thereby eliminating or suppressing fluctuations in the zero-phase sequence voltage, making it possible to more effectively suppress the excitation of switching ripple.

[0052] In this case, as in the eighth aspect of the present invention, the output voltage command generating unit generates the zero-phase voltage command value v z ref and the zero-phase voltage v of the secondary inverter z2 The sum of the above is added to the primary side output voltage vector command value.

[0053] Furthermore, as in the invention of claim 9, the primary side modulation unit and the secondary side modulation unit perform pulse width modulation to match the timing of the rise and fall of the phase voltage of a specific phase with the rise and fall of the phase voltage of another phase to generate a switching signal, thereby eliminating or suppressing fluctuations in the zero-phase voltage and effectively suppressing the excitation of switching ripple.

[0054] In this case, as in the invention of claim 10, the output voltage command generating unit adds a zero-phase voltage command value v to the primary side output voltage vector command value and the secondary side output voltage vector command value. z ref and adding them, and as in the invention of claim 11, there is a case where the gain M1 distributed to the primary side output voltage vector command value is 1 and the gain M2 distributed to the secondary side output voltage vector command value is 0, and as in the invention of claim 12, there is a case where the gain M1 distributed to the primary side output voltage vector command value is 0 and the gain M2 distributed to the secondary side output voltage vector command value is 1, thereby increasing the degree of freedom of control. However, if the gain distributed to the inverter with a higher DC link voltage is 1 and the gain distributed to the inverter with a lower DC link voltage is 0, the zero-phase sequence voltage command value v z ref This has the effect of ensuring a certain operating range by reducing the burden on the driver.

[0055] Furthermore, as in the invention of claim 13, the primary side modulation unit generates a switching signal by executing either pulse width modulation that outputs only odd voltage vectors during one control period, or pulse width modulation that outputs only even voltage vectors during one control period, or pulse width modulation that switches between pulse width modulation that outputs only odd voltage vectors during one control period and pulse width modulation that outputs only even voltage vectors during one control period according to a voltage vector command value, thereby eliminating or suppressing fluctuations in the zero-phase sequence voltage and effectively suppressing the excitation of switching ripple.

[0056] In addition, as in the invention of claim 14, the secondary side modulation unit generates a switching signal by performing pulse width modulation that matches the timing of the rise and fall of the phase voltage of a specific phase with the rise and fall of the phase voltage of another phase, thereby eliminating or suppressing fluctuations in the zero-phase sequence voltage, making it possible to more effectively suppress the excitation of switching ripple.

[0057] In this case, as in the invention of claim 15, the output voltage command generating unit generates the zero-phase voltage command value v z ref and the zero-phase voltage v of the primary inverter z1 The sum of the above is added to the secondary side output voltage vector command value.

[0058] 1 is an electrical circuit diagram of a power conversion device with a common-GND dual inverter according to an embodiment of the present invention. FIG. 1 is a block diagram of a control device for the power conversion device of FIG. 1. FIG. 1 is a diagram illustrating an equivalent circuit of a zero-phase-sequence component of the power conversion device of FIG. 1. FIG. 2 is a diagram modeling secondary-side voltage control of the power conversion device of FIG. 1. FIG. 3 is a detailed block diagram of an output voltage command generation unit, a primary-side modulation unit, and a secondary-side modulation unit of the control device of FIG. 2 (Embodiment 1). FIG. 5 is a diagram illustrating a linear output region and a PWM pattern of RSPWM using an odd voltage vector performed by the secondary-side modulation unit of FIG. 5. FIG. 6 is a diagram illustrating a linear output region and a PWM pattern of RSPWM using an even voltage vector performed by the secondary-side modulation unit of FIG. 5. FIG. 7 is a diagram illustrating an effect of the control device in the case of FIG. 5. FIG. 8 is a diagram illustrating a zero-phase-sequence current by the control device in the case of FIG. 5. FIG. 9 is a detailed block diagram of an output voltage command generation unit, a primary-side modulation unit, and a secondary-side modulation unit of another embodiment of the control device of FIG. 2 (Embodiment 2). FIG. 10 is a diagram illustrating a PWM pattern of ZFCPWM performed by the primary-side modulation unit of FIG. 11. FIG. 11 is a diagram illustrating an effect of the control device in the case of FIG. 10. 17 is an electrical circuit diagram of a power conversion device of a conventional common power supply type dual inverter. 18 is an electrical circuit diagram of a power conversion device of a conventional isolated power supply type dual inverter. 19 is an electrical circuit diagram of a power conversion device of a conventional floating capacitor type dual inverter. 20 is a block diagram for explaining a drive control method for the floating capacitor type dual inverter of FIG. 21. 21 is a diagram for explaining a charge mode and a discharge mode of the capacitors in the floating capacitor type dual inverter of FIG. 21.

[0059] 1 is an electrical circuit diagram of a power conversion device 1 of a common ground type dual inverter according to an embodiment of the present invention. The power conversion device 1 of the embodiment converts a DC voltage V dc (for example, DC 500V) is converted into a three-phase AC voltage (AC output) and supplied to the motor M. The motor M in this embodiment is a three-phase permanent magnet synchronous motor (Interior Permanent Magnet Synchronous Motor) that is composed of a stator with windings and a rotor with a built-in magnet that rotates inside the stator, and drives an electric compressor used in an air conditioning system for an electric vehicle such as an electric car or a hybrid car. dc is the primary voltage V dc1 is.

[0060] In this embodiment, the power conversion device 1 is configured with a three-phase primary side inverter INV1 consisting of upper and lower arm switching elements 3A to 3F, a three-phase secondary side inverter INV2 consisting of upper and lower arm switching elements 4A to 4F, a control device 6 (FIG. 2), etc. In this embodiment, each of the switching elements 3A to 3F and 4A to 4F is configured with an insulated gate bipolar transistor (IGBT) incorporating a MOS structure in the gate portion.

[0061] In this invention, the DC voltage V of the DC power source 2 is generated by the primary side inverter INV1 and the secondary side inverter INV2. dc is converted into a three-phase AC voltage (AC output), and the differential voltage between inverters INV1 and INV2 is applied to windings (stator windings) 7U, 7V, and 7W of motor M. Here, windings 7U, 7V, and 7W of motor M have an open-ended structure that is not bundled at the neutral point.

[0062] (2) Primary-Side Inverter INV1 The primary-side inverter INV1 has a U-phase half-bridge circuit 9U, a V-phase half-bridge circuit 9V, and a W-phase half-bridge circuit 9W, and each of the half-bridge circuits 9U to 9W for each phase has the above-mentioned upper-arm switching elements 3A to 3C and lower-arm switching elements 3D to 3F. Furthermore, each of the switching elements 3A to 3F has a built-in freewheeling diode connected in antiparallel.

[0063] The collector electrodes of the upper arm switching elements 3A to 3C, which are the positive input terminals of the primary side inverter INV1, are connected to a positive side power supply line 11 (HV+) of the DC power supply 2. On the other hand, the emitter electrodes of the lower arm switching elements 3D to 3F, which are the negative input terminals of the primary side inverter INV1, are connected to a negative side power supply line 12 (HV-) of the DC power supply 2. In the figure, reference numeral 13 denotes a capacitor connected between the positive side power supply line 11 and the negative side power supply line 12, which constitutes a noise filter.

[0064] In the primary inverter INV1, the emitter electrode of the upper arm switching element 3A and the collector electrode of the lower arm switching element 3D of the U-phase half-bridge circuit 9U are connected, and their connection point (arm midpoint: output end of the primary inverter INV1) is connected to one end of the U-phase winding 7U of the motor M.

[0065] In addition, the emitter electrode of the upper arm switching element 3B and the collector electrode of the lower arm switching element 3E of the V-phase half bridge circuit 9V are connected, and their connection point (arm midpoint: output terminal of the primary side inverter INV1) is connected to one end of the V-phase stator winding 7V of the motor M.

[0066] Furthermore, the emitter electrode of the upper arm switching element 3C and the collector electrode of the lower arm switching element 3F of the W-phase half-bridge circuit 9W are connected, and their connection point (arm midpoint: output terminal of the primary side inverter INV1) is connected to one end of the W-phase stator winding 7W of the motor M.

[0067] (3) Secondary-side inverter INV2 The secondary-side inverter INV2 also includes three half-bridge circuits 16U, 16V, and 16W corresponding to the phases U, V, and W. The neutral points of the phases of the motor M are not bundled together, and the windings 7U to 7W of the motor M are sandwiched between the half-bridge circuits 9U to 9W of the primary-side inverter INV1 and the half-bridge circuits 16U to 16W of the secondary-side inverter INV2.

[0068] The half-bridge circuits 16U to 16W of the secondary-side inverter INV2 also have upper-arm switching elements 4A to 4C and lower-arm switching elements 4D to 4F, respectively. Each of the switching elements 4A to 4F also incorporates a free-wheeling diode connected in antiparallel.

[0069] A capacitor C is connected between the collector electrodes of the upper arm switching elements 4A to 4C, which are the positive input terminals of the secondary inverter INV2, and the emitter electrodes of the lower arm switching elements 4D to 4F, which are the negative input terminals of the secondary inverter INV2.

[0070] However, the collector electrodes of the upper arm switching elements 4A to 4C, which are the positive input terminals of the secondary side inverter INV2, are not connected to but separated from the positive power supply line 11 of the DC power supply 2. On the other hand, in the present invention, the emitter electrodes H of the lower arm switching elements 4D to 4F, which are the negative input terminals of the secondary side inverter INV2, are connected to the negative power supply line 12 of the DC power supply 2.

[0071] The emitter electrode of the upper arm switching element 4A and the collector electrode of the lower arm switching element 4D of the U-phase half-bridge circuit 16U are connected, and their connection point (arm midpoint: output terminal of the secondary side inverter INV2) is connected to the other end of the U-phase winding 7U of the motor M.

[0072] In addition, the emitter electrode of the upper arm switching element 4B and the collector electrode of the lower arm switching element 4E of the V-phase half bridge circuit 16V are connected, and their connection point (arm midpoint: output terminal of the secondary side inverter INV2) is connected to the other end of the V-phase stator winding 7V of the motor M.

[0073] Furthermore, the emitter electrode of the upper arm switching element 4C and the collector electrode of the lower arm switching element 4F of the W-phase half bridge circuit 16W are connected, and their connection point (arm midpoint: output end of the secondary side inverter INV2) is connected to the other end of the W-phase stator winding 7W of the motor M. In other words, the power conversion device 1 of the present invention has the above-mentioned common GND type dual inverter configuration.

[0074] (4) Control Device 6 Next, Fig. 2 shows a block diagram of the control device 6. The control device 6 of the embodiment is composed of a microcomputer having a processor, and receives a speed command value ω from the ECU of the electric vehicle. rm ref and the secondary voltage command value v dc2 ref and the phase current of the motor M is input from a current sensor (not shown), and based on these, the ON / OFF states of the switching elements 3A to 3F and 4A to 4F of the primary inverter INV1 and the secondary inverter INV2 are controlled (switching).

[0075] Specifically, it controls the gate voltages applied to the gates of the switching elements 3A to 3F and 4A to 4F. The control device 6 of the embodiment also includes a speed control unit 21, a dqz-axis current control unit 22 serving as the z-axis current control unit of the present invention, an output voltage command generation unit 23, a primary-side modulation unit 24, a secondary-side modulation unit 26, a secondary-side voltage control unit 27, etc.

[0076] The speed control unit 21 performs PI calculation and q-axis current i q and torque, the q-axis current command value i q ref The secondary voltage control unit 27 calculates and outputs the secondary voltage command value v dc2 ref Based on the zero-phase current command value i z ref Generate.

[0077] The dqz-axis current control unit 22 calculates the d-axis voltage command value v by PI calculation and non-interference control. d ref and the q-axis voltage command value v q refIn this case, the d-axis current control unit 22 basically calculates and outputs the d-axis current command value i d ref and d-axis current i d (estimated value), q-axis current command value i q ref and q-axis current i q The d-axis voltage command value v in the direction that eliminates the deviation from the (estimated value) d ref and the q-axis voltage command value v q ref is calculated.

[0078] In addition, the dqz-axis current control unit 22 as the z-axis current control unit in the present invention controls the secondary voltage V dc2 is the secondary voltage command value V dc2 ref The secondary DC link current required for controlling the z The zero-phase voltage command value v z ref Generate and output.

[0079] Then, the output voltage command generating unit 23 calculates the d-axis current i d (estimated value) and q-axis current i q (estimated value), the d-axis voltage command value v output by the dqz-axis current control unit 22 d ref and the q-axis voltage command value v q ref , the zero-phase voltage command value v z ref to the primary side output voltage command value v for switching each of the switching elements 3A to 3F of the primary side inverter INV1. u1 ref , v v1 ref , v w1 ref and a secondary-side output voltage command value v for switching the respective switching elements 4A to 4F of the secondary-side inverter INV2. u2 ref , v v2 ref , v w2 refThe operations of the secondary side voltage control unit 27, the dqz axis current control unit 22, and the output voltage command generation unit 23 will be described in detail later.

[0080] The primary side modulator 24 outputs a primary side output voltage command value v u1 ref , v v1 ref , v w1 ref The secondary side modulator 26 generates and outputs a primary side inverter switching signal (PWM signal) for switching (PWM control) each of the switching elements 3A to 3F of the primary side inverter INV1. u2 ref , v v2 ref , v w2 ref The primary side modulator 24 and the secondary side modulator 26 generate and output secondary side inverter switching signals (PWM signals) for switching (PWM control) the switching elements 4A to 4F of the secondary side inverter INV2. The operations of the primary side modulator 24 and the secondary side modulator 26 will also be described in detail later.

[0081] (5) Configurations and Operations of the dqz-axis current control unit 22, output voltage command generation unit 23, primary side modulation unit 24, and secondary side modulation unit 25 Next, the operations of the dqz-axis current control unit 22, output voltage command generation unit 23, primary side modulation unit 24, and secondary side modulation unit 25 of this embodiment will be described with reference to Figures 3 to 9. Here, the equivalent circuit of the zero-phase sequence component of the power conversion device 1 of Figure 1 is shown in the lower part of Figure 3. This equivalent circuit in the lower part of Figure 3 can be regarded as a chopper circuit of a step-down chopper and a step-up chopper.

[0082] In addition, V in the figure dc1 is the primary voltage, which is the DC voltage of the DC power supply 2. dc2 is the capacitance of the capacitor C, and L z is the self-inductance of the z-axis of the motor M, R a is the internal resistance.

[0083] From this equivalent circuit, the zero-phase voltage v of the primary inverter INV1 z1 and the zero-phase voltage v of the secondary inverter INV2 z2The zero-phase voltage V is the potential difference z (=v z1 -v z2 ) to calculate the zero-phase current i z By controlling the secondary voltage v dc2 Therefore, the control device 6 of the present invention can control the zero-phase current i z is used as an index of the secondary DC link current. This is the zero-phase current i z is approximately equal to the secondary DC link current.

[0084] As described above, assuming that the zero-phase component (z-axis) of the power conversion device 1 can be expressed equivalently to a chopper circuit, the controller and the controlled object (plant model) of the power conversion device 1 in FIG. 1 are modeled as shown in FIG. 4. In this case, the plant model of the controlled object is expressed as the voltage (v z -e z ) to the current (zero-phase current i z ) model (shown as 31 in the figure), an LR load is assumed as in the dq axis, and the zero-phase current i z to the secondary voltage V dc2 The model for ω (shown as 32 in the figure) is modeled as a simple capacitor. re is the electrical angular velocity, θ re is the rotor position.

[0085] Regarding the control system, the voltage PI control system 33 that controls the secondary side voltage controls the zero-phase current command value i z ref and outputs the zero-phase current i z The current PI control system 34 controls the zero-phase voltage command value v z ref The induced voltage component e is generated on the z-axis by the rotation of the motor M. z The non-interfering voltage command value e z est The zero-phase voltage command value v z ref Add to the zero-phase voltage v z This makes it possible to design a controller for the power conversion device 1 of FIG.

[0086] In addition, the secondary voltage V dc2 The control of the zero-phase current i z In the model of Figure 4, PI control is applied, so the secondary voltage V dc2 However, the non-interacting voltage command value e z est Due to parameter errors, etc., the zero-phase current i z When a ripple occurs in the secondary voltage V dc2 Since ripples are also excited, advanced current control can be achieved by adopting a controller that suppresses AC components (periodic disturbance components) from the viewpoint of voltage quality, such as periodic disturbance suppression control, sine wave tracking control, or repetitive control.

[0087] A block diagram of the output voltage command generating unit 23 designed by the above modeling is shown in Fig. 5. The voltage PI control system 33 in Fig. 4 constitutes the secondary side voltage control unit 27 in Fig. 2, and this secondary side voltage control unit 27 controls the zero-phase current command value i z ref (Secondary voltage V dc2 4 constitutes a part of the dqz-axis current control unit 22 in FIG. 2, and this dqz-axis current control unit 22 controls the zero-phase current i z (secondary DC link current) is expressed as the zero-phase current command value i z ref (secondary DC link current command value) z ref Output.

[0088] In this embodiment, the current PI control system 34 constituting the dqz axis current control unit 22 receives the zero-phase current command value i z ref From the zero-phase current i z The current PI control system 34 of the dqz-axis current control unit 22 receives the zero-phase current command value i z ref and zero-phase current i z The zero-phase voltage command value v in the direction of eliminating the difference zref Calculate the zero-phase current i z The zero-phase current command value i z ref Control to.

[0089] The block shown on the left side of Fig. 5 is the output voltage command generating unit 22 of Fig. 2. The basic control block of this output voltage command generating unit 22 is the same as that of the floating capacitor type dual inverter power conversion device shown in Fig. 22. That is, the output voltage command generating unit 22 of this embodiment has an active / inactive decomposition unit 36, and this active / inactive decomposition unit 36 ​​calculates the d-axis voltage command value v d ref and the q-axis voltage command value v q ref , the effective voltage V real and reactive voltage V imag This effective voltage V real is an example of a primary side output voltage vector command value in the present invention, and the reactive voltage V imag is an example of a secondary side output voltage vector command value in the present invention.

[0090] The output voltage command generating unit 22 calculates the effective voltage V decomposed as above using the above-mentioned formula (II). real is the d-axis voltage command value v of the primary inverter INV1. d1 ref and the q-axis voltage command value v of the primary inverter INV1 q1 ref is set to 0. In addition, the d-axis voltage command value v d2 ref is set to 0, and the reactive voltage V imag The q-axis voltage command value v of the secondary inverter INV2 q2 ref Let's say.

[0091] Next, the dq-uvw conversion unit 37 of the output voltage command generation unit 22 converts the d-axis voltage command value v d1 ref and the q-axis voltage command value v of the primary side inverter INV1. q1 refinto output voltage command values ​​for each phase u, v, and w of the primary inverter INV1. Also, a dq-uvw conversion unit 38 converts the d-axis voltage command value v d2 ref and the q-axis voltage command value v of the secondary inverter INV2. q2 ref , the secondary output voltage command value v u2 ref , v v2 ref , v w2 ref Output as

[0092] In addition, the output voltage command generating unit 22 adds the zero-phase voltage command value v outputted from the dqz-axis current control unit 22 to the adder 42. z ref and the zero-phase voltage v of the secondary-side inverter INV2 from the secondary-side conversion unit 26. z2 The sum is added by an adder 41 to the output voltage command values ​​of the uvw phases of the primary inverter INV1 output by the dq-uvw conversion unit 37, to obtain the primary output voltage command value v u1 ref , v v1 ref , v w1 ref Output as

[0093] That is, the effective voltage V as the primary side output voltage vector command value real (Actually, this is converted into the output voltage command value for each phase uvw) and the zero-phase voltage command value v z ref and the zero-phase voltage v of the secondary inverter INV2 z2 By adding the total value of the above, the primary side output voltage command value v u1 ref , v v1 ref , v w1 ref Generate and output.

[0094] In this way, the output voltage command generating unit 22 calculates the d-axis voltage command value v d ref and the q-axis voltage command value v q ref From this, the effective voltage V as the primary side output voltage vector command valuereal and the reactive voltage V as the secondary output voltage vector command value imag and generates an effective voltage V real The above-mentioned sum value is added to the primary side output voltage command value v for switching the primary side inverter INV1. u1 ref , v v1 ref , v w1 ref and generates a reactive voltage V imag The secondary side output voltage command value v for switching the secondary side inverter INV2 is u2 ref , v v2 ref , v w2 ref Generate and output.

[0095] In this embodiment, the primary side modulator 24 modulates the primary side output voltage command value v outputted by the output voltage command generator 22. u1 ref , v v1 ref , v w1 ref A primary side inverter switching signal (PWM signal) S is generated to perform general three-phase modulation (hereinafter referred to as CPWM) to switch (PWM control) each of the switching elements 3A to 3F of the primary side inverter INV1. u1 , S v1 , S w1 Generate and output.

[0096] (5-1) RSPWM by odd voltage vector On the other hand, the secondary side modulation unit 26 of this embodiment uses the secondary side output voltage command value v u2 ref , v v2 ref , v w2 ref From the above, a secondary inverter switching signal (PWM signal) S for switching (PWM control) each of the switching elements 4A to 4F of the secondary inverter INV2 by RSPWM (Remote State PWM) using odd voltage vectors is generated. u2 , S v2 , S w2 Generate and output.

[0097] The RSPWM using odd voltage vectors is a pulse width modulation in which only the odd voltage vectors V1, V3, and V5 shown in FIG. 6 are output during one control period among the eight voltage vectors (output basic vectors) V0 to V7. u is the U-phase voltage of the motor M, v v is the V phase voltage, v w is the W-phase voltage, and v z is the zero-phase voltage (in this embodiment, since RSPWM is performed in the secondary side modulation unit 26, the zero-phase voltage v of the secondary side inverter INV2 is actually z2 )

[0098] (5-2) RSPWM by Even Voltage Vectors In addition to the RSPWM by odd voltage vectors as described above, the secondary side modulation unit 26 may also use the secondary side output voltage command value v u2 ref , v v2 ref , v w2 ref From this, a secondary side inverter switching signal (PWM signal) S for switching (PWM control) each of the switching elements 4A to 4F of the secondary side inverter INV2 by RSPWM using an even voltage vector is generated. u2 , S v2 , S w2 may be generated and output.

[0099] RSPWM using even voltage vectors is pulse width modulation in which, of the eight voltage vectors V0 to V7, only the even voltage vectors V2, V4, and V6 shown in Fig. 7 are output during one control period. In addition to the RSPWM using even voltage vectors and the RSPWM using odd voltage vectors described above, RSPWM using all voltage vectors may be executed by switching between RSPWM using odd voltage vectors and RSPWM using even voltage vectors according to a voltage vector command value.

[0100] According to the RSPWM described above, the zero-phase voltage v z However, the fluctuation of the zero-phase voltage v due to the modulation method can be eliminated. z (In this embodiment, the zero-phase voltage vz2 ) is a fixed value, so as mentioned above, the zero-phase voltage v of the secondary inverter INV2 z2 and the zero-phase voltage command value v z ref The sum of these values ​​is calculated as the zero-phase voltage v of the primary inverter INV1. z1 and the adder 41 adds it to the output voltage command values ​​of the uvw phases of the primary inverter INV1 outputted from the dq-uvw conversion unit 37 to obtain the primary output voltage command value v u1 ref , v v1 ref , v w1 ref Let's say.

[0101] The top row of FIG. 8 shows the zero-phase voltage v of the primary inverter INV1 in this embodiment. z1 , the second row from the top is the zero-phase voltage v of the secondary inverter INV2 z2 , the third row from the top is the zero-phase voltage v z1 -Zero-phase voltage v z2 The zero-phase voltage V is derived from z , the bottom row is the zero-phase current i z The zero-phase voltage v of the primary inverter INV1 is shown. z1 fluctuates six times due to CPWM, but the zero-phase voltage v of the secondary inverter INV2 z2 The fluctuation of the zero-phase voltage v of the primary inverter INV1 driven by CPWM is completely suppressed and becomes a constant value due to the effect of RSPWM. z1 The zero-phase current i z The switching ripple is excited.

[0102] FIG. 9 shows the zero-phase current i z Here, when CPWM is performed in both the primary side modulation unit 24 and the secondary side modulation unit 26, the values ​​corresponding to the values ​​shown in FIG. 8 are as shown in FIG. 17, and the zero-phase current i z The amplitude of the zero-phase voltage v of the primary inverter INV1 is as shown in FIG. z1 and the zero-phase voltage v of the secondary inverter INV2 z2 fluctuates six times due to CPWM, and four levels of fluctuation amplitude are excited, so the zero-phase current i z18 is 100%, the amplitude in the case of FIG. 9 is suppressed to 83.6%.

[0103] As described above, the power conversion device 1 of the present invention is a common-GND dual inverter, and by charging the capacitor C from the DC power source 2 via the motor M by the secondary-side inverter INV2, and applying an AC output to the motor M using the voltage charged in the capacitor C by the secondary-side inverter INV2, it is possible to expand the drivable range in response to changes in the input voltage.

[0104] Furthermore, since the positive input terminal of the secondary side inverter INV2 is not connected to the positive power supply line 11 of the DC power supply 2 and the negative input terminal of the secondary side inverter INV2 is connected to the negative power supply line 12 of the DC power supply 2, the secondary side inverter INV2 does not have a floating potential and the reference voltages of the primary side inverter INV1 and the secondary side inverter INV2 become the same.

[0105] This eliminates the need to create a gate drive power supply for the switching elements 4A to 4F of the secondary inverter INV2 separate from the gate drive power supply for the switching elements 3A to 3F of the primary inverter INV1, and since the switching elements 3A to 3F of the primary inverter INV1 and the switching elements 4A to 4F of the secondary inverter INV2 can be switched using a common gate drive power supply, it becomes possible to expand the drivable range in response to changes in input voltage without using a separate boost converter, etc.

[0106] Furthermore, since there is no longer a need to create a gate drive power supply for each inverter, the increase in component mounting area can be suppressed, and miniaturization and cost reduction can be achieved. This is extremely effective for devices such as automotive electric compressors, where the input voltage varies greatly, there is a strong demand for cost reduction, and the board area is limited, requiring the power conversion device to be miniaturized.

[0107] In particular, in the present invention, the control device 6 controls the zero-phase voltage v z1 and the zero-phase voltage v of the secondary inverter INV2 z2The zero-phase voltage V is the potential difference z Since the fluctuation of the secondary voltage V dc2 When driven in a boosted state, it is possible to minimize the excitation of switching ripple and reduce the current amplitude, preventing an increase in loss and conduction noise and making it unnecessary to increase the maximum rated current.

[0108] In addition, the control device 6 determines the secondary side voltage command value v dc2 ref Based on the zero-phase current command value i z ref and a secondary side voltage control unit 27 that generates a zero-phase current command value i z ref Based on the zero-phase voltage command value v z ref and a dqz-axis current control unit 22 that generates a dq-axis voltage command value v d ref , v q ref to the effective voltage V real (primary side output voltage vector command value) and reactive voltage V imag (secondary side output voltage vector command value) and generates an effective voltage V real The primary side output voltage command value v for switching the primary side inverter INV1 is u1 ref , v v1 ref , v w1 ref and generates a reactive voltage V imag The secondary side output voltage command value v for switching the secondary side inverter INV2 is u2 ref , v v2 ref , v w2 ref Since the output voltage command generating unit 23 is configured to generate a stable secondary side voltage V dc2 and the drive control of the motor M.

[0109] In this embodiment, the secondary side modulation unit 26 performs one of pulse width modulation (RSPWM) for outputting only odd voltage vectors during one control period, pulse width modulation (RSPWM) for outputting only even voltage vectors during one control period, and pulse width modulation (all vectors RSPWM) for switching between pulse width modulation for outputting only odd voltage vectors during one control period and pulse width modulation for outputting only even voltage vectors during one control period according to a voltage vector command value, to generate a switching signal S u2 , S v2 , S w2 Since it is generated, the zero-phase voltage v z Therefore, the fluctuation of the switching frequency is suppressed, and the excitation of the switching ripple can be effectively suppressed.

[0110] (6) Configurations and operations of the output voltage command generating unit 23, the primary side modulator 24, and the secondary side modulator 25 (another embodiment) Next, a power conversion device 1 according to another embodiment of the present invention will be described with reference to Fig. 10 to Fig. 13. In this embodiment, the primary side modulator 24 of the embodiment shown in Fig. 5 executes ZFCPWM as shown in Fig. 10, rather than the general three-phase modulation (CPWM) described above. Other configurations in Fig. 10 are the same as those in Fig. 5.

[0111] (6-1) ZFCPWM That is, the primary side modulation unit 24 of this embodiment modulates the secondary side output voltage command value v outputted by the output voltage command generation unit 22. u2 ref , v v2 ref , v w2 ref From the primary inverter INV1, a primary inverter switching signal (PWM signal) S is generated to switch (PWM control) each of the switching elements 3A to 3F of the primary inverter INV1 by ZFCPWM (Zero Voltage Fluctuation Cancel PWM). u1 , S v1 , S w1 Generate and output.

[0112] This ZFCPWM is a pulse width modulation that synchronizes the rising and falling timing of the phase voltage of a specific phase with the rising and falling timing of the phase voltage of another phase. In other words, the switching timing is synchronized in opposite phase twice during one carrier period. For example, as shown in FIG. 11, when the U-phase voltage v u At the rising timing of V phase voltage v v By matching the falling timing of the U phase voltage v u At the falling edge of w Adjust the timing of the rising edge.

[0113] In ZFCPWM, the zero-phase voltage v z (In this example, v z1 ) fluctuations up to two times, the average zero-phase voltage v z can be arbitrarily specified, the modulatable region becomes the entire hexagon (vector) depending on the PWM output degree of freedom.

[0114] The top row of FIG. 12 shows the zero-phase voltage v of the primary inverter INV1 in this embodiment. z1 , the second row from the top is the zero-phase voltage v of the secondary inverter INV2 z2 , the third row from the top is the zero-phase voltage v z1 -Zero-phase voltage v z2 The zero-phase voltage V is derived from z , the bottom row is the zero-phase current i z The zero-phase voltage v of the primary inverter INV1 is shown. z1 The number of fluctuations (two fluctuations) and the fluctuation level are significantly reduced by ZFCPWM compared to CPWM in the case of FIG.

[0115] In this embodiment, the zero-phase voltage v of the secondary inverter INV2 z2 The fluctuation of the zero-phase voltage v of the primary inverter INV1 driven by ZFCPWM is completely suppressed and becomes a constant value due to the effect of RSPWM. z1 The zero-phase current i z The switching ripple is excited.

[0116] FIG. 13 shows the zero-phase current i zIn this example, the amplitude of the zero-phase current i z The excitation of the switching ripple in is excited only by the ZFCPWM of the primary side inverter INV1, but since the fluctuation level and number of fluctuations are small, if the amplitude in the method shown in FIG. 18 is taken as 100%, it can be seen that the amplitude in the case of FIG. 13 is greatly suppressed to 14.7%, which is an improvement.

[0117] (7) Configurations and operations of the output voltage command generating unit 23, the primary side modulator 24, and the secondary side modulator 25 (another embodiment) Next, a power conversion device 1 according to another embodiment of the present invention will be described with reference to Figs. 14 to 16. In this embodiment, the secondary side modulator 26 of the embodiment shown in Fig. 10 executes the ZFCPWM described above as shown in Fig. 14, rather than the RSPWM described above. In Fig. 14, components denoted by the same reference numerals as those in Fig. 10 perform the same or similar functions.

[0118] That is, both the primary side modulator 24 and the secondary side modulator 26 execute ZFCPWM. In this embodiment, both the primary side modulator 24 and the secondary side modulator 26 operate in the zero-phase-sequence mode. z Since it is possible to output any desired zero-phase voltage v z Therefore, the zero-phase voltage command value v z ref is distributed to the primary side inverter INV1 and the secondary side inverter INV2.

[0119] (7-1) Zero-phase-sequence voltage distribution unit 39 That is, in the power conversion device 1 of this embodiment, the output voltage command generation unit 23 has a zero-phase-sequence voltage distribution unit 39 (FIG. 14). This zero-phase-sequence voltage distribution unit 39 divides the zero-phase-sequence voltage command value v z ref is distributed to the inverters INV1 and INV2 with a gain M1 for distributing to the primary side inverter INV1 and a gain M2 for distributing to the secondary side inverter INV2.

[0120] Then, an adder 41 multiplies the output voltage command values ​​of the uvw phases of the primary side inverter INV1 output from the dq-uvw conversion unit 37 by a gain M1 to obtain a zero-phase voltage command value v z refand the primary side output voltage command value v u1 ref , v v1 ref , v w1 ref and an adder 42 multiplies the output voltage command values ​​of the uvw phases of the secondary inverter INV2 output from the dq-uvw conversion unit 38 by a gain M2 to obtain a zero-phase voltage command value v z ref and the secondary output voltage command value v u2 ref , v v2 ref , v w2 ref Output as

[0121] That is, the effective voltage V as the primary side output voltage vector command value real and the reactive voltage V as the secondary output voltage vector command value imag (Actually, these are converted into output voltage command values ​​for each phase of u, v, and w) z ref By distributing and adding, the primary side output voltage command value v u1 ref , v v1 ref , v w1 ref and the secondary output voltage command value v u2 ref , v v2 ref , v w2 ref Generate and output.

[0122] Here, gain M1 + gain M2 = 1. The zero-phase voltage distribution unit 39 has two cases: gain M1 = 1 and gain M2 = 0, and gain M1 = 0 and gain M2 = 1. Therefore, the zero-phase voltage command value v is added to the output voltage command values ​​of the u, v, and w phases of either one of the inverters INV1 and INV2. z ref In some cases, one is added to the other and the other is not.

[0123] Basically, the zero-phase voltage command value v z refHowever, since the zero-phase voltage is added, it is better that the inverter with the higher voltage should bear the burden, considering the operating range, etc. When the voltage of the primary side inverter INV1 is high, the zero-phase voltage distribution unit 39 sets the gain M1=1 and the gain M2=0, and the zero-phase voltage command value v is added only to the primary side inverter INV1. z ref to impose a burden.

[0124] The zero-phase voltage command value v multiplied by the gain M2 added in the adder 42 is z ref The reason why the sign of is set to - (i.e., subtraction) is that the zero-phase voltage v z This is because it is necessary to create a potential difference of v, and the adder 42 actually becomes a subtractor, and the zero-phase voltage command value v multiplied by the gain M2 is z ref The negative value of is added to the output voltage command value of each phase uvw.

[0125] That is, the output voltage command generating unit 22 of this embodiment generates the d-axis voltage command value v d ref and the q-axis voltage command value v q ref From this, the effective voltage V as the primary side output voltage vector command value real and the reactive voltage V as the secondary output voltage vector command value imag and generates an effective voltage V real and reactive voltage V imag The zero-phase voltage command value v z ref By distributing and adding these, the primary side output voltage command value v for switching the primary side inverter INV1 is obtained. u1 ref , v v1 ref , v w1 ref and a secondary-side output voltage command value v for switching the secondary-side inverter INV2. u2 ref , v v2 ref , v w2ref Generate and output.

[0126] The primary side modulator 24 modulates the primary side output voltage command value v outputted by the output voltage command generator 23. u1 ref , v v1 ref , v w1 ref From ZFCPWM, a primary side inverter switching signal (PWM signal) S for switching (PWM control) each of the switching elements 3A to 3F of the primary side inverter INV1 is generated. u1 , S v1 , S w1 The secondary-side modulator 26 also generates and outputs the secondary-side output voltage command value v u2 ref , v v2 ref , v w2 ref to generate a secondary inverter switching signal (PWM signal) S for switching (PWM control) each of the switching elements 4A to 4F of the secondary inverter INV2. u2 , S v2 , S w2 Generate and output.

[0127] The top row of FIG. 15 shows the zero-phase voltage v of the primary inverter INV1 in this embodiment. z1 , the second row from the top is the zero-phase voltage v of the secondary inverter INV2 z2 , the third row from the top is the zero-phase voltage v z1 -Zero-phase voltage v z2 The zero-phase voltage V is derived from z , the bottom row is the zero-phase current i z The zero-phase voltage v of the primary inverter INV1 is shown. z1 and the zero-phase voltage v of the secondary inverter INV2 z2 However, since the number of fluctuations and the amplitude are small, the switching ripple is small.

[0128] FIG. 16 shows the zero-phase current i z In this example, the amplitude of the zero-phase current i zThe excitation of the switching ripple is generated by ZFCPWM in both the primary side inverter INV1 and the secondary side inverter INV2, but since the number of fluctuations and the amplitude are small, if the amplitude in the method shown in Figure 18 is taken as 100%, it can be seen that the amplitude in the case of Figure 16 is greatly suppressed to 10.2%, which is an improvement.

[0129] Here, in the above-described first embodiment, the primary side modulation unit 24 executes CPWM and the secondary side modulation unit 26 executes RSPWM, but this is not limiting, and the primary side modulation unit 24 may execute RSPWM and the secondary side modulation unit 26 may execute CPWM.

[0130] However, in that case, contrary to the case of the first embodiment, the zero-phase voltage v z1 and the zero-phase voltage command value v z ref The sum of these values ​​is calculated as the zero-phase voltage v of the secondary inverter INV2. z2 and add it to the output voltage command values ​​of the uvw phases of the secondary inverter INV2 output by the dq-uvw conversion unit 38, to obtain the secondary output voltage command value v u2 ref , v v2 ref , v w2 ref Let's say.

[0131] In addition, in the above-mentioned second embodiment, the primary side modulation unit 24 executes ZFCPWM and the secondary side modulation unit 26 executes RSPWM, but this is not limiting, and the primary side modulation unit 24 may execute RSPWM and the secondary side modulation unit 26 may execute ZFCPWM.

[0132] However, in this case, contrary to the case of the second embodiment, the zero-phase voltage v z1 and the zero-phase voltage command value v z ref The sum of these values ​​is calculated as the zero-phase voltage v of the secondary inverter INV2. z2 and add it to the output voltage command values ​​of the uvw phases of the secondary inverter INV2 output by the dq-uvw conversion unit 38, to obtain the secondary output voltage command value v u2 ref , v v2 ref , vw2 ref Let's say.

[0133] In each embodiment, the zero-phase voltage v z However, depending on the specific conditions such as the allowable operating range, the fluctuation of the zero-phase voltage v z It goes without saying that it is also possible to eliminate the fluctuations in the

[0134] In the embodiment, the output voltage command generating unit 23 generates an effective voltage V as the primary side output voltage vector command value and the secondary side output voltage vector command value. real and reactive voltage V imag However, the inventions of claims 1 to 3 are not limited to this, and the method of determining the primary side output voltage vector command value and the secondary side output voltage vector command value can be changed in various ways. That is, the zero-phase sequence voltage command value v z ref As one of the methods for determining the primary side output voltage vector command value and the secondary side output voltage vector command value before adding real and reactive voltage V imag There is a method to divide it into

[0135] Furthermore, although the switching elements in the embodiments are IGBTs, MOSFETs may also be used. In particular, the specific configurations and numerical values ​​shown in the embodiments are not limited to these and may be changed within the scope of the present invention.

[0136] REFERENCE SIGNS LIST 1 Power conversion device 2 DC power supply 3A to 3F, 4A to 4F Switching element 6 Control device 7U, 7V, 7W Winding 11 Positive power supply line 12 Negative power supply line 21 Speed ​​control section 22 dqz axis current control section (z axis current control section) 23 Output voltage command generation section 24 Primary side modulation section 26 Secondary side modulation section 27 Secondary side voltage control section 36 Effective / reactive decomposition section 39 Zero-phase voltage distribution section C Capacitor INV1 Primary side inverter INV2 Secondary side inverter M Motor

Claims

1. A power conversion device comprising a primary-side inverter connected to one end of a winding having an open-end structure of a motor, and a secondary-side inverter connected to the other end of the winding, and applying a differential voltage between the primary-side inverter and the secondary-side inverter to the motor, wherein the primary-side inverter is connected to a DC power supply, the secondary-side inverter is connected to a capacitor, a negative power supply line of the primary-side inverter and a negative power supply line of the secondary-side inverter are shared, and a control device for canceling or suppressing fluctuations in the zero-phase voltage, which is the potential difference between the zero-phase voltage of the primary-side inverter and the zero-phase voltage of the secondary-side inverter, is provided.

2. The primary-side inverter and the secondary-side inverter are each composed of a plurality of switching elements. A positive input terminal of the primary-side inverter is connected to a positive power supply line of the DC power supply, a negative input terminal of the primary-side inverter is connected to the negative power supply line of the DC power supply, an output terminal of the primary-side inverter is connected to one end of the winding, an output terminal of the secondary-side inverter is connected to the other end of the winding, the capacitor is connected between a positive input terminal and a negative input terminal of the secondary-side inverter, the positive input terminal of the secondary-side inverter is not connected to the positive power supply line of the DC power supply, the negative input terminal of the secondary-side inverter is connected to the negative power supply line of the DC power supply, and an AC output is generated from the DC power supply by switching each of the switching elements by the control device. The power conversion device according to claim 1, characterized in that 3. The control device includes a secondary-side voltage control unit that generates a zero-phase current command value based on a secondary-side voltage command value, a z-axis current control unit that generates a zero-phase voltage command value based on the zero-phase current command value, and an output voltage command generation unit that generates a primary-side output voltage vector command value and a secondary-side output voltage vector command value from dq-axis voltage command values, generates a primary-side output voltage command value for switching the primary-side inverter from the primary-side output voltage vector command value, and generates a secondary-side output voltage command value for switching the secondary-side inverter from the secondary-side output voltage vector command value. The power conversion device according to claim 1, characterized in that it has 4. The output voltage command generation unit decomposes the dq-axis voltage command value into an active voltage and a reactive voltage, uses the active voltage as the primary-side output voltage vector command value, and uses the reactive voltage as the secondary-side output voltage vector command value. The power conversion device according to claim 3, characterized in that.

5. The control device includes a primary-side modulation unit that generates a switching signal for the primary-side inverter from the primary-side output voltage command value, and a secondary-side modulation unit that generates a switching signal for the secondary-side inverter from the secondary-side output voltage command value. The power conversion device according to claim 3, characterized by having.

6. The secondary-side modulation unit performs pulse width modulation that outputs only odd voltage vectors during one control period, or pulse width modulation that outputs only even voltage vectors during one control period, or pulse width modulation that outputs only odd voltage vectors during one control period according to the voltage vector command value and switches between pulse width modulation that outputs only even voltage vectors during one control period to generate the switching signal, thereby eliminating or suppressing fluctuations in the zero-phase voltage. The power conversion device according to claim 5, characterized in that.

7. The primary-side modulation unit generates the switching signal by performing pulse width modulation that matches the rise and fall timings of the phase voltages of other phases with the rise and fall of the phase voltage of a specific phase, thereby eliminating or suppressing fluctuations in the zero-phase voltage. The power conversion device according to claim 6, characterized in that.

8. The output voltage command generation unit adds the sum of the zero-phase voltage command value and the zero-phase voltage of the secondary-side inverter to the primary-side output voltage vector command value. The power conversion device according to claim 6 or claim 7, characterized in that.

9. The primary-side modulation unit and the secondary-side modulation unit generate the switching signal by performing pulse width modulation that matches the rise and fall timings of the phase voltages of other phases with the rise and fall of the phase voltage of a specific phase, thereby eliminating or suppressing fluctuations in the zero-phase voltage. The power conversion device according to claim 5, characterized in that.

10. The output voltage command generation unit distributes and adds the zero-phase voltage command value to the primary-side output voltage vector command value and the secondary-side output voltage vector command value. The power conversion device according to claim 9, characterized in that.

11. The power conversion device according to claim 10, wherein the output voltage command generation unit has a case where a gain distributed to the primary side output voltage vector command value is 1 and a gain distributed to the secondary side output voltage vector command value is 0.

12. The power conversion device according to claim 10, wherein the output voltage command generation unit has a case where a gain distributed to the primary side output voltage vector command value is 0 and a gain distributed to the secondary side output voltage vector command value is 1.

13. The primary side modulation unit generates the switching signal by executing any one of pulse width modulation that outputs only odd voltage vectors during one control period, pulse width modulation that outputs only even voltage vectors during one control period, or pulse width modulation that switches between pulse width modulation that outputs only odd voltage vectors during one control period according to the voltage vector command value and pulse width modulation that outputs only even voltage vectors during one control period, thereby eliminating or suppressing the fluctuation of the zero-phase voltage. The power conversion device according to claim 5.

14. The secondary side modulation unit generates the switching signal by executing pulse width modulation that matches the rising and falling timings of the phase voltages of other phases with the rising and falling of the phase voltage of a specific phase, thereby eliminating or suppressing the fluctuation of the zero-phase voltage. The power conversion device according to claim 13.

15. The power conversion device according to claim 13 or claim 14, wherein the output voltage command generation unit adds the sum of the zero-phase voltage command value and the zero-phase voltage of the primary side inverter to the secondary side output voltage vector command value.

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