Simulation method for generating pulse patterns for controlling an inverter

Abstract

The invention relates to a simulation method for generating pulse patterns defined by switching angles, for a synchronous PWM modulation method for controlling an inverter which feeds an electric motor, the simulation method comprising the following steps: a. providing an inverter model which reflects inverter losses of the inverter during execution of the synchronous PWM modulation method for operating the electric motor; b. providing a motor model of the electric motor which reflects motor losses of the electric motor during execution of the synchronous PWM modulation method for operating the electric motor; c. determining at least one switching angle of the synchronous PWM modulation method in order to define a starting pulse pattern, and ascertaining the inverter losses and motor losses in view of the starting pulse pattern on the basis of the inverter model and the motor model; d. searching for switching angles suitable for the operating of the electric motor, by changing the at least one switching angle in order to obtain varied pulse patterns and evaluating the inverter losses and motor losses ascertained for each of the varied pulse patterns; and e. storing the switching angles found by the searching, for controlling the inverter corresponding to the inverter model in order to operate the electric motor corresponding to the motor model.

Description

[0001]202201068 1 Description Simulation method for generating pulse patterns The invention relates to a simulation method for generating pulse patterns defined by switching angles, as well as a corresponding computer program. Approaches for optimizing switching angles can be found in the prior art. For example, a genetic algorithm in combination with selective harmonic elimination (SHE) is used to select which order of the harmonic voltages is to be eliminated with regard to minimal distortion. With this approach, the number of possibilities increases exponentially with the increasing number of switching angles, whereby the algorithm requires a large number of iterations, which results in long computing times. In addition, the position of the switching angles cannot be influenced because no starting angles are required. Thus, this approach is not suitable for delivering realistic results.In addition, approaches to optimizing switching angles using particle swarm optimization exist. Due to the poor convergence to a global optimum and the high number of iterations, this approach also does not lead to the desired results. Against this background, the object of the invention is to create a simulation method that delivers realistic results in order to determine optimized pulse patterns. At the very least, the object of the invention is to create an alternative to the prior art. This object(s) is(are) achieved by a simulation method according to patent claim 1. Preferred embodiments are the subject of the dependent patent claims. The simulation method according to the invention for generating pulse patterns defined by switching angles for a synchronous PWM modulation method for controlling an inverter that feeds an electric motor has the following steps: 202201068 2 a.Providing an inverter model that reflects inverter losses of the inverter when executing the synchronous PWM modulation method for operating the electric motor; b. Providing a motor model of the electric motor that reflects motor losses of the electric motor when executing the synchronous PWM modulation method for operating the electric motor; c. Determining at least one switching angle of the synchronous PWM modulation method to establish a starting pulse pattern and determining the inverter losses and motor losses taking into account the starting pulse pattern based on the inverter model and the motor model; d. Searching for switching angles suitable for operating the electric motor by changing the at least one switching angle to obtain varied pulse patterns and evaluating the inverter and motor losses determined for the varied pulse patterns; and e.Storing the switching angles found by the search for controlling the inverter corresponding to the inverter model in order to operate the electric motor corresponding to the motor model. The simulation method is in particular a computer-implemented method that is carried out with computer support. The simulated motor is, for example, a permanent magnet synchronous motor. The search carried out in step d. is preferably implemented by an algorithm that operates according to a system corresponding to the “patternsearch” algorithm in MATLAB. In this way, the algorithm finds globally optimized switching angles. The storing carried out in step e. takes place in a motor control system, preferably in a modulation block for controlling an inverter corresponding to the inverter model, which in turn feeds the motor that corresponds to the motor model. 202201068 3 A number of the values ​​found in step c.The determined switching angle depends on the clock rate p, wherein the respective switching angles are changed in step d. Furthermore, in step c., a starting edge of the switching angle can be set positively or negatively. The switching angles found by the simulation method and the pulse patterns defined thereby form a center pulse modulation or edge pulse modulation. Reference is made to the corresponding explanations in the preferred embodiment. The simulation method is preferably designed such that in step c. the at least one switching angle of the synchronous PWM modulation method is determined for a quarter or half period of the starting pulse pattern and / or in step d. the switching angle for obtaining the varied pulse patterns is changed for a quarter or half period of the respective pulse pattern; wherein the inverter model has a switching angles block that maps the switching angle of the quarter or half period to a full period.To determine the inverter losses, the inverter model preferably takes into account a type of switching element of the inverter corresponding to the inverter model and corresponding switching and / or conduction losses of the switching elements. The inverter losses determined in step d. relate to switching and / or conduction losses during switching operations that occur taking into account currents flowing at that time, and / or the inverter losses determined in step d. relate to switching and / or conduction losses during switching operations that occur taking into account averaged currents averaged over a corresponding period resulting from the quarter or half period. The types of switching elements are, for example, transistors such as IGBTs or MOSFETs. The embodiments in the preferred embodiment of the invention with regard to the inverter model can be considered in isolation in connection with the inverter model.In the simulation method according to the invention, the motor model preferably takes into account iron, magnet and / or copper losses, preferably divided between the rotor and stator of the motor, to determine the motor losses. 202201068 4 The embodiments in the preferred embodiment of the invention with regard to the motor model can also be considered in isolation in connection with the motor model. The simulation method is preferably designed such that a plurality of operating points of the motor corresponding to the motor model are defined, and for each operating point of the plurality of operating points, steps c. and d. are carried out for each cycle number (p) of a plurality of cycle numbers, wherein for a considered operating point of the plurality of operating points, in step c. a number of switching angles is determined as a function of the cycle number (p), then in step d.the switching angles determined in number are changed to obtain varied pulse patterns, preferably by calculation, and after the changes according to step d. have been carried out, steps c. and d. are repeated for a different cycle number of the plurality of cycle numbers. The cycle numbers p are in particular odd, p = 3, 5, 7, ..., 23. Each of the operating points is preferably defined by the speed n and torque Tq of the motor. Alternatively, each of the operating points can also be defined by the modulation degree m and torque Tq. Each of the PWM pulse patterns, which is defined by the switching angles found in step d. and stored in step e., preferably corresponds exactly to one operating point.The simulation method according to the invention can be applied, for example, to 120 V drive architectures with Si-MOSFETS, HV inverters with IGBTs, railway technology, wind power, energy supply, photovoltaic inverters, micromobility, or even to high-voltage vehicle drives with at least 400 V or 800 V nominal voltage or a nominal power of at least 100 kW, 200 kW or more. Furthermore, it is proposed to operate an electric drive according to the result of the simulation method. In particular, it is proposed that an electric drive be operated which has an electric motor and an inverter. In this case, those switching angles which are determined or stored as described here are called up from a memory. The inverter is operated according to the switching angles. The inverter thereby generates 202201068 5 a (multi-phase) current which is supplied to the motor. The current supplied to the motor is based on the switching angles.The inverter corresponds to the inverter model. The electric motor corresponds to the motor model. A preferred embodiment of the simulation method according to the invention is explained below with reference to the attached figures. The figures serve to illustrate exemplary embodiments of the methods described here and the corresponding computer program. Figure 1 shows the conceptual structure of a control loop for controlling an electric motor, wherein the essential part of the invention relates to the control of the inverter feeding the motor using a synchronous PWM modulation method; Figures 2a and 2b show a circuit diagram of the inverter and a corresponding equivalent circuit; Figure 3 shows an exemplary PWM pulse pattern of a phase U, V, W. Figure 4 shows a block diagram of a simulation model used to implement the simulation method according to the invention.Figures 5a and 5b show the calculation of the inverter losses and the motor losses in detail. Figures 6a and 6b show two options for center and edge pulse modulation. Figure 1 shows a conceptual structure of a control system for an electric motor 1, such as a permanent-magnet synchronous motor, which can preferably be operated in motor mode, but also in generator mode as intended. The electric motor 1 is preferably an electric drive unit of a motor vehicle or motorcycle. The simulation method according to the invention, which is described in detail below, serves to generate loss-optimized synchronous PWM pulse patterns for the subsequently implemented control of the motor 1. 202201068 6 The control of the motor 1 will now first be explained for a better understanding of the invention, with particular attention to the synchronous PWM pulse patterns used.The control loop of the control system shown detects, at a first block T1, the alternating current intensities iu, iv, iw flowing through phases U, V, W supplying motor 1. Block T1 preferably detects only the current intensities of phases U and V in Figure 1. The remaining alternating current value of phase W is calculated from the detected values. The alternating currents, in particular, have a sinusoidal shape and are preferably phase-shifted by more than 120° from one another, with the same applying to the corresponding phase voltages. Block T1 transforms, for example, the detected three alternating currents flowing in phases U, V, W into stator- and / or rotor-fixed current values ​​using a Clarke transformation and / or a Park transformation. These transformations are known.The obtained current variables can be compared with externally specified reference variables Ref on a comparison element VG and then modeled in a desired manner via the control block RB shown, which is, for example, a PI controller. For example, the externally specified reference variables Ref correspond to acceleration requests issued by a driver. Received output variables of the control block RB form an operating point of motor 1, which is to be approached, and input variables of a modulation block 3. The operating point is defined in particular by a desired speed n and a desired torque Tq of motor 1. As an alternative to the speed, a modulation degree can be used to define the operating point. Preferably, additional parameters can also be used to define the operating point.202201068 7 Modulation block 3 controls an inverter 2 for generating target AC voltages (with amplitudes ûsoll) at phases U, V, and W, which are phase-shifted from one another and supply motor 1 to approach the operating point. The modulation block 3 shown controls inverter 2 by simulating or modeling each of the target AC voltages (U, V, W with amplitudes ûsoll) using synchronous PWM pulse width modulation. Modulation block 3 implements a synchronous modulation method, explained in more detail below, which does not calculate the required synchronous PWM pulse pattern in real time (online), but accesses a stored PWM pulse pattern generated offline for the respective operating point. For this purpose, the modulation block 3 has a memory in which the necessary PWM pulse patterns for possible operating points are stored.The present invention is directed to obtaining the necessary synchronous PWM pulse patterns using a simulation method and storing them in the memory of the specific control loop corresponding to the simulation. Figure 2a shows a circuit diagram of inverter 2. Inverter 2, which feeds motor 1, has a central task. In the control loop shown in Figure 1, it converts a direct voltage Ud from a battery (not shown) (high-voltage battery) into the three required nominal alternating voltages U, V, W (with amplitudes ûsoll ) for motor 1. The three phases U, V, W of motor 1 are connected to one another in a star connection in Figure 2a, for example. A delta connection is also possible. To convert the direct voltage Ud, inverter 2 operates with clocked switching states. Instead of a continuous energy flow, the energy is transmitted in individual packets using the synchronous PWM modulation method.A high-frequency sequence of these packets transforms the divided energy flow into an approximately continuous output signal that feeds the corresponding phase of motor 1. 202201068 8 The inverter 2 shown in Figure 2a is preferably a B6C bridge circuit with three bridge arms, each bridge arm corresponding to one of the phases U, V, W. Two switches 20, 21, and 22 are located in each of the three bridge arms, with the respective associated phase U, V, W being tapped between the corresponding switches 20, 21, and 22. The switches 20, 21, and 22 are controlled by means of the aforementioned synchronous PWM modulation method in order to generate the alternating variables in the phases U, V, and W. Each of the three bridge branches thus feeds one of the phases U, V, W, whereby the three-phase alternating voltage for motor 1 can be realized with variable frequency and amplitude using the synchronous PWM modulation method.This allows any operating point—defined by speed n or modulation depth m and torque Tq—to be approached within the operating range of motor 1. To achieve the required high switching frequencies, semiconductor switches such as IGBTs (Insulated Gate Bipolar Transistors) or MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) are usually used for switches 20, 21, and 22. However, these power semiconductors limit the switching frequency, as their switching losses increase with increasing switching frequency and there is a limitation on the dissipated power loss. A compromise must therefore be found between good modulation of the phase voltages and thus low losses due to the harmonics of the phase currents and not excessively high switching losses. The simulation method explained later finds this compromise. The bridge circuit shown in Figure 2a, corresponding to inverter 2, can be represented ideally.Figure 2b shows the corresponding equivalent circuit in which switches 20, 21, and 22 are replaced by simple switches. A virtual center point of the intermediate circuit is introduced, but not connected to a load star point N, as otherwise it would no longer be a symmetrical three-phase voltage system. This is the case for both the aforementioned star connection and the delta connection. 202201068 9 In the equivalent circuit, the battery voltage Ud is divided into two identical voltage sources. The three phases U, V, and W can thus be switched to the two available voltages +Ud / 2 and -Ud / 2 in order to simulate the desired sinusoidal voltage waveform. The resulting currents iU, iV, and iW ideally have the same sinusoidal shape. The same relationship applies to the magnetic flux.In addition, the equivalent circuit diagram shown in Figure 2b shows the center-point voltages UU0, UV0, and UW0, which can be derived from the respective switch position. These are calculated as follows: The neutral point voltage UN0 is also used to calculate the phase voltages. For a symmetrical load from motor 1 without a common-mode component, this is: Due to the symmetrical voltage system without a neutral conductor, the sum of the phase currents iU, iV, and iW must also be zero. Thus, a phase current can be calculated from the remaining two. From the three bridge branches, 2 3 = 8 discrete switching states. The switching states and the corresponding voltages are summarized in the following table. 202201068 10 The eight different states 0 Z to 7 Z can still be converted into stress-forming states ( 1 Z to 6Z) and zero states ( 0 Z and 7 Z). The latter each result in a phase voltage of zero, since the three phases are short-circuited via inverter 2. As can be seen from the table above, in the state 0 Z all switches in position -1 and consequently in the state 7Z in the +1 position. In this context, the modulation block 3 shown in Figure 1 is configured to control the switches 20, 21, 22 according to the read-out synchronous PWM pulse pattern, which is merely phase-shifted for the phases U, V, W, in order to generate the corresponding alternating currents in the respective phases. In particular, the modulation block 3 uses the voltage-generating states to generate the respective PWM pulse patterns. In general, the states according to the table above can be represented as space vectors in the α-β coordinate system, with the modulation block 3 using these space vectors, in particular according to the voltage-generating states, to control the inverter 2. The modulation method used is, as already mentioned, a synchronous modulation method. 202201068 11 Here, the selected switching frequency is always synchronous with a fundamental frequency of the output voltage, and the zero crossings are synchronized.A duty cycle nT is an integer and a switching frequency is variable. The duty cycle nT specifies a ratio of the switching frequency fsw to the frequency of the output voltage, where the former switching frequency fsw is a measure of the number of switching cycles of a switch per unit of time. The temporal characteristic of the switching frequency is identical over each electrical period, which means that when transformed into the frequency domain, no frequencies lower than the fundamental frequency occur. Undershoots, which occur with asynchronous modulation, are avoided. In addition to synchronicity, symmetries of the phase voltages U, V, W to be controlled are used. In addition to half-period symmetry, quarter-period symmetry is used because it reduces current harmonics. This symmetry in the phase voltages means that no even-numbered harmonics occur.Furthermore, it is only necessary for modulation block 3 to store switching angles for the first quarter period of each PWM pulse pattern. From this, the full PWM pulse pattern can be developed. Figure 3 shows an example synchronous PWM pulse pattern for one of the phases U, V, W, which is simply phase-shifted for the others. The second half period forms the inverse switching function of the first half period. In addition, the switching angle curves are axially symmetric to one another, with the axes of symmetry at π / 2 and 3π / 2. The entire information of the pulse pattern is thus contained in the first quarter period and can be used as the basis for developing the full PWM pulse pattern by modulation block 3. The harmonic voltage overtones have the same phase position as the fundamental oscillation or are in antiphase with it. Quarter-period symmetry also requires an odd clock cycle number p and an existing switching edge at ωt = 0 and ωt = π.Instead of calculating the switching angles in real time (online), they are calculated in advance depending on specific parameters corresponding to the respective operating point and stored in the memory of modulation block 3. The determination and optimization of the switching edges and thus the 202201068 12 pulse patterns are carried out using the simulation method explained below. In order to approach a specific operating point, a modulator (not shown) is provided which reads the pulse pattern matching the input parameters (operating point) from the memory of modulation block 3. Preferably, for each modulation level m, the PWM modulation patterns with their switching angles and switching states are stored in said memory; these are converted into discrete switching times depending on the speed n. The modulation block 3 then passes the control signals on to inverter 2, which directly controls motor 1.Due to the dependence of the PWM pulse pattern on the fundamental frequency, the switching frequency increases proportionally with increasing fundamental frequency. Due to the maximum switching frequency limited by inverter 2, the clock frequency p is varied across the entire operating range. The different clock frequencies p and the optimization options in the simulation process are discussed below to explain the different possibilities for generating synchronous modulation or synchronous PWM pulse patterns. Triple clocking is the easiest clock frequency p to implement, since a voltage block per half period is only interrupted by an intermediate pulse. The number of freely selectable switching angles q that occur in a quarter period can be determined using the following equation. The clock frequency p describes the number of intermediate pulses over one electrical period.Using the triple clocking example, six switching operations correspond to three intermediate pulses. Using the triple clocking example mentioned above, this results in only one free switching angle in a quarter period. 202201068 13 For synchronous clocking, at least two designs are possible: center-pulse modulation and edge-pulse modulation. They differ in the position of the intermediate pulse, either in the center or on the edges of a voltage block of length π. All switching angles of a pulse pattern together determine the modulation depth. The modulation depth m describes the standardized length of the voltage space vector and is calculated from the ratio between the amplitude of the fundamental output voltage and the intermediate circuit voltage divided by the square root of three. In this application, this always refers to the modulation depth determined by motor variables.For both center pulse and edge pulse modulation, there are two ways to achieve the desired modulation depth m. These differ in a positive or negative switching edge at time ωt = 0. The following figures show these two possibilities, with the pulse patterns achieving the same modulation depth. Type 1 denotes a positive starting edge in phase U, which is accompanied by a positive sign for the Fourier coefficients of the phase voltage amplitude. Type 2 represents a negative starting edge with a correspondingly negative sign. As shown in the figures mentioned above, a positive sign means that the switching pattern starts in the on state. A negative sign means that the switching pattern starts in the off state. When determining the pulse patterns in the simulation method according to the invention, two solutions therefore arise, both of which are calculated. Not all solutions are possible for every modulation depth m; some are only available for a sub-range. 202201068 14 As already mentioned, with edge pulse modulation, the pulses at which the voltage edge changes are located on the edges of the half-period. Likewise, the corresponding switching angle can be set proportionally to the desired modulation depth m. With edge pulse modulation, the maximum modulation depth is also limited by the inverter dead time.Overall, center-pulse modulation has a greater influence on the amplitude of the fundamental oscillation than edge-pulse modulation due to the position of the intermediate pulse in the center of the switching pattern. However, edge-pulse modulation has advantages at large modulation levels because the switching angles are in the range of small voltage amplitudes. In general, the goal of the synchronous modulation method is to minimize the amplitudes of the harmonics in the current. This primarily reduces the harmonic losses in the motor. On the other hand, the switching losses in the inverter must be limited, which fundamentally increase due to more complex modulation and the associated higher effective switching frequencies. According to the invention and the simulation method, the switching angles (α1,..., αq) of the quarter periods are optimized directly based on simulated inverter and motor losses Pʋ. The following boundary conditions apply: where the amplitude of the target AC voltage ûsoll corresponds to the amplitude of the fundamental wave ûʋ=1 resulting from all switching angles. Firstly, the simulation method finds the optimum defined from all free switching angles, which in this case is a minimum. Secondly, the required target voltage amplitude ûsoll and thus the desired modulation depth m, which correlates with the speed at the operating point, is derived from all switching angles. 202201068 15 Finally, due to quarter-period symmetry, no switching angle may be smaller than 0 rad or greater than π / 2, and all switching angles are preferably sorted in ascending order. The PWM pulse patterns found are finally saved so that they are available for the operation of the real inverter 2 and motor 1 belonging to the simulation and can be stored in the memory of the modulation block 3.A simulation model used to implement the method according to the invention is explained below. The simulation method based on the simulation model results in the optimized PWM pulse patterns, in particular the optimized switching angles of the quarter period, from which the full PWM pulse pattern can be developed. According to the simulation method, the switching angles are optimized for the total losses in order to calculate a consumption-optimized result. For this purpose, a simulation model for motor and inverter losses that is as realistic as possible is used. Based on these loss models, various synchronous modulation types, center and / or edge pulse modulation, as well as clock frequencies are first simulated at a fixed operating point.The switching angles are preferably also determined as a function of the modulation depth m; this enables, for example, verification of the simulation results on a test setup in which the simulation results are compared with a state-of-the-art asynchronous space vector modulation (SVPWM). Figure 4 shows a block diagram of the simulation model used and the integration of the switching angle optimization. The simulation model provides an inverter model IM and a motor model MM for the simulation process, which are used to calculate the total losses. The simulation process using the simulation model is, for example, a computer program that is preferably executed on two parallel information-processing computers. 202201068 16 First, the operating points to be simulated are selected at simulation block SB1 and their input parameters are specified.The essential and variable input parameters for the simulation process are torque Tq and speed n. The other input parameters are preferably constant to reduce computational effort. This is followed by transfer to the actual switching angle optimization in simulation block SB2. SB2 represents the main program, which is structured, for example, as a MATLAB script. All other blocks are also preferably defined as MATLAB functions, since they are accessed multiple times. The optimization is performed individually for each operating point based on the calculation of the total losses, including inverter and motor losses. In the process, a cycle rate p and a switching angle (starting angle) are specified for the operating point in SB2. The optimization loop shown in Figure 4 refers to an iteration in which the total losses are calculated for a specified cycle rate p and switching angle.The calculation is then repeated by changing / calculating the switching angle for the operating point while maintaining the cycle rate p. Once all switching angles have been calculated for the cycle rate p, the cycle rate p is increased in the process and the switching angles for this cycle rate are run through / calculated again. The number of switching angles varied for each cycle rate depends on the cycle rate. These loops are preferably run through for p = 3, 5, 7, ..., 23. Only when the switching angle has been varied using a solver until the total losses are minimal is the next operating point optimized. For each operating point, the different losses as well as the switching angle and preferably the modulation depth are stored in appropriately large, multi-dimensional matrices. 202201068 17 The blocks used for the simulation process are explained in more detail in the following sections.In block SB1, the operating points are determined and their input parameters are defined. The following table contains parameters that are kept constant throughout the entire simulation process, as well as the aforementioned variable parameters. In addition to the parameters Tq and n, the battery voltage and temperature could also be varied, although this would significantly increase the number of solutions. After parameterizing the operating points, the desired motor model MM and inverter model IM are loaded. Furthermore, the desired output parameters of the motor model can be selected, since only a fraction of the simulated values ​​are required per iteration. This serves to minimize computing time. The solver, for example, PatternSearch in MATLAB, is then initialized. This is described in more detail below. An outer for loop calculates the switching angles from the minimum to the maximum selected cycle rate. Depending on the cycle rate, the corresponding number of switching angles (starting angles) is initialized. 202201068 18 The simulation procedure is configured to distinguish between center-pulse and edge-pulse modulation. The inverter model is considered first.The area IM delineated in Figure 4 simulates the power electronics of inverter 2. The following section explains the properties of the inverter model IM and the simulation of its losses. In this application, an inverter 2 with SiC MOSFETs with the following properties was simulated. The simulation of inverter 2 can preferably be divided into three large blocks, which are represented by functions. Switching Angles IM1 Due to the quarter-period symmetry, the optimized angles are only defined and optimized for a quarter period. For further calculations, these are mapped to an entire electrical period in block IM1. Furthermore, in this block IM1 you can preferably select whether the switching angles start with a positive or negative switching edge. This block IM1 is only preferred. Alternatively, the switching angles can be stored for the entire period. Effective Switching Angles IM2 202201068 19 The stator current resulting from the operating point is calculated from the DC quantities Id and Iq and preferably converted to an effective current. Since these DC quantities are motor quantities, a conversion to the phase quantities is carried out to determine the inverter current.If the simulated electric motor is connected in a delta, the effective current value must be multiplied by the square root of three. The effective current serves as the starting value for the further inverter loss calculation. The inverter loss calculation preferably initially refers to the losses during switching operations at the time of maximum current amplitude. For this purpose, the effective current value is converted to a pure sinusoidal oscillation. The current amplitudes at the switching times are then determined from this sinusoidal current, taking into account the phase shift of an operating point. Block IM2 preferably calculates the average current over a period from the current amplitudes, which would flow at all switching times. This—normalized to the maximum current occurring—results in the correction factor. It can assume values ​​between 0 and 1, whereby the value 0 would correspond exclusively to switching times in the phase responses of the current.The value 1 corresponds to the case where all switching angles occur at times of maximum current in the inverter. Neither of these limiting cases occurs in reality, so the value of the correction factors is always in between. This preferred approximation requires significantly less computing time than the alternative comparison of the actual current curve with the switching pattern, since the inverter losses must be recalculated in each iteration. The sharp increase in computing time with increasing iteration number outweighs the slight inaccuracy of this approximation. Inverter loss calculation IM3 The preferred inverter loss calculation includes all losses that occur in inverter 2 when converting the DC link voltage to the multi-phase system. The majority of the inverter losses are caused by the switching and conduction losses, which are considered preferentially.By saving 202201068 20 switching operations in certain operating ranges, synchronous PWM can achieve advantages over asynchronous SVPWM. The simulation method achieves a short computing time for the loss calculation in order to cope with the large number of iterations, while simultaneously achieving the most realistic result possible. Switching and conduction losses are calculated separately for the MOSFETs and the diodes per bridge branch. Figure 5a shows a schematic representation of the calculation of the inverter losses according to IM3. Ieff and the initially defined boundary conditions must be passed to the functions. Switching losses IM31 Switching losses occur during the switching on and off processes of the MOSFETs during commutation of the output current.For both the MOSFET and the diode, the switching energy loss is interpolated from a table using the maximum current amplitude in inverter 2, the intermediate circuit voltage, and the temperature. The temperature is assumed to be the motor temperature. Multiplying this value by the number of switching elements yields the maximum switching losses for the entire inverter 2 for all three phases. The maximum switching losses are then calculated using the previously determined correction factor and the effective switching frequency. Forward losses IM32 The forward losses are calculated in the same way as the switching losses. They occur in the semiconductor elements during current-carrying operation. In the case of the MOSFET, the drain-source voltage of the MOSFET UDS is interpolated from a table using the maximum current amplitude in inverter 2 and the temperature.This voltage – multiplied by the same maximum current – ​​results in the theoretically maximum achievable forward losses. The maximum forward losses thus calculated are in turn offset against the quadratic correction factor, since both the interpolation and the multiplication by the maximum current occur. 202201068 21 For the diode, the forward losses are calculated from the interpolated freewheeling voltage Uf of the diode, multiplied by the maximum current. The resulting power loss is also corrected using the quadratic correction factor. In addition, the calculation of dead time losses is omitted from this block. Since these account for a large proportion of inverter losses, especially at high switching frequencies, their calculation cannot be neglected. In the case of a negative phase current, these losses are attributable to the body diode of the MOSFET. These losses are added to the total inverter losses.Reverse losses. Furthermore, other losses exist in switching transistors and diodes, such as reverse or drive losses. Due to the low leakage currents, reverse losses are very small compared to the main losses – forward and switching losses – and can, but do not have to, be considered. Drive losses. Drive losses are generally small due to the short, pulse-like drive currents in IGBTs and MOSFETs and can also be neglected. SMPS, PCC, and Busbar Losses. In addition to switching and forward losses, switched-mode power supply (SMPS), busbar, and power capacitor chip (PCC) losses are also preferentially calculated. These together represent a smaller proportion than the switching and forward losses, but can be considered preferentially. SMPS losses arise in the power supplies that supply the MOSFETs.A distinction is made between low voltage (LV) and high voltage (HV) switched-mode power supply losses, which together make up the total SMPS losses. Both switched-mode power supplies must provide a constant gate-source voltage. Since they are also dependent on the switching frequency, lower losses are achieved by selecting a lower clock speed at an operating point. The effective switching frequency fsw,eff used is calculated from the electrical frequency fel multiplied by the clock speed p. 202201068 22 Busbar losses originate from the busbars between the inverter and motor and are divided into losses caused by the alternating component and the direct component of the flowing current. Both components result from the dq currents for the respective operating point from the machine loss model. The effective current Ieff is used for the alternating component.The DC component of the current IDC is calculated using cos(φ) and the respective modulation depth m. The individual currents are multiplied by the AC and DC resistance of the busbar of all three phases, which were obtained from a measurement. The AC and DC losses add up to the total busbar losses. For the PCC losses, the losses occurring in the intermediate circuit capacitor must be calculated. To do this, the flowing current is first calculated using the effective current, cos(φ), and the modulation depth m. Using the determined current and the equivalent series resistance of the capacitor (ESR), the corresponding power loss at the respective operating point can then be determined. The motor model is examined below. The area MM delineated in Figure 4 represents the simulated electric motor and its loss calculation.The properties of the motor model MM and the simulation of the losses are explained in more detail below with reference to Figure 5b. The following properties were chosen for electric motor 1. The simulation of the motor can be divided into four large, preferred blocks, which are represented by functions. 202201068 23 PWM Pulse Pattern MM1 In block MM1, the pulse pattern is generated from the input variable of the switching angle with the information whether a falling or rising edge is present. To simulate the corresponding PWM driver, its parameters are also specified. These include the clock frequency of the installed CPU and the number of periods to be simulated. One switching angle can be realized per PWM period. Since in synchronous PWM each period is symmetrical with the electrical frequency due to the synchronized zero crossings, simulating one electrical period is sufficient. PWM Spectrum List MM2 The generated pulse pattern is passed to block MM2, which also receives the desired number of harmonic orders.For example, the 150 orders with the largest absolute voltage amplitudes are always used. Since the orders are not automatically in ascending order, a vector must be generated that contains the information about the order of the harmonic voltage phasors. The complex voltage phasors of the harmonics are then generated for all three phases. This is done using a fast Fourier transform (FFT), which switches from the image domain to the frequency domain. Since the pulse pattern changes with each iteration, the complex spectrum must also be generated in each iteration. Udq Spectra MM3 In block MM3, the complex harmonic voltages are preferably converted into the dq coordinate system using an inverse Clarke-Park transformation. This is preferred because the motor loss calculation can be performed in these coordinates and the fundamental current is also available as Id or Iq.202201068 24 The most important output variable of the motor model is its loss (block MM4). For each operating point in block SB1, the complex voltage spectrum is generated from the switching angles via the pulse pattern (MM41). The fundamental oscillation of the currents Idq,fund is determined by optimization according to the maximum torque per ampere (MTPA) control. From the harmonic complex voltages, the harmonic harmonics of the d and q currents are also calculated using machine equations (MM42). These, together with the fundamental oscillation, are included in the loss calculation for the harmonic losses. The harmonic iron, magnet, and copper losses are calculated – divided by stator and rotor. After adding the harmonic losses, the losses of the fundamental oscillation are added together, since these only need to be calculated once for each operating point. In contrast, the harmonic losses differ with each change in the switching angle.The simulation method finds a global optimum and enables parallelized calculation. Various search algorithms are generally possible for finding the optimum. The MATLAB algorithm "patternsearch" is preferred, enabling the global loss optimum to be found and the calculation to be performed in parallel. Starting values ​​for the switching angles can be specified to more quickly find a solution with switching angles on the flanks or in the middle of the half-periods. The applied algorithm creates a pattern of potential solutions around the starting point (switching angle), the so-called mesh. Its size is initialized at the beginning and changed over subsequent iterations. The new points are derived from the starting value plus the size and direction of the grid. The calculated solutions for the points are compared, and the best solution is selected as the new starting value.Results that do not meet the selected boundary conditions are not considered. The boundary conditions used are the formulas 202201068 25 explained above and presented below. used. In addition, a minimum distance between the switching angles can preferably be inserted, which corresponds to the dead time. This is necessary to ensure that the inverter switches are not switched on at the same time. For the new potential solutions, the procedure is repeated and the mesh size is increased until no better solution occurs. The mesh is then reduced in each step, for example, halved. The solver stops when a minimum is iterated and the termination criteria are met. No information about the gradients is required to perform pattern search, which means that this solution method is also suitable for discontinuous functions. With regard to switching angle optimization, the switching angles obtained for the respective operating point are saved and used as the starting value for the next operating point.This favors a fast iteration to the optimum, since with the low discretization used the optimal switching angles are close to the previous operating point.

Claims

202201068 26 patent claims 1. Simulation method for generating pulse patterns defined by switching angles for a synchronous PWM modulation method for controlling an inverter that feeds an electric motor, the simulation method comprising the following steps: a. Providing an inverter model that reflects inverter losses of the inverter when executing the synchronous PWM modulation method for operating the electric motor; b. Providing a motor model of the electric motor that reflects motor losses of the electric motor when executing the synchronous PWM modulation method for operating the electric motor; c. Determining at least one switching angle of the synchronous PWM modulation method for establishing a start pulse pattern and determining the inverter losses and motor losses taking into account the start pulse pattern on the basis of the inverter model and the motor model; d.Searching for switching angles suitable for operating the electric motor by changing the at least one switching angle to obtain varied pulse patterns and evaluating the inverter and motor losses determined for each varied pulse pattern; and (e) storing the switching angles found by the search for controlling the inverter corresponding to the inverter model in order to operate the electric motor corresponding to the motor model.

2. Simulation method according to claim 1, wherein in step c. the at least one switching angle of the synchronous PWM modulation method is determined for a quarter or half period of the starting pulse pattern and / or in step d. the switching angle is changed for a quarter or half period of the respective pulse pattern to obtain the varied pulse patterns; and the inverter model has a switching angles block that maps the switching angle of the quarter or half period to a full period.Simulation method according to claim 1 or 2, wherein for determining the inverter losses the inverter model comprises a type of switching elements of the inverter corresponding to the inverter model and. 202201068 27 corresponding switching and / or conduction losses of the switching elements are taken into account, and the inverter losses determined in step d. relate to switching and / or conduction losses during switching operations that occur taking into account currents flowing at that time.

4. Simulation method according to claim 1 or 2, wherein, to determine the inverter losses, the inverter model takes into account a type of switching element of the inverter corresponding to the inverter model and corresponding switching and / or conduction losses of the switching elements, and the inverter losses determined in step d. relate to switching and / or conduction losses during switching operations that occur taking into account averaged currents averaged over a corresponding period resulting from the quarter or half period. 5.Simulation method according to claim 3 or 4, wherein, in addition to the corresponding switching and / or conduction losses of the switching elements, the inverter model takes the following losses into account to determine the inverter losses: -switched-mode power supply losses (SBT); - busbar losses; and / or - power capacitor chip losses (PCC).

6. Simulation method according to one of claims 1 to 5, wherein, in order to determine the motor losses, the motor model takes iron, magnet, and / or copper losses of the motor corresponding to the motor model into account.

7. Simulation method according to claim 6, wherein, in order to determine the motor losses, the motor model takes the iron, magnet, and / or copper losses divided between the rotor and stator of the motor into account.

8. Simulation method according to one of claims 1 to 7, wherein, for each operating point of the plurality of operating points, steps c. and d. are carried out.for each bar number (p) of a plurality of bar numbers, whereby. 202201068 28 for a considered operating point of the plurality of operating points, in step c. a number of switching angles is determined as a function of the cycle rate (p), then in step d. the number of switching angles determined are changed to obtain varied pulse patterns, and after the changes according to step d. have been carried out, steps c. and d. are repeated for a different cycle rate of the plurality of cycle rates.

9. Simulation method according to claim 8, wherein to determine the motor losses the motor model takes into account the iron, magnet and / or copper losses of the motor and supplies these separately for a fundamental oscillation and harmonics with regard to the considered operating point, and for each cycle rate the iron, magnet and / or copper losses of the motor are determined only once in the corresponding steps c. and d. for the fundamental oscillation and for the harmonics for each change in the switching angles according to d.

10. A computer program configured, when executed on a computer, to carry out the simulation method according to any one of the preceding claims 1 to 9.

11. Operating an electric drive having an electric motor and an inverter, wherein those switching angles stored according to the method according to any one of claims 1 to 9 are retrieved from a memory, and wherein the inverter is operated according to the switching angles and generates a current that is supplied to the motor.

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