Rotary machine control device, rotary machine drive system, and electric vehicle

Abstract

According to the present invention, a power conversion unit converts a DC voltage of a DC power supply into an AC voltage and applies the AC voltage to a rotary machine, a current detection unit detects a current supplied from the power conversion unit to the rotary machine, and a power conversion control unit issues a switching command to the power conversion unit, generates a first voltage command to be applied to the rotary machine on the basis of the detected current, calculates, on the basis of a d-axis voltage on a d-axis and a q-axis voltage on a q-axis, which are synchronized with the rotation of the rotary machine, the phase of a second voltage command so as to be equal to the phase of the first voltage command, and determines, as the amplitude of the second voltage command, a voltage limit value corresponding to the DC voltage of the DC power supply.

Description

Rotating machine control device, rotating machine drive system, and electric vehicle

[0001] The present disclosure relates to a rotating machine control device, a rotating machine drive system, and an electric vehicle.

[0002] In the control of rotating machines such as synchronous machines and induction motors, a voltage vector is determined based on the current flowing through the rotating machine, and a voltage command is derived from the determined voltage vector. The voltage vector has voltage amplitude and voltage phase as elements. The current supplied to the rotating machine is controlled by switching based on the derived voltage command value. One method for controlling the rotating machine current is voltage phase control. Voltage phase control fixes the amplitude of the voltage command to a constant value and manipulates the voltage phase. One example of voltage phase control is square wave control. Square wave control fixes the amplitude of the voltage command to the maximum voltage and sets the number of pulses per electrical angle cycle to one. The number of pulses corresponds to the number of changes in the switching operation of the semiconductor switching elements of the inverter. Square wave control is also called one-pulse control. For convenience, in this application, voltage phase control refers to a method in which the number of pulses per electrical angle cycle is greater than one, and is distinguished from square wave control.

[0003] When the induced voltage of a rotating machine is sufficiently smaller than the maximum voltage it can output, current vector control is performed. When the difference between the induced voltage and the maximum voltage becomes small, the control may switch to voltage phase control or square wave control. Current vector control does not fix the voltage amplitude to a limit value. Voltage phase control or square wave control fixes the voltage amplitude to a constant value and controls only the voltage phase. As a result, the electrical resonance components of the rotating machine cannot be sufficiently suppressed, resulting in large current pulsations. In particular, square wave control has a smaller number of pulses per electrical angle cycle than voltage phase control, which increases the proportion of harmonic components in the voltage input to the rotating machine, resulting in significant current pulsations. While control is performed to suppress the pulsations by feeding back the pulsating current, the pulsations may not be canceled out and may continue, or the control may become unstable. Ensuring the stability of the control system requires reducing the responsiveness of the current controller, which may ultimately result in reduced current and torque responsiveness.

[0004] For example, the motor control device described in Patent Document 1 aims to improve the torque response performance in rectangular wave driving of a synchronous motor. More specifically, the motor control device calculates a d-axis voltage command v on a two-axis orthogonal coordinate system of d-axis and q-axis synchronized with the rotation speed of a rotating machine. d1 and the q-axis voltage command v q1 The d-axis voltage command v is generated so that the voltage amplitude output to the inverter becomes the maximum voltage that the inverter can output. d1 The difference between the d-axis voltage command v d2 Calculate the maximum power and the d-axis voltage command v d1 The difference between the q-axis voltage command v q2 The command tracking of the d-axis current that manages the induced voltage is improved, so there is no delay in the response of the current and torque.

[0005] Japanese Patent Publication No. 2008-5629 Japanese Patent No. 4556572 Japanese Patent No. 5584794

[0006] The sensor detection values, including the current information of the rotating machine, contain not only fundamental wave components based on the operation command, but also pulsation components caused by elements such as the rotating machine, inverter, and sensor, as well as their structure. When controlling the rotating machine current by feedback, a voltage command is generated to cancel out the pulsation component, so the d-axis voltage command v d1 contains a pulsating component. In some cases, the pulsating component is q1 The d-axis voltage command v, which includes a pulsating component, is also included. d1 If the voltage phase for square wave control is calculated based only on the voltage phase, the pulsating component contained in the voltage phase obtained by the calculation may be amplified.

[0007] Furthermore, when square wave control is performed using the technique described in Patent Document 1, the voltage phase value differs between current vector control and square wave control. When switching between current vector control and square wave control occurs, the voltage phase value fluctuates abruptly, causing pulsation in the voltage command. This ultimately leads to a problem of large pulsation in the current of the rotating machine and the torque generated by the rotating machine. The same problem occurs when voltage phase control is used instead of square wave control.

[0008] A first aspect of the present disclosure is a rotating machine control device including a power conversion unit that converts a DC voltage of a DC power supply into an AC voltage and applies the AC voltage to a rotating machine, a current detection unit that detects a current supplied from the power conversion unit to the rotating machine, and a power conversion control unit that gives a switching command to the power conversion unit, wherein the power conversion control unit has a voltage command generation unit that generates a first voltage command to be applied to the rotating machine based on the current, and a voltage command correction unit that corrects the first voltage command to a second voltage command, and the voltage command correction unit calculates a phase of the second voltage command based on a d-axis voltage on a d-axis and a q-axis voltage on a q-axis that are synchronized with rotation of the rotating machine, so that the phase of the second voltage command is equal to the phase of the first voltage command, and determines a voltage limit value corresponding to the DC voltage of the DC power supply as the amplitude of the second voltage command.

[0009] A second aspect of the present disclosure may be a rotating machine drive system including the rotating machine and the rotating machine control device.

[0010] A third aspect of the present disclosure may be an electric vehicle including the rotary machine drive system.

[0011] According to the rotating machine control device disclosed in the present application, pulsation of the current flowing through the rotating machine and pulsation of the torque generated in the rotating machine are suppressed.

[0012] 1 is a block diagram showing a first configuration example of a rotating machine drive system according to embodiment 1. FIG. 2 is a block diagram showing a second configuration example of a rotating machine drive system according to embodiment 1. FIG. 3 is a diagram showing a hardware configuration example of a power conversion control unit according to embodiment 1. FIG. 4 is a block diagram showing a functional configuration example of a power conversion control unit according to embodiment 1. FIG. 5 is a diagram showing an operation example of a pulse width modulation signal generation unit according to embodiment 1. FIG. 6 is a block diagram showing an example of a configuration of a voltage command generation unit according to embodiment 1. FIG. 7 is a block diagram showing an example of a configuration of a voltage command correction unit according to embodiment 1. FIG. 8 is a diagram illustrating voltage phases according to embodiment 1. FIG. 9 is a diagram illustrating voltage phases according to a comparative example. FIG. 10 is a comparison diagram illustrating time changes in voltage phases. FIG. 11 is a comparison diagram illustrating voltage phase pulsation. FIG. 12 is a comparison diagram illustrating axial currents. FIG. 13 is a diagram illustrating the operation of a voltage phase calculator according to the comparative example. FIG. 14 is a block diagram showing an example of a configuration of a voltage command generation unit according to embodiment 2. FIG. 15 is a block diagram showing an example of a configuration of a voltage command regulator according to embodiment 2. FIG. 16 is a comparison diagram illustrating configuration examples of a voltage command generation unit in a rotating machine drive system. FIG. 17 is a comparison diagram illustrating configuration examples of a voltage command generation unit and a voltage command correction unit in a rotating machine drive system. FIG. 18 is a Bode diagram illustrating an open-loop transfer function of a rotating machine drive system. FIG. 19 is a block diagram showing an example of a configuration of a voltage command correction unit according to embodiment 3. FIG. 10 is a diagram illustrating an example of the configuration of an electric vehicle according to a fourth embodiment.

[0013] Hereinafter, embodiments of the present disclosure will be described with reference to the drawings. Elements common to or corresponding to each drawing are designated by the same reference numerals, and the description thereof will be incorporated unless otherwise specified. <Embodiment 1> First, embodiment 1 of the present disclosure will be described. FIG. 1 is a schematic block diagram showing a first configuration example of a rotating machine drive system S1 according to this embodiment. The rotating machine drive system S1 according to this embodiment includes a rotating machine control device 1 and a rotating machine 6. The rotating machine control device 1 includes a power conversion control unit 2, a power conversion unit 4, and a current detection unit 5.

[0014] The power conversion control unit 2 includes a controller that generates a switching command based on the operation command and the current value of the AC current and outputs the generated switching command to the power conversion unit 4. The operation command indicates a target value for one or more items of the torque, rotational speed, current, voltage, etc. of the rotating machine 6. The operation command is input from outside the rotating machine drive system S1, for example, from a vehicle controller 204 (described later). The current value is input from the current detection unit 5. The switching command is represented by an electrical signal consisting of a series of pulse commands. The power conversion control unit 2 can also be considered as an AC power regulator that mainly controls the operation of the power conversion unit 4. An example configuration of the power conversion control unit 2 will be described later.

[0015] The power conversion unit 4 is connected to the DC power supply 3 and the rotating machine 6 via wiring. The power conversion unit 4 converts the DC voltage supplied from the DC power supply 3 into an AC voltage based on a switching command input from the power conversion control unit 2. The power conversion unit 4 applies the converted AC voltage to the rotating machine 6, supplying AC power to the rotating machine 6. The power conversion unit 4 is, for example, an inverter or a converter having an inverter function. The power conversion unit 4 includes, for example, a plurality of switching elements that form a full-bridge circuit.

[0016] The current detection unit 5 detects the AC current of each phase supplied from the power conversion unit 4 to the rotating machine 6, and notifies the voltage command generation unit 11 of the current value of the detected AC current. When three-phase AC power is supplied from the power conversion unit 4 to the rotating machine 6, the u-phase current i u , v-phase current i v and w-phase current i w is detected. When the rotation speed of the rotating machine 6 approaches zero, the AC component contained in the AC voltage supplied from the power conversion unit 4 to the rotating machine 6 decreases and approaches a DC voltage. In that case, the AC current supplied from the power conversion unit 4 to the rotating machine 6 also approaches a DC current. The AC voltage and the AC current may contain a DC offset component. Instead of detecting the currents of some or all of the phases supplied to the rotating machine 6, the current detection unit 5 may notify the voltage command generation unit 11 of a predetermined current command value as a current value.

[0017] The rotating machine control device 1 may include or use a current estimation unit (not shown) instead of the current detection unit 5. The current estimation unit detects the AC voltage applied to each phase by the power conversion unit 4, and calculates a current estimation value for each phase based on the detected AC voltage and preset rotating machine characteristics. When calculating the current estimation value, the current estimation unit uses, for example, the relationship between AC current and AC voltage shown in Equation (1). The current estimation unit may detect a DC current supplied from the DC power source 3 to the power conversion unit 4, and calculate a current estimation value for each phase from the detected DC current. The current estimation unit notifies the voltage command generation unit 11 of the calculated current estimation value as a current value of the AC current.

[0018]

[0019] Equation (1) is a voltage equation that gives the voltage generated in the rotating machine 6 in uvw coordinates, which are fixed coordinates. In equation (1), v u , v v , v w indicate the voltages of the u, v, and w phases, respectively. R and s indicate the winding resistance and the Laplace operator, respectively. L u , L v , L w are the self-inductances of the u, v, and w phase windings, respectively. M uv , M vw , M wu are the mutual inductances between the uv phases, the vw phases, and the wu phases, respectively. re , φ mag , θ re indicate the electrical angular velocity, the permanent magnet magnetic flux, and the electrical angular position, respectively. The current estimator may be substituted for the current detection unit 5 to reduce production costs, or may be substituted for the current detection unit 5 regardless of production costs. The current estimator may be substituted when the accuracy of the current detected by the current detection unit 5 is low. In that case, the current detection accuracy is improved. The current estimator may be substituted when the responsiveness of the current detected by the current detection unit 5 is low. In that case, a transiently changing current is observed.

[0020] The rotating machine 6 consumes AC power supplied from the power conversion unit 4 to rotate its rotor. The rotating machine 6 may be, for example, a three-phase synchronous motor, a three-phase induction motor, a double-winding three-phase motor, etc. The rotating machine 6 may also be a motor other than a three-phase motor such as a five-phase motor, such as a field-winding motor, a synchronous reluctance motor, or a switched reluctance motor.

[0021] Next, a second configuration example of the rotating machine drive system S1 according to the present disclosure will be described. Fig. 2 is a schematic block diagram showing the second configuration example of the rotating machine control device according to the present embodiment. The rotating machine drive system S1 illustrated in Fig. 2 includes a rotating machine control device 1 and a rotating machine 6, as well as a DC voltage detection unit 7 and a rotor position detection unit 8. That is, the rotating machine drive system S1 according to this configuration example differs from the rotating machine drive system S1 according to the first configuration example in that it further includes the DC voltage detection unit 7 and the rotor position detection unit 8.

[0022] The DC voltage detection unit 7 detects the DC voltage supplied from the DC power supply 3 to the power conversion unit 4. The DC voltage detection unit 7 transmits the detected DC voltage to the power conversion control unit 2. The DC voltage detection unit 7 includes, for example, a voltage sensor. The rotor position detection unit 8 detects rotor position information indicating the position of the rotor provided in the rotating machine 6. For example, the rotation angle is detected as the rotor position. The rotor position detection unit 8 transmits the detected rotor position information to the power conversion control unit 2. The rotor position detection unit 8 may include, for example, any of sensors such as a resolver, an encoder, or a Hall element.

[0023] Next, an example of the hardware configuration of the power conversion control unit 2 will be described. FIG. 3 is a block diagram showing an example of the hardware configuration of the power conversion control unit 2 according to this embodiment. The power conversion control unit 2 includes a processor 100 and a storage device 101. The storage device 101 includes a volatile storage device and an auxiliary storage device (not shown). The volatile storage device includes, for example, a random access memory. The non-volatile auxiliary storage device includes, for example, a flash memory. The non-volatile auxiliary storage device may include, for example, a hard disk drive, a solid state disk drive, or the like, instead of or in addition to the flash memory. The power conversion control unit 2 may be configured, for example, as a microcontroller including a computer system, or as an integrated circuit.

[0024] Next, a description will be given of an example of the functional configuration of the power conversion control unit 2. Fig. 4 is a block diagram showing an example of the functional configuration of the power conversion control unit 2 according to this embodiment. The power conversion control unit 2 includes a voltage command generation unit 11, a voltage command correction unit 12, a carrier signal generation unit 13, and a pulse width modulation signal generation unit 14.

[0025] The voltage command generator 11 generates a voltage command 1 based on the operation command, AC current, rotor position, rotational speed, DC voltage, and voltage saturation amount. The voltage command generator 11 outputs the generated voltage command 1 to the voltage command corrector 12 and the carrier signal generator 13. The operation command is input to the voltage command generator 11 from outside the rotating machine drive system S1. The current value of the AC current is notified to the voltage command generator 11 from the current detector 5.

[0026] If the rotating machine drive system S1 includes a rotor position detector 8, the rotor position is input from the rotor position detector 8 to the voltage command generator 11. The voltage command generator 11 calculates the rotation speed by differentiating the rotor position. If the rotating machine drive system S1 does not include a rotor position detector 8, the power conversion controller 2 may estimate the rotor position using, for example, a preset estimation model from the current value of the AC current. The estimated value of the rotor position obtained by estimation and the rotation speed obtained by differentiating the estimated value can be used to generate the voltage command 1.

[0027] When the rotating machine drive system S1 is provided with a DC voltage detection unit 7, the DC voltage is input from the DC voltage detection unit 7 to the voltage command generation unit 11. When the rotating machine drive system S1 does not have a DC voltage detection unit 7, a set value of the DC voltage may be set in advance in the voltage command generation unit 11, and this set value may be used to generate the voltage command 1.

[0028] The voltage command corrector 12 calculates the voltage saturation amount based on the DC voltage, the voltage command 1, and the rotor position, and corrects the voltage command 1 to a voltage command 2. The voltage command corrector 12 outputs the voltage command 2 obtained by the correction to the pulse width modulation signal generator 14, and outputs the calculated voltage saturation amount to the voltage command generator 11. The voltage command 1 is input from the voltage command generator 11.

[0029] If the rotating machine drive system S1 includes a DC voltage detection unit 7, the DC voltage is input from the DC voltage detection unit 7 to the voltage command correction unit 12. If the rotating machine drive system S1 does not include a DC voltage detection unit 7, the voltage command correction unit 12 may set a DC voltage in advance and use this set value to correct the voltage command 1. If the rotating machine drive system S1 includes a rotor position detection unit 8, the rotor position is input from the rotor position detection unit 8 to the voltage command correction unit 12. If the rotating machine drive system S1 does not include a rotor position detection unit 8, the voltage command correction unit 12 may use the above-mentioned estimated value of the rotor position to correct the voltage command 1.

[0030] The carrier signal generating unit 13 calculates the phase of one of the phases (for example, the u-phase) based on the voltage command 1 and the rotor position, and generates a triangular wave having a frequency that is an integer multiple of the rotational speed as a carrier signal based on the calculated phase. The voltage command 1 is input from the voltage command generating unit 11. If the rotating machine drive system S1 is provided with a rotor position detecting unit 8, the rotor position is input from the rotor position detecting unit 8 to the carrier signal generating unit 13. If the rotating machine drive system S1 does not have the rotor position detecting unit 8, the carrier signal generating unit 13 may use the above-mentioned estimated value of the rotor position to generate the carrier signal.

[0031] A pulse width modulation (PWM) method based on a carrier signal generated using a voltage command 1 and rotor position is called synchronous PWM. PWM based on a carrier signal having a frequency 3, 6, 9, or 15 times the rotation speed is called synchronous 3P, synchronous 6P, synchronous 9P, or synchronous 15P, respectively. The amplitude of the carrier signal may be normalized to fall within a predetermined range. The range may be, for example, 0 to 1, or -0.5 to 0.5. The range of the carrier signal may be set to be equal to the range of the output voltage.

[0032] Note that carrier signal generator 13 may generate a carrier signal without using voltage command 1 and rotor position. PWM based on a carrier signal generated without using voltage command 1 and rotor position is called asynchronous PWM. Carrier signal generator 13 outputs the generated carrier signal to pulse width modulation signal generator 14.

[0033] The pulse-width modulation signal generator 14 generates a switching command based on the DC voltage, voltage command 2, and carrier signal. More specifically, the pulse-width modulation signal generator 14 generates a modulation signal having an amplitude equal to a normalized voltage obtained by dividing voltage command 2 by the DC voltage. The range of values ​​is, for example, 0 to 1, or -0.5 to 0.5. The pulse-width modulation signal generator 14 compares the amplitude of the generated modulation signal with the amplitude of the carrier signal to determine which is larger. Generally, the larger the amplitude of the modulation signal, the less the amplitudes of the fundamental wave components of the modulation signal and the switching command match. Before determining the minimum amplitude relationship, the pulse-width modulation signal generator 14 may correct the amplitude of the modulation signal so that the amplitudes of the fundamental wave components match.

[0034] The pulse-width modulation signal generation unit 14 generates a binary switching command indicating either H (High) or L (Low) depending on the determined magnitude relationship. For example, when the amplitude of the modulated signal is equal to or greater than the amplitude of the carrier signal, H is set as the voltage value of the switching command, and when the amplitude of the modulated signal is less than the amplitude of the carrier signal, L is set as the voltage value of the switching command. When the pulse-width modulation signal generation unit 14 is configured according to positive logic, H and L correspond to 1 and 0, respectively. H is a voltage value significantly higher than L. When the pulse-width modulation signal generation unit 14 is configured according to negative logic, H and L correspond to 0 and 1, respectively.

[0035] When synchronous 3P is applied to current vector control or voltage phase control, the number of times the switching command changes per electrical cycle is three. That is, the number of times the voltage value becomes H, the number of times the voltage value becomes L, and the number of pulses in one electrical cycle are each three. When synchronous 6P is applied, the number of times the switching command changes per electrical cycle is six. That is, the number of times the voltage value becomes H, the number of times the voltage value becomes L, and the number of pulses in one electrical cycle are each six. In square wave control, the number of times the switching command changes per electrical cycle is one. That is, the number of times the voltage value becomes H, the number of times the voltage value becomes L, and the number of pulses in one electrical cycle are each one.

[0036] In general, the more times the switching command changes per electrical angle cycle, the greater the switching loss generated in the semiconductor switching elements. In other words, the fewer times the switching command changes, the smaller the switching loss in the power conversion unit 4, and therefore, in the case of square wave control, the switching loss is minimized. However, the fewer times the switching command changes, the greater the proportion of harmonic components contained in the output from the power conversion unit 4. Therefore, pulsations in voltage and current due to harmonics become more pronounced, and control of these pulsations tends to become more difficult.

[0037] In synchronous PWM, when the frequency of the carrier signal is an integer multiple of the rotation speed, the waveforms of the switching commands for the u-phase phase from 0 to 180 degrees and from 180 to 360 degrees can be maintained point-symmetrically about the 180-degree phase. Furthermore, when the frequency of the carrier signal is an odd multiple of the rotation speed, the waveforms of the switching commands for the 0 to 90 degrees and from 90 to 180 degrees can be maintained line-symmetrically about the 90-degree phase. Furthermore, the waveforms of the switching commands for the 180 to 270 degrees and from 270 to 360 degrees can be maintained line-symmetrically about the 270-degree phase. This reduces the error between the commanded AC voltage output to the rotating machine 6 and the fundamental component of the actual AC voltage, and suppresses unnecessary harmonic components. In contrast, with square-wave control, the phase specified by a single switching command tends to have a significant effect on the AC error. In this respect, synchronous PWM is desirable.

[0038] When the carrier signal has a triangular waveform, the relationship between the command AC voltage output to the rotating machine 6 and the amplitude of the fundamental component of the AC voltage actually supplied can be approximately linear. The waveform of the carrier signal is not limited to a triangular wave and may be other shapes, such as a sine wave or a rectangular wave. Even in such cases, a correlation can be obtained between the command AC voltage output to the rotating machine 6 and the amplitude of the fundamental component of the AC voltage actually supplied. In general, the larger the amplitude of the modulation signal, the more likely it is that the amplitudes of the fundamental components of the modulation signal and the switching command will not match. Before determining the minimum amplitude relationship, the pulse-width modulation signal generator 14 may correct the amplitude of the modulation signal so that the amplitudes of the fundamental components of the modulation signal and the switching command match.

[0039] Next, an example of the operation of the pulse-width modulation signal generator 14 according to this embodiment will be described. FIG. 5 is a diagram showing an example of the operation of the pulse-width modulation signal generator 14 according to this embodiment. FIG. 5 illustrates an example of the modulation signal, carrier signal, and switching command for one electrical angle period when rectangular wave control is performed under synchronous 3P and positive logic. The carrier signal has three periods of triangular waves in one electrical angle period. The amplitude of the modulation signal corresponds to a value normalized by dividing the voltage command 2 by the DC voltage. The amplitude of the carrier signal is normalized so that its minimum value is 0 and its maximum value is 1.

[0040] In the example of Figure 5, in the section where the phase is from 0 degrees to 180 degrees, the amplitude of the modulated signal is greater than the amplitude of the carrier signal. In this section, the value of the switching command is set to 1. In the section where the phase is from 180 degrees to 360 degrees, the amplitude of the modulated signal is smaller than the amplitude of the carrier signal. In this section, the value of the switching command is set to 0. Therefore, the number of changes in the switching command is one.

[0041] In the square wave control illustrated in FIG. 5 , the maximum and minimum amplitudes of the modulation signal are greater than the maximum and smaller than the minimum amplitudes of the carrier signal, respectively, so the number of changes in the switching command is one. Here, if the amplitude of the modulation signal is made smaller than the amplitude of the carrier signal, the number of intersections between the waveforms of the modulation signal and the carrier signal increases. This increases the number of changes in the switching command per electrical angle cycle. Therefore, by manipulating the amplitude of the modulation signal, it is possible to switch between square wave control and voltage phase control or current vector control.

[0042] For example, when the induced voltage of the rotating machine 6 is lower than a predetermined reference value of the induced voltage, the pulse-width-modulated signal generator 14 adjusts the amplitude range of the modulated signal so that it is narrower than the amplitude range of the carrier signal. In this case, the number of changes in the switching command per one electrical angle cycle depends on the cycle of the carrier signal, so voltage phase control or current vector control is realized. When the induced voltage of the rotating machine 6 is equal to or greater than the reference value, the pulse-width-modulated signal generator 14 adjusts the amplitude range of the modulated signal so that it is wider than the amplitude range of the carrier signal. In this case, the number of changes in the switching command per one electrical angle cycle is limited to one, so square-wave control is realized.

[0043] Next, a description will be given of an example of the configuration of the voltage command generating unit 11. Fig. 6 is a block diagram showing an example of the configuration of the voltage command generating unit 11 according to this embodiment. The voltage command generating unit 11 includes a current command generator 21, a coordinate converter 22, and a current controller 23.

[0044] The current command generator 21 generates current commands i for the d-axis and q-axis of the rotating machine 6 so that losses in the rotating machine 6 and the power conversion unit 4 are minimized, that is, reduced as much as possible, within the range that the rotating machine 6 can output, based on the operation command, the rotation speed, and the DC voltage. d * , i q * The current command generator 21 generates the current command i so that the loss in the rotating machine 6 and the power conversion unit 4 is maximized, that is, as large as possible. d * , i q * The current command generator 21 may generate the generated current command i d * , i q * is output to the current controller 23.

[0045] The coordinate converter 22 calculates the d-axis current i based on the AC current and the rotor position. d and q-axis current i q The coordinate converter 22 calculates the current i of each phase according to, for example, equation (2). u , i v , i w is the rotor position θre Based on the d-axis current i d and q-axis current i q The coordinate converter 22 converts the converted d-axis current i d and q-axis current i q is output to the current controller 23.

[0046]

[0047] The current controller 23 calculates the d-axis current i d and q-axis current i q is the current command i input from the current command generator d * , i q * The current controller 23 outputs the generated voltage command 1 to the voltage command corrector 12 and the carrier signal generator 13.

[0048] The current controller 23 calculates the d-axis current i d and q-axis current i q and the current command i d * , i q * Based on the d-axis voltage command v d1 and the q-axis voltage command v q1 A vector [v d1 , v q1 ] T ( T is calculated as the voltage command 1. The first term of the formula (3) relates to PI control, and the second term relates to decoupling control. In PI control, the current i d , i q and current command i d * , i q * The main purpose of decoupling control is to reduce the induced voltage components that interfere with the d-axis voltage and the q-axis voltage. In equation (3), K pd , K pq indicate the P gain for the d-axis and q-axis, respectively. K id , K iqindicates the I gain for the d-axis and q-axis, respectively. s indicates the Laplace operator. v ddec , v qdec indicate the induced voltages generated on the d-axis and q-axis, respectively. The induced voltages are proportional to the rotation speed. K pd , K pq , K id , K iq are determined based on the specifications of the rotating machine 6. ddec , v qdec is calculated based on the product of the magnetic flux and the rotational speed of the rotating machine 6.

[0049]

[0050] Note that the current controller 23 does not necessarily need to use equation (3) when determining the voltage command 1, as long as the required specifications can be met. For example, instead of PI control, any of P control, I control, PD control, PID control, I-PD control, and two-degree-of-freedom control may be used. Furthermore, the current controller 23 may determine the voltage command 1 using the deviation between the current of each axis and the current command, without using the induced voltage component. When I control is included in the process of determining the voltage command 1, the current controller 23 may manipulate the accumulated value stored in an integrator that performs I control based on the voltage saturation amount. This manipulation is called anti-windup processing. According to the anti-windup processing, the accumulated value is adjusted so that the voltage command 1 does not exceed the voltage saturation amount.

[0051] Next, a description will be given of an example configuration of the voltage command corrector 12. Fig. 7 is a block diagram showing an example configuration of the voltage command corrector 12 according to this embodiment. The voltage command corrector 12 includes a voltage phase calculator 31, a voltage amplitude calculator 32, a control switch determiner 33, a voltage amplitude selector 35, an orthogonal coordinate converter 36, a voltage saturation calculator 37, and a coordinate converter 38.

[0052] The voltage phase calculator 31 calculates the voltage phase θ of the voltage command 1. v1 Calculate the calculated voltage phase θ v1 The voltage phase θ of voltage command 2 v2 The voltage command v of each axis, which is the element of voltage phase 1, is defined as d1 , v q1 and voltage phase θv1 ,θ v2 The relationship between these is exemplified in equation (4). In equation (4), tan -1 (...) indicates the arccosine function of .... The voltage phase calculator 31 outputs the calculated voltage phase to the Cartesian coordinate converter 36.

[0053]

[0054] The voltage amplitude calculator 32 calculates the voltage amplitude for each control method based on the voltage command 1 and the DC voltage. For vector control, the voltage amplitude calculator 32 calculates the voltage command v for each axis that constitutes the voltage command 1 as shown in equation (5). d1 , v d2 The square root of the sum of the squares of the voltage amplitude v amp2 However, the voltage amplitude v calculated using equation (5) is amp2 is the voltage limit value v lim If the voltage amplitude calculator 32 determines that the voltage limit value v lim The voltage amplitude v amp2 Set as.

[0055]

[0056] The voltage amplitude calculator 32 always uses the voltage limit value v lim The voltage amplitude v amp2 The voltage amplitude calculator 32 multiplies the DC voltage by the voltage utilization rate to obtain the voltage limit value v lim The voltage utilization rate differs depending on the control method. Therefore, a different voltage utilization rate is set in advance in the voltage amplitude calculator 32 for each modulation method employed in each control method, and the voltage limit value v is calculated using the voltage utilization rate corresponding to the modulation method. lim For example, the voltage utilization factor for sinusoidal modulation is √(3 / 8), for third-order harmonic superposition or two-phase modulation it is 1 / √2, and for square wave control it is √6 / π. In other words, the voltage utilization factors for current vector control and voltage phase control are both less than √6 / π, which is lower than the voltage utilization factor for square wave control, √6 / π.

[0057] For square wave control, the voltage amplitude calculator 32 is set to a value slightly larger than the voltage utilization rate, and the DC voltage is multiplied by the set value to obtain the voltage limit value v lim This ensures that the range of the amplitude of the modulating signal is wider than the amplitude of the carrier signal, thereby ensuring the realization of rectangular wave control. In other words, the generation of short pulses near the peaks of the carrier signal is prevented. The voltage amplitude calculator 32 is set to a value slightly smaller than the voltage utilization rate for current vector control and voltage phase control, and the DC voltage is multiplied by this set value to obtain the voltage limit value v lim As a result, the range of the amplitude of the modulated signal is reliably narrower than the amplitude of the carrier signal, making it possible to reliably achieve current vector control and voltage phase control. The voltage amplitude calculator 32 outputs the calculated voltage amplitude to the voltage amplitude selector 35.

[0058] The control switching determiner 33 switches the control method between current vector control and voltage phase control or square wave control based on the voltage command 1 and the DC voltage. The control switching determiner 33 notifies the voltage amplitude selector 35 of a switching signal indicating the control method selected by the switching. More specifically, the control switching determiner 33 calculates a voltage utilization rate from the voltage command 1 and the DC voltage. The voltage utilization rate corresponds to the ratio of the maximum voltage command value for each phase that constitutes the voltage command 1 to the DC voltage. The control switching determiner 33 compares the calculated voltage utilization rate with a predetermined reference value for the voltage utilization rate to determine whether the calculated voltage utilization rate is larger or smaller than the reference value. The control switching determiner 33 selects the control method corresponding to the determined magnitude relationship and outputs a switching flag indicating the selected control method to the voltage amplitude selector 35.

[0059] For example, when the voltage utilization rate is greater than a reference value, the control switching determiner 33 determines the control method to be voltage phase control or square wave control. By determining voltage phase control or square wave control, it is possible to increase the output voltage. When square wave control is selected, it is possible to minimize the number of switching operations per electrical angle cycle, thereby reducing switching loss in the power conversion unit 4. When the voltage utilization rate is equal to or less than the reference value, the control switching determiner 33 selects current vector control. By selecting current vector control, it is possible to maintain the switching frequency at a certain level or higher, reduce harmonic voltages output from the power conversion unit 4, and maintain high efficiency of the rotating machine 6.

[0060] The control switching determiner 33 may calculate the modulation factor by further dividing the voltage utilization factor calculated from the voltage command 1 and the DC voltage by the maximum voltage utilization factor related to the modulation method. The control switching determiner 33 may determine whether to select the control method from voltage phase control, square wave control, or current vector control based on whether the calculated modulation factor is greater than a predetermined reference value of the modulation factor. By applying the control method selected in this way depending on the voltage utilization factor or modulation factor, the efficiency of the rotating machine 6 and the power conversion unit 4 can be maintained high.

[0061] The voltage amplitude selector 35 selects, from the voltage amplitudes input from the voltage amplitude calculator 32, a voltage amplitude corresponding to the control method indicated by the switching signal input from the control switching determiner 33. For example, when the switching flag input from the control switching determiner 33 indicates current vector control, the voltage amplitude selector 35 selects the voltage amplitude calculated for current vector control. When the switching flag input from the control switching determiner 33 indicates voltage phase control or square wave control, the voltage amplitude selector 35 selects the voltage amplitude calculated for voltage phase control or square wave control. The voltage amplitude selector 35 outputs the selected voltage amplitude to the Cartesian coordinate converter 36.

[0062] The orthogonal coordinate converter 36 converts the voltage amplitude v input from the voltage amplitude selector 35 into amp2 and the voltage phase θ input from the voltage phase calculator 31 v2The orthogonal coordinate converter 36 converts the polar coordinate values ​​into the d-axis voltage command v d2 and the q-axis voltage command v q2 The orthogonal coordinate converter 36 converts the voltage command v for each axis into an orthogonal coordinate value including v as an element. d2 , v q2 is output to the coordinate converter 38.

[0063] The voltage saturation amount calculator 37 determines the voltage saturation amount based on the voltage amplitude input from the voltage amplitude selector 35 and the voltage command 1. More specifically, the voltage saturation amount calculator 37 determines the voltage saturation amount based on the voltage amplitude v amp2 and the voltage amplitude of voltage command 1, the voltage amplitude of voltage command 1 is greater than the voltage amplitude v amp2 If it is larger than the voltage amplitude of voltage command 1 and the voltage amplitude v amp2 The difference between the voltage amplitude of voltage command 1 and the voltage amplitude v amp2 If the following holds, the voltage saturation amount is determined to be zero. The voltage saturation amount calculator 37 outputs the determined voltage saturation amount to the voltage command generator 11.

[0064] The coordinate converter 38 converts the voltage command v for each axis input from the Cartesian coordinate converter 36 into d2 , v q2 is the rotor position θ re More specifically, the coordinate converter 38 converts the voltage command v of each phase into a voltage command 2 in accordance with the equation (6). u , v v , v w The coordinate converter 38 calculates a voltage command 2 including the above as an element. The coordinate converter 38 outputs the converted voltage command 2 to the pulse width modulation signal generator 14.

[0065]

[0066] Next, an example of the voltage phase according to this embodiment will be described. FIG. 8 is a diagram illustrating the voltage phase according to this embodiment. In each of FIGS. 8(a) and 8(b), the vertical axis indicates the q-axis voltage, and the horizontal axis indicates the d-axis voltage. The voltage command 1 starts at the origin and ends at v dq1 The voltage limit is represented by the radius of the dashed arc.

[0067] FIG. 8(a) shows the amplitude v of the voltage command 1. dq1 8(b) shows a case where the amplitude v of the voltage command 1 is smaller than the voltage limit value. This case can be used in voltage phase control. dq1 8(a) and 8(b), the voltage phase calculator 31 calculates the phase θ of the voltage command 1. v1 is the voltage phase θ of voltage command 2 v2 Even when switching between current vector control and voltage phase control is performed, the voltage phase is equal between voltage command 2 and voltage command 1, so that the shock caused by the switching can be reduced.

[0068] As a comparative example, according to an embodiment described in Patent Document 1, the corrected voltage phase θ v2 is calculated according to equation (7). In equation (7), sign(...) denotes the sign function, and v lim indicates the voltage limit value. The sign function outputs 1 when the input value is a positive value, and outputs 0 when the input value is a negative value. Therefore, in the comparative example, the corrected voltage phase θ v2 is the voltage phase before correction θ v1 is different.

[0069]

[0070] Next, an example of a voltage phase according to a comparative example will be described. Fig. 9 is a diagram illustrating a voltage phase according to a comparative example. Fig. 9(a) shows the amplitude v of the voltage command 1. d1 This example is sometimes used in voltage phase control. d1 This example shows a case where the voltage phase θ is greater than the voltage limit value. This example may be used in current vector control or voltage phase control. v2 is generally the voltage phase θ v1 is different.

[0071] However, the voltage phase θ v1 When the voltage command 1 is directed to the q-axis, the voltage phase θ v2and θ v1 are the same. Therefore, when switching from one of current vector control and voltage phase control to the other, the voltage phase changes. A shock may occur when switching the control method. A voltage phase calculator that employs the voltage phase control method based on equation (5) is also called a d-axis voltage-priority voltage limiter. A voltage limiter that employs a control method different from the d-axis voltage-priority voltage limiter (for example, a q-axis voltage-priority voltage limiter) also has the same issues as the d-axis voltage-priority voltage limiter.

[0072] Next, the time changes in the voltage phase obtained by the comparative example and this embodiment will be described. FIG. 10 is a diagram illustrating the time changes in the voltage phase. In FIG. 10(a), the vertical axis shows the voltage phase calculated according to the comparative example. In FIG. 10(b), the vertical axis shows the voltage phase calculated by the voltage phase calculator 31 of this embodiment. Both FIGS. 10(a) and 10(b) illustrate the time changes in the voltage phase observed when the control method is switched from current vector control to voltage phase control. The dashed line indicates the time when the control method is switched.

[0073] FIG. 10(a) shows the voltage phase θ v1 and voltage phase θ v2 The voltage phase θ v1 and voltage phase θ v2 On the other hand, FIG. 10(b) shows that the voltage phase θ v1 and voltage phase θ v2 The voltage phase θ v1 and voltage phase θ v2 This indicates that the present embodiment can suppress the pulsation of the voltage phase that accompanies switching of the control method.

[0074] Furthermore, the voltage phase calculator 31 according to this embodiment has the advantage of being less susceptible to voltage pulsation caused by pulsation components due to feedback control of the d-axis and q-axis currents, and pulsation components mixed into the amount detected by the sensor. Fig. 11 is a diagram illustrating pulsation occurring in the voltage phase obtained by a comparative example and this embodiment. Fig. 11(a) shows the d-axis voltage pulsation Δv d is mixed into the voltage command 1 and generates a voltage phase θ v1 11(b) shows the d-axis voltage pulsation Δθv1. d The voltage phase θ calculated by the comparative example from the voltage command 1 mixed with v2 In this example, the voltage phase θ v1 The voltage phase θ v2 This shows that the pulsation Δθv2 of the voltage phase is larger than that of the voltage phase.

[0075] FIG. 11(c) shows the d-axis voltage pulsation Δv d The voltage phase θ obtained by the voltage phase calculator 31 according to this embodiment from the voltage command 1 mixed with v2 In this embodiment, the pulsation Δθv2 occurring in the voltage phase θ v1 Pulsation Δθv1 and voltage phase θ v2 By making the pulsation Δθv2 of the voltage phase constant equal to each other, the pulsation of the voltage phase is not amplified as in the comparative example. This shows that the present embodiment can suppress the pulsation occurring in the voltage phase more effectively than the comparative example.

[0076] FIG. 12 illustrates the time variation of the q-axis current obtained by executing square wave control. FIG. 12(a) shows the waveform of the q-axis current observed in the comparative example for both the detected current value and the current command (command). In the example of FIG. 12(a), the detected value and the command diverge, and the q-axis current pulsates significantly over time. FIG. 12(b) shows the waveform of the q-axis current observed in this embodiment. In the example of FIG. 12(b), both the deviation between the detected value and the command and the q-axis current pulsation are significantly suppressed compared to the example of FIG. 12(a). This also demonstrates that this embodiment suppresses pulsation in the q-axis current by not amplifying the pulsation occurring in the voltage phase.

[0077] The comparative example may also have the following problem. Fig. 13(a) illustrates a case where the d-axis component of the voltage command 1 is greater than the voltage limit value. In this case, the voltage phase θ calculated based on the equation (5) v2 When the d-axis component of the voltage command 1 is close to the voltage limit value and is slightly pulsating, the voltage phase θ v2 changes suddenly between an arbitrary value and 90 degrees. This change can cause significant pulsation in the voltage phase. FIG. 13(b) illustrates a case where voltage pulsation on the q axis crosses the d axis and is mixed into the voltage command 1. In this case, the voltage phase θ calculated based on equation (5) v2 The pulsation Δθv2 of the voltage phase θ v1 11 and 12, the voltage phase pulsation is amplified. On the other hand, according to the voltage phase calculator 31 of this embodiment, the voltage phase θ v2 is the voltage phase θ v1 Therefore, the pulsation of the voltage phase is not amplified.

[0078] As described above, the rotating machine control device 1 according to this embodiment includes the power conversion unit 4 that converts a DC voltage of a DC power supply into an AC voltage and applies the AC voltage to the rotating machine 6, the current detection unit 5 that detects a current supplied from the power conversion unit 4 to the rotating machine 6, and the power conversion control unit 2 that issues a switching command to the power conversion unit 4. The power conversion control unit 2 includes a voltage command generation unit 11 that generates a first voltage command (e.g., voltage command 1) to be applied to the rotating machine 6 based on the detected current, and a voltage command correction unit 12 that corrects the first voltage command to a second voltage command (e.g., voltage command 2). The voltage command correction unit 12 calculates the phase of the second voltage command based on a d-axis voltage on the d-axis and a q-axis voltage on the q-axis that are synchronized with the rotation of the rotating machine 6 so that the phase of the second voltage command is equal to the phase of the first voltage command, and determines a voltage limit value corresponding to the DC voltage of the DC power supply as the amplitude of the second voltage command. This embodiment may also be realized as a rotating machine drive system S1 that includes the rotating machine control device 1 and the rotating machine 6.

[0079] According to this configuration, even if pulsation is superimposed on the voltage command, the pulsation in the voltage phase is not amplified. Therefore, the pulsation in the current supplied to the rotating machine 6 can be suppressed. Consequently, the pulsation in the torque generated by the rotating machine 6 is suppressed. That is, even if pulsation is superimposed on the voltage command for each axis, the pulsation is not amplified during the calculation process of the voltage phase, regardless of the control method. Therefore, the pulsation generated in the current and torque can be suppressed. Furthermore, according to this configuration, the pulsation generated in the current and torque can be suppressed, and therefore the responsiveness of the current and torque can be improved. Furthermore, using this configuration, the control can be stabilized even in square wave control, which tends to be more affected by current pulsation than voltage phase control. Stable operation of square wave control reduces switching loss in the power conversion unit 4.

[0080] Alternatively, the voltage command corrector 12 may set the voltage limit value by multiplying the DC voltage by a value equal to or greater than √6 / π. This configuration more reliably achieves square wave control, which tends to be more susceptible to current pulsation than voltage phase control. It also prevents short pulses from occurring near the peaks of the carrier signal.

[0081] Alternatively, the power conversion control unit 2 may change the switching command once per electrical angle period. This configuration enables square wave control to be achieved, and switching loss in the power conversion unit 4 can be reduced.

[0082] Furthermore, the voltage command corrector 12 may correct the first voltage command based on the control method indicated by the switching signal. This configuration can suppress current pulsation due to voltage fluctuations that may occur when switching between control methods (e.g., between current vector control and square wave control), thereby improving control stability. For example, equalizing the voltage phase between current vector control and square wave control can mitigate shocks associated with switching between control methods. Suppressing pulsation in the current and torque enhances the control stability of the rotating machine 6 and improves the responsiveness of the current and torque. Furthermore, voltage phase control can also achieve the same effects as square wave control.

[0083] Furthermore, the power conversion control unit 2 may generate a switching command based on the second voltage command and the carrier signal, and the frequency of the carrier signal may be set to an integer multiple of the rotational speed of the rotating machine 6. This configuration achieves synchronous PWM and maintains the symmetry of the waveform of the switching signal. As a result, it is possible to reduce the error between the AC voltage command input to the rotating machine 6 and the fundamental component of the AC voltage actually supplied, and further suppress unnecessary harmonic components.

[0084] Second Embodiment Next, a second embodiment of the present disclosure will be described. The following description will focus mainly on differences from the first embodiment. For commonalities with the first embodiment, the description of the first embodiment will be used unless otherwise specified. The rotating machine drive system S1 according to this embodiment includes a voltage command generator 11A in place of the voltage command generator 11 ( FIG. 6 ) in the rotating machine control device 1.

[0085] 14 is a schematic block diagram showing an example of the configuration of a voltage command generation unit 11A according to this embodiment. The voltage command generation unit 11A includes a current command generator 21, a coordinate converter 22, a current controller 23, a correction coefficient calculator 24, and a voltage command regulator 25. The voltage command generation unit 11A differs from the voltage command generation unit 11 ( FIG. 6 ) according to the first embodiment in that it further includes the correction coefficient calculator 24 and the voltage command regulator 25.

[0086] The current controller 23 according to this embodiment calculates the voltage command 1 according to the first embodiment as a voltage command 0, and outputs the calculated voltage command 0 to the voltage command regulator 25. That is, the current controller 23 calculates the current command i of each axis according to the equation (8). d , iq, current i for each axis d , i q and the voltage command v of each axis, which becomes an element of the voltage command 0 based on the rotation speed. d0 , v q0 The equation (8) shows a calculation that is substantially the same as the equation (3). That is, the voltage command v in the equation (8) is calculated as follows: d0 , v q0 is the voltage command v in equation (3). d1 , v q1 In this embodiment, the induced voltage v ddec , v qdecis calculated based on the product of the magnetic flux and the rotational speed of the rotating machine 6.

[0087]

[0088] The correction coefficient calculator 24 calculates a correction coefficient based on the operation command input to the rotating machine control device 1 and the rotation speed of the rotating machine 6. The correction coefficient calculator 24 outputs the calculated correction coefficient to the voltage command regulator 25. The correction coefficient is used when correcting the voltage command 0 obtained from the current controller 23 to calculate the voltage command 1. The correction coefficient is calculated based on the matrix F shown in equation (9) or its element F 11 , F 12 , F 21 , F 22 The matrix F is the inverse matrix of the matrix F'. The matrix F' is a transfer function that indicates the input / output relationship at the equilibrium point of the voltage command correction unit 12. That is, in the steady state, the product obtained by multiplying the voltage command 1 by the matrix F' corresponds to the voltage command 2. Therefore, the matrix F is the inverse matrix of the voltage limit value v lim , voltage command v of each axis that constitutes voltage command 2 d2 , v q2 However, the voltage command v d2 , v q2 depends on the operating point of the rotating machine 6.

[0089]

[0090] Therefore, a data map indicating the matrix F for each value range of a set of an operation command and a rotation speed is set in advance in the correction coefficient calculator 24. The correction coefficient calculator 24 identifies a correction coefficient corresponding to a value range including the operation command and the rotation speed input from the data map, and outputs the identified correction coefficient to the voltage command regulator 25.

[0091] Instead of referring to the data map, the correction coefficient calculator 24 may calculate a matrix that gives a voltage command 2 as a multiplication value for a voltage command 0 at a time earlier than the latest cycle (for example, the next new cycle) as a correction coefficient. d , i qA function may be set in advance to calculate the current i at that time using the input value as the correction coefficient as the output value. d , i q The correction coefficient may be calculated from

[0092] The voltage command regulator 25 calculates a voltage command 1 by correcting the voltage command 0 input from the current controller 23 so as to reduce the voltage error included in the voltage command 2 calculated by the voltage command correction unit 12. Here, the voltage command regulator 25 calculates the voltage command 1 using a correction coefficient input from the correction coefficient calculator 24 for the voltage command 0. As exemplified in equation (10), the voltage command regulator 25 calculates the voltage command 1 by multiplying the voltage command 0, which is a vector, by a correction coefficient F, which is a matrix. FIG. 15 shows an example of the configuration of the voltage command regulator 25 for executing the calculation exemplified in equation (10). In the example of FIG. 15, the voltage command regulator 25 calculates the voltage command v d0 Correction coefficient F 11 , F 12 and the q-axis voltage command v q0 Correction coefficient F 21 , F 22 and a correction coefficient F 11 and a correction coefficient F 21 The products obtained from each multiplier are added together to obtain the corrected d-axis voltage command v d1 and an adder for calculating the correction coefficient F 12 and a correction coefficient F 22 The products obtained from each multiplier are added together to obtain the corrected q-axis voltage command v q1 and an adder for calculating:

[0093]

[0094] Next, the effects obtained by the voltage command regulator 25 will be described. Here, differences in the configuration of the rotating machine drive system among the comparative example, the first embodiment, and this embodiment will be described with reference to Fig. 16. In Fig. 16, (a) shows the comparative example, (b) shows the first embodiment, and (c) shows the second embodiment.

[0095] The comparative example, embodiment 1, and this embodiment are common in that they all include a voltage command generation unit, a voltage command correction unit, and a rotating machine. A voltage command correction unit 12A according to the comparative example calculates a voltage phase from a voltage command 1 input from a voltage command generation unit 11 using equation (7), and outputs a voltage command 2 having the calculated voltage phase to the rotating machine 6. In contrast, embodiment 1 and this embodiment differ from the comparative example in that the voltage phase is calculated using equation (4). This embodiment also differs from embodiment 1 or the comparative example in that the voltage command generation unit 11A includes a voltage command regulator 25.

[0096] Next, using FIG. 17 , differences in the operation of the voltage command correction unit among the comparative example, embodiment 1, and this embodiment will be described. The voltage command correction units 12 according to the comparative example, embodiment 1, and embodiment 2 are all common in that they correct a voltage command 1 and output a voltage command 2 obtained by the correction to the rotating machine 6. The transfer function F″ of the voltage command correction unit 12 according to the comparative example represents input / output characteristics at an equilibrium point. The transfer function F′ of the voltage command correction unit 12 according to embodiments 1 and 2 also represents input / output characteristics at an equilibrium point, but corresponds to the transfer function F′ exemplified in equation (9). None of the transfer functions F″ and F′ are unit matrices, and therefore the voltage command 2 output from the voltage command correction unit has a different value from the voltage command 1.

[0097] In this regard, the voltage command generation unit 11A according to the second embodiment has a voltage command regulator 25. A transfer function F indicating the input / output characteristics of the voltage command regulator 25 corresponds to the inverse matrix of the transfer function F'. Since the transfer function of the part where the voltage command regulator 25 and the voltage command correction unit 12 are connected in series is an identity matrix, the voltage command 2 output from the voltage command correction unit 12 approximates the voltage command 0 output from the current controller 23. Therefore, the amplification of pulsation by the voltage command correction unit 12 is suppressed.

[0098] Next, the control characteristics of the rotary machine drive system S1 according to this embodiment will be described. Fig. 18 is a Bode diagram illustrating an example of an open-loop transfer function of the rotary machine drive system S1. Fig. 18 shows the current command i input to the rotary machine drive system S1 for each of this embodiment and embodiment 1. d The current command i output from d18 shows an open loop transfer function up to ω. The vertical axis of FIG. 18 indicates gain in the upper part and phase in the lower part. The horizontal axis of FIG. 18 indicates frequency. The dashed line indicates the gain or phase observed from the first embodiment. The solid line indicates the gain or phase observed from the second embodiment. In both the first and second embodiments, the design value of the crossover angular frequency is set to ω. c It was set as follows.

[0099] In both the first and second embodiments, the gain decreases monotonically as the frequency increases. The change in gain due to frequency change is more gradual in this embodiment than in the first embodiment. In the second embodiment, the crossover angular frequency at which the gain becomes 0 dB is the design value ω c In contrast, in the first embodiment, the crossover angular frequency is approximately equal to the design value ω c The phase characteristics are significantly different between the first and second embodiments. In the first embodiment, the phase in the frequency band near the crossover angular frequency is significantly lower than the phase in other frequency bands. In contrast, in the second embodiment, the phase change due to the change in frequency is much gentler than in the first embodiment. In the first embodiment, the phase margin at the crossover angular frequency is only about 10 degrees, whereas in the second embodiment, the phase margin is about 90 degrees.

[0100] As described above, in the rotating machine control device 1 according to this embodiment, the voltage command generating unit 11A includes the voltage command regulator 25. The voltage command regulator 25 corrects the first voltage command (e.g., voltage command 0) so as to reduce a voltage error that occurs in the process of calculating the second voltage command (e.g., voltage command 2). This configuration not only suppresses current and torque pulsation but also improves the phase margin. This improves control stability and suppresses current pulsation due to a decrease in stability. Furthermore, it alleviates a decrease in the crossover angular frequency, thereby obtaining a response closer to the design value. This suppresses a decrease in current and torque responsiveness.

[0101] Furthermore, the passband limiter 34B may determine a correction coefficient using a predetermined data map or function based on one or more of the torque of the rotating machine 6, the rotational speed of the rotating machine 6, the past first and second voltage commands, and the current supplied to the rotating machine 6, and correct the first voltage command based on the correction coefficient. With this configuration, the voltage command is adjusted according to the operating point at that time. This improves the stability of control and further enhances the effect of suppressing current pulsation.

[0102] Third Embodiment Next, a third embodiment of the present disclosure will be described. The following description will mainly focus on differences from the first or second embodiment. For commonalities with the first or second embodiment, the description thereof will be used unless otherwise specified. The rotating machine drive system S1 according to this embodiment includes a voltage command correction unit 12B in place of the voltage command correction unit 12 ( FIG. 7 ) in the rotating machine control device 1.

[0103] 19 is a schematic block diagram showing an example of the configuration of a voltage command correction unit 12B according to this embodiment. The voltage command correction unit 12B includes a voltage phase calculator 31, a voltage amplitude calculator 32, a control switch determiner 33, a passband limiter 34B, a voltage amplitude selector 35, an orthogonal coordinate converter 36, a voltage saturation amount calculator 37, and a coordinate converter 38. The voltage command correction unit 12B according to this embodiment differs from the voltage command correction unit 12 according to the first embodiment in that it includes the passband limiter 34B.

[0104] The passband limiter 34B suppresses specific harmonic components contained in the voltage phase input from the voltage phase calculator 31 and outputs the processed voltage phase to the Cartesian coordinate converter 36. The passband limiter 34B includes, for example, a notch filter, a low-pass filter, a high-pass filter, and an all-pass filter. A notch filter attenuates the amplitude of specific frequency components while not attenuating the amplitude of other frequency components. A notch filter is also called a band-stop filter. A low-pass filter passes frequency components lower than a predetermined first cutoff frequency and blocks higher frequency components. A high-pass filter passes frequency components higher than a predetermined second cutoff frequency and blocks lower frequency components. The second cutoff frequency is set to be higher than the first cutoff frequency. An all-pass filter delays the phase of the input and does not attenuate or amplify the amplitude.

[0105] Due to its impedance characteristics, the rotating machine 6 has a resonance characteristic at an electrical angular frequency (1f) on the rotational coordinate system or a frequency close to the electrical angular frequency. Therefore, the bandwidth of each filter is set so that the gain at frequency 1f is significantly smaller than the gain at other frequencies. However, the bandwidth of each filter may also be set so that the gain at frequencies other than frequency 1f is smaller than the gain at other frequencies. Frequencies other than frequency 1f may include, for example, frequencies that are integer multiples of frequency 1f, frequency bands other than the passband, and other frequencies of unwanted components that are desired to be removed. Furthermore, the passband limiter 34B may be set so as to function as little as possible when the pulsation of the voltage phase input from the voltage phase calculator 31 is relatively small.

[0106] As described above, the rotating machine control device 1 according to this embodiment includes the passband limiter 34B. The passband limiter 34B corrects the phase of the first voltage command (for example, voltage command 1) by removing a specific frequency component. The specific frequency is set to an integer multiple of the rotational frequency of the rotating machine 6. This configuration makes it possible to suppress current and torque pulsations, such as harmonic components, that cannot be suppressed in the first and second embodiments. Consequently, it is possible to improve the responsiveness of the current and torque.

[0107] <Fourth Embodiment> Next, a fourth embodiment of the present disclosure will be described. The following description will mainly focus on differences from the first to third embodiments. This embodiment differs from the first to third embodiments in that, for example, the use of the rotating machine 6 is specialized as a power source for the wheels 201 of the electric vehicle 200. For points in common with the first to third embodiments, the description thereof will be used unless otherwise specified. The rotating machine control device 1 according to this embodiment may have the same configuration as the rotating machine control device 1 according to any of the first to third embodiments.

[0108] 20 is a diagram showing an example of the configuration of an electric vehicle 200 according to this embodiment. The electric vehicle 200 includes a rotating machine control device 1, a DC power supply 3, a rotating machine 6, wheels 201, a differential gear 202, a transmission 203, and a vehicle controller 204.

[0109] The two wheels 201 are connected to both ends of a differential gear 202, and each rotates to drive the electric vehicle 200. The differential gear 202 transmits the driving force of the rotating machine 6 to the wheels 201. An input shaft of the transmission 203 is connected to the rotating shaft of the rotating machine 6, and an output shaft is connected to the two wheels 201 via the differential gear 202. Note that the transmission 203 may be omitted from the electric vehicle 200. In that case, the rotating shaft of the rotating machine 6 may be directly connected to the differential gear 202, or the rotating shaft of the rotating machine 6 may be directly connected to the wheels 201.

[0110] Generally, from the standpoint of quietness and driving feel, the electric vehicle 200 is expected to have small vibrations and torque pulsations caused by the rotating machine 6. The concept of driving feel includes so-called steering feel, ride comfort, and the like. Furthermore, high acceleration performance is desirable when driving on highways or uphill. The electric vehicle 200 according to this embodiment includes the rotating machine control device 1. Therefore, current and torque pulsations and vibrations of the rotating machine 6 can be suppressed, thereby improving driving performance such as quietness and driving feel. Furthermore, suppressing current pulsations and improving control stability can improve current and torque responsiveness. Therefore, the acceleration performance of the electric vehicle 200 can be improved. Furthermore, when the rotating machine control device 1 performs square wave control, the number of switching operations of the power conversion unit 4 per one electrical angle cycle of the rotating machine 6 can be minimized. This reduces switching loss in the power conversion unit 4 and reduces power consumption.

[0111] Although the embodiments of the present disclosure have been described above, the present disclosure is not limited to these embodiments and their variations. Addition, omission, substitution, and other modifications of the configuration are possible without departing from the spirit of the present disclosure. The direction of arrows shown in block diagrams and other drawings is for the convenience of explanation and does not limit the direction of the flow of information, data, signals, etc. during implementation. Furthermore, the present disclosure is not limited by the above description, but is limited only by the appended claims.

[0112] For example, in the above description, the power conversion control unit 2 includes a general-purpose computer system having the processor 100 and the storage device 101, but this is not limiting. Part or all of the power conversion control unit 2 may be configured to include dedicated hardware, for example, various integrated circuits and other circuit elements. The power conversion control unit 2 according to the third embodiment may include a voltage command generation unit 11A (FIG. 14) instead of the voltage command generation unit 11 (FIG. 6). In the first to third embodiments, the rotating machine 6 may be used for purposes other than driving the electric vehicle 200, for example, for steering assistance.

[0113] The rotating machine control device, rotating machine drive system, and electric vehicle according to the present disclosure suppress pulsation in the current flowing through the rotating machine and pulsation in the torque generated in the rotating machine. Suppressing the pulsation improves the stability of control of the rotating machine, and ultimately improves the responsiveness of the current and torque.

[0114] 1...Rotating machine control device, 2...Power conversion control unit, 3...DC power supply, 4...Power conversion unit, 5...Current detection unit, 6...Rotating machine, 7...DC voltage detection unit, 8...Rotor position detection unit, 11...Voltage command generation unit, 12, 12A, 12B...Voltage command correction unit, 13...Carrier signal generation unit, 14...Pulse width modulation signal generation unit, 21...Current command generator, 22...Coordinate converter, 23...Current controller, 24...Correction coefficient calculator, 25...Voltage command regulator, 31...Voltage phase calculator, 32...Voltage amplitude calculator, 33...Control switching determiner, 34, 34B...Passband limiter, 35...Voltage amplitude selector, 36...Cartesian coordinate converter, 37...Voltage saturation amount calculator, 38...Coordinate converter, 100...Processor, 101...Storage device, 200...Electric vehicle, 201...Wheel, 202...Differential gear, 203...Transmission, 204...Vehicle controller

Claims

1. A rotating machine control device comprising: a power conversion unit that converts a DC voltage of a DC power supply into an AC voltage and applies the AC voltage to a rotating machine; a current detection unit that detects a current supplied from the power conversion unit to the rotating machine; and a power conversion control unit that gives a switching command to the power conversion unit, wherein the power conversion control unit has: a voltage command generation unit that generates a first voltage command to be applied to the rotating machine based on the current; and a voltage command correction unit that corrects the first voltage command to a second voltage command, the voltage command correction unit calculates the phase of the second voltage command based on a d-axis voltage on a d-axis and a q-axis voltage on a q-axis that are synchronized with rotation of the rotating machine, so that the phase of the second voltage command is equal to the phase of the first voltage command, and determines a voltage limit value corresponding to the DC voltage of the DC power supply as the amplitude of the second voltage command.

2. The rotating machine control device according to claim 1, wherein the voltage command correction unit sets the voltage limit value by multiplying the DC voltage by a value equal to or greater than √6 / π.

3. A rotating machine control device according to claim 1 or 2, wherein the power conversion control unit changes the switching command once per period of the electrical angle.

4. A rotating machine control device according to claim 2 or 3, wherein the voltage command correction unit corrects the first voltage command based on a control method indicated by a switching signal.

5. A rotating machine control device according to any one of claims 1 to 4, wherein the power conversion control unit generates the switching command based on the second voltage command and a carrier signal, and the frequency of the carrier signal is an integer multiple of the rotational speed of the rotating machine.

6. A rotating machine control device according to any one of claims 1 to 5, wherein the voltage command correction unit has a passband limiter, the passband limiter corrects the phase of the first voltage command by removing a specific frequency component, and the specific frequency is an integer multiple of the rotational frequency of the rotating machine.

7. A rotating machine control device according to any one of claims 1 to 6, wherein the voltage command generation unit has a voltage command regulator, and the voltage command regulator corrects the first voltage command so as to reduce a voltage error occurring in the process of calculating the second voltage command.

8. A rotating machine control device according to claim 7, wherein the voltage command regulator determines a correction coefficient using a predetermined data map or function based on one or more of the torque of the rotating machine, the rotational speed of the rotating machine, the past first voltage command and second voltage command, and the current supplied to the rotating machine, and corrects the current first voltage command based on the correction coefficient.

9. A rotating machine drive system comprising: a rotating machine control device according to any one of claims 1 to 8; and the rotating machine.

10. An electric vehicle equipped with the rotary machine drive system according to claim 9.

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