Radar device and target detection method

Abstract

A transmission signal generation unit (11) of a radar device (1) generates a plurality of FMCW-type modulation signals that have center frequencies shifted from each other by a frequency shift amount based on a pulse period or a plurality of modulation signals that have been shifted in a time domain according to mutually different time shift amounts based on the pulse period, and generates a plurality of transmission signals by pulsing each modulation signal on the basis of the pulse period. A beat signal generation unit (16) generates a plurality of beat signals for each of reception signals that have been generated from received radio waves, by passing a mixing signal that has been obtained by mixing each modulation signal and each reception signal through a BPF (43-1, ..., 43-N). A range Doppler data generation unit (17) converts each beat signal into a frequency domain and generates a plurality of items of range Doppler data. A target detection unit (18) detects a target on the basis of a beam that has been formed from the plurality of items of range Doppler data.

Description

Radar device and target detection method

[0001] This disclosure relates to a radar system and a target detection method.

[0002] One type of radar system for detecting targets is the MIMO (Multiple-Input Multiple-Output) system, which has multiple transmitting antennas and multiple receiving antennas. An example of this type of radar system is disclosed in Patent Document 1. The MIMO system disclosed in Patent Document 1 mixes the transmitted signal with the received signal obtained by receiving the signal transmitted from the transmitting antenna and reflected by the target with the receiving antenna. The radar system generates range Doppler data, which is distance and velocity axis data, by performing a Fourier transform on the mixed result, and detects the target based on the beam generated from the range Doppler data.

[0003] Special Publication No. 2023-527226

[0004] In the radar system disclosed in Patent Document 1, the FMICW (Frequency Modulated Interrupted Continuous Wave) method may be used as the modulation method for the transmitted signal to protect the receiver when sufficient isolation between the transmitting and receiving antennas cannot be ensured. In this case, the range Doppler data includes not only the autocorrelation component necessary for target detection, but also a cross-correlation component with a frequency close to that of the autocorrelation component. As a result, the radar system may mistakenly detect a target based on the cross-correlation component.

[0005] This disclosure is made in view of the circumstances described above and aims to provide a radar device and a target detection method with high target detection accuracy.

[0006] To achieve the above objective, the radar system of this disclosure is a MIMO (Multiple-Input Multiple-Output) radar system for detecting targets, comprising a transmit signal generation unit, a plurality of transmit antennas, a plurality of receive antennas, a beat signal generation unit, a range Doppler data generation unit, and a target detection unit. The transmit signal generation unit generates a plurality of modulation signals of the FMCW (Frequency Modulation Continuous Wave) type, in which the frequency changes periodically according to the chirp rate, which indicates the amount of frequency change for each chirp period, and whose center frequencies are shifted from each other by a frequency shift amount based on the pulse period, or a plurality of modulation signals that are shifted in the time domain according to different time shift amounts based on the pulse period, and generates a plurality of transmit signals by pulsing each of the modulation signals based on the pulse period. The plurality of transmit antennas radiate transmit radio waves based on different transmit signals. The plurality of receive antennas receive transmit radio waves reflected by the target and generate a received signal based on the received radio waves. The beat signal generation unit generates multiple beat signals for each received signal by mixing the received signal and the modulated signal, and passing the resulting mixed signal through a band-pass filter. The range Doppler data generation unit generates multiple range Doppler data indicating the distance difference and speed difference to the target by converting each beat signal to the frequency domain. The target detection unit forms a beam from the multiple range Doppler data and detects the target from the formed beam.

[0007] The radar system of this disclosure generates multiple modulated signals whose center frequencies are shifted from each other by a frequency shift amount based on the pulse period, or multiple modulated signals that are shifted in the time domain according to different time shift amounts based on the pulse period, and generates multiple transmission signals by pulsing the modulated signals. For each received signal, the radar system generates a beat signal by mixing the received signal with each modulated signal and passing the resulting mixed signal through a band-pass filter. By generating multiple modulated signals based on a frequency shift amount or time shift amount based on the pulse period, the frequency difference between the autocorrelation component and the cross-correlation component in the mixed signal widens, making it possible to reduce the cross-correlation component in the band-pass filter, and thus a radar system with high target detection accuracy can be obtained.

[0008] A block diagram showing the configuration of the radar device according to Embodiment 1. A diagram showing the hardware configuration of the radar device according to Embodiment 1. A diagram showing an example of radio waves emitted or received by the radar device according to Embodiment 1. A flowchart showing an example of the operation of the target detection process performed by the radar device according to Embodiment 1. A diagram showing an example of each signal generated by the radar device according to Embodiment 1. A diagram showing an example of a transmitted signal generated by the radar device according to Embodiment 1. A diagram showing an example of a received signal generated by the radar device according to Embodiment 1. A diagram showing an example of a transmitted signal and a received signal generated by the radar device according to Embodiment 1. A diagram showing an example of frequency components included in the mixing signal generated by the radar device according to Embodiment 1. A diagram showing an example of each signal generated by the radar device according to Embodiment 2. A diagram showing an example of a transmitted signal generated by the radar device according to Embodiment 2. A diagram showing an example of a transmitted signal and a received signal generated by the radar device according to Embodiment 2. A diagram showing a modified example of the hardware configuration of the radar device according to the embodiment.

[0009] The radar device and target detection method according to the embodiments of this disclosure will be described in detail below with reference to the drawings. In the drawings, the same or equivalent parts are denoted by the same reference numerals.

[0010] (Embodiment 1) A radar device 1 according to Embodiment 1 will be described using a MIMO (Multiple-Input Multiple-Output) radar device equipped with multiple transmitting antennas and receiving antennas as an example. The radar device 1 shown in Figure 1 includes a transmitter 10 that emits multiple transmitted radio waves based on the FMICW (Frequency Modulation Interrupted Continuous Wave) method, and a receiver 14 that receives transmitted radio waves reflected by a target.

[0011] The transmitter 10 includes a transmission signal generation unit 11 that generates multiple transmission signals based on the FMICW method, multiple transmission antennas 12_1, ..., 12_N that radiate transmission radio waves based on different transmission signals, and a reference signal generation circuit 13 that generates a reference signal for driving the transmission signal generation unit 11.

[0012] The transmission signal generation unit 11 includes N DDS (Direct Digital Synthesizers) 21_1, ..., 21_N that match the MIMO multiplexing number N, N switches 22_1, ..., 22_N, and N amplifiers 23_1, ..., 23_N. In Embodiment 1, the multiplexing number N is set to match the number of transmission antennas 12_1, ..., 12_N.

[0013] The receiver 14 includes a plurality of receiving antennas 15_1, ..., 15_M that receive transmitted radio waves reflected by a target and generate a received signal; a beat signal generation unit 16 that generates a plurality of beat signals based on the mixing of the received signal and the modulated signal; a range Doppler data generation unit 17 that generates a plurality of range Doppler data by converting the beat signals to the frequency domain; a target detection unit 18 that forms a beam from the range Doppler data and detects a target from the formed beam; and N ADCs (Analog-to-Digital Converters) 19_1, ..., 19_N that match the multiplexing number N.

[0014] The beat signal generation unit 16 has the same number of receiving signal processing circuits 31_1, ..., 31_M as the number of receiving antennas 15_1, ..., 15_M. Each receiving signal processing circuit 31_1, ..., 31_M has an ADC 41, N mixers 42_1, ..., 42_N corresponding to the multiplexing number N, and N BPFs (Band Pass Filters) 43_1, ..., 43_N.

[0015] The range Doppler data generation unit 17 has the same number of FFT (Fast Fourier Transform) circuits 32_1, ..., 32_M as the number of receiving antennas 15_1, ..., 15_M. Each of the FFT circuits 32_1, ..., 32_M has N range FFTs 44_1, ..., 44_N and N Doppler FFTs 45_1, ..., 45_N.

[0016] Figure 2 shows the hardware configuration of the radar device 1 having the above configuration. As shown in Figure 2, the radar device 1 is implemented by a processing circuit 81. The processing circuit 81 of the radar device 1 shown in Figure 2 is connected via an interface circuit 82 to external devices, such as devices that utilize the position and speed of a target detected by the target detection unit 18, GPS (Global Positioning Satellite) / GNSS (Global Navigation Satellite System) signal receivers, etc.

[0017] If the processing circuit 81 is dedicated hardware, it may include, for example, a single circuit, a composite circuit, a processor, an ASIC (Application Specific Integrated Circuit), an FPGA (Field Programmable Gate Array), or a combination thereof. Each part of the radar device 1 may be implemented by an individual processing circuit 81, or each part of the radar device 1 may be implemented by a common processing circuit 81. The interface circuit 82 has one or more interface modules conforming to standards, depending on the connection destination.

[0018] The radar device 1 having the above configuration detects targets using N transmitting antennas 12_1, ..., 12_N and M receiving antennas 15_1, ..., 15_M. As an example, as shown in Figure 3, the radar device 1 includes two transmitting antennas 12_1, 12_2 and M receiving antennas 15_1, ..., 15_M. As shown by the solid arrows in Figure 3, when the transmitted radio wave W1 radiated by the transmitting antenna 12_1 reaches the target 100, it is reflected by the target 100. The transmitted radio wave W1 reflected by the target 100 becomes the reflected radio wave W1'. As shown by the dashed arrows in Figure 3, the reflected radio wave W1' is received by each of the receiving antennas 15_1, ..., 15_M.

[0019] Similarly, as shown by the dotted arrows in Figure 3, when the transmitted radio wave W2 radiated by the transmitting antenna 12_2 reaches the target 100, it is reflected by the target 100. The transmitted radio wave W2 reflected by the target 100 becomes the reflected radio wave W2'. As shown by the dashed arrows in Figure 3, the reflected radio wave W2' is received by the receiving antennas 15_1, ..., 15_M, respectively.

[0020] The radar device 1 detects the target 100 based on the modulated signal and the received signals based on the received reflected radio waves W1' and W2'. Details of the radar device 1 are described below. The radar device 1 having the above configuration performs the target detection process shown in Figure 4 at predetermined timings, for example, at regular intervals, while in operation. The transmission signal generation unit 11 of the transmitter 10 of the radar device 1 generates a plurality of modulated signals that are FMCW (Frequency Modulation Continuous Wave) modulated signals in which the frequency changes periodically, and whose center frequencies are shifted from each other by a frequency shift amount based on a predetermined pulse period (step S11). The transmission signal generation unit 11 generates a plurality of transmission signals by pulsing each of the modulated signals based on the pulse period (step S12).

[0021] In detail, the reference signal generation circuit 13 shown in Figure 1 acquires GPS / GNSS signals from a receiver that receives GPS / GNSS signals from satellites. The reference signal generation circuit 13 controls an oscillator in synchronization with the time acquired from the GPS / GNSS signals to generate a reference signal, which is a periodic signal, such as a sine wave signal.

[0022] The transmission signal generation unit 11 generates multiple transmission signals, which are intermittent signals whose frequency changes periodically and are driven by a reference signal. Specifically, the DDS 21_1, ..., 21_N of the transmission signal generation unit 11 are each FMCW modulated signals whose frequency changes periodically and are driven by a reference signal output by the reference signal generation circuit 13, and generate multiple modulated signals whose center frequencies are shifted from each other by a frequency shift amount based on a predetermined pulse period. The pulse period indicates the pulse repetition period of the FMICW transmission signal generated by the transmission signal generation unit 11. It is assumed that each of the DDS 21_1, ..., DDS 21_N has information about the pulse period stored in advance. Since the configurations of DDS 21_1, ..., 21_N are similar to each other, DDS 21_1 will be described.

[0023] The DDS21_1 generates an FMCW analog modulated signal in which the frequency changes periodically according to the chirp rate (unit: Hz / s), which indicates the amount of frequency change for each chirp period that is longer than the pulse period. As an example, the center frequency of the modulated signal generated by the DDS21_1 is fc １ The DDS21_1 has a center frequency fc depending on the chirp rate k. １ A modulated signal is generated by changing the frequency based on a reference value.

[0024] As a result, a modulated signal is generated in which the frequency changes periodically according to the chirp rate, as shown in Graph A of Figure 5. In Graph A of Figure 5, the horizontal axis is time and the vertical axis is frequency. In the example of Graph A of Figure 5, the frequency of the modulated signal is the chirp period T. ｃOver time, it increases linearly from the minimum frequency to the maximum frequency. When the frequency of the modulation signal reaches the maximum frequency, it is reset and starts increasing again from the minimum frequency. The frequency of the modulation signal repeats the above-described variation every chirp period T ｃ . The average value of the frequency of the modulation signal in the chirp period T ｃ is the center frequency fc １ .

[0025] As an example, the frequency of the modulation signal starts increasing from the minimum frequency at time T0, and when it reaches the maximum frequency at time T1, it is reset and starts increasing again from the minimum frequency. After that, when the frequency of the modulation signal reaches the maximum frequency at time T2 after the chirp period T ｃ has elapsed from time T1, it is reset and starts increasing again from the minimum frequency. The frequency of the modulation signal changes periodically every chirp period T ｃ .[[ID=I4]]

[0026] Let the center frequencies of the modulation signals generated by DDS21_1, 21_2,..., DDS21_N be fc １ , fc ２ ,..., fc Ｎ . The center frequencies fc １ , fc ２ ,..., fc Ｎ are different from each other.

[0027] The switches 22_1,..., 22_N perform a switching operation every pulse period T ｐｒｉ to pulse the modulation signals output by DDS21_1,..., 21_N. In other words, the outputs of the switches 22_1,..., 22_N are signals in the FMICW format.

[0028] Graph B in FIG. 5 shows the rectangular pulse signal with the pulse period T ｐｒｉ . The switches 22_1,..., 22_N perform a switching operation of repeating on and off like the rectangular pulse signal shown in graph B to generate a transmission signal that is an intermittent signal whose frequency changes periodically as shown in graph C in FIG. 5. The horizontal axis of graph B in FIG. 5 is time, and the vertical axis is amplitude. The rectangular pulse signal has a pulse period Tｐｒｉ This is a continuous pulse signal that rises repeatedly with each pulse. In Graph C of Figure 5, the horizontal axis is time and the vertical axis is frequency. In Graph C of Figure 5, the frequency of the modulated signal is shown by a dashed line. The transmitted signal corresponds to the signal obtained by multiplying the modulated signal by the rectangular pulse signal. The transmitted signal is the pulse width T when the rectangular pulse signal is on. ｗ This is a signal that is transmitted only during that period.

[0029] Amplifiers 23_1, ..., 23_N amplify the power of the transmission signal output by switches 22_1, ..., 22_N, and send the amplified transmission signal to the transmitting antennas 12_1, ..., 12_N. In other words, the output of amplifiers 23_1, ..., 23_N is a high-power FMICW analog modulated signal.

[0030] The transmitting antennas 12_1, ..., 12_N are provided in a one-to-one correspondence with the amplifiers 23_1, ..., 23_N. Preferably, the transmitting antennas 12_1, ..., 12_N are provided at positions far enough apart from each other to sufficiently reduce interference of the transmitted radio waves immediately after radiation. As shown in Figure 4, the transmitting antennas 12_1, ..., 12_N generate transmitted radio waves based on different transmitted signals and radiate the transmitted radio waves toward the target 100 (step S13). In detail, the transmitting antennas 12_1, ..., 12_N generate transmitted radio waves from the analog transmitted signals output from the corresponding amplifiers 23_1, ..., 23_N and radiate the transmitted radio waves toward the target 100.

[0031] In order to suppress false detection of target 100 by radar device 1, as described above, the center frequency of the modulated signal generated by DDS 21_1, ..., 21_N is the pulse period T ｐｒｉ The frequency shifts are based on the following frequency shift amount. As an example, let's assume that the center frequencies of the modulated signals generated by DDS21_1, ..., 21_N increase in order. In this case, the signal components of the modulated signals generated by DDS21_1, ..., 21_N are arranged in order in the frequency domain. For example, the center frequencies of the two modulated signals generated by DDS21_1 and 21_2 are adjacent, and the center frequencies of the two modulated signals generated by DDS21_2 and 21_3 are adjacent.

[0032] Figure 6 shows an example of the frequency of the transmission signal output by switches 22_1 and 22_2. In Figure 6, the horizontal axis represents time, and the vertical axis represents frequency. The center frequencies of the modulated signals generated by DDS 21_1 and 21_2 are fc, respectively. １ ,fc ２ In Figure 6, the frequencies of the modulated signals generated by DDS 21_1 and 21_2 are shown by the dashed lines. In Figure 6, the frequencies of the transmitted signals output by switches 22_1 and 22_2 are shown by the solid lines. The frequency of the modulated signal generated by DDS 21_1 changes over time, with a center frequency fc １ The chirp period T is defined as increasing linearly from the minimum frequency to the maximum frequency, centered around a certain point. ｃ Repeat each time. Similarly, the frequency of the modulated signal generated by DDS21_2 changes over time, with a center frequency fc ２ The chirp period T is defined as increasing linearly from the minimum frequency to the maximum frequency, centered around a certain point. ｃ Repeat each time.

[0033] The shift in the center frequency of the modulated signal generated by DDS21_1, ..., 21_N is determined by the difference between adjacent center frequencies within a range that satisfies the conditions corresponding to the pulse period. １ <fc ２ <...<fc Ｎ Assume that the following holds true.

[0034] For example, the center frequencies of the modulation signals generated by DDS21_i and 21_i+1 are fc ｉ ,fc ｉ＋１ Therefore, the difference in center frequencies fc ｉ＋１ -fc ｉ The following equation (1) is satisfied, where i is any natural number between 1 and N-1. In equation (1) below, the first coefficient A ｉT is a non-negative integer determined according to the bandwidth of the transmitting antennas 12_1, ..., 12_N, the bandwidth of the receiving antennas 15_1, ..., 15_M, and the MIMO multiplexing number N. In the example of radar device 1 shown in Figure 1, the multiplexing number N in equation (1) below is equal to the number of transmitting antennas 12_1, ..., 12_N. In equation (1) below, the second coefficient a is a natural number less than N that is relatively prime to N. In equation (1) below, T ｐｒｉ This is the pulse period.

[0035]

[0036] DDS21_1, ..., 21_N generate multiple modulated signals whose center frequencies satisfy the conditions of equation (1) above. The first coefficient A in equation (1) above ｉ The center frequency is fc １ , ..., fc Ｎ Each of these is set within a range that satisfies the condition that it is included in the bandwidth of the transmitting antennas 12_1, ..., 12_N.

[0037] By having DDS21_1, ..., 21_N each perform the above-described processing, the transmission signal generation unit 11 becomes capable of generating multiple transmission signals with mutually different center frequencies.

[0038] The transmitted radio waves emitted from the transmitter 10 having the above configuration and reflected by the target 100 are received by the receiver 14. The receiving antennas 15_1, ..., 15_M of the receiver 14 have a center frequency fc １ , ..., fc Ｎ It is preferable that they be positioned at a distance from each other accordingly. As shown in Figure 4, each receiving antenna 15_1, ..., 15_M receives transmitted radio waves radiated from transmitting antennas 12_1, ..., 12_N and reflected by target 100, and generates a received signal based on the received radio waves (step S14).

[0039] ADCs 19_1, ..., 19_N are provided in one-to-one correspondence with DDSs 21_1, ..., 21_N. ADCs 19_1, ..., 19_N perform A / D (Analog-to-Digital) conversion of the modulated signal output by DDSs 21_1, ..., 21_N and output the A / D converted modulated signal. Specifically, ADC 19_1 performs A / D conversion of the modulated signal output from DDS 21_1 and outputs the A / D converted modulated signal to the mixer 42_1 provided in each of the receiving signal processing circuits 31_1, ..., 31_M. Similarly, ADC 19_N performs A / D conversion of the modulated signal output from DDS 21_N and outputs the A / D converted modulated signal to the mixer 42_N provided in each of the receiving signal processing circuits 31_1, ..., 31_M.

[0040] The beat signal generation unit 16 generates multiple beat signals for each received signal based on the mixing of the received signal with each of the modulated signals generated by DDS 21_1, ..., 21_N (step S15).

[0041] The received signal processing circuits 31_1, ..., 31_M of the beat signal generation unit 16 are provided in one-to-one correspondence with the receiving antennas 15_1, ..., 15_M. Since the configurations of the received signal processing circuits 31_1, ..., 31_M are similar to each other, the received signal processing circuit 31_1 will be described. The ADC 41 of the received signal processing circuit 31_1 converts the received signal to digital data and generates a received signal.

[0042] As described above, each of the receiving antennas 15_1, ..., 15_M receives transmitted radio waves radiated from the transmitting antennas 12_1, ..., 12_N and reflected by the target 100. Therefore, the received signal contains components originating from multiple transmitted signals. As shown in Figure 3, when the radar device 1 is equipped with two transmitting antennas 12_1, 12_2, as shown in Figure 7, the received signal generated by each antenna 15_1, ..., 15_M contains components originating from the transmitted signals generated by the DDS 21_1, 21_2, specifically, the delayed transmitted signals. The received signal has a center frequency of fc １ The signal component is such that the center frequency is fc２ It includes the signal component. In Figure 7, the horizontal axis represents time, and the vertical axis represents frequency.

[0043] The center frequency is fc １ The signal component and center frequency are fc ２ The frequency of the signal component begins to increase from its minimum frequency at time T0', which is later than time T0, and then begins to increase again from its minimum frequency at time T1'. After that, the center frequency is fc １ The signal component and center frequency are fc ２ The frequency of the signal component is determined from time T1' to chirp period T ｃ At time T2', the frequency starts increasing again from the minimum frequency. The center frequency of the received signal is fc １ The signal component and center frequency are fc ２ The frequency of the signal component is the chirp period T, as described above. ｃ It changes periodically.

[0044] The mixers 42_1, ..., 42_N shown in Figure 1 are provided in one-to-one correspondence with the DDS 21_1, ..., 21_N. Mixer 42_1 receives the modulated signal output from DDS 21_1 and A / D converted by ADC 19_1, and receives the received signal, which is digital data, from ADC 41. Mixer 42_1 mixes the received signal and the modulated signal and outputs the resulting mixed signal to BPF 43_1. The same applies to mixers 42_2, ..., 42_N.

[0045] BPFs 43_1, ..., 43_N are provided in one-to-one correspondence with mixers 42_1, ..., 42_N. The output of mixer 42_1 is input to BPF 43_1. The beat signal, which is the output of BPF 43_1, is a signal in which only the frequency components included in a defined passband are extracted from the output of mixer 42_1. The same applies to BPFs 43_2, ..., 43_N.

[0046] The mixed signal obtained by mixer 42_1 includes an autocorrelation component showing the correlation between the modulated signal obtained from DDS 21_1 and the component of the transmitted signal based on said modulated signal included in the received signal, and a crosscorrelation component showing the correlation between the modulated signal obtained from DDS 21_1 and the component of the transmitted signal based on modulated signals other than said modulated signal included in the received signal. The same applies to mixers 42_2, ..., 42_N.

[0047] As shown in Figure 3, when the radar device 1 is equipped with two transmitting antennas 12_1 and 12_2, the received signal supplied to the mixer 42_2 from the ADC 41 includes components resulting from the transmitted signals generated by the DDS 21_1 and 21_2, specifically, delayed components of each transmitted signal. The received signal has a center frequency of fc １ The signal component is such that the center frequency is fc ２ The signal includes the following signal components. The frequencies of the components resulting from the transmitted signal generated by DDS 21_1 and 21_2 change periodically. Figure 8 shows one period of the periodically changing received and transmitted signals. In Figure 8, the horizontal axis represents time, and the vertical axis represents frequency. In Figure 8, the frequency of the modulated signal generated by DDS 21_2 is shown by a dashed line, and the frequency of the transmitted signal output by switch 22_2 is shown by a solid line. In Figure 8, the frequencies of the signal components included in the received signal are shown by dotted lines.

[0048] Mixer 42_2 mixes the modulated signal output from DDS 21_2 and A / D converted by ADC 19_2 with the received signal acquired from ADC 41. As shown in Figure 9, the output signal of mixer 42_2 includes an autocorrelation component shown in white and a cross-correlation component shown in black and diagonal lines in the frequency domain. In Figure 9, the horizontal axis represents frequency and the vertical axis represents amplitude. Figure 9 shows the output signal of mixer 42_2 when the multiplexing number N = 2, that is, when the radar device 1 is equipped with transmitting antennas 12_1 and 12_2.

[0049] The received signal generated from the transmitted radio waves reflected by target 100 lags behind the transmitted signal, so an autocorrelation component appears at a position shifted in the negative direction from the origin. A number of cross-correlation components appear depending on the multiplexing number N; specifically, for each mixing signal, N-1 cross-correlation components appear. In the example of the mixing signal shown in Figure 9, since the multiplexing number N=2, one cross-correlation component appears, i.e., a cross-correlation component with the component caused by the modulated signal generated by DDS21_1. In Figure 9, the blacked-out and hatched areas represent the spectra of the cross-correlation components. Of the spectra of the cross-correlation components, the blacked-out spectrum represents the difference in center frequency fc from the autocorrelation component. ２ -fc １ It appears at a position shifted by only that much. Also, within the spectrum of the cross-correlation components, the spectrum shown in diagonal lines is derived from the cross-correlation component shown in black, with a pulse period T. ｐｒｉ The reciprocal of is 1 / T ｐｒｉ This is a replica component that appears periodically at intervals of . As mentioned above, since the transmitted signal is an intermittent signal, the spectrum of the cross-correlation component is shown in black and diagonal lines. When the FMCW method is used as the modulation method for the transmitted signal, only the spectrum of the cross-correlation component shown in black appears.

[0050] As an example, when the number of multiplexers N=3, the beat frequency F of the autocorrelation component included in the output of mixers 42_1, 42_2, and 42_3 ａｃ This is expressed by the following equation (2). In the following equation (2), c is the speed of light (unit: m / s), k is the chirp rate (unit: Hz / s), and R is the distance between the radar device 1 and the target 100 (unit: m). As shown in the following equation (2), the beat frequency F ａｃ This changes depending on the distance to the target of 100. In other words, the position where the autocorrelation component appears changes depending on the distance to the target of 100.

[0051]

[0052] When the number of multiplexers N=3, the center frequency of the modulated signal generated by DDS21_1, 21_2, and 21_3 is fc １ ,fc ２ ,fc ３The beat frequency F of the cross-correlation component included in the output of mixer 42_1 is defined as follows. ｃ２ｃ１ , F ｃ３ｃ１ This is expressed by equations (3) and (4) below. In equations (3) and (4) below, the coefficient b is a non-negative integer represented by b = 0, 1, 2, ...

[0053] At the output of mixer 42_1, the cross-correlation component in the received signal between the component caused by the modulated signal generated by DDS21_2 and the modulated signal generated by DDS21_1 is the beat frequency F when b=0. ｃ２ｃ１ And the beat frequency F when b = 0 ｃ２ｃ１ Centered around 1 / T ｐｒｉ They appear at positions shifted by one increment. At the output of mixer 42_1, the cross-correlation component between the component caused by the modulated signal generated by DDS21_3 and the modulated signal generated by DDS21_1, which is included in the received signal, is the beat frequency F when b=0. ｃ３ｃ１ And the beat frequency F when b = 0 ｃ３ｃ１ Centered around 1 / T ｐｒｉ They appear in positions that are slightly offset from each other.

[0054]

[0055]

[0056] Similarly, the beat frequency F of the cross-correlation component included in the output of mixer 42_2 ｃ１ｃ２ , F ｃ３ｃ２ This is expressed by equations (5) and (6) below. The beat frequency F when b = 0 in equation (5) below ｃ１ｃ２ This is the cross-correlation component shown in black in Figure 9. In equation (5) below, the beat frequency F when b = 1, 2, ... ｃ１ｃ２ This represents the cross-correlation component shown by the shaded area in Figure 9.

[0057]

[0058]

[0059] Similarly, the beat frequency F of the cross-correlation component included in the output of mixer 42_3 ｃ１ｃ３ , F ｃ２ｃ３This is expressed by equations (7) and (8) below.

[0060]

[0061]

[0062] For each mixer 42_1, ..., 42_N, the absolute value of the beat frequency difference between the replica component (indicated by a shaded area) closest to the autocorrelation component and the autocorrelation component is calculated, and the minimum value of the calculated value is taken as Δf. When the distance R between the radar device 1 and the target 100 is less than or equal to the reference distance D expressed in equation (9) below, the autocorrelation component appears in the range where the beat frequency is -Δf or greater and 0 or less, and none of the cross-correlation components appear in the range where the beat frequency is -Δf or greater and 0 or less. In equation (9) below, c represents the speed of light (unit: m / s), and k represents the chirp rate (unit: Hz / s). When the distance between the radar device 1 and the target 100 is greater than the reference distance D, the replica component, which is the cross-correlation component, exists in the range where the beat frequency is -Δf or greater and 0 or less, but when the reference distance D is sufficiently large, the cross-correlation component itself may be considered not to be observed due to the distance attenuation of radio waves. In other words, when the reference distance D is sufficiently large, the range where the beat frequency is greater than or equal to -Δf and less than or equal to 0 is a region where only autocorrelation components exist.

[0063]

[0064] By maximizing the beat frequency difference Δf, the reference distance D becomes sufficiently large, and in the range where the beat frequency is greater than or equal to -Δf and less than or equal to 0, there are no replica components, and only autocorrelation components exist, thereby suppressing the target detection unit 18 from mistakenly detecting the target 100 based on the replica components.

[0065] The replica component is located at a position shifted from the autocorrelation component by the difference in center frequency, with a pulse period T. ｐｒｉ The reciprocal of 1 / T ｐｒｉ It appears at a shifted position each time. As shown in Figures 8 and 9, the difference in the center frequency of the transmitted signal is fc ２ -fc １It is represented by. By setting the difference in the center frequencies, that is, the frequency shift amount, to the value obtained by the above equation (1), the beat frequency difference Δf becomes maximum.

[0066] Δf which is the maximum value of the beat frequency difference Δf ｍａｘ is represented by the following equation (10). When the difference in the center frequencies of the modulation signals is represented by the above equation (1), the beat frequency difference Δf is equal to Δf ｍａｘ is equal to.

[0067]

[0068] At this time, the passband widths of BPFs 43_1,..., 43_N may be set to Δf ｍａｘ or more. Specifically, the lower limit value of the passband of BPFs 43_1,..., 43_N may be set to -Δf ｍａｘ and the upper limit value may be set to 0. Thereby, the replica components are removed from the outputs of mixers 42_1,..., 42_N, and the beat signals output from BPFs 43_1,..., 43_N do not include replica components. Note that the lower limit value of the passband of BPFs 43_1,..., 43_N may be set to -Δf ｍａｘ or less, and the upper limit value may be set to 0 or more. At this time, for the replica components having a frequency less than -Δf ｍａｘ or greater than 0 and passing through BPFs 43_1,..., 43_N, a process of ignoring them may be performed in the range Doppler data generation unit 17.

[0069] As an example, when the radar device 1 includes two transmission antennas 12_1 and 12_2, in the above equation (1), substituting the first coefficient A ｉ = 0, 1, 2,..., the second coefficient a = 1, and N = 2, we get fc ２ -fc １ = 0.5 / T ｐｒｉ , 1.5 / T ｐｒｉ , 2.5 / T ｐｒｉ ,... and so on. At this time, the beat frequency difference Δf is maximized, and Δf ｍａｘ = 0.5 / T ｐｒｉ is obtained. Based on the above equation (10), the passband of BPF 43_2 is set to -0.5 / T ｐｒｉIf the value is above and below the range of 0, the replica component will not be included in the passband of BPF43_2. Therefore, the beat signal output by BPF43_2 will not contain the replica component, making it possible to suppress false detection of target 100.

[0070] It is preferable that the output of BPF43_1, ..., 43_N is subjected to decimation processing and the result is supplied to the range Doppler data generation unit 17. By reducing the number of samples through decimation processing, it is possible to speed up subsequent processing.

[0071] As shown in Figure 4, the range Doppler data generation unit 17 generates multiple range Doppler data indicating the distance difference and the speed difference with the target 100 by performing a conversion to the frequency domain, specifically a Fourier transform, for each beat signal (step S16). The FFT circuits 32_1, ..., 32_M provided in the range Doppler data generation unit 17 are associated one-to-one with the received signal processing circuits 31_1, ..., 31_M.

[0072] Since the configurations of the FFT circuits 32_1, ..., 32_M are the same, we will describe FFT circuit 32_1. The range FFTs 44_1, ..., 44_N of FFT circuit 32_1 are provided in one-to-one correspondence with BPFs 43_1, ..., 43_N. Since the configurations of the range FFTs 44_1, ..., 44_N are the same, we will describe range FFT 44_1. Range FFT 44_1 converts the beat signal output by BPF 43_1 into a range axis signal by dividing the received signal into chirp period units and performing FFT processing.

[0073] The Doppler FFTs 45_1, ..., 45_N of the FFT circuit 32_1 are provided in one-to-one correspondence with the range FFTs 44_1, ..., 44_N. Since the configurations of the Doppler FFTs 45_1, ..., 45_N are similar to each other, the Doppler FFT 45_1 will be described. The Doppler FFT 45_1 converts the output of the range FFT 44_1 into a Doppler axis signal by performing an FFT operation on the sweep direction of the received signal. As a result, multiple range Doppler data indicating the distance difference and the speed difference with respect to the target 100 are generated.

[0074] The frequency conversion processing for range FFTs 44_1, ..., 44_N and Doppler FFTs 45_1, ..., 45_N is an existing frequency conversion processing method, for example, the range dimension and Doppler dimension FFTs disclosed in Patent Document 1, specifically, FFT processing along the beat frequency axis direction and FFT processing along the sweep direction of the received signal.

[0075] The target detection unit 18 forms a beam from multiple range Doppler data and detects a target from the formed beam (step S17). The target detection process of the target detection unit 18 is an existing beamforming process, for example, the MIMO beamforming method disclosed in Patent Document 1. When the process in step S17 is completed, the radar device 1 terminates the target detection process shown in Figure 4.

[0076] As described above, the radar device 1 according to Embodiment 1 emits transmitted radio waves based on multiple transmitted signals generated from multiple modulated signals whose center frequencies are shifted from each other by a frequency shift amount based on the pulse period, and receives the transmitted radio waves reflected by the target 100 to generate multiple received signals. For each received signal, the radar device 1 generates multiple beat signals by mixing the received signal with each modulated signal. Since the frequency shift amount is determined according to the pulse period, the frequency difference between the autocorrelation component and the cross-correlation component included in the mixed signal widens, making it possible to reduce the cross-correlation component in the band-pass filter. As a result, erroneous detection of the target 100 based on the replica component is suppressed. In other words, a radar device 1 with high target detection accuracy can be obtained.

[0077] (Embodiment 2) The method for generating a modulation signal is not limited to the above example. A radar device 1 that generates a plurality of modulation signals shifted in the time domain according to different time shift amounts based on a pulse period will be described in Embodiment 2. The configuration of the radar device 1 according to Embodiment 2 is the same as the configuration of the radar device 1 according to Embodiment 1. The target detection process performed by the radar device 1 according to Embodiment 2 is the same as the target detection process shown in FIG. 4 performed by the radar device 1 according to Embodiment 1. The radar device 1 according to Embodiment 2 will be described focusing on the differences from Embodiment 1.

[0078] The DDS21_2 generates a signal with its frequency changed based on the chirp rate k while using the center frequency fc １ as a reference, and shifts the signal by a time shift amount ΔT ２ in the time domain to generate a modulation signal.

[0079] As described above, the frequency of the modulation signal generated by the DDS21_2 starts to rise with a delay of the time shift amount ΔT ２ compared to the modulation signal generated by the DDS21_2 in Embodiment 1, as shown in graph A of FIG. 10. The horizontal axis of graph A in FIG. 10 is time, and the vertical axis is frequency. In the example of graph A in FIG. 10, the frequency of the modulation signal linearly increases from the minimum frequency to the maximum frequency over the chirp period T ｃ . When the frequency of the modulation signal reaches the maximum frequency, it is reset and starts increasing again from the minimum frequency. The frequency of the modulation signal repeats the above-described variation every chirp period T ｃ . The average value of the frequency of the modulation signal in the chirp period T ｃ is the center frequency fc １ .

[0080] As an example, the frequency of the modulation signal starts to increase from the minimum frequency at time T10, is reset when it reaches the maximum frequency at time T11, and starts increasing again from the minimum frequency. After that, the frequency of the modulation signal is from time T11 to the chirp period T ｃWhen the maximum frequency is reached at time T12, it is reset and starts increasing again from the minimum frequency. The frequency of the modulated signal is determined by the chirp period T as described above. ｃ It changes periodically.

[0081] Switch 22_1, similar to Embodiment 1, has a pulse period T ｐｒｉ By performing a switching operation for each switch, the modulated signals output by DDS21_1, ..., 21_N are pulsed. In other words, the outputs of switches 22_1, ..., 22_N are FMICW signals.

[0082] Graph B in Figure 10 shows the pulse period T. ｐｒｉ This shows a rectangular pulse signal. Switches 22_1, ..., 22_N perform switching operations, repeatedly switching on and off as shown in the rectangular pulse signal in Graph B, thereby generating a transmission signal, which is an intermittent signal with a periodically changing frequency, as shown in Graph C of Figure 10. The rectangular pulse signal shown in Graph B of Figure 10 is the same as the rectangular pulse signal shown in Graph B of Figure 5. The rectangular pulse signal has a pulse period T ｐｒｉ The process of rising and falling is repeated. In Graph C of Figure 10, the horizontal axis is time and the vertical axis is frequency. In Graph C of Figure 10, the frequency of the modulated signal is shown by the dashed line. The transmitted signal corresponds to the signal obtained by multiplying the modulated signal by the rectangular pulse signal. The transmitted signal is the pulse width T when the rectangular pulse signal is on. ｗ This is a signal that is transmitted only during that period.

[0083] In order to suppress false detection of target 100 by radar device 1, the modulated signals generated by DDS 21_1, ..., 21_N have a pulse period T ｐｒｉ Based on this, they are shifted in the time domain by different time shift amounts. The time shift amount used when DDS21_1, ..., 21_N generates the modulation signal is the pulse period T. ｐｒｉDifferent time shift amounts are defined for each. DDS21_1, ..., 21_N perform time shifts based on the defined time shift amounts. As a result, DDS21_1, ..., 21_N generate multiple modulated signals with different time shift amounts corresponding to the pulse period. For example, the time shift amounts defined for DDS21_1, ..., 21_N are assumed to increase in order. In this case, the timing at which the frequencies of the modulated signals generated by DDS21_1, ..., 21_N begin to change is sequential in the time domain. For example, the timing at which the frequencies of the two modulated signals generated by DDS21_1 and 21_2 begin to change, specifically the timing at which they first begin to increase from the minimum frequency, are adjacent in the time domain. Similarly, the timing at which the frequencies of the two modulated signals generated by DDS21_2 and 21_3 begin to change is adjacent in the time domain.

[0084] Figure 11 shows an example of the transmission signals output by switches 22_1 and 22_2. The time shift amounts defined in DDS 21_1 and 21_2 are ΔT, respectively. １ ΔT ２ Let's assume the time shift amount ΔT. １ = 0, and the time shift amount ΔT ２ Assume > 0. In Figure 11, the frequencies of the modulated signals generated by DDS21_1 and 21_2 are shown by dashed lines. In Figure 11, the frequencies of the transmitted signals output by switches 22_1 and 22_2 are shown by solid lines. In Figure 11, the modulated signals generated by DDS21_1 and 21_2 are shown as modulated signal #1 and modulated signal #2, respectively. In Figure 11, the transmitted signals generated by DDS21_1 and 21_2 are shown as transmitted signal #1 and transmitted signal #2, respectively.

[0085] The frequency of the modulated signal generated by DDS21_1 changes over time, with a center frequency fc １ The chirp period T is defined as increasing linearly from the minimum frequency to the maximum frequency, centered around a certain point. ｃ Repeat each time. Similarly, the frequency of the modulated signal generated by DDS21_2 changes over time, with a center frequency fc １ The chirp period T is defined as increasing linearly from the minimum frequency to the maximum frequency, centered around a certain point.ｃ This is repeated each time. The time T20 at which the frequency of the modulated signal generated by DDS21_2 begins to change is shifted by a time amount ΔT from the time T0 at which the frequency of the modulated signal generated by DDS21_1 begins to change. ２ It's just slow.

[0086] The time shift amount ΔT used to generate the modulated signal in DDS21_1, ..., 21_N １ ,・・・,ΔT Ｎ Each of these is determined within a range that satisfies the conditions corresponding to the pulse period, where the difference between two time shift amounts corresponding to two adjacent modulated signals whose frequency begins to change in the time domain is within a range that satisfies the conditions corresponding to the pulse period. １ = 0, and ΔT １ <ΔT ２ <...<ΔT Ｎ Assume that the following holds true.

[0087] For example, the time shift amounts used to generate the modulated signals in DDS21_i and 21_i+1 are ΔT, respectively. ｉ ΔT ｉ＋１ Therefore, the time shift amount ΔT ｉ ΔT ｉ＋１ The difference ΔT ｉ＋１ -ΔT ｉ The following equation (11) is satisfied, where i is any natural number between 1 and N-1. The third coefficient B in equation (11) below. ｉ This is the chirp period T ｃ It is a non-negative integer determined according to the polynomial N. Specifically, ΔT Ｎ -ΔT １ <T ｃ Within the range that satisfies the conditions, ΔT １ ,・・・,ΔT Ｎ The third coefficient B for determining ｉ The second coefficient in equation (11) below is the same as the second coefficient in equation (1) above. In equation (11) below, k is the chirp rate. The difference ΔT of the time shift amount is obtained by equation (12) below, which is obtained by rearranging equation (11) below. ｉ＋１ -ΔT ｉ This will be decided.

[0088]

[0089]

[0090] The center frequency fc used by DDS21_1, ..., 21_N １ This is predetermined according to the bandwidth of the transmitting antennas 12_1, ..., 12_N.

[0091] Switches 22_1, ..., 22_N perform switching operations to generate a transmission signal, similar to Embodiment 1. Transmitting antennas 12_1, ..., 12_N generate and radiate radio waves from the transmission signal generated by switches 22_1, ..., 22_N and amplified by amplifiers 23_1, ..., 23_N. When the radar device 1 is equipped with two transmitting antennas 12_1, 12_2, as shown in Figure 3, the received signal generated by receiving antennas 15_1, ..., 15_M includes a component originating from the transmission signal generated by DDS 21_1, 21_2, specifically, the delayed transmission signal. The received signal has a time shift amount ΔT １ The signal component is 0, and the time shift amount ΔT ２ Includes the signal component and time shift amount ΔT １ Signal component and time shift amount ΔT = 0 ２ All of the signal components have a center frequency of fc １ That is the case.

[0092] Mixer 42_2 mixes the modulated signal output from DDS 21_2 and A / D converted by ADC 19_2 with the received signal acquired from ADC 41. The output signal of mixer 42_2 includes an autocorrelation component and a cross-correlation component in the frequency domain, similar to the example of Embodiment 1 shown in Figure 9.

[0093] As an example, when the number of multiplexers N=3, the beat frequency F of the autocorrelation component included in the output of mixers 42_1, 42_2, and 42_3 ａｃ This is expressed by equation (2) above, similar to Embodiment 1. As an example, when the number of multiplexers N = 3, the center frequency of the modulation signal generated by DDS21_1, 21_2, 21_3 is fc １ The beat frequency F of the cross-correlation component included in the output of mixer 42_1 is defined as follows. ｃ２ｃ１ , F ｃ３ｃ１This is expressed by equations (13) and (14) below. In equations (13) and (14) below, ΔT １ ΔT ２ ΔT ３ This is the time shift amount used when generating the modulated signal in DDS21_1, 21_2, and 21_3, and the coefficient b is a non-negative integer represented by b = 0, 1, 2, ...

[0094] At the output of mixer 42_1, the cross-correlation component in the received signal between the component caused by the modulated signal generated by DDS21_2 and the modulated signal generated by DDS21_1 is the beat frequency F when b=0. ｃ２ｃ１ And the beat frequency F when b = 0 ｃ２ｃ１ Centered around 1 / T ｐｒｉ They appear at positions shifted by one increment. At the output of mixer 42_1, the cross-correlation component between the component caused by the modulated signal generated by DDS21_3 and the modulated signal generated by DDS21_1 is the beat frequency F when b=0. ｃ３ｃ１ And the beat frequency F when b = 0 ｃ３ｃ１ Centered around 1 / T ｐｒｉ They appear in positions that are slightly offset from each other.

[0095]

[0096]

[0097] Similarly, the beat frequency F of the cross-correlation component included in the output of mixer 42_2 ｃ１ｃ２ , F ｃ３ｃ２ This is expressed by equations (15) and (16) below. Similarly, the beat frequency F of the cross-correlation component included in the output of mixer 42_3 ｃ１ｃ３ , F ｃ２ｃ３ This is expressed by equations (17) and (18) below.

[0098]

[0099]

[0100]

[0101]

[0102] As mentioned above, pulse period T ｐｒｉ Time shift amount ΔT based on this １ ,・・・,ΔT Ｎ If set, the beat frequency difference Δf is the maximum value Δf ｍａｘ This is the result. At this time, the lower limit of the passband of BPF43_1, ..., 43_N is -Δf ｍａｘ The upper limit can be set to 0. This removes the replica component from the output of mixers 42_1, ..., 42_N, and the beat signal output by BPFs 43_1, ..., 43_N will no longer contain the replica component. Note that the lower limit of the passband of BPFs 43_1, ..., 43_N should be -Δf ｍａｘ The following applies, and the upper limit may be set to 0 or greater. In this case, -Δf ｍａｘ For replica components with frequencies less than or greater than 0 that have passed through BPF 43_1, ..., 43_N, the range Doppler data generation unit 17 can simply ignore them.

[0103] As described above, the radar device 1 according to Embodiment 2 emits transmitted radio waves based on multiple transmitted signals generated from multiple modulated signals that are shifted in the time domain according to different time shift amounts based on the pulse period, and receives transmitted radio waves reflected by the target 100 to generate multiple received signals. For each received signal, the radar device 1 generates multiple beat signals by mixing the received signal with each modulated signal. Since the amount of time shift is determined according to the pulse period, the frequency difference between the autocorrelation component and the cross-correlation component included in the mixed signal widens, making it possible to reduce the cross-correlation component in the band-pass filter. As a result, erroneous detection of the target 100 based on the replica component is suppressed. In other words, a radar device 1 with high target detection accuracy can be obtained.

[0104] This disclosure is not limited to the embodiments described above. In embodiments 1 and 2, multiple transmission signals were generated using a reference signal generation circuit, multiple DDSs, multiple switches, and multiple amplifiers, but transmission signals may be generated with different configurations. For example, a transmitter comprising a reference signal generation circuit, a pulse signal generation circuit, multiple modulation circuits, multiple DACs (Digital-to-Analog Converters), and multiple amplifiers may generate multiple modulated signals whose center frequencies are shifted from each other by a frequency shift amount based on the pulse period, or multiple modulated signals shifted in the time domain according to different time shift amounts based on the pulse period, and multiple transmission signals may be generated by multiplying each of the modulated signals by a rectangular pulse signal.

[0105] This disclosure is not limited to the embodiments described above. For example, the number of transmitting antennas and the number of receiving antennas may be the same or different.

[0106] In embodiments 1 and 2, the MIMO multiplexing number N was shown to be the same as the number of transmitting antennas 12_1, ..., 12_N. However, the multiplexing number N does not have to be the same as the number of transmitting antennas. For example, the multiplexing number N may be the same as the number of transmitting antennas 12_1, ..., 12_2N.

[0107] In embodiments 1 and 2, the time-dependent change in the frequency of the modulation signal generated by DDS21_1, ..., 21_N is represented by a sawtooth wave waveform, but it may also be represented by a triangular wave waveform.

[0108] The target detection unit 18 determines from among the multiple range Doppler data that the range is Δf obtained by equation (10) in equation (9) above. ｍａｘ A beam may also be formed from data at or below the reference distance D obtained by substituting the values.

[0109] The hardware configuration described above is an example and can be changed and modified as needed. As another example of a hardware configuration, a modified example of the hardware configuration of radar device 1 is shown in Figure 13. Radar device 1 comprises a processor 84, a memory 85, and an interface 86. The processor 84, memory 85, and interface 86 are connected to each other by a bus 83. The processor 84 includes any electronic circuit including transistors and can be considered a circuit or processor circuit.

[0110] The functions of the radar device 1 are realized by software, firmware (software embedded in electronic equipment), or a combination of software and firmware. The software is written as a program and stored in memory 85. The processor 84 reads and executes the program stored in memory 85, thereby realizing the functions of each of the above-mentioned parts. In other words, memory 85 stores a program for executing the processing of the radar device 1.

[0111] Memory 85 includes, for example, non-volatile or volatile semiconductor memories such as RAM (Random Access Memory), ROM (Read-Only Memory), flash memory, EPROM (Erasable Programmable Read Only Memory), and EEPROM (Electrically Erasable and Programmable Read-Only Memory), as well as magnetic disks, flexible disks, optical disks, compact disks, minidiscs, DVDs (Digital Versatile Discs), etc.

[0112] In Figure 13, one processor 84 and one memory 85 are shown, but the radar device 1 may be implemented with multiple processors 84 and multiple memory 85. In this case, the various functions of the radar device 1 can be executed by the cooperation of the multiple processors 84 and multiple memory 85.

[0113] Some functions of the radar device 1 may be implemented by dedicated hardware, while others may be implemented by software or firmware. For example, in the radar device 1, the processing of the transmission signal generation unit 11, the reference signal generation circuit 13, ADC 19_1, ..., ADC 19_N, and the beat signal generation unit 16 may be implemented by the processing circuit 81 shown in Figure 2, for example, an FPGA, and the range Doppler data generation unit 17 and the target detection unit 18 may be implemented by the processor 84 shown in Figure 13 reading and executing a program stored in memory 85. By supplying the result of decimation processing on the beat signal generated by the beat signal generation unit 16 to the range Doppler data generation unit 17, the processing of the range Doppler data generation unit 17 and the target detection unit 18 can be processed in real time, for example, on a computer server.

[0114] The various aspects of this disclosure are summarized below as appendices. (Appendix 1) A MIMO (Multiple-Input Multiple-Output) radar device for detecting a target, comprising: a transmission signal generation unit that generates a plurality of modulation signals in the FMCW (Frequency Modulation Continuous Wave) method, in which the frequency changes periodically according to a chirp rate indicating the amount of frequency change for each chirp period, wherein the center frequencies are shifted from each other by a frequency shift amount based on the pulse period, or a plurality of modulation signals shifted in the time domain according to different time shift amounts based on the pulse period, and generates a plurality of transmission signals by pulsing each of the modulation signals based on the pulse period; a plurality of transmission antennas that radiate transmission radio waves based on different transmission signals; a plurality of receiving antennas that receive the transmission radio waves reflected by the target and generate a received signal based on the received radio waves; and a beat signal generation unit that generates a plurality of beat signals for each of the received signals by mixing the received signal and each of the modulation signals and passing the resulting mixed signal through a band-pass filter. A radar device comprising: a range Doppler data generation unit that generates a plurality of range Doppler data indicating the distance difference to the target and the speed difference to the target by converting each of the beat signals to the frequency domain; and a target detection unit that forms a beam from the plurality of range Doppler data and detects the target from the formed beam. (Note 2) The radar device according to Note 1, wherein the transmission signal generation unit generates a plurality of modulated signals whose center frequencies are shifted from each other by the frequency shift amount, and the difference in center frequencies of two adjacent modulated signals is determined based on the result of adding a value obtained by dividing a second coefficient, which is a natural number relatively prime to the multiplexing number and less than the multiplexing number, by the multiplexing number, to a first coefficient, which is a non-negative integer determined according to the bandwidth of the transmission antenna, the bandwidth of the reception antenna, and the multiplexing number of MIMO, by the multiplexing number, and then multiplying the result by the reciprocal of the pulse period.(Note 3) The radar device according to Note 1, wherein the transmitting signal generation unit generates a plurality of modulated signals that are shifted in the time domain according to the time shift amount, and the difference between two time shift amounts corresponding to two adjacent modulated signals whose timings for starting to change in the time domain are determined based on the result of adding a value obtained by dividing a second coefficient, which is a natural number relatively prime to the multiplexing number and less than the multiplexing number, by the multiplexing number, to a third coefficient which is a non-negative integer determined according to the chirp period and the MIMO multiplexing number, and then multiplying the result by the reciprocal of the pulse period and the reciprocal of the chirp rate. (Note 4) The radar device according to any one of Notes 1 to 3, wherein the bandwidth of the passband of the bandpass filter is determined based on the result of multiplying the reciprocal of the MIMO multiplexing number and the reciprocal of the pulse period. (Note 5) The radar device according to Note 4, wherein the lower limit of the passband of the band-pass filter is a negative value, the absolute value of the lower limit is determined based on the result of multiplying the reciprocal of the multiplexing number and the reciprocal of the pulse period, and the upper limit of the passband of the band-pass filter is 0. (Note 6) The radar device according to any one of Notes 1 to 5, wherein the target detection unit forms the beam from data among a plurality of range Doppler data, the range of which is less than or equal to a reference distance determined based on the result of multiplying the result of dividing the speed of light by 2 by the reciprocal of the MIMO multiplexing number, the reciprocal of the pulse period, and the reciprocal of the chirp rate.(Note 7) A target detection method performed by a MIMO (Multiple-Input Multiple-Output) radar system for detecting targets, wherein the modulation signal is of the FMCW (Frequency Modulation Continuous Wave) type, in which the frequency changes periodically according to the chirp rate, which indicates the amount of frequency change for each chirp period, and the center frequencies of the multiple modulation signals are shifted from each other by a frequency shift amount based on the pulse period, or the multiple modulation signals are shifted in the time domain according to different time shift amounts based on the pulse period, and each of the modulation signals is pulsed based on the pulse period to generate multiple transmission signals, and the transmission radio waves based on the different transmission signals are radiated from multiple transmitting antennas, the transmission radio waves reflected by the target are received by multiple receiving antennas, and a received signal is generated based on the received radio waves, and for each of the received signals, the received signal and each of the modulation signals are mixed to obtain a mixed signal which is then passed through a band-pass filter to generate multiple beat signals for each of the received signals, A target detection method comprising: converting each of the aforementioned beat signals to the frequency domain to generate a plurality of range Doppler data indicating the distance difference and the velocity difference with respect to the target; forming a beam from the plurality of range Doppler data; and detecting the target from the formed beam.

[0115] This disclosure allows for various embodiments and modifications without departing from the broad spirit and scope of this disclosure. Furthermore, the embodiments described above are for illustrative purposes only and do not limit the scope of this disclosure. In other words, the scope of this disclosure is indicated by the claims, not by the embodiments. Various modifications made within the scope of the claims and the equivalent significance of the disclosure are considered to be within the scope of this disclosure.

[0116] This application is based on Japanese Patent Application No. 2024-170058, filed on 30 September 2024. The entire specification, claims, and drawings of Japanese Patent Application No. 2024-170058 are incorporated herein by reference.

[0117] 1 Radar device, 10 Transmitter, 11 Transmit signal generation unit, 12_1, ..., 12_N Transmitting antenna, 13 Reference signal generation circuit, 14 Receiver, 15_1, ..., 15_M Receiving antenna, 16 Beat signal generation unit, 17 Range Doppler data generation unit, 18 Target detection unit, 19_1, ..., 19_N ADC, 21_1, ..., 21_N DDS, 22_1, ..., 22_N Switch, 23_1, ..., 23_N Amplifier, 31_1, ..., 31_M Receiving signal processing circuit, 32_1, ..., 32_M FFT circuit, 41 ADC, 42_1, ..., 42_N Mixer, 43_1, ..., 43_N BPF, 44_1, ..., 44_N Range FFT, 45_1, ..., 45_N Doppler FFT, 81 Processing circuit, 82 Interface circuit, 83 Bus, 84 Processor, 85 Memory, 86 Interface, 100 Target, W1, W2 Transmitted radio waves, W1', W2' Reflected radio waves

Claims

1. A MIMO (Multiple-Input Multiple-Output) radar system for detecting targets, comprising: a transmission signal generation unit that generates a plurality of modulation signals in the FMCW (Frequency Modulation Continuous Wave) method, in which the frequency changes periodically according to a chirp rate indicating the amount of frequency change for each chirp period, wherein the center frequencies are shifted from each other by a frequency shift amount based on the pulse period, or a plurality of modulation signals shifted in the time domain according to different time shift amounts based on the pulse period, and generates a plurality of transmission signals by pulsing each of the modulation signals based on the pulse period; a plurality of transmission antennas that radiate transmission radio waves based on different transmission signals; a plurality of receiving antennas that receive the transmission radio waves reflected by the target and generate a received signal based on the received radio waves; and a beat signal generation unit that generates a plurality of beat signals for each of the received signals by mixing the received signal and each of the modulation signals and passing the resulting mixed signal through a band-pass filter. A radar device comprising: a range Doppler data generation unit that generates a plurality of range Doppler data indicating the distance difference and the speed difference with respect to the target by converting each of the beat signals to the frequency domain; and a target detection unit that forms a beam from the plurality of range Doppler data and detects the target from the formed beam.

2. The radar device according to claim 1, wherein the transmitting signal generation unit generates a plurality of modulated signals whose center frequencies are shifted from each other by the frequency shift amount, and the difference in the center frequencies of two adjacent modulated signals is determined based on the result of adding a value obtained by dividing a second coefficient, which is a natural number relatively prime to the multiplexing number and less than the multiplexing number, by the multiplexing number, to a first coefficient, which is an integer of 0 or more determined according to the bandwidth of the transmitting antenna, the bandwidth of the receiving antenna, and the multiplexing number of MIMO, by the multiplexing number, and then multiplying the result by the reciprocal of the pulse period.

3. The radar device according to claim 1, wherein the transmitting signal generation unit generates a plurality of modulated signals that are shifted in the time domain according to the time shift amount, and the difference between two time shift amounts corresponding to two adjacent modulated signals whose timings in the time domain begin to change is determined based on the result of adding a value obtained by dividing a second coefficient, which is a natural number relatively prime to the multiplexing number and less than the multiplexing number, by the multiplexing number, to a third coefficient which is a non-negative integer determined according to the chirp period and the MIMO multiplexing number, and then multiplying the result by the reciprocal of the pulse period and the reciprocal of the chirp rate.

4. The radar apparatus according to any one of claims 1 to 3, wherein the bandwidth of the passband of the bandpass filter is determined based on the result of multiplying the reciprocal of the MIMO multiplexing number by the reciprocal of the pulse period.

5. The radar device according to claim 4, wherein the lower limit of the passband of the band-pass filter is a negative value, the absolute value of the lower limit is determined based on the result of multiplying the reciprocal of the multiplexing number by the reciprocal of the pulse period, and the upper limit of the passband of the band-pass filter is 0.

6. The radar apparatus according to any one of claims 1 to 5, wherein the target detection unit forms the beam from data among a plurality of range Doppler data, the range of which is less than or equal to a reference distance determined based on the result of multiplying the result of dividing the speed of light by 2 by the reciprocal of the MIMO multiplexing number, the reciprocal of the pulse period, and the reciprocal of the chirp rate.

7. A target detection method performed by a MIMO (Multiple-Input Multiple-Output) radar system for detecting a target, comprising: generating a plurality of modulation signals in the FMCW (Frequency Modulation Continuous Wave) method, the frequency of which changes periodically according to a chirp rate indicating the amount of frequency change for each chirp period, wherein the center frequencies of the plurality of modulation signals are shifted from each other by a frequency shift amount based on the pulse period, or a plurality of modulation signals that are shifted in the time domain according to different time shift amounts based on the pulse period; generating a plurality of transmission signals by pulsing each of the modulation signals based on the pulse period; radiating transmission radio waves based on the different transmission signals from a plurality of transmitting antennas; receiving the transmission radio waves reflected by the target with a plurality of receiving antennas, and generating a received signal based on the received radio waves; generating a plurality of beat signals for each of the received signals by mixing the received signal and each of the modulation signals and passing the resulting mixed signal through a band-pass filter; and generating a plurality of range Doppler data indicating the distance difference to the target and the speed difference to the target by converting each of the beat signals to the frequency domain. A target detection method comprising forming a beam from multiple range Doppler data and detecting the target from the formed beam.

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