Phase measurement for high-precision distance measurement

Abstract

In at least one embodiment, a method for measuring a distance between a first communication device including a first local oscillator and a second communication device including a second local oscillator includes unwarping N phase values ​​to produce N unwarped phase values. N is an integer greater than 1. Each of the N phase values ​​indicates an instantaneous phase of a received signal. The method includes averaging the N unwarped phase values ​​to produce an average phase value. The method also includes warping the average phase value to produce a final phase measurement of the first local oscillator relative to the second local oscillator.

Description

Technical Field

[0001] The present disclosure relates generally to communication systems, and more particularly to radio frequency (RF) communication systems and related methods for measuring distance using phase measurements. Background Art

[0002] Typically, the position of a wireless device in a plane (i.e., two dimensions) can be determined using triangulation (e.g., using two angle measurements), trilateration (e.g., using three distance measurements), or a combination thereof. Ranging techniques include received signal strength indicator (RSSI)-based, time-based, and phase-based distance measurements. Since RSSI-based measurements are susceptible to multipath fading and complex noise interference in indoor applications, time-based or phase-based distance measurements are preferably used in short-range radio frequency communication systems (e.g., Bluetooth compliant). TM ,Bluetooth TM Low energy (BLE), Zigbee TM or other LAN protocol standard systems).

[0003] Bluetooth Low Energy is an exemplary communication protocol designed for low power and low latency applications. BLE devices (i.e., devices that conform to the BLE standardized communication protocol) consume substantially less power than traditional Bluetooth (i.e., Classic Bluetooth) devices (i.e., devices that conform to the Bluetooth standardized communication protocol). Exemplary BLE devices can start data transmission faster than traditional Bluetooth devices. Therefore, BLE devices can be turned on continuously or turned on and off frequently so that they can communicate with other devices intermittently. BLE communication devices implement phase-based distance measurement. Phase-based distance measurement relies on a phase shift φ introduced to the radio signal by a pure line-of-sight radio channel, which phase shift φ is a linear function of frequency f and distance R, i.e.,

[0004]

[0005] Where c is the speed of light in a vacuum. Therefore, a BLE device determines the distance between itself and another BLE device by measuring the slope of the phase as a function of frequency.

[0006] The BLE device that starts the ranging technique is called the initiator. The other BLE device that responds to the initiator is called the responder. After the frequency calibration phase, the initiator uses the first channel to send the local oscillator signal (i.e., with a frequency f k The responding party measures the phase φ of the received carrier. R To determine the phase of the initiator's local oscillator as seen at the responder:

[0007]

[0008] Among them, Δ t is the time offset (i.e., propagation time) between the initiator and the reactant, and θ is the phase difference between the local oscillator signal of the initiator and the local oscillator signal of the reactant. R Depends on the local oscillator of the initiator, the local oscillator of the responder, and the distance between them. However, in some embodiments of the BLE communication system, the frequency (2.4 GHz) corresponds to a wavelength that causes phase wrapping (e.g., a wavelength of approximately 12 cm), resulting in ambiguity in the distance measurement. This ambiguity can be resolved by measuring the phase shift using two (or more) different tones (e.g., tones with a 1 MHz difference):

[0009]

[0010] The phase still wraps around, but it happens relative to the frequency corresponding to f1-f2 (e.g., 1 MHz). Similarly, the reactant sends a continuous wave to the initiator over the same channel so that the initiator can measure the phase φ I To determine the phase of the responder's local oscillator as seen at the initiator:

[0011]

[0012]

[0013] The communication device may repeat this process for at least one additional channel in the frequency band to reduce the effects of multipath fading and other impairments on range measurements.

[0014] The reactor sends a phase measurement to the initiator. The initiator calculates the round-trip phase φ by adding the two phase measurements. RT :

[0015]

[0016] The initiator calculates the round-trip distance R:

[0017]

[0018] where φ RT1 =φ I1 +φ R1 And φ RT2 =φ I2 +φ R2 Note that when using two frequency measurement ranges R, the constant phase offset θ between the transmitter and receiver local oscillators of the responding party cancels out.

[0019] In at least one embodiment, for a target maximum distance measurement of approximately 50m and a maximum initiator speed relative to the reactant of 5km / h, the distance calculation complies with the BLE standardized communication protocol specification in high-accuracy operating mode if the distance calculation is within ±10% for distances greater than 5m, and within ±0.50m for distances less than 5m. Phase measurement performance at each device is critical to the round-trip distance calculation. Noise or any frequency offset or drift between the initiator's local oscillator and the target frequency, and between the reactant's local oscillator and the target frequency, can introduce errors into the distance measurement. Therefore, techniques are desired to reduce or eliminate the effects of noise, frequency offset, or frequency drift on phase measurements. Summary of the Invention

[0020] In at least one embodiment, a method for measuring a distance between a first communication device including a first local oscillator and a second communication device including a second local oscillator includes unwarping N phase values ​​to produce N unwarped phase values. N is an integer greater than 1. Each of the N phase values ​​indicates an instantaneous phase measurement of a received signal. The method includes averaging the N unwarped phase values ​​to produce an average phase value. The method includes warping the average phase value to produce a final phase measurement of the first local oscillator relative to the second local oscillator.

[0021] In at least one embodiment, a communication system includes a first communication device configured to receive a signal using a local oscillator. The first communication device includes a phase value generator configured to generate N phase values ​​based on N corresponding samples of the received signal. The first communication device includes a phase unwrapping circuit configured to unwrap the N phase values ​​to generate N unwrap phase values. N is an integer greater than 1. Each of the N phase values ​​indicates an instantaneous phase measurement of the received signal. The first communication device includes an averaging circuit configured to average the N unwrap phase values ​​to generate an averaged phase value. The first communication device includes a phase warping circuit configured to warp the averaged phase value to generate a final phase measurement of a first local oscillator relative to a second local oscillator.

[0022] In at least one embodiment, a method for measuring a distance between a first communication device including a first local oscillator and a second communication device including a second local oscillator includes averaging N in-phase components of N samples of a received signal to produce an averaged in-phase component. N is an integer greater than 1. The method includes averaging N quadrature components of the N samples to produce an averaged quadrature component. The method includes generating a wrap count indicating a number of revolutions of a unit circle of the phase of the N samples. The method includes generating a phase measurement of the first local oscillator relative to the second local oscillator based on the averaged in-phase component, the averaged quadrature component, and the wrap count. The wrap count is based on a subinterval of a symbol period of the received signal. BRIEF DESCRIPTION OF THE DRAWINGS

[0023] The present invention may be better understood, and its numerous objects, features, and advantages made apparent to those skilled in the art by referencing the accompanying drawings.

[0024] Figure 1 A functional block diagram of an exemplary wireless communication system is shown.

[0025] Figure 2 A functional block diagram of a receiver of an exemplary wireless communication device is shown.

[0026] Figure 3 A functional block diagram of an exemplary angle measurement technique that includes integrating samples prior to angle calculation for range measurement by a wireless communication device is shown.

[0027] Figure 4 A functional block diagram of an exemplary angle measurement technique including averaging calculated angles for use in distance measurement by a wireless communication device in accordance with at least one embodiment of the present invention is shown.

[0028] Figure 5 A functional block diagram of an exemplary angle measurement technique including integrating samples prior to angle calculation and tracking phase wrapping for range measurement by a communication device according to at least one embodiment of the present invention is shown.

[0029] Figure 6 A functional block diagram of an exemplary angle measurement technique that integrates samples over intervals that avoid phase wrapping and averages the calculated angles for range measurement by a wireless communication device in accordance with at least one embodiment of the present invention is shown.

[0030] Figure 7 An exemplary timing diagram illustrating communications between wireless communication devices used in distance measurement according to at least one embodiment of the present invention is shown.

[0031] Figure 8 A functional block diagram of portions of an exemplary receiver for estimating and compensating for DC offset in a signal used in distance measurements according to at least one embodiment of the present invention is shown.

[0032] Figure 9 A functional block diagram of portions of an exemplary receiver for measuring phase and measuring and compensating for frequency offset in a signal used in distance measurements according to at least one embodiment of the present invention is shown.

[0033] Figure 10 Exemplary information and control flows are shown for compensating for frequency offset during a frequency compensation interval in communications between wireless communication devices in accordance with at least one embodiment of the present invention.

[0034] Figure 11 Exemplary information and control flows are shown for compensating for residual frequency offset and measuring phase using packet-switched and tone-switched communications between wireless communication devices in accordance with at least one embodiment of the present invention.

[0035] The use of the same reference numbers in different drawings indicates similar or identical items. DETAILED DESCRIPTION

[0036] In at least one embodiment, reference Figure 1, the wireless communication system includes a wireless communication device 102 (including a transmitter 104, a receiver 106, data processing circuitry 107, a memory 103, and a local oscillator 105) and a wireless communication device 112 (including a transmitter 114, a receiver 116, data processing circuitry 138, a memory 136, and a local oscillator 115). Wireless communication device 102 is located at a distance R from wireless communication device 112. Although wireless communication device 102 and wireless communication device 112 are shown as each including only one antenna, in other embodiments of wireless communication system 100, wireless communication device 102 and wireless communication device 112 each include multiple antennas. Wireless communication system 100 conforms to the BLE standardized communication protocol designed for low-power and low-latency applications. However, in other embodiments, wireless communication system 100 conforms to other wireless communication protocols (e.g., Classic Bluetooth, Zigbee, or other short-range radio frequency protocol standards). Local oscillators 105 and local oscillators 115 provide signals used in the transceiver functions of wireless communication device 102 and wireless communication device 112, respectively. If wireless communication device 102 and wireless communication device 112 are manufactured by different vendors, the frequency of local oscillator 105 may be substantially different from the frequency of local oscillator 115. However, for phase measurements, the transmitted continuous wave signals transmitted by the initiator or the reacter are nominally the same. If not properly accounted for, random frequency errors (e.g., ±40 ppm on each side) between the initiator and reacter continuous wave signals may introduce errors into the measurement of distance R made by wireless communication system 100.

[0037] Figure 2 An exemplary embodiment of a receiver that can be included in wireless communication device 102 or wireless communication device 112 is shown. Antenna 101 provides an RF signal to a passive network 120, which provides impedance matching, filtering, and electrostatic discharge protection. Passive network 120 is coupled to a low noise amplifier (LNA) 122, which amplifies the RF signal without substantially degrading the signal-to-noise ratio and provides the amplified RF signal to a mixer 124. Mixer 124 uses a reference or local oscillator (LO) signal provided by local oscillator 115 to perform frequency conversion or shifting of the RF signal. For example, in at least one operating mode of receiver 116, mixer 124 frequency converts the RF signal to a baseband frequency centered about DC (i.e., a zero intermediate frequency (ZIF) in the ZIF operating mode). In another operating mode, receiver 116 is configured as a low intermediate frequency (LIF) receiver (i.e., in the LIF operating mode), and mixer 124 converts the RF signal to a low intermediate frequency (e.g., 100 to 200 kHz) to avoid the DC offset and 1 / f noise issues of a ZIF receiver.

[0038] Mixer 124 provides converted output signals as a set of two signals, namely, an in-phase (I) signal and a quadrature (Q) signal. The I and Q signals are analog time-domain signals. In at least one embodiment of receiver 116, analog amplifier and filter 128 provides amplified and filtered versions of the I and Q signals to analog-to-digital converter (ADC) 130, which converts these versions of the I and Q signals into digital I and Q signals (i.e., I and Q samples). Exemplary embodiments of ADC 130 use various signal conversion techniques (e.g., delta-sigma (i.e., sigma-delta) analog-to-digital conversion). ADC 130 provides the digital I and Q signals to signal processing circuitry 132. Typically, signal processing circuitry 132 performs processing on the digital I and Q signals (e.g., demodulation, frequency conversion (e.g., using mixer 131), filtering, or signal correction). In at least one embodiment, signal processing circuitry 132 includes a demodulator 141 that retrieves or extracts information from the digital I and Q signals (e.g., data signals modulated by a transmitter (not shown) and provided as RF signals to antenna 101). In at least one embodiment, one or more circuits of signal processing circuitry 132 convert the digital I and Q signals from a Cartesian representation to a polar representation (i.e., instantaneous phase and instantaneous amplitude) for use by frequency correction circuitry 142 or phase averaging circuitry 143. In at least one embodiment, signal processing circuitry 132 (described further below) generates at least one correction value for application to local oscillator 115 or other circuitry of receiver 116.

[0039] In at least one embodiment, signal processing circuitry 132 provides information, such as demodulated data or phase measurements, to data processing circuitry 138. Data processing circuitry 138 can perform various functions (e.g., logic, arithmetic, etc.). For example, data processing circuitry 138 can use the demodulated data in a program, routine, or algorithm (whether in software, firmware, hardware, or a combination thereof) to perform desired control or data processing tasks. In at least one embodiment, data processing circuitry 138, including memory 136, controls other circuits, subsystems, or systems (not shown).

[0040] In at least one embodiment, phase averaging circuit 143 or other circuitry within signal processing circuit 132 calculates a phase value (i.e., an angle value) for a filtered version of each pair of digital I and Q signals received from digital filter 140 during a measurement interval of receiver operation and generates a phase measurement for the phase measurement interval. The phase measurement interval can have different values ​​through negotiation between the initiator and the reflector. In an exemplary embodiment, a phase measurement interval of 80 μs is used. This interval is relatively long enough to include averaging of the measurements, which increases the accuracy of the phase measurement used for distance measurement. Figure 3An exemplary prior art technique for calculating a phase measurement is shown by integrating the digital I and Q baseband (ZIF) signals during a measurement interval using respective integrator circuits 302 and 304 (e.g., averaging N I samples and N Q samples, where N is an integer greater than 1 and indicates the number of samples in the measurement interval), and then providing the integrated digital I signal and the integrated digital Q signal to an inverse tangent circuit 306, which calculates a single angle measurement, namely the inverse tangent of the integrated digital Q signal divided by the integrated digital I signal. Performing the integration prior to calculating the angle measurement suppresses noise and frequency offset. In at least one embodiment of the receiver, Figure 3 The technique is sufficient to meet specifications in systems with low frequency offsets (e.g., ±4 ppm). However, in environments with larger frequency offsets (e.g., systems including initiators and reacters with frequency offsets up to 80 ppm), errors of ±π (i.e., 3.14159 radians) can occur. To achieve 0.5 m accuracy for the corresponding distance measurement, the sum of the phase errors from all sources must be less than 0.1 radians, so it is desirable to improve the techniques used to perform phase measurements.

[0041] Figure 4 shows a technique that uses Figure 3 Compared to phase measurements using the α-axis phase measurement technique, this technique performs phase measurements with an improved signal-to-noise ratio and reduces or eliminates errors due to phase boundary transitions. The inverse tangent circuit 306 calculates N phase values ​​based on N I and Q baseband (ZIF) samples during a measurement interval (e.g., 80 microseconds). In at least one embodiment of a wireless communication system, the local oscillator operates at 2.4 GHz with a wavelength of approximately 12 cm, and due to frequency offset, the phase delay between the initiator transmitter (LO) and the reacting receiver (LO) increases (i.e., accumulates) over the time interval. The accumulated phase delay is proportional to the distance, and in response to ensuring that all appropriate multiples of 2π are included in the sum, it may be necessary to unwrap the phase. The unwrap circuit 308 counts the number of cycles of 2π by detecting phase jumps at the π to -π boundary in the digital representation of the angle. The cycle count is added to the instantaneous angle to form the unwrap angle. After each angle measurement is calculated, averaging is performed to suppress noise and frequency offset. In at least one embodiment, a target worst-case frequency offset of ±80 ppm at 2.4 GHz corresponds to phase wrap every 5.2 microseconds.

[0042] In at least one embodiment, the inverse tangent circuit 306 performs an inverse tangent function with a range of 0 to 2π (i.e., all four quadrants). In at least one embodiment, the inverse tangent circuit 306 first calculates the angle assuming that the I and Q samples are in the first quadrant (e.g., angle = arctan(abs(Q) / abs(I)). Next, the inverse tangent circuit 306 checks the sign of the Q sample and the sign of the I sample to place the wrapped angle in the appropriate quadrant (first, second, third, or fourth) by adding the corresponding multiple of π to the angle calculated assuming that both the Q sample and the I sample are positive. The inverse tangent circuit 306 provides a wrapped angle value. The unwrapping circuit 308 observes the wrapped input angle value and adds 2π to the input angle value as needed to unwrap the input angle value. Typically, the wrapped angle refers to an angle value contained in the range between -π and π radians. The unwrapping circuit 308 provides the unwrapped angle value to the averaging circuit 310. Typically, the unwrapping The unwrapping circuit adds an appropriate multiple of 2π to each angle input to recover the original phase value. In at least one embodiment, the unwrapping circuit 308 adds M×2×π to the angle value. The averaging circuit 310 averages the N unwrapped angle values, where N is the number of samples in the averaging interval, and provides the average angle value to the warping circuit 312. In at least one embodiment, the warping circuit 312 performs a modulo 2π operation on the average value to provide a warped average angle value between -π and π or between 0 and 2π. In at least one embodiment, the averaging circuit 310 accumulates the angle values ​​using a fixed-point representation of a single binary word having an integer portion and a fractional portion. The integer portion represents the number of cycles of 2π, and the fractional portion represents the residual phase. The accumulated fixed-point representation is truncated to the fractional portion to warp the angle back to between -π and π. Figure 3 This technique results in a reduction in phase measurement error compared to the technique of Figure 4 The proposed method achieves higher signal-to-noise ratio measurements, increases the tolerance to frequency errors between the initiator and the reactant, and does not lead to erroneous phase measurements because phase wrapping is taken into account.

[0043] Figure 3 and Figure 4 Hybrid variants of technologies such as Figure 5 and Figure 6 shown. Figure 5 and Figure 6 The technique provides digital I and Q signals to integrators 302 and 304, respectively, which integrate the digital I and Q signals before the inverse tangent calculation. Figure 3 Similar to the technology, Figure 5 The technique performs a single angle calculation. Figure 3 The technique of ignoring winding is different, Figure 5The technique monitors the sign of the digital I and Q signals to detect when the samples wrap around by 2π and mitigates the potentially large errors that could result from this wraparound. However, the noise filtering characteristics are not as good as Figure 4 The noise filtering characteristics of the technology.

[0044] Figure 6 shows a technique for integrating P samples of the digital I and Q signals before performing the angle calculation, thus, Figure 4 Compared with the N angle calculations of the technology, N / P angle calculations are performed. Figure 6 The technique uses a reset signal to implement an integrate and dump operation on I and Q samples, followed by a series of phase values ​​and their corresponding averages over an interval (e.g., 4 microseconds) that is a fraction of the measurement interval to prevent wraparound and reduce the number of inverse tangent calculations. Integrators 302 and 304 integrate P I and Q samples (e.g., 10, which is a fraction of the number of phase values ​​generated during the phase measurement interval) and provide N / P integrated I and Q samples to inverse tangent circuit 306, followed by dewarping circuit 308 and averaging circuit 310, which averages the N / P dewarped phase values. Wrapping circuit 312 generates a phase measurement based on the averaged dewarped phase value. Figure 3 This technique reduces the number of required inverse tangent calculations and reduces errors compared to the technique of

[0045] In at least one embodiment, the inverse tangent circuit 306 is implemented using a coordinate rotation digital computer (CORDIC), which can be dedicated to phase measurement implementation or shared with other operations of the receiver. Typically, a CORDIC implements known techniques to perform calculations, including trigonometric functions and complex multiplications, without using multipliers. The only operations used by the CORDIC are addition, subtraction, shifts, and table lookup operations to implement the inverse tangent function. In other embodiments, a digital signal processor that executes firmware or inverse tangent circuitry is used. In at least one embodiment, the communication system stores the resulting angle measurement in a memory for use in distance calculations. In at least one embodiment, the communication system applies a phase correction term (PCT) to the resulting angle measurement and stores the phase-corrected angle measurement in a memory for use in distance measurement.

[0046] refer to Figure 1 Typically, PCT is applied to the received samples to compensate for the propagation delay between the initiator antenna and the reactant antenna. In at least one embodiment, PCT is characterized at the antenna, even though the signal is observed at different points in the receiver path. R Applied to the received samples at the reacting party to compensate for the propagation delay t between the initiator antenna and the reacting party antenna p Similarly, PCTI Applied to the received samples at the initiator to compensate for the propagation delay t between the initiator antenna and the reacter antenna p .For example:

[0047] PCT R (t)=

[0048] 2π(f initiator -f reflector )t reflector +2πf reflector t p +θ intiator -θ reflector +n reflector (t), and

[0049] PCT I (t)=

[0050] 2π(f reflector -f initiator )t initiator +2πf initiator t p +θ reflecto r-θ initiator +n initiator (t).

[0051] In at least one embodiment of the wireless communication system 100, the PCT measurement is:

[0052]

[0053]

[0054] Under ideal conditions, PCT R =PCT I , and the noise terms of both are averaged to negligible values. Therefore,

[0055] PCT R +PCT I =2πf reflector t p +2πf initiator t p .

[0056] Phase measurements at the initiator and reactor must be offset by the same amount of time

[0057] 2π(f initiator -f reflector )t reflector +2π(f initiator -f reflector )t initiator .

[0058] However, if the frequencies of the initiator and the responder do not match, the frequency offset (ie, f EST =f reflector -f initiator ) must be estimated and compensated to reduce the measurement time dependency:

[0059] 2π(f reflector -f initiator -f EST )t initiator =0.

[0060] refer to Figure 1 and Figure 7 In at least one embodiment, the wireless communication system 100 implements the BLE communication protocol, which performs frequency compensation at the initiator to reduce the frequency offset (i.e., f) between the frequency of the initiator's local oscillator and the frequency of the responder's local oscillator. initiator -f reflector ). However, frequency compensation implemented by an initiator manufactured by another vendor may not be sufficient to reduce the frequency offset to a level that provides distance measurements within a target accuracy specification (e.g., ±4 ppm). Therefore, in at least one embodiment of the wireless communication system 100, the reacter implements a frequency compensation technique to establish communications that conform to the target communication protocol, which occurs during a frequency compensation interval of an electronic handshake process performed by the communicating devices.

[0061] Although the wireless communication device 102 is configured as an initiator that performs frequency compensation during a first interval (e.g., a frequency compensation interval), the wireless communication device 112 that is configured as a reactant also performs frequency compensation during the first interval, but before the initiator frequency offset compensation. During the first interval, the wireless communication system 100 operates the receiver in LIF mode. The initiator transmits a packet during the initiator interval 702, and the reactant receives the packet during the reactant interval 722. The reactant uses a portion of the packet received from the initiator during the reactant interval 722 to determine the frequency offset f ESTand generates an associated adjustment value. An exemplary packet format includes a preamble with a predefined data pattern that the receiver can use to detect and determine its control loop, for example, an eight-bit sequence of alternating ones and zeros and a payload of length zero. In at least one embodiment, the packet includes a preamble, a 32-bit synchronization word (e.g., a PN sequence pn[31:0]), and a 96-bit probe sequence. In one embodiment, the PN sequence is a 32-bit pseudo-noise sequence (also known as a pseudo-random binary sequence (PRBS)) that is known to both the initiator and the reactant. In one embodiment, the synchronization word is an address or other identifier associated with the receiver that marks the start of the probe sequence. In other embodiments, the reactant determines the frequency offset f based on the packet received in the ZIF mode of operation. EST .

[0062] The reactant uses successive samples of the preamble, sync word, or a combination thereof to determine a frequency offset and applies an adjustment based on the frequency offset to the reactant local oscillator 115. This adjustment is effective for packets sent during reactant interval 724 and the remainder of the first interval, as well as at least during the second interval used to perform distance measurements (e.g., an interval including the packet exchange subinterval and the phase measurement subinterval). The initiator receives the packet during initiator interval 704 and subsequently receives successive waves that were sent by the reactant during reactant interval 726 and received during initiator interval 706. In an exemplary embodiment of wireless communication system 100, the packet received by the initiator during interval 704 and the successive waves received by the initiator during interval 706 will have a reduced frequency offset (e.g., a frequency offset of 5 kHz or less). The initiator uses the received packet and the received successive waves to determine the frequency offset f EST2 (eg, using the estimation techniques described herein or other techniques for estimating frequency offset), and applying associated adjustments to the initiator local oscillator 105 if necessary.

[0063] exist Figure 7 During the second interval, the reactant compensates for the frequency offset between the initiator and reactant to a negligible value, making the PCT measurement start point and duration irrelevant. In an exemplary embodiment of wireless communication system 100, the second interval includes a packet exchange subinterval and a phase measurement subinterval. In the packet exchange subinterval, the initiator transmits a packet during initiator interval 708, and the reactant receives the packet during reactant interval 728. Similarly, the reactant transmits a packet during reactant interval 730, and the initiator receives the packet during initiator interval 710. In other embodiments of wireless communication system 100, the packet exchange subinterval is omitted.

[0064] In at least one embodiment, the wireless communication system 100 includes a ZIF mode of operation for use during the phase measurement subinterval. In the ZIF mode of operation, the initiator and the reacter transmit baseband intermediate frequency (i.e., ZIF) signals. Since the baseband signal is at or near DC, the ZIF mode reduces or eliminates phase shifts in the received signal through baseband circuitry, ADCs, and digital filters, which may affect distance measurements based on phase measurements. Therefore, there is no need to know or calibrate the phase shifts through these circuits, and using the ZIF mode of operation simplifies phase measurements. In at least one embodiment of the wireless communication system 100, the reacter transmits a continuous wave CWT during the reacter interval 732. f1 , and the initiator uses continuous Boeing CWT during the initiator interval 712 f1 The receiving version measures the phase φ I Similarly, the initiator sends continuous Boeing CWT during the initiator interval 714 f2 , and the reactant uses continuous Boeing CWT during reactant interval 734 f2 The receiving version measures the phase φ R The wireless communication system 100 repeats the phase measurement for additional values ​​of i, where 1≤i≤I (e.g., I=70 and corresponds to 70 spaced carriers on an 80 MHz channel at 2.4 GHz). In at least one embodiment of the wireless communication system 100, the reacting party uses a continuous wave CWT received in ZIF mode. fi The residual frequency offset f is determined by ESTR In addition, an adjustment based on the residual frequency offset is applied to the PCT at the input of the demodulator to compensate for the frequency rotation of the phase correction value. The initiator determines the round-trip phase φ after the second interval RT .

[0065] In the ZIF mode of operation, self-mixing can cause DC offset, which is a significant source of error. In an exemplary embodiment of the wireless communication system 100, for received signals near the sensitivity level, the analog I and Q signals have a peak-to-peak level of approximately 20 mV. The residual DC offset is specified to be much less than 1 mV DC to reduce or eliminate phase errors caused by the DC offset shifting the I and Q signals. Because information is transmitted in DC in the ZIF mode of operation, traditional techniques for attenuating DC offset (e.g., high-pass filtering using series AC coupling capacitors) are not feasible. In at least one embodiment, the wireless communication system 100 includes a calibration mode of operation that estimates the DC offset, generates DC offset compensation values, and stores these DC offset compensation values ​​in a memory element for later use during normal operation.

[0066] Figure 8Receiver 116 is shown configured in a calibration mode of operation, wherein the receiver input is coupled to a reference node (e.g., ground) to zero the receiver input signal. In the calibration mode of operation, receiver 116 is also configured in a ZIF mode of operation. In at least one embodiment of the digital filter 140, during the calibration mode of operation, the switch is configured in position 1, resulting in the low-pass filter output being subtracted from the input signal. After a predetermined amount of time, the controller 139 stores the final DC estimate for later use during normal operation mode. In at least one embodiment, the analog circuit (e.g., a programmable gain amplifier or an analog-to-digital converter) has multiple settings, and the controller 139 configures the receiver 116 to estimate the DC offset of each setting of the analog circuit and stores each estimate in a corresponding storage element or storage location (e.g., in a lookup table).

[0067] In at least one embodiment of the digital filter 140, in a calibration mode of operation, during calculation of the low-pass filter output, the controller 139 configures the switch in position 1 to couple the differential node to receive the corresponding current low-pass filter output value. During calculation of an updated value of the low-pass filter output, the corresponding digital signal is compensated by the current low-pass filter output value, and the controller 139 compares the compensated value (e.g., the output of the differential node) to a predetermined value (e.g., 0). If the difference is less than or equal to the predetermined value, the controller causes the low-pass filter output to be stored as a final DC estimate for later use during normal operation. In at least one embodiment, the analog circuit (e.g., a programmable gain amplifier or an analog-to-digital converter) has multiple settings, and the controller 139 configures the receiver 116 to estimate the DC offset of each setting of the analog circuit and store each estimate in a corresponding storage element or storage location (e.g., in a lookup table).

[0068] In at least one embodiment, for each setting of the analog circuit, the controller 139 first operates the low-pass filter using a first coefficient (e.g., k=k1) to cause the low-pass filter to have a first time constant of the low-pass filter, and after a predetermined number of samples, the controller 139 updates the low-pass filter to use a second coefficient (e.g., k=k2) to cause the low-pass filter to have a second time constant, wherein the second time constant is longer than the first time constant (e.g., |k1|>|k2|) to accelerate convergence of the low-pass filter output value. In an alternative embodiment, the controller 139 starts the low-pass filter, which is configured to operate with the first time constant of the low-pass filter, and in response to the compensation signal being below a threshold (i.e., indicating convergence), the controller 139 changes the value of k to a second value k2 associated with the second time constant of the low-pass filter. In at least one embodiment, the additional value of k is used to further control the convergence of the DC estimate. The low-pass filter can be a single-pole infinite impulse response low-pass filter circuit, a moving average circuit, or other finite impulse response low-pass filter circuit, or a variable-order infinite impulse response low-pass filter circuit.

[0069] After completing the calibration mode for each setting of the analog circuitry, in a normal operating mode in which the receiver 116 is configured in the ZIF operating mode, the controller 139 configures the switches in position 2 to couple the stored DC estimates corresponding to the active settings of the analog circuitry (e.g., the active settings of the PGA) to the corresponding differential nodes to compensate for DC offset in the digital I and Q signals received by the digital filter 140. For example, in the ZIF mode of the normal operating mode, the estimated value of the DC offset is subtracted from the digital I and Q signals, and the difference is used as the DC offset compensated digital I and Q signals (i.e., I C and Q C ) is provided to CORDIC 137 or other digital processing circuitry for further processing. In a normal operating mode where receiver 116 is configured for LIF operating mode, controller 139 configures the switch in position 1 and digital filter 140 is configured as a high-pass filter that blocks DC for the LIF operating mode.

[0070] refer to Figure 9 In at least one embodiment, the receiver 116 implements the Figure 4 In addition, the receiver 116 estimates the frequency offset f EST and the residual frequency offset f ESTR , and these operations share CORDIC 137. In at least one embodiment, CORDIC 137 receives samples (e.g., filtered versions of the digital I and Q signals) from digital filter 140 and converts the digital I and Q signals from Cartesian representation to polar representation using the Pythagorean theorem to calculate the magnitude (e.g., ) and an inverse tangent operation (e.g., arctan(Q / I)) to calculate a phase value used by frequency correction circuit 142 and phase averaging circuit 143 to calculate a frequency offset estimate f EST , residual frequency offset estimation f ESTR and phase measurement φ.

[0071] In at least one embodiment, as described above, receiver 116 is configured as a ZIF receiver in a normal operating mode that includes a phase averaging circuit 143 that averages the phase measurement during the phase measurement interval. The unwrapping circuit 144 receives the phase value, adds / subtracts M×2×π to / from the phase value to generate an unwrapped phase value, and provides the unwrapped phase value to the averaging circuit 150. The unwrapping circuit counts the number of π / -π boundary crossings to determine the cycle count M. The averaging circuit 310 averages N unwrapped phase values, where N is an integer number of samples in the measurement interval. The averaging circuit 150 provides the averaged unwrapped value to the warping circuit 154. The warping circuit 154 performs a modulo 2π operation on the averaged unwrapped value to provide a warped phase measurement φ to the PCT circuit 158. In at least one embodiment, the warping circuit 154 warps the phase to within a digital representation of ±π. The PCT circuit 158 ​​applies frequency and delay corrections to the phase measurement. to calculate the phase correction term according to the PCT described above. In at least one embodiment, the reactant stores the corrected phase measurement for use in the distance measurement performed by the data processing circuit 138. In other embodiments, the reactant sends the phase-corrected phase measurement to the initiator for use in the distance measurement calculated at the initiator.

[0072] In at least one embodiment, in normal operating mode, receiver 116 is selectively configured as a LIF receiver (e.g., a digital mixer (not shown) is active in the receiver path) or a ZIF receiver. Frequency offset estimator 148 receives the phase value from CORDIC 137 and generates a frequency offset estimate f EST (i.e., f EST =f reflector -f initiator For example, the frequency offset estimator 148 calculates the discrete time phase difference value (eg, φ[n]-φ[n-1]=f X ) to differentiate the phase output of the CORDIC). The frequency offset estimator 148 subtracts a value corresponding to an expected phase difference value predetermined and stored in memory from the incoming discrete-time phase difference value, accumulates the error, and divides by a predetermined number of symbols (e.g., N symbols, 8≤N≤32) to form an estimate of the frequency error. The expected phase difference value is based on the specified frequency deviation f of the physical layer. DEV(e.g., ±250kHz).

[0073] For example, for frequency shift keying, when sending a "1", the transmitter sends a frequency of f RF +f DEV When sending "0", the transmitter sends a frequency of f RF -f DEV Tone. In an exemplary embodiment where the eight symbols of the preamble are used to estimate the frequency error, expected values ​​ξ1, ξ2, ξ3, -ξ1, -ξ2, -ξ3 corresponding to the expected instantaneous frequency deviation (i.e., the phase difference across the symbol) are stored in a memory. The values ​​of ξ1, ξ2, and ξ3 vary with the system BT parameters (e.g., bandwidth × bit time = 0.5), which determine the effects of transmitter pulse shaping and receiver filtering bandwidth. In at least one embodiment, the effects of filtering in the receiver and transmitter pulse shaping from one bit can affect subsequent bits, which is known as inter-symbol interference (ISI). If consecutive symbols contain relatively long "1s", a full frequency deviation (e.g., ξ1) is expected. If consecutive symbols alternate between "1" and "0", the expected frequency deviation is less than the full frequency deviation (e.g., ±ξ3) due to the effects of filtering. Each expected value corresponds to a different three-bit data pattern (i.e., b n 、b n-1 , and b n-2 ). After receiver filtering, exemplary values ​​of ξ1, ξ2, and ξ3 correspond to expected deviations f of 250 kHz, 173 kHz, and 92 kHz, respectively. DEV However, the expected deviation will vary depending on the filtering characteristics implemented in the receiver. If consecutive symbols include relatively long "1s", a full frequency deviation (e.g., 250 kHz) is expected, and if the exemplary transmitter has a 50 kHz offset, the frequency of the transmitted signal is f RF +f DEV +50kHz. Therefore, the frequency offset estimator 148 will calculate the average frequency offset value as follows:

[0074]

[0075] The frequency offset estimator 148 uses the average frequency offset value as the frequency offset estimate f EST The frequency correction circuit 160 is provided once per packet. In at least one embodiment, the frequency correction circuit 160 estimates the frequency offset based on the frequency offset f ESTA frequency adjustment value is generated (e.g., by negating the average frequency offset value) and combined with other control values ​​(FREQ or IFREQ) to generate LOCONTROL. For example, LOCONTROL controls the local oscillator 126 (e.g., controls the fractional-N phase-locked loop 164) so ​​that the reacting local oscillator frequency matches the initiating local oscillator frequency within a target specification (e.g., ±4 ppm).

[0076] In at least one embodiment, although receiver 116 is configured as a ZIF receiver that receives a continuous wave with the initiator local oscillator frequency, dewarping circuit 144 is shared with frequency correction circuit 142 and provides the dewarped phase value to phase-to-frequency circuit 146, which generates a frequency estimate based on the dewarped phase value (e.g., by calculating a discrete-time phase difference value (e.g., φ[n]-φ[n-1])). If the initiator local oscillator is perfectly matched to local oscillator 126, the output of mixer 124 is a DC signal and the output of phase-to-frequency circuit 146 is zero. If the initiator local oscillator and the reactor local oscillator are not matched, the output of mixer 124 is not a DC signal, and the frequency estimate output of phase-to-frequency circuit 146 is an estimate of the residual frequency offset between the initiator local oscillator and the reactor local oscillator. In at least one embodiment of the phase-to-frequency circuit 146, rather than using adjacent unwarped phase values, the phase-to-frequency circuit 146 uses non-adjacent unwarped phase values ​​(e.g., φ[n]-φ[n-2]) to calculate the discrete-time phase difference value to improve accuracy. The use of non-adjacent values ​​improves accuracy by increasing the expected phase difference above the system quantization limit. The phase-to-frequency circuit 146 provides a frequency offset estimate to an averaging circuit 152, which averages the N values ​​and provides the average phase difference (i.e., the average frequency, e.g., the average frequency offset when the receiver 116 is receiving a continuous wave with the initiator's local oscillator frequency) to a warping circuit 156. The warping circuit 156 warps the average phase difference (i.e., the frequency error) to within the limits of the output format (e.g., within ±400 kHz). In an exemplary embodiment, since the frequency offset estimate f is based on the frequency offset estimate f, the average phase difference (i.e., the frequency error) is calculated based on the frequency offset estimate f. EST The applied frequency correction brings the frequency offset into a narrower range (e.g., 15 kHz) so that the average frequency offset does not exceed the limits of the output format. The extra margin accommodates the maximum allowable frequency drift (e.g., 20 Hz / μsec). The warping circuit 156 uses the warped average frequency value as the residual frequency offset estimate f ESTR Provided to the frequency correction circuit 160. In some embodiments, the warping circuit 156 converts the residual frequency offset estimate f ESTR The residual frequency offset estimate f is provided to the PCT circuit 158, which converts the residual frequency offset estimate f into a value based on the measurement period. ESTRConvert to radians (for example, f ESTR × Figure 7 1 / 2 of the period of the interval 734). Return to reference Figure 9 , although in the embodiment the frequency correction circuit 142 is described as generating the residual frequency offset estimate f during the time when the receiver 116 is configured in the ZIF mode of operation ESTR , but in other embodiments, during the period when the receiver 116 is configured in the LIF mode of operation (e.g., Figure 7 During the frequency compensation performed in the interval 728), the frequency correction circuit 142 generates a residual frequency offset estimate f ESTR . Return to reference Figure 9 In at least one embodiment, the frequency correction circuit 160 uses the residual frequency offset to estimate f ESTR In at least one embodiment, the frequency correction circuit 160 estimates the residual frequency offset f ESTR Generates frequency compensation value.

[0077] In at least one embodiment, the frequency correction circuit 160 estimates the residual frequency offset based on the ESTR The compensation value is based on the frequency offset estimation f EST , and another compensation value of , to generate a control signal for adjusting the frequency of local oscillator 126. In at least one embodiment, when available, frequency correction circuit 160 calculates the residual frequency offset estimate f ESTR and frequency offset estimate f EST In some embodiments, when available, the frequency correction circuit 160 calculates the residual frequency offset estimate f ESTR and frequency offset estimate f EST The frequency correction circuit 160 is implemented in accordance with the embodiment of the present invention. ...

[0078] refer to Figure 1In at least one embodiment, the wireless communication system 100 compensates for the frequency offset between the wireless communication device 102 (configured as an initiator operating in LIF mode) and the wireless communication device 112 (configured as a responder operating in LIF mode). In other embodiments of the wireless communication system 100, the wireless communication device 112 is configured as a responder and the wireless communication device 102 is configured as an initiator, and the wireless communication system 100 uses Figure 10 sequence to compensate for the associated frequency offset. Figure 1 and Figure 10 During interval T1, the initiator's receiver 106 and the reacter's receiver 116 each generate analog DC offset compensation values ​​for each setting of the analog circuit (e.g., each gain level of the programmable gain amplifier) ​​according to the above-described techniques. After DC offset compensation, the initiator's transmitter 104 transmits the packet to the reacter, which operates its receiver in LIF mode. During interval T2, the reacter's receiver 116 estimates the frequency offset using the packet's preamble to estimate the frequency offset f EST Next, during interval T3, the initiator switches from the transmit mode of operation to the receive mode of operation, the reacter switches from the receive mode of operation to the transmit mode of operation, and the reacter configures its local oscillator (eg, oscillator 115) to compensate for the frequency offset f EST In the interval T4, the responding party compensates for the frequency offset f EST The initiator sends a packet to the local oscillator 115 of the initiator. The initiator uses this packet to estimate and compensate for the first initiator frequency offset, thereby reducing any frequency offset. The initiator and the reactant switch from packet communication mode to continuous wave communication mode during interval T5. The reactant sends a continuous wave to the initiator during interval T6, which the initiator uses to estimate and compensate for the fine initiator frequency offset, thereby further reducing any frequency offset.

[0079] In at least one embodiment, after a sequence of intervals T1 to T6 in which the initiator and reactant perform frequency estimation and compensation, the wireless communication system 100 performs packet exchange and tone exchange communications that include phase measurements at the initiator and reactant and residual frequency offset estimation and compensation by the reactant. Figure 1 and Figure 11 During interval T7, the initiator switches from the receive mode of operation to the transmit mode of operation, the reacter switches from the transmit mode of operation to the receive mode of operation, the reacter operates its receiver in LIF mode, and the initiator and reacter adjust their respective local oscillators to operate at the target frequency f iThe initiator transmits a synchronization packet including a probe sequence or other information, which is received by the reacting party during interval T8, and the reacting party performs a synchronization operation. During interval T9, the initiator switches from the transmit mode of operation to the receive mode of operation, and the reacting party switches from the receive mode of operation to the transmit mode of operation. During interval T10, the reacting party transmits a synchronization packet including a probe sequence or other information, which is received by the initiator, and the initiator performs a synchronization operation. Next, during interval T11, the wireless communication system 100 switches from the packet mode to the continuous wave mode and from the LIF mode of operation to the ZIF mode of operation, with the automatic gain control fixed to a predetermined setting.

[0080] In the interval T12, the responder sends a frequency of f i , where i is an integer, 1≤i≤I, and I≥2. As described above, the initiator receives the continuous wave and measures the phase in the ZIF mode of operation. During the interval T13, the wireless communication system 100 switches the reacting device from the transmit mode of operation to the receive mode of operation, and switches the initiator device from the receive mode of operation to the transmit mode of operation. The initiator transmits at a frequency of f i The reacting party measures the phase based on the received continuous wave and determines the residual frequency offset estimate f ESTR The wireless communication system 100 repeats the sequence performed during the interval T12 to T14 for the next value of I (i.e., the sequence is performed for no less than two different frequencies). In at least one embodiment, after taking i phase measurements, the reacting party sends the phase measurements to the initiating party to generate a distance calculation. Note that Figure 10 and Figure 11 The sequence is merely exemplary, and the techniques described herein may be applicable to other communication sequences.

[0081] Thus, techniques for measuring the distance between a first communication device and a second communication device using phase measurements have been disclosed. The description of the invention set forth herein is illustrative and is not intended to limit the scope of the invention as set forth in the following claims. For example, although the invention has been implemented in Figure 4 The phase measurement technique shown in FIG is described in detail in the embodiment of the receiver, but those skilled in the art will understand that the teachings herein can be used with other devices that implement Figure 5 or Figure 6In addition, although the present invention is described in the context of an embodiment using a wireless communication device coupled to an antenna, those skilled in the art will understand that the teachings herein can be used with communication devices coupled to other radio frequency sources (e.g., coaxial cables). Unless the context clearly indicates otherwise, the terms "first", "second", "third", etc. used in the claims are intended to distinguish different items in the claims and do not otherwise indicate or imply any time, position, or quality order. For example, "a first received network signal", "a second received network signal" does not indicate or imply that the first received network signal occurs before the second received network signal. The embodiments disclosed herein may be changed and modified based on the description set forth herein without departing from the scope of the invention as set forth in the following claims.

Claims

1. A method for measuring a distance between a first communication device including a first local oscillator and a second communication device including a second local oscillator, the method comprising: generating N phase values ​​by converting N corresponding samples of the received signal from a Cartesian representation to a polar representation comprising N phase values; unwarping the N phase values ​​by combining each of the N phase values ​​with a respective multiple of 2π to produce N unwarped phase values, where N is an integer greater than 1, each of the N phase values ​​indicating an instantaneous phase measurement of the received signal; averaging the N unwrapped phase values ​​to generate an average phase value; as well as The average phase value is warped by performing a modulo 2π operation on the average phase value to produce a final phase measurement of the first local oscillator relative to the second local oscillator.

2. The method according to claim 1, wherein The N phase values ​​are generated by calculating an inverse tangent of a quadrature component of a zero intermediate frequency (ZIF) signal based on the N corresponding samples of the received signal divided by an in-phase component of the ZIF signal.

3. The method according to claim 1, wherein Unwinding involves converting ±M i ×2×π is added to each of the N phase values, where M i is a count value corresponding to phase value i among the N phase values, where i is another integer, 1≤i≤N.

4. The method according to claim 3, further comprising: Determine the M corresponding to each phase value i in the N phase values i .

5. The method according to claim 1, wherein Warping includes subtracting ±M×2×π from the average phase value to limit the average phase value to between ±π, where M is a count value corresponding to the average phase value.

6. The method according to claim 1, further comprising: An unmodulated carrier wave is transmitted by the first communication device using the first local oscillator.

7. The method of claim 1, 2, 3, 4, 5 or 6, further comprising: Based on the final phase measurement and another phase measurement of the second local oscillator generated by the first communications device, a measurement of the distance between the first communications device and the second communications device is generated.

8. A communication system comprising: A first communication device is configured to receive a signal using a local oscillator, where the signal is generated using a second local oscillator, the first communication device comprising: a converter circuit configured to generate N phase values ​​by converting N corresponding samples of the received signal from a Cartesian representation to a polar representation comprising the N phase values; a phase unwrapping circuit configured to combine each of the N phase values ​​with a corresponding multiple of 2π to generate N unwrapped phase values, where N is an integer greater than 1; an averaging circuit configured to average the N unwrapped phase values ​​to generate an average phase value; and A phase warping circuit is configured to perform a modulo 2π operation on the average phase value to produce a final phase measurement of the second local oscillator relative to the local oscillator.

9. The communication system according to claim 8, further comprising: The second communication device is configured to generate the signal using a remote oscillator, the signal being received by the first communication device from the second communication device.

10. The communication system according to claim 8, wherein: The signal is an unmodulated carrier signal.

11. The communication system according to claim 8, wherein: The converter circuit is configured to generate the N phase values ​​based on an in-phase component of a zero intermediate frequency (ZIF) signal and a quadrature component of the ZIF signal.

12. The communication system according to claim 11, further comprising: a first integrator configured to provide N averaged in-phase samples to the converter circuit; as well as A second integrator is configured to provide N averaged quadrature samples to the converter circuit.

13. The communication system according to claim 11, wherein: The converter circuit includes a coordinate rotation digital computer (CORDIC) configured to calculate an arc tangent of a quadrature component of the ZIF signal divided by an in-phase component of the ZIF signal.

14. The communication system according to claim 8, 9, 10, 11, 12 or 13, further comprising: A processor is configured to execute instructions to generate a measurement of a distance between the first communications device and the second communications device based on the final phase measurement and another phase measurement generated by the second communications device.

15. The communication system according to claim 8, 9, 10, 11, 12 or 13, in, The phase unwrapping circuit is configured to convert ±M i ×2×π is added to each of the N phase values, where M i is a count value corresponding to phase value i among the N phase values, where i is another integer, 1≤i≤N, and The phase wrapping circuit is configured to subtract ±L×2×π from the average phase value to limit the average phase value to between ±π, where L is a count value corresponding to the average phase value.

16. A method for measuring a distance between a first communication device comprising a first local oscillator and a second communication device comprising a second local oscillator, the method comprising: averaging N in-phase components of N samples of the received signal to produce an average in-phase component, where N is an integer greater than 1; Averaging the N orthogonal components of the N samples to generate an averaged orthogonal component; generating a wrap count indicating a number of revolutions of a unit circle of the phase of the N samples; as well as generating a phase measurement of the first local oscillator relative to the second local oscillator based on the average in-phase component, the average quadrature component, and the wrap count, The wrap count is based on a subinterval of a symbol period of the received signal.

17. The method according to claim 16, wherein Generating the phase measurement includes calculating an inverse tangent of the average quadrature component divided by the average in-phase component, and generating the phase measurement based on the inverse tangent and the wrap count.

18. The method according to claim 16 or 17, further comprising: Based on the phase measurement and another phase measurement of the second local oscillator generated by the first communication device, a measurement of the distance between the first communication device and the second communication device is generated.

Images