Interleaved boost chopper circuit and high-frequency switching power supply

Abstract

The invention discloses an interleaved boost chopper circuit and a high-frequency switching power supply, the circuit is used for performing boost processing on an input voltage to output a DC output voltage, and the circuit comprises a driving module and at least one boost module connected with the driving module; when a plurality of boosting modules are arranged, all the boosting modules are connected in parallel, and the driving module is used for alternately driving the plurality of boosting modules; the boosting module comprises a low-frequency current loop and a high-frequency current loop which are mutually coupled; the low-frequency current loop is used for boosting the input voltage and outputting the low-frequency current in the interleaved boost chopper circuit from the positive end and the negative end of the input voltage; the high-frequency current loop is an internal closed loop and is used for enabling high-frequency switch alternating current generated when the driving module drives the boosting module to flow in a closed mode. And the high-frequency conducted interference in an input loop and the radiated interference to the surrounding space are greatly and obviously reduced.

Description

Technical Field

[0001] This invention relates to the field of power supply technology, and in particular to an interleaved boost chopper circuit and a high-frequency switching power supply. Background Technology

[0002] In the field of high-frequency switching power supplies, power factor correction (PFC) circuits are widely used. Their core purpose is to ensure that the input power factor and input current harmonics of the switching power supply converter meet mandatory specifications. Currently, BOOST boost topology circuits using pulse width modulation (PWM) are widely used in PFC circuit design.

[0003] However, existing BOOST boost topology PFC circuits have significant technical defects: First, the AC ripple amplitude of the input current switching frequency is large, directly causing conducted interference (CE) in electromagnetic interference (EMC) to exceed the standard; second, the large fluctuation current generated by high-frequency switching needs to be transmitted through the input line, forming a large high-frequency current loop, which in turn causes serious radiated interference (RE) problems. Under these circumstances, without the addition of an additional low-pass filter, the finished switching power supply will fail to pass the relevant safety certifications and lose its market access qualification. Therefore, if a multi-stage inductor-capacitor low-pass filter is added to the AC side of the rectifier bridge BD1 to meet the mandatory EMC conducted and radiated interference standards, the power converter efficiency will be significantly reduced due to device conduction losses and magnetic losses, while also significantly increasing hardware costs.

[0004] In summary, there is an urgent need for an interleaved boost chopper circuit that can effectively reduce electromagnetic interference, does not affect power efficiency, and does not increase costs, in order to overcome the current technological bottlenecks. Summary of the Invention

[0005] In view of the shortcomings of the prior art described above, the purpose of this application is to provide an interleaved boost chopper circuit and a high-frequency switching power supply to solve the problems of large electromagnetic interference in the current boost topology PFC circuit, the inability of the finished switching power supply to obtain various safety certifications, and the efficiency reduction and cost increase caused by adding a low-pass filter.

[0006] The first aspect of this application provides an interleaved boost chopper circuit for boosting an input voltage to output a DC output voltage, including a driving module and at least one boost module connected to the driving module; when there are multiple boost modules, all the boost modules are connected in parallel, and the driving module alternately drives the multiple boost modules. The boost module includes a low-frequency current loop and a high-frequency current loop that are coupled to each other. The low-frequency current loop is used to boost the input voltage under the drive of the drive module, and to allow the low-frequency current in the interleaved boost chopper circuit to flow into the positive terminal and output from the negative terminal of the input voltage. The high-frequency current loop is an internal closed loop, used to ensure the closed flow of high-frequency switching AC current generated when the drive module drives the boost module.

[0007] In some embodiments of this application, the boost module includes a dual-winding inductor, a power switching unit, a diode, and a filtering unit. The dual-winding inductor includes a first winding inductor and a second winding inductor, and the filtering unit includes an auxiliary inductor and a filtering capacitor. The first end of the first winding inductor is connected to the second end of the auxiliary inductor. The second end of the first winding inductor is connected to the first current-conducting terminal of the power switching unit and the anode of the diode. The first end of the auxiliary inductor is connected to the positive terminal of the input voltage. The second current-conducting terminal of the power switching unit is connected to the negative terminal of the DC output voltage. The cathode of the diode is connected to the positive terminal of the DC output voltage. The drive control terminal of the power switching unit is connected to the drive module. The first end of the second winding inductor is connected to the second end of the filter capacitor, the second end of the second winding inductor is connected to the first current conduction terminal of the power switching unit and the anode of the diode, and the first end of the filter capacitor is connected to the negative terminal of the input voltage. The first winding inductance and the second winding inductance have the same number of turns.

[0008] In some embodiments of this application, the boost module further includes a current-limiting resistor and a protection resistor. The current-limiting resistor is connected between the drive module and the drive control terminal of the power switch unit, and the protection resistor is connected between the drive control terminal of the power switch unit and the negative terminal of the DC output voltage.

[0009] In some embodiments of this application, the interleaved boost chopper circuit further includes a first capacitor connected between the positive and negative terminals of the input voltage and a second capacitor connected between the positive and negative terminals of the DC output voltage.

[0010] In some embodiments of this application, the interleaved boost chopper circuit further includes a current sampling resistor connected between the negative terminal of the input voltage and the second current-on terminal of the power switching unit.

[0011] In some embodiments of this application, the interleaved boost chopper circuit further includes a full-bridge rectifier circuit, wherein the positive output terminal of the full-bridge rectifier circuit is connected to the positive terminal of the input voltage, and the negative output terminal of the full-bridge rectifier circuit is connected to the negative terminal of the input voltage.

[0012] In some embodiments of this application, the driving module controls the power switching units in the multiple boost modules to alternately conduct by outputting switching signals with different phases and the same pulse width to the driving control terminals of the power switching units in the multiple boost modules respectively. The sum of the phases of all the switching signals is 360 degrees.

[0013] In some embodiments of this application, the interleaved boost chopper circuit includes two boost modules. The driving module controls the alternating conduction of the power switching units in the two boost modules by outputting switching signals with 180-degree phase and the same pulse width that are interleaved to the driving control terminals of the power switching units in the two boost modules.

[0014] A second aspect of this application provides a high-frequency switching power supply, including multiple series-connected interleaved boost chopper circuits, wherein the interleaved boost chopper circuit is the interleaved boost chopper circuit.

[0015] A third aspect of this application provides a high-frequency switching power supply, including multiple parallel-connected interleaved boost chopper circuits, wherein the interleaved boost chopper circuit is the interleaved boost chopper circuit.

[0016] Compared with the prior art, one or more embodiments of the above solutions may have the following advantages or beneficial effects: The interleaved boost chopper circuit provided in this invention, by setting separate low-frequency and high-frequency current loops, allows most of the low-frequency current in the circuit to flow in and out through the low-frequency current loop, thus locking most of the high-frequency switching current in the high-frequency closed loop. This significantly reduces high-frequency conducted interference in the input loop and radiated interference to the surrounding space. Using interleaved phase drive signals to alternately drive multiple boost modules greatly reduces the peak current of the power switching unit, further reducing the amplitude and energy of conducted and radiated interference. When applied to AC / DC switching power supplies, it effectively reduces the number and stages of electromagnetic compatibility filters at the neutral and live wire AC input front-ends, significantly lowering the cost of the switching power supply. Without increasing cost, the overall size is reduced simply by increasing the frequency of the drive signal, while still meeting safety certification electromagnetic compatibility specifications, improving economic applicability. Furthermore, it can generate significant economic and social benefits and has the potential for large-scale application.

[0017] Other features and advantages of the invention will be set forth in the description which follows, and will be apparent in part from the description, or may be learned by practicing the invention. The objects and other advantages of the invention may be realized and obtained by means of the structures particularly pointed out in the description, claims, and drawings. Attached Figure Description

[0018] The accompanying drawings are provided to further illustrate the invention and form part of the specification. They are used in conjunction with the embodiments of the invention to explain the invention and do not constitute a limitation thereof. In the drawings: Figure 1 The diagram shown is a schematic diagram of the interleaved boost chopper circuit described in an embodiment of this application.

[0019] Figure 2 The diagram shown is a first specific structural schematic of the interleaved boost chopper circuit described in an embodiment of this application.

[0020] Figure 3 The diagram shown is a second specific structural schematic of the interleaved boost chopper circuit described in the embodiments of this application.

[0021] Figure 4 The diagram shown is a third specific structural schematic of the interleaved boost chopper circuit described in the embodiments of this application.

[0022] Figure 5 The diagram shows the waveform timing between the poles of the power switching unit when two switching signals are output to two boost modules in the interleaved boost chopper circuit described in this application embodiment.

[0023] Figure 6 The diagram shows a comparison of the high-frequency current waveforms of the main components in a high-frequency period of the low-frequency current waveform of the interleaved boost chopper circuit described in this application embodiment.

[0024] Figure 7 The diagram shows a comparison of the high-frequency voltage waveforms of the main components in a short high-frequency period of the low-frequency current waveform of the interleaved boost chopper circuit described in this application embodiment.

[0025] Figure 8 The diagram shows a comparison of the high-frequency current waveforms of the main components in the interleaved boost chopper circuit described in this application embodiment during a long low-frequency period of the low-frequency current waveform.

[0026] Figure 9 The diagram shown is a structural schematic of a high-frequency switching power supply according to an embodiment of this application.

[0027] Figure 10 The diagram shown is another structural schematic of the high-frequency switching power supply described in the embodiments of this application.

[0028] Specific element symbol explanation: 1-Interleaved boost chopper circuit, 10-Drive module, 11-Boost module, 2-High frequency switching power supply. Detailed Implementation

[0029] To make the technical problems, technical solutions, and beneficial effects to be solved by this application clearer, the following detailed description is provided in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative and are not intended to limit the scope of this application.

[0030] It should be noted that when a component is referred to as being "set on" another component, it can be directly on or indirectly on that other component. When a component is referred to as being "connected to" another component, it can be directly connected to or indirectly connected to that other component.

[0031] Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of technical features indicated. Thus, a feature defined as "first" or "second" may explicitly or implicitly include one or more of that feature. In the description of this application, "multiple" means two or more, unless otherwise explicitly specified.

[0032] Existing BOOST power factor correction (PFC) circuits suffer from significant electromagnetic compatibility (EMC) defects and related efficiency and cost issues. Specifically, the high-frequency fluctuations in the switching frequency current of the dual-winding inductor, combined with the low-frequency current from the mains power supply, are directly conducted to the input L and N lines through the power lines. This causes the conducted interference (CE) and radiated interference (RE) generated by the high-frequency components of the switching frequency current flowing through the L and N lines to exceed the limits of mandatory EMC standards such as EN55022 CLASS B. Therefore, without the addition of a low-pass filter, the finished switching power supply cannot pass safety certification and loses its market access qualification.

[0033] Currently, most power supply products limit the operating frequency of the BOOST circuit to, for example, below 75kHz to pass the aforementioned mandatory EMC tests. This ensures that the harmonic interference of the switching frequency remains below the minimum limit for conducted interference testing, such as 150kHz, thus reducing the difficulty of passing the tests. However, the upper limit of the frequency prevents the inductor from being further reduced in size by increasing the frequency, making it difficult to reduce the size of the power converter and significantly lower the cost. At the same time, to meet the mandatory EMC conducted and radiated standards, a multi-stage inductor-capacitor low-pass filter module needs to be added to the AC side of the rectifier bridge BD1. The conduction loss and magnetic loss of this module significantly reduce the power supply efficiency and greatly increase the hardware cost, making it difficult to improve the efficiency of existing power converters and further reduce costs. Therefore, the current BOOST boost circuit front-end all rely on multi-stage low-pass filters to meet EMC requirements, inherently resulting in low efficiency and high cost.

[0034] Based on this, this application innovates the interleaved boost chopper circuit and high-frequency switching power supply in related technologies.

[0035] refer to Figure 1 As shown, the interleaved boost chopper circuit 1 in this embodiment is mainly used to boost the input voltage to output a DC output voltage. Typically, the input voltage of the interleaved boost chopper circuit 1 is a DC input voltage. If applied between the live wire and the neutral wire, the interleaved boost chopper circuit 1 may also include a rectifier section to convert the AC power to DC power before boosting it.

[0036] The interleaved boost chopper circuit 1 in this embodiment specifically includes a driver module 10 and at least one boost module 11 connected to the driver module 10. When the interleaved boost chopper circuit 1 includes only one boost module 11, the driver module 10 can boost the input voltage to obtain a DC output voltage by outputting a pulse switching signal of arbitrary phase and pulse width to the power switching unit in the boost module 11. However, this method has problems such as large ripple and poor dynamic response. The driver module 10 can be a driver chip.

[0037] When the interleaved boost chopper circuit 1 includes multiple boost modules 11, all boost modules 11 are connected between the positive terminal BD+ and the negative terminal BD- of the input voltage, and all boost modules 11 are connected in parallel. At this time, the drive module 10 can alternately drive the multiple boost modules 11 through multiple switching signals with different phases but the same pulse width. This method can effectively reduce the high-frequency voltage ripple and high-frequency current peak of the input voltage.

[0038] Specifically, the boost module 11 includes a low-frequency current loop and a high-frequency current loop coupled to each other. The low-frequency current loop is used to boost the input voltage under the drive of the drive module, and directs the low-frequency current in the circuit from the positive terminal of the input voltage to the negative terminal. The high-frequency current loop is designed as an internal closed loop to control the closed-loop flow of the high-frequency switching AC current generated when the drive module 10 drives the boost module 11. The boost module 11 effectively separates the paths of the high-frequency switching AC current and the low-frequency current of the input voltage, allowing the low-frequency current to flow to the positive and negative terminals of the input voltage, while confining the high-frequency switching AC current within a limited internal closed loop.

[0039] In one embodiment, the sum of the phases of multiple switching signals can be set to 360 degrees. This setting can further reduce the output voltage ripple and increase the frequency, while also improving voltage stability and regulation accuracy.

[0040] This embodiment separates the low-frequency and high-frequency current loops, ensuring that most of the low-frequency current is output through the low-frequency loop, while the high-frequency switching current is locked within the high-frequency closed loop, significantly reducing high-frequency conducted interference and spatial radiated interference in the input loop. Simultaneously, interleaved phase drive signals are used to alternately drive multiple boost modules 11, significantly reducing the peak current of the power switching unit and further weakening the amplitude and energy of conducted and radiated interference. When applied to AC / DC switching power supplies, this effectively reduces the number and stages of EMC filters at the neutral and live wire input front-ends, significantly lowering power supply costs.

[0041] To facilitate a clearer explanation of the interleaved boost chopper circuit 1, the following embodiments will use a first boost module 11-1 (the overall structure is not shown in the figure) and a second boost module 11-2 (the overall structure is not shown in the figure) as examples. The driving module 10 controls the power switching units in the two boost modules 11 to alternately conduct by outputting switching signals with 180-degree phase shifts and the same pulse width to the drive control terminals G of the power switching units in the two boost modules 11. When the interleaved boost chopper circuit 1 includes other numbers of boost modules 11, its operation is the same as that of the circuit with two boost modules 11, and therefore will not be listed individually.

[0042] refer to Figure 2As shown, in one embodiment, the first boost module 11 in the interleaved boost chopper circuit 1 specifically includes a dual-winding inductor L1 (e.g., a dual-winding differential-mode inductor), a power switch unit Q1, a diode D1, and a filter unit. The dual-winding inductor L1 includes a first winding inductor L1-N1 and a second winding inductor L1-N2. The filter unit includes an auxiliary inductor L3 and a filter capacitor C3. The first end of the first winding inductor L1-N1 is connected to the second end of the auxiliary inductor L3. The second end of the first winding inductor L1-N1 is connected to the first current-conducting terminal of the power switch unit Q1 and the anode of the diode D1, respectively. The first end of the auxiliary inductor L3 is connected to the positive terminal BD+ of the input voltage. The cathode of the diode D1 is connected to the positive terminal VO of the DC output voltage. The second current-conducting terminal of the power switch unit Q1 is connected to the negative terminal VO-GND of the DC output voltage. The drive control terminal G of the power switch unit Q1 is connected to the drive module 10. The first end of the second winding inductor L1-N2 is connected to the second end of the filter capacitor C3. The second end of the second winding inductor L1-N2 is connected to the first current conduction terminal of the power switch unit Q1 and the anode of the diode D1, respectively. The first end of the filter capacitor C3 is connected to the negative terminal BD- of the input voltage and the negative terminal VO-GND of the DC output voltage.

[0043] In the circuit described above, the second terminals of the first winding inductor L1-N1 and the second winding inductor L1-N2, the first current-conducting terminal of the power switch unit Q1, and the anode of diode D1 are all connected together. The cathode of diode D1 is connected to the positive terminal VO of the DC output voltage. Furthermore, the number of turns of the first winding inductor L1-N1 and the second winding inductor L1-N2 are set to be equal.

[0044] refer to Figure 3 As shown, in one embodiment, the boost module 11 further includes a current-limiting resistor R1 and a protection resistor R2. The current-limiting resistor R1 is connected between the drive module 10 and the drive control terminal G of the power switch unit Q1, and the protection resistor R2 is connected between the drive control terminal G of the power switch unit Q1 and the negative terminal VO-GND of the DC output voltage, so as to limit the current of the circuit input to the unit drive control terminal G of the power switch Q1 and protect the power switch unit Q1.

[0045] The structure of the second boost module 11 in the interleaved boost chopper circuit 1 is the same as that of the first boost module 11. The dual-winding inductor is L2, the first winding inductor of the dual-winding inductor L2 is L2-N1, the second winding inductor is L2-N2, the auxiliary inductor is L4, the filter capacitor is C4, the current-limiting resistor is R3, the protection resistor is R4, and the diode is D2. The specific connection method will not be described in detail here.

[0046] Based on the above structure, the interleaved boost chopper circuit 1 in this embodiment also includes a first capacitor C1 and a second capacitor C2. The first capacitor C1 is connected between the positive terminal BD+ and the negative terminal BD- of the input voltage to filter the input voltage. The second capacitor C2 is connected between the positive terminal VO and the negative terminal VO-GND of the DC output voltage to filter the DC output voltage.

[0047] In one embodiment, the interleaved boost chopper circuit 1 further includes a current sampling resistor RS1, which is connected between the negative terminal BD- of the input voltage and the second current-on terminal S of the power switch unit Q1 of the second boost module 11. In one embodiment, the power switch unit is an NMOS transistor; however, other types of power switching devices are not fixed here.

[0048] refer to Figure 4 As shown, in one embodiment, when the interleaved boost chopper circuit 1 further includes a rectification section, the rectification section is a full-bridge rectifier circuit BD1. The positive output terminal of the full-bridge rectifier circuit BD1 is connected to the positive terminal BD+ of the input voltage, and the negative output terminal of the full-bridge rectifier circuit BD1 is connected to the negative terminal BD- of the input voltage.

[0049] Figure 5 This is a schematic diagram showing the voltage waveform timing between the gate and source, and between the source and drain, in the power switching unit of the interleaved boost chopper circuit 1 of this application, when the drive module 10 outputs switching signals to the power switching units of the two boost modules 11. (Reference) Figure 5 and Figure 3 As shown, based on the above structure, the specific working principle of the interleaved boost chopper circuit 1 of this application is as follows: each boost module 11 has two working modes, which work in a cyclic manner. The first boost module 11-1 is used as an example for explanation.

[0050] When the driving voltage (i.e., the gate and source voltage) of power switch unit Q1 changes from low level to high level, power switch unit Q1 changes from the off state to the on state. At this time, power switch unit Q1 is in the high-frequency operating switching cycle T. The time period is set to the 1-1 mode.

[0051] Since the number of turns of the first winding inductor L1-N1 and the number of turns of the second winding inductor L1-N2 of the dual-winding inductor L1 are equal, that is... Furthermore, the leakage inductance between the first winding inductor L1-N1 and the second winding inductor L1-N2 is usually negligible compared to the inductance of the dual-winding inductor L1. Therefore, the voltage across the first winding inductor L1-N1 and the voltage across the second winding inductor L1-N1 are approximately equal. The inductance of the auxiliary inductor L3 is also very small compared to that of the dual-winding inductor L1. Because the impedance between the drain (D) and source (S) of the power switch unit Q1 is extremely low when it is turned on, i.e. Therefore, when the power switch unit Q1 is turned on, the voltage drop between its drain D and source S can be ignored, i.e. Because the resistance of the current sampling resistor RS1 is extremely small, that is... Therefore, the voltage across the current sampling resistor RS1 can be ignored. .

[0052] The positive direction of the induced voltage of the first winding inductor L1-N1 of the dual-winding inductor L1 is defined as follows: when the end of the first winding inductor L1-N1 connected to the drain D of the power switching unit Q1 induces a negative voltage, and the end of the first winding inductor L1-N1 connected to the auxiliary inductor L3 induces a positive voltage, this is defined as the positive voltage direction of the first winding inductor L1-N1. In operating mode 1-1, the induced voltage of the first winding inductor L1-N1 is a positive voltage: the end of the first winding inductor L1-N1 connected to the auxiliary inductor L3 induces a positive voltage, and the end of the first winding inductor L1-N1 connected to the drain D of the power switching unit Q1 induces a negative voltage.

[0053] Since the common connection point of the first winding inductor L1-N1 and the second winding inductor L1-N2 with the drain D of Q1 is the same as the induced voltage terminal, the positive direction of the induced voltage of the second winding inductor L1-N2 is defined as follows: when the end of the second winding inductor L1-N2 connected to the drain D of the power switch unit Q1 induces a negative voltage, and the end of the second winding inductor L1-N2 connected to the filter capacitor C3 induces a positive voltage, this is defined as the positive voltage direction of the second winding inductor L1-N2. In operating mode 1-1, the induced voltage of the second winding inductor L1-N2 is a positive voltage: the end of the second winding inductor L1-N2 connected to the filter capacitor C3 induces a positive voltage, and the end of the second winding inductor L1-N2 connected to the drain D of the power switch unit Q1 induces a negative voltage.

[0054] With the negative terminal BD- of the input voltage as the reference zero potential, the following exists in this mode: (1) Under operating mode 1-1, due to the instantaneous value of the induced voltage across the first winding inductor L1-N1 Instantaneous value of induced voltage across the second winding inductor L1-N2 They are approximately equal, and their value is approximately equal to the instantaneous value of the voltage across the filter capacitor C3. At the same time, the instantaneous voltage across the first capacitor C1... Instantaneous voltage across auxiliary inductor L3 The difference between , and They are all approximately equal, and the instantaneous voltage values ​​are... , and All are AC voltages. In operating mode 1-1, both negative and positive voltage values ​​exist. and It maintains a positive voltage value under operating mode 1-1.

[0055] Therefore, a quasi-resonance based on the high-frequency switching period T will occur between the auxiliary inductor L3 and the filter capacitor C3, thereby affecting the voltage value. An oscillating voltage based on the high-frequency switching period T is superimposed on the low-frequency voltage value, and the voltage value is... This becomes an AC oscillating voltage based on the high-frequency operating switching period T. This is due to the voltage across the first winding inductor L1-N1 and the second winding inductor L1-N2. and Always equal to voltage value Furthermore, the dual-winding inductor L1 does not participate in the quasi-resonant process of the aforementioned high-frequency operating switching period T.

[0056] In operating mode 1-1, the low-frequency current loop includes: the positive terminal BD+ of the input voltage, the first capacitor C1, the auxiliary inductor L3, the first winding inductance L1-N1, the drain D and source S of the power switching unit Q1, the current sampling resistor RS1, and the negative terminal BD- of the input voltage. The high-frequency current loop includes: the filter capacitor C3, the second winding inductance L1-N2, the drain D and source S of the power switching unit Q1, and the current sampling resistor RS1.

[0057] When the auxiliary inductor L3 in the low-frequency current loop is set to have a high high-frequency impedance At this time, the high-frequency current loop has a very low (approximately zero ohms) high-frequency impedance. In this case, the low-frequency current loop is mainly used to pass low-frequency current, while the high-frequency current loop is mainly used to pass high-frequency current.

[0058] When the interleaved boost chopper circuit 1 of this application is used for the power factor correction (PFC) function of a high-frequency switching power supply, the low-frequency current is mainly the AC power frequency input current from the mains grid (approximately 50-60Hz), while the high-frequency current is mainly the current from the power switching unit Q1 operating at its operating frequency. When switching is performed, a high-frequency sawtooth wave current is generated in the dual-winding inductor L1.

[0059] In operating mode 1-1, the induced voltage direction of the first winding inductor L1-N1 is as follows: the end of the first winding inductor L1-N1 connected to the auxiliary inductor L3 (opposite-named terminal) is positive, and the end of the first winding inductor L1-N1 connected to the drain D of the power switching unit Q1 (same-named terminal) is negative. At the same time, the induced voltage direction of the second winding inductor L1-N2 is as follows: the end of the second winding inductor L1-N2 connected to the filter capacitor C3 (opposite-named terminal) is positive, and the end of the second winding inductor L1-N2 connected to the drain D of the power switching unit Q1 (same-named terminal) is negative.

[0060] At this time, the current flowing through each component has the following relationship: (2) From equation (2), it can be seen that the voltage across the filter capacitor C3 is... The driving voltage of power switch unit Q1 is approximately a fixed value within a high-frequency operating cycle T. Opening time The time period T is a fixed value within a high-frequency operating cycle. Therefore, under this operating mode 1-1, the current in the second winding inductor L1-N2 in the high-frequency current loop is... rate of change and the charging and discharging current of auxiliary capacitor C3 rate of change A fixed rate of change that is approximately positive over a high-frequency operating cycle T (current direction defined as: when the current...). The current flows out of the positive terminal of the filter capacitor C3 (causing C3 to discharge) in the positive direction; when the current flows out... When the current flows into the positive terminal of the filter capacitor C3 (charging C3), the current direction is negative. and From the minimum current value (negative current value, The current (charged by the filter capacitor C3) increases linearly with a fixed positive slope to the maximum current value (positive current value). (Discharge the filter capacitor C3). In this operating mode 1-1, the dual-winding inductor L1 begins to store energy.

[0061] Simultaneously, the drain (D) and source (S) currents of the power switching unit Q1 are... The current flowing through the first winding inductor L1-N1 The current in the second winding inductor L1-N2 The sum of. Because of the current. It is mainly composed of direct current, while the current is... In this operating mode 1-1, the main wave is the rising segment of an AC sawtooth wave with a linearly changing positive slope, so the current... When the current in the dual-winding inductor L1 is in continuous state, it is a trapezoidal wave (during the first boost module 11-1 operating mode 1-2 period). (Zero amperes).

[0062] In this operating mode 1-1, the aforementioned low-frequency current loop and high-frequency current loop are parallel circuits passing through the first winding inductor L1-N1 and the second winding inductor L1-N2 of the dual-winding inductor L1, respectively, and the high-frequency impedance of the auxiliary inductor L3 in the low-frequency current loop is... The impedance is much larger than that of the leakage inductance in a high-frequency current loop. Therefore, when the power switch unit Q1 switches from the off state to the on state, most of the high-frequency current fluctuation component generated in the dual-winding inductor L1 will flow through the second winding inductor L1-N2, causing the current in the second winding inductor L1-N2 to... and the current of filter capacitor C3 This generates large-amplitude high-frequency fluctuations, and the current flowing through the first winding inductor L1-N1... and the current of auxiliary inductor L3 The mid-to-high frequency fluctuation component will be significantly reduced.

[0063] As can be seen from the above, in operating mode 1-1, the low-frequency current loop of the boost module 11 in the interleaved boost chopper circuit 1 of this application flows to the external power supply line through the positive terminal BD+ of the input voltage. and The main current is low-frequency, while the high-frequency current loop is an internally closed circuit, and its high-frequency current... and It does not transmit to external circuits. Therefore, this circuit significantly reduces the amplitude of high-frequency conducted interference (EMI) in the lines flowing to the positive terminal BD+ and the negative terminal BD- of the input voltage.

[0064] refer to Figure 5 and Figure 3 As shown, the driving voltage U(GS-Q1) of power switch unit Q1 (where the gate (G) of power switch unit Q1, VG-Q1, is relative to its source (VS-Q1, reference zero potential)) changes from high to low, and power switch unit Q1 changes from the on state to the off state. At this time, power switch unit Q1 is in the high-frequency operating switching cycle T. The time period is set to work mode 1-2.

[0065] Because the impedance between the drain D and source S of the power switch unit Q1 is extremely high when it is turned off, i.e. Therefore, when the power switch unit Q1 is turned off, the leakage current between its drain D and source S can be ignored. In operating mode 1-2, the energy stored in the dual-winding inductor L1 begins to be released. The induced voltage of its first winding inductor L1-N1 is a negative voltage: the end where the first winding inductor L1-N1, the second winding inductor L1-N2, the drain D of the power switch unit Q1, and the anode of the diode D1 are connected together is the induced positive voltage, and the end where the first winding inductor L1-N1 is connected to the auxiliary inductor L3 is the induced negative voltage.

[0066] At this time, the induced voltage of the second winding inductor L1-N2 is a negative voltage: the end of the second winding inductor L1-N2 connected to the filter capacitor C3 has an induced negative voltage, and the end of the second winding inductor L1-N2 connected to the drain D of the power switch unit Q1 has an induced positive voltage. Taking the negative terminal BD- of the input voltage as the reference zero potential, we know that: (3) As can be seen from equation (3), in this operating mode 1-2, due to the instantaneous value of the induced voltage across the inductor L1-N1 of the first winding... Instantaneous value of induced voltage across the second winding inductor L1-N2 The voltage across the first capacitor C1 is approximately equal, therefore the instantaneous value of the voltage across the first capacitor C1 is approximately equal. Instantaneous value of voltage across auxiliary inductor L3 The difference is related to the instantaneous voltage value across the filter capacitor C3. They are approximately equal. At the same time, it can be determined that the instantaneous voltage across the second capacitor C2 is... With instantaneous voltage value (or The sum of () and the instantaneous value of the voltage. They are approximately equal. The instantaneous voltage value is... , and AC voltage, In this operating mode 1-2, the voltage is positive. and The voltage is negative in this operating mode 1-2.

[0067] And due to voltage value With voltage value The voltage value is approximately fixed within one high-frequency switching cycle T. Therefore, the auxiliary inductor L3 and the filter capacitor C3 generate a quasi-resonance based on the high-frequency operating switching cycle T, thereby causing the voltage value to... An oscillating voltage based on the high-frequency switching period T is superimposed on the low-frequency voltage value, and the voltage value is... This becomes the AC oscillation voltage based on the high-frequency switching period T. As can be seen above, the voltage across the first winding inductor L1-N1... Voltage across the second winding inductor L1-N2 Approximately equal to voltage value With voltage value The difference, therefore the voltage and The voltage value is approximately fixed within a high-frequency switching cycle T, so the dual-winding inductor L1 does not participate in the quasi-resonant process of the aforementioned high-frequency operating switching cycle T.

[0068] refer to Figure 3 As shown, in this operating mode 1-2, there are two current loops. The low-frequency current loop includes: the positive terminal BD+ of the input voltage, the first capacitor C1, the auxiliary inductor L3, the first winding inductor L1-N1, the diode D1, the second capacitor C2, the current sampling resistor RS1, and the negative terminal BD- of the input voltage. The high-frequency current loop includes: the filter capacitor C3, the second winding inductor L1-N2, the diode D1, and the current sampling resistor RS1.

[0069] As mentioned above, the auxiliary inductor L3 has a high high-frequency impedance in the low-frequency current loop. The high-frequency current loop has a very low (approximately zero ohms) high-frequency impedance compared to the low-frequency current loop. Therefore, the low-frequency current loop is mainly used to carry low-frequency current, while the high-frequency current loop is mainly used to carry high-frequency current. When the interleaved boost chopper circuit 1 of this application is used for the power factor correction (PFC) function of a high-frequency switching power supply, the low-frequency current is mainly the AC power frequency input current of the mains grid (approximately 50-60Hz), while the high-frequency current is mainly the current from the power switching unit Q1 at its operating frequency. When switching is performed, a high-frequency sawtooth wave current is generated in the dual-winding inductor L1.

[0070] In operating mode 1-2, the induced voltage direction of the first winding inductor L1-N1 is as follows: the end of the first winding inductor L1-N1 connected to the auxiliary inductor L3 (opposite-named terminal) is negative, and the end of the first winding inductor L1-N1 connected to the drain D of the power switching unit Q1 (same-named terminal) is positive. At the same time, the induced voltage direction of the second winding inductor L1-N2 is as follows: the end of the second winding inductor L1-N2 connected to the filter capacitor C3 (opposite-named terminal) is negative, and the end of the second winding inductor L1-N2 connected to the drain D of the power switching unit Q1 (same-named terminal) is positive.

[0071] The currents flowing through each component in modes 1-2 have the following relationship: (4) From equation (4), it can be seen that the voltage across the filter capacitor C3 is... The driving voltage of power switch unit Q1 is approximately a fixed value within a high-frequency operating cycle T. shutdown time The inductor current of the second winding in the high-frequency current loop is a fixed value within a high-frequency operating cycle T. rate of change and the charging and discharging current of filter capacitor C3 rate of change A fixed rate of change that is approximately negative over a high-frequency operating cycle T (current direction defined as: when the current...). The current flows out of the positive terminal of the filter capacitor C3 (causing C3 to discharge) in the positive direction; when the current flows out... When the current flows into the positive terminal of the filter capacitor C3 (charging C3), the current direction is negative. and From the maximum current value (positive current value, Discharging the filter capacitor C3) linearly reduces the current value to the minimum value (negative current value) with a fixed negative slope. (Charging the filter capacitor C3). In this operating mode 1-2, the energy stored in the dual-winding inductor L1 begins to be released. Simultaneously, based on equation (4), it can also be seen that the drain D and source S currents of the power switching unit Q1... It is zero amperes.

[0072] As can be seen from the above, under operating modes 1-2, the aforementioned low-frequency current loop and high-frequency current loop are parallel circuits passing through the first winding inductor L1-N1 and the second winding inductor L1-N2 of the dual-winding inductor L1, respectively, and the high-frequency impedance of the auxiliary inductor L3 in the low-frequency current loop is... The impedance is much larger than that of the leakage inductance in a high-frequency current loop. Therefore, when the power switch unit Q1 changes from the on state to the off state, most of the high-frequency current fluctuation component generated in the dual-winding inductor L1 will flow through the second winding inductor L1-N2, causing the current in the second winding inductor L1-N2 to... and the current of filter capacitor C3 This generates large-amplitude high-frequency fluctuations, and the current flowing through the first winding inductor L1-N1... and the current of auxiliary inductor L3 The mid-to-high frequency fluctuation component will be significantly reduced.

[0073] In summary, under operating modes 1-2, the low-frequency current loop in the first boost module 11-1 of the interleaved boost chopper circuit 1 of this application flows to the external power supply line through the positive terminal BD+ of the input voltage. and The main current is low-frequency, while the high-frequency current loop is an internally closed circuit, and its high-frequency current... and It does not transmit to external circuits. Therefore, the interleaved boost chopper circuit 1 of this application significantly reduces the amplitude of high-frequency conducted interference (EMI) in the lines flowing to the positive terminal BD+ and the negative terminal BD- of the input voltage.

[0074] The second boost module 11-2 of the interleaved boost chopper circuit 1 in this application also includes two operating modes. In this mode, the positive terminal VG-Q2 of the driving voltage U(GS-Q2) of the power switch unit Q2 changes from a low level to a high level relative to its negative terminal VS-Q2 (reference zero potential), and the power switch unit Q2 changes from an off state to an on state. This is during the high-frequency operating switching period T of the power switch unit Q2. Time period; this operating mode is set as operating mode 2-1. Since the phase difference between the driving voltages of the first boost module 11-1 and the second boost module 11-2 is 180°, and other parameters (amplitude and frequency, etc.) are exactly the same, and the first and second boost modules operate independently, in parallel, and alternately, the operating process and principle of the second boost module 11-2 in operating mode 2-1 can be referred to as the second boost module 11 in operating mode 1-1, and will not be repeated here.

[0075] Simultaneously, the positive terminal VG-Q2 of the driving voltage U(GS-Q2) of power switch unit Q2 changes from high to low relative to its negative terminal VS-Q2 (reference zero potential), and power switch unit Q2 changes from the on state to the off state, which is the high-frequency operating switching period T of power switch unit Q2. Time period; set this working mode as working mode 2-2. As mentioned above, the working process and principle of the second boost module 11-2 working mode 2-2 can be referred to the first boost module 11-1 working mode 1-2, and will not be repeated here.

[0076] In the interleaved boost chopper circuit 1 of this application, the driving voltages U(GS-Q1) and U(GS-Q2) of the two sets of power switching units Q1 and Q2 have a phase difference of 180°, which makes the operating current and voltage waveforms of the power switching units Q1 and Q2 exhibit a periodic alternating high and low complementary working state. At the same time, it greatly reduces the peak value of the high-frequency current, greatly reduces the high-frequency radiation and conducted interference generated by the high-frequency switching circuit, and finally greatly reduces the high-frequency voltage ripple across the second capacitor C2.

[0077] In the interleaved boost chopper circuit 1 of this application, most of the high-frequency current flows in each operating mode ( and The high-frequency current loops are all closed loops within the circuit, and their high-frequency current does not flow to external circuits. Meanwhile, in the high-frequency range, there are high-impedance low-frequency current loops that flow through the positive terminal BD+ of the input voltage to the external power supply line. and The main operating current is low frequency. Therefore, the interleaved boost chopper circuit 1 of this application significantly reduces the amplitude of high-frequency conducted interference CE and high-frequency radiated interference RE in the lines flowing to the positive terminal BD+ and the negative terminal BD- of the input voltage from multiple aspects.

[0078] To provide a more detailed explanation of the interleaved boost chopper circuit 1 of this application, it is applied to form an AC input voltage and current power factor correction (PFC) boost converter. For example, the following component specifications are selected: Input AC voltage VIN = 110V (AC). Output DC voltage U(VO,VO-GND)=400V (DC). The output DC current I(VO) = ​​1.03A (DC). High-frequency switching transistors Q1 and Q2 are model SPP20N60C3. Diodes D1 and D2 are model number MUR860. The capacitance of capacitor C1 is: C1 = 0.01uF. The capacitances of capacitors C3 and C4 are: C3 = C4 = 2.2uF. The inductance values ​​of auxiliary inductors L3 and L4 are: L3 = L4 = 5uH. The inductance of coils N1 and N2 of the main inductor L1: L(L1-N1) = L(L1-N2) = 200uH. The capacitance of output capacitor C2 is: C2 = 330uF. The turns ratio of coil N1 to coil N2 in the main inductor L1 is n = N1 / N2 = 1. The frequencies of the high-frequency drive voltage signals U(GS-Q1) and U(GS-Q2) are: f = 200 kHz. The control circuit 10 uses existing technology control ICs and common existing technology control circuits: UCC28070.

[0079] When the interleaved boost chopper circuit 1 of this application operates in a state where the currents of the dual-winding inductor L1 and the dual-winding inductor L2 are continuous, a high-frequency period of the low-frequency current waveform is selected, and the comparison diagram of the high-frequency current waveforms of its main components is shown in the figure below. Figure 6 As shown, Figure 6The high-frequency current waveform flowing through the first winding inductor L1-N1 of the dual-winding inductor L1 is IL1-N1; the high-frequency current waveform flowing through the second winding inductor L1-N2 of the dual-winding inductor L1 is IL1-N2; the high-frequency current waveform flowing through the first winding inductor L2-N1 of the dual-winding inductor L2 is IL2-N1; the high-frequency current waveform flowing through the second winding inductor L2-N2 of the dual-winding inductor L2 is IL2-N2; the high-frequency current waveform between the drain D and source S of the power switching unit Q1 is I-Q1; the high-frequency current waveform between the drain D and source S of the power switching unit Q2 is I-Q2; the forward high-frequency current waveform flowing through diode D1 is I-D1; the forward high-frequency current waveform flowing through diode D2 is I-D2; and the high-frequency current waveform flowing through the current sampling resistor RS1 is I-RS1.

[0080] Depend on Figure 6 It can be seen that when the interleaved boost chopper circuit 1 of this application is in operating mode 1-1, that is, during the high-frequency operating switching period T of the power switching unit Q1... During the time period, the current flowing through the components of the first boost module 11-1 is as follows: the high-frequency current waveform I-Q1 between the drain D and source S of the power switch unit Q1 is a trapezoidal wave, and the forward high-frequency current waveform I-D1 of the diode D1 is an inverted trapezoidal wave. Both are composite waveforms of DC current components and AC current components.

[0081] During operating mode 1-1, the current flowing through the components of the first boost module 11-1 is as follows: the high-frequency short-term current waveform IL1-N1 of the first winding inductor L1-N1 of the dual-winding inductor L1 is mainly composed of DC current (approximately 2.75A), with a secondary component being a quasi-resonant AC current waveform (approximately ±0.05A). The high-frequency short-term current waveform IL1-N2 of the second winding inductor L1-N2 of the dual-winding inductor L1 is an AC triangular wave current (approximately ±1.15A). It can be seen that in the above example, during operating mode 1-1, the high-frequency AC current flowing through the second winding inductor L1-N2 of the dual-winding inductor L1 is more than twenty times that flowing through the first winding inductor L1-N1. Therefore, the high-frequency AC current is essentially confined within the aforementioned high-frequency current loop, significantly and effectively reducing the conducted interference (EMI) of the high-frequency circuit to the input line, and significantly reducing the high-frequency current loop area, thereby also significantly and effectively reducing radiated interference (EMI).

[0082] At the same time Figure 6It is also known that the current flowing through the components of the second boost module 11-2: the high-frequency current waveform IL2-N1 of the first winding inductor L2-N1 of the dual-winding inductor L2, the high-frequency current waveform IL2-N2 of the second winding inductor L2-N2 of the dual-winding inductor L2, the high-frequency current waveform I-Q2 between the drain D and source S of the power switch unit Q2, and the forward high-frequency current waveform I-D2 of the diode D2, are 180° out of phase with the current waveforms of the corresponding components of the first boost module 11-1, and the waveforms, amplitudes, and frequency parameters of the high-frequency currents of the corresponding components in the two circuits are basically the same.

[0083] The common component current flowing through the alternating boost of the first boost module 11-1 and the second boost module 11-2 is: the high-frequency AC component amplitude in the waveform of the current sampling resistor RS1, I-RS1, is approximately ±0.58A. The AC component amplitude in this alternating current is significantly reduced to about half the amplitude of the high-frequency AC currents IL1-N2 and IL2-N2 in the two circuits mentioned above. Therefore, it can be seen that the current I-RS1 in this application is about one-quarter of the high-frequency AC current amplitude of a single-channel boost circuit in the prior art with the same voltage and power parameters. Due to the significant reduction in the peak value of the high-frequency AC current, and the fact that most of the high-frequency current flows in a closed loop, the high-frequency conducted and radiated EMI interference of the circuit in this application is significantly reduced compared to the prior art.

[0084] When the interleaved boost chopper circuit 1 of this application operates with continuous current in the dual-winding inductor L1 and dual-winding inductor L2, a high-frequency short period of the low-frequency voltage waveform is selected, and the comparison diagram of the high-frequency voltage waveforms of its main components is shown below. Figure 7 As shown, Figure 7 The high-frequency voltage waveform between the drain D and source S of the medium-power switching unit Q1 is VD-Q1; the high-frequency voltage waveform between the drain D and source S of the power switching unit Q2 is VD-Q2; the high-frequency voltage waveform across the filter capacitor C3 is V-C3; the high-frequency voltage waveform across the filter capacitor C4 is V-C4; the high-frequency voltage waveform across the auxiliary inductor L3 is V-L3; the high-frequency voltage waveform across the auxiliary inductor L4 is V-L4; the high-frequency voltage waveform of the first winding inductor L1-N1 of the dual-winding inductor L1 is VL1-N1; the high-frequency voltage waveform of the first winding inductor L2-N1 of the dual-winding inductor L2 is VL2-N1; and the high-frequency voltage waveform across the first capacitor C1 is V-C1.

[0085] like Figure 7As shown, the high-frequency voltage waveforms VL1-N1 of the first winding inductor L1 and VL2-N1 of the first winding inductor L2 are both high-frequency square waves, and their voltage amplitudes remain basically constant within a high-frequency switching cycle T. The high-frequency voltage waveforms V-L3 and V-L4 across the auxiliary inductor L3 and L4 are quasi-resonant high-frequency AC voltages with small amplitudes. Therefore, the auxiliary inductors L3 and L4 effectively suppress the high-frequency current in the aforementioned low-frequency current loop. The high-frequency AC component voltage amplitude of the high-frequency voltage waveform V-C1 across the first capacitor C1 is significantly lower than that of the high-frequency AC component voltage amplitudes of the high-frequency voltage waveforms V-C3 and V-C4 across the filter capacitor C3 and C4, respectively. The low-frequency voltage amplitudes of the first capacitor C1 are basically the same as those of the filter capacitors C3 and C4.

[0086] When the filter capacitor of this application operates with continuous current in the dual-winding inductors L1 and L2, a comparison of the low-frequency voltage and current waveforms of its main components over a long period of time is shown in the figure below. Figure 8 As shown. Figure 8 The low-frequency voltage waveform of the AC voltage between the L and N lines of the AC input is VIN; the low-frequency voltage waveform of the DC voltage output is VO; the low-frequency voltage waveform across the first capacitor C1 is V-C1; the low-frequency current waveform flowing through the current sampling resistor RS1 is I-RS1; the low-frequency current waveform between the drain D and source S of the power switch unit Q2 is I-Q2; the low-frequency current waveform flowing through the first winding of the dual-winding inductor L1 is... The low-frequency current waveform of L1-N1 is IL1-N1, the low-frequency current waveform flowing through the winding L1-N2 of the double-winding inductor L1 is IL1-N2, the low-frequency current waveform flowing through the first winding inductor L2-N1 of the double-winding inductor L2 is IL2-N1, the low-frequency current waveform flowing through the second winding inductor L2-N2 of the double-winding inductor L2 is IL2-N2, the forward low-frequency current waveform flowing through diode D1 is I-D1, and the forward low-frequency current waveform flowing through diode D2 is I-D2.

[0087] refer to Figure 8 It can be seen that the open low-frequency current loops IL1-N1 and IL2-N1, which are connected to the AC input lines L and N, mainly carry low-frequency DC current, while the internal closed high-frequency current loops IL1-N2 and IL2-N2, which are isolated from the AC input lines L and N, mainly carry high-frequency AC current.

[0088] In summary, the interleaved boost chopper circuit 1 of this application locks most of the high-frequency switching current in a closed loop and allows most of the low-frequency current to pass through the open input power supply loop. Therefore, compared with the BOOST circuit in the prior art, the interleaved boost chopper circuit 1 of this application significantly reduces high-frequency conduction in the input loop and radiation interference to the surrounding space.

[0089] The interleaved boost chopper circuit 1 of this invention adopts a dual-path topology circuit and a control mode with interleaved 180° phase drive signals, which further significantly reduces the peak value of high-frequency switching current, thereby further significantly reducing the amplitude and energy of conducted and radiated interference.

[0090] As described above, because the interleaved boost chopper circuit 1 of this application can significantly reduce the amplitude and energy of conducted and radiated interference, when the circuit of this application is used as the power factor correction (PFC) circuit module of the AC input line of an AC / DC switching power supply, compared with the BOOST circuit in the prior art, it can effectively reduce the number and quantity of electromagnetic compatibility (EMC) filter stages at the L / N line AC input front end, thereby further reducing costs.

[0091] Because the interleaved boost chopper circuit 1 of this application can significantly reduce the amplitude and energy of conducted and radiated interference, when this invention is used in a small-sized switching power supply, compared with the technical method in the prior art BOOST circuit that generally limits the switching frequency to less than 75KHz to pass the relevant safety certification EMC specifications, the circuit of this application can further reduce the size of the dual-winding inductors L1 and L2 by increasing the switching frequency, and can still meet the relevant safety certification EMC specifications without increasing the cost, thereby further meeting the new needs of economic and social development.

[0092] In summary, the interleaved boost chopper circuit 1 of this application can be widely used in a large number of switching power supplies, and can further reduce the number of EMC filters to reduce costs or further increase the switching frequency to reduce size. Therefore, it can generate significant economic and social benefits and has the prospect of large-scale application.

[0093] Furthermore, in order to better implement the interleaved boost chopper circuit 1 in any of the above embodiments, based on the interleaved boost chopper circuit 1 described above, this application embodiment also provides a high-frequency switching power supply 2. The high-frequency switching power supply 2 includes multiple interleaved boost chopper circuits 1 connected in series, or includes multiple interleaved boost chopper circuits 1 connected in parallel. The interleaved boost chopper circuit 1 is the interleaved boost chopper circuit 1 as described in the above embodiments.

[0094] refer to Figure 9As shown, when multiple interleaved boost chopper circuits 1 are connected in series, the negative terminal BD1- of the input voltage of the adjacent high-voltage side interleaved boost chopper circuit 1 (taking the first interleaved boost chopper circuit 1 as an example) serves as the positive terminal BD2+ of the input voltage of the adjacent low-voltage side interleaved boost chopper circuit 1 (taking the second interleaved boost chopper circuit 1), and the negative terminal VO1-GND of the DC output voltage of the adjacent high-voltage side interleaved boost chopper circuit 1 serves as the positive terminal VO2 of the DC output voltage of the adjacent low-voltage side interleaved boost chopper circuit 1.

[0095] refer to Figure 10 As shown, when multiple interleaved boost chopper circuits 1 are connected in parallel, each group of interleaved boost chopper circuits 1 has the same positive terminal BD+ and negative terminal BD- of the input voltage, as well as the same positive terminal VO and positive terminal VO-GND of the DC output voltage.

[0096] In the above embodiments, the descriptions of each embodiment have different focuses. For parts that are not described in detail or recorded in a certain embodiment, please refer to the relevant descriptions of other embodiments.

[0097] The basic concepts have been described above. Obviously, for those skilled in the art, the detailed disclosure above is merely illustrative and does not constitute a limitation of this application. Although not explicitly stated herein, those skilled in the art may make various modifications, improvements, and corrections to this application. Such modifications, improvements, and corrections are suggested in this application, and therefore remain within the spirit and scope of the exemplary embodiments of this application.

[0098] Furthermore, this application uses specific terms to describe embodiments of the application. For example, "an embodiment," "one embodiment," and / or "some embodiments" refer to a particular feature, structure, or characteristic associated with at least one embodiment of the application. Therefore, it should be emphasized and noted that "an embodiment," "one embodiment," or "an alternative embodiment" mentioned twice or more in different locations in this specification do not necessarily refer to the same embodiment. In addition, certain features, structures, or characteristics in one or more embodiments of the application can be appropriately combined.

[0099] Similarly, it should be noted that, in order to simplify the description of the present application and thus aid in the understanding of one or more embodiments of the invention, the foregoing description of the embodiments of the present application sometimes combines multiple features into a single embodiment, drawing, or description thereof. However, this disclosure method does not imply that the subject matter of the application requires more features than those mentioned in the claims. In fact, the embodiments contain fewer features than all the features of the single embodiments disclosed above.

[0100] The above-described embodiments are only used to illustrate the technical solutions of this application, and are not intended to limit them. Although this application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of this application, and should all be included within the protection scope of this application.

Claims

1. An interleaved boost chopper circuit for boosting an input voltage to output a DC output voltage, characterized in that, The drive module and at least one voltage boosting module connected with the drive module; when the voltage boosting module is multiple, all the voltage boosting modules are connected in parallel, and the drive module alternately drives multiple voltage boosting modules; The voltage boosting module includes a low-frequency current loop and a high-frequency current loop coupled with each other; The low-frequency current loop is used for boosting the input voltage under the drive of the drive module, and the low-frequency current in the interleaved voltage boosting chopper circuit flows from the positive terminal of the input voltage and outputs from the negative terminal; The high-frequency current loop is an internal closed loop, which is used for closing the flow of high-frequency switching alternating current generated when the drive module drives the voltage boosting module.

2. The interleaved boost buck converter of claim 1, wherein, The voltage boosting module includes a double-winding inductor, a power switch unit, a diode, and a filter unit, the double-winding inductor includes a first winding inductor and a second winding inductor, and the filter unit includes an auxiliary inductor and a filter capacitor; The first end of the first winding inductor is connected with the second end of the auxiliary inductor, the second end of the first winding inductor is connected with the first current conduction end of the power switch unit and the anode of the diode respectively, the first end of the auxiliary inductor is connected with the positive terminal of the input voltage, the second current conduction end of the power switch unit is connected with the negative terminal of the direct current output voltage, the cathode of the diode is connected with the positive terminal of the direct current output voltage, and the drive control end of the power switch unit is connected with the drive module; The first end of the second winding inductor is connected with the second end of the filter capacitor, and the second end of the second winding inductor is connected with the first current conduction end of the power switch unit and the anode of the diode respectively, and the first end of the filter capacitor is connected with the negative terminal of the input voltage. The number of turns of the first winding inductor and the second winding inductor is equal.

3. The interleaved boost buck converter of claim 2, wherein, The voltage boosting module further includes a current limiting resistor and a protection resistor, the current limiting resistor is connected between the drive module and the drive control end of the power switch unit, and the protection resistor is connected between the drive control end of the power switch unit and the negative terminal of the direct current output voltage.

4. The interleaved boost buck converter of claim 1, wherein, The interleaved voltage boosting chopper circuit further includes a first capacitor connected between the positive terminal and the negative terminal of the input voltage and a second capacitor connected between the positive terminal and the negative terminal of the direct current output voltage.

5. The interleaved boost buck converter of claim 1, wherein, The interleaved voltage boosting chopper circuit further includes a current sampling resistor connected between the negative terminal of the input voltage and the second current conduction end of the power switch unit.

6. The interleaved boost buck converter of claim 1, wherein, The full-bridge rectifier circuit further includes a full-bridge rectifier circuit, the positive output end of the full-bridge rectifier circuit is connected with the positive terminal of the input voltage, and the negative output end of the full-bridge rectifier circuit is connected with the negative terminal of the input voltage.

7. The interleaved boost buck converter of claim 2, wherein, The drive module outputs switching signals with different phases and the same pulse width to the drive control ends of the power switch units in multiple voltage boosting modules respectively, so as to control the power switch units in multiple voltage boosting modules to be alternately turned on. The sum of the phases of all the switching signals is 360 degrees.

8. The interleaved boost buck converter of claim 2, wherein, The drive module controls the alternate conduction of the power switch units in the two boost modules by outputting switching signals with 180-degree staggered phase and same pulse width to the drive control terminals of the power switch units in the two boost modules.

9. A high frequency switching power supply, characterized by The application provides an interleaved boost chopper circuit, comprising a plurality of interleaved boost chopper circuits connected in series.

10. A high frequency switching power supply, characterized by comprising: The application provides an interleaved boost chopper circuit, comprising a plurality of interleaved boost chopper circuits connected in parallel. The application provides an interleaved boost chopper circuit, comprising a plurality of interleaved boost chopper circuits connected in parallel.

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