Junction temperature compensated gate driver

Abstract

The invention relates to junction temperature compensated gate drivers. A vehicle driveline includes a power inverter and a controller. The power inverter includes a load switch monolithically integrated with a mirror switch, a current sensor configured to measure a current flowing through the load switch to provide current feedback for vector control of a motor. The controller can be configured to operate the power inverter to reduce a temperature of the load switch in response to a difference between a mirror current of the mirror switch and an output of the current sensor resulting from a temperature change of the load switch.

Description

Technical Field

[0001] The present application generally relates to a gate driver for a solid-state switch of a power module in which a comparison between an output from a current sensor of the power module and a signal from a current mirror of the solid-state switch is used to regulate the operation of the solid-state switch to reduce the temperature of the solid-state switch. Background Art

[0002] Electrified vehicles, including hybrid electric vehicles (HEVs), plug-in hybrid electric vehicles (PHEVs), and battery electric vehicles (BEVs), rely on a traction battery to power the traction motor for propulsion, and a power inverter between the traction battery and the traction motor to convert direct current (DC) power to alternating current (AC). A typical AC traction motor is a three-phase motor that can be powered by three sinusoidal signals, each driven with 120 degrees of phase separation. The traction battery is configured to operate within a specific voltage range and provide maximum current. The traction battery is also optionally referred to as a high-voltage battery.

[0003] Furthermore, many electrified vehicles include a DC-DC converter (also known as a variable voltage converter (VVC)) to convert the voltage of the traction battery to the operating voltage level of the electric machines. The electric machines, which may include traction motors, can require high voltages and high currents. Due to the voltage, current, and switching requirements, solid-state switches, such as insulated-gate bipolar junction transistors (IGBTs), are often used to generate signals in the power inverter and VVC. Summary of the Invention

[0004] A vehicle powertrain system includes a power inverter and a controller. The power inverter includes a load switch monolithically integrated with a mirror switch; and a current sensor configured to measure current flowing through the load switch to provide current feedback for vector control of a motor. The controller may be configured to operate the power inverter to reduce the temperature of the load switch in response to a difference between a mirror current of the mirror switch and an output of the current sensor due to temperature variations of the load switch.

[0005] According to one embodiment of the present invention, operating the power inverter to reduce the temperature of the load switch includes: reducing the switching frequency of the load switch.

[0006] A method for controlling a vehicle powertrain includes disabling an inverter switch monolithically integrated with a mirror switch that enables the mirror current to flow, in response to a mirror current exceeding a threshold. The method further controls the vehicle powertrain based on a signal from an inverter current sensor, and in response to a difference between the mirror current and the signal resulting from a temperature change of the inverter switch, operates the vehicle powertrain to reduce the temperature of the inverter switch.

[0007] According to one embodiment of the present invention, operating the vehicle powertrain system to reduce the temperature of the inverter switch includes reducing a switching frequency of a load switch of the vehicle powertrain system.

[0008] A vehicle powertrain inverter includes an insulated gate bipolar junction transistor (IGBT), a current sensor, and a controller. The IGBT is monolithically integrated with a mirror IGBT. The current sensor is configured to measure current flowing through the IGBT. The controller can be configured to operate the IGBT to reduce the temperature of the IGBT in response to a difference between a mirror current of the mirror IGBT and an output of the current sensor due to temperature changes of the IGBT.

[0009] According to an embodiment of the invention, the output is an analog output and the difference is adjusted by a scaling factor applied to the output.

[0010] According to one embodiment of the present invention, operating the IGBT to reduce the temperature of the IGBT includes reducing a pulse width modulation duty cycle of the IGBT.

[0011] According to one embodiment of the present invention, the controller operates the IGBT based on a model-based junction temperature estimation model that accumulates error over time, and is further configured to adjust the model-based junction temperature estimation model based on the difference to minimize the error.

[0012] According to one embodiment of the present invention, the adjustment of the model-based junction temperature estimation model is periodic. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] Figure 1 is a diagram of a hybrid vehicle showing a typical powertrain and energy storage components with a variable voltage converter and a power inverter therebetween.

[0014] Figure 2 This is a schematic diagram of an on-board variable voltage converter.

[0015] Figure 3 This is a schematic diagram of the vehicle motor inverter.

[0016] Figure 4 is a schematic diagram of a control circuit having a temperature compensation circuit.

[0017] Figure 5 is a schematic diagram of an insulated gate bipolar junction transistor with a current mirror and a sense resistor.

[0018] Figure 6It is a graphical representation of the collector current of an IGBT versus the voltage between the collector and emitter, and the mirror current of a mirror device monolithically integrated with the IGBT versus the voltage between the collector and emitter. DETAILED DESCRIPTION

[0019] Embodiments of the present disclosure are described herein. However, it will be understood that the disclosed embodiments are merely examples, and that other embodiments may take various forms and alternative forms. The figures are not necessarily drawn to scale; some features may be exaggerated or minimized to show details of particular components. Therefore, the specific structural and functional details disclosed herein should not be interpreted as limiting, but merely as a representative basis for teaching those skilled in the art to utilize the invention in various forms. As will be understood by those of ordinary skill in the art, the various features shown and described with reference to any one of the figures may be combined with features shown in one or more other figures to produce embodiments that are not explicitly shown or described. The combinations of features shown provide representative embodiments for typical applications. However, various combinations and variations of features consistent with the teachings of the present disclosure may be desired for specific applications or implementations.

[0020] When considering an xEV such as an HEV, PHEV, or BEV, operating conditions may vary significantly over the life of the xEV. For example, power devices operating in a powertrain system (such as transistors in a converter, the converter comprising a DC-DC converter or a DC-AC converter) may operate over a wide temperature range, and therefore, the power devices in the converter may have junction temperatures (Tj) that vary from extremely low temperatures (e.g., -40°C) to very high temperatures (e.g., 150°C). As the operating voltage changes, the breakdown voltage (VB) of the power device also changes because it is a function of Tj. Here, a circuit is disclosed for automatically adjusting for changes in the junction temperature of a device to better avoid reverse voltage breakdown of the device. Typically, room temperature is 25°C, low temperature is any temperature below room temperature, and very low temperature is any temperature below 0°C (i.e., the temperature at which water freezes). Typically, the breakdown voltage of a switch is measured at room temperature (i.e., 25°C). The breakdown of an IGBT can be specified as the collector-to-emitter breakdown voltage (BVces) with the gate shorted to the emitter, while limiting the conditions associated with that specification. For example, the specification may limit the breakdown condition to a temperature of 25°C with a collector current of 1 mA and Vge of 0 V. However, when used in harsh environments, the breakdown voltage BVces may drop by another 5% at -25°C and another 7% at -50°C.

[0021] Therefore, electrical modules in vehicles (such as DC-DC converters or DC-AC converters) may exhibit voltage peaks that are below the breakdown voltage when Tj is greater than room temperature and may exceed the breakdown voltage at low temperatures. The voltage peak depends on the gate current amplitude, the rate of change of the current through the switch, and the amplitude of the current. The breakdown voltage of a switch is typically determined by the circuit topology and manufacturing process. For a given current capacity, switches with higher breakdown voltages typically have higher costs, and in some instances, switches with higher breakdown voltages are not feasible due to material limitations of the switch. Circuit designers utilizing switches typically desire to operate the switch close to the breakdown voltage without exceeding it. Therefore, to meet the requirements and constraints of the switch, the system is typically designed with a minimum breakdown voltage across the entire operating temperature range. However, switches typically operate only for a short period of time at very low or extremely low temperatures (e.g., -50°C, -40°C, -35°C, -25°C, -15°C, or -5°C), after which the component heats up (either internally or through the use of external heaters) and the breakdown voltage increases. To reduce cost and improve efficiency, methods and circuits are disclosed for regulating the gate current of a switch based on the temperature of the switch to proportionally regulate the load current when the switch operates at low or very low temperatures.

[0022] Generally, solid-state devices (SSDs), such as insulated-gate bipolar junction transistors (IGBTs), metal-oxide-semiconductor field-effect transistors (MOSFETs), or bipolar junction transistors (BJTs), are widely used in various automotive and industrial applications, such as electric motor drives, power inverters, DC-DC converters, and power modules. The operation of IGBTs and MOSFETs is voltage-controlled, where the operation is based on the voltage applied to the gate of the IGBT or MOSFET, while the operation of BJTs is current-controlled, where the operation is based on the current applied to the base of the BJT. The use of IGBTs will be discussed herein, but the structure and methods are applicable to other SSDs, for example, insulated-gate SSDs including both IGBTs and MOSFETs. The operation of the IGBT is controlled by a gate voltage provided by a gate driver. Conventional gate drivers are typically based on a voltage greater than a threshold voltage applied to the gate of the IGBT with a current-limiting resistor. The gate driver typically consists of a switchable voltage source and a gate resistor. Low gate resistance results in fast switching speeds and low switching losses, but can also lead to higher stresses (e.g., overvoltage stress) on the semiconductor device. Therefore, the gate resistor is selected to achieve a compromise between switching losses, switching delay, and stress. When the IGBT is turned off, the gate resistor reduces the current flowing from the gate, thereby increasing the IGBT's off time. Furthermore, IGBTs have unequal losses during the on and off periods, so a gate driver that provides an on-resistance different from the off-resistance is used.

[0023] Disclosed herein is a system and method for monitoring IGBT junction temperature that can be implemented with high accuracy and low cost. This system and method derives the switch's Tj based on data from at least one inverter output current sensor and an IGBT current mirror sensor. The primary principle exploits the fact that the output of the IGBT current mirror sensor is sensitive to changes in Tj, while the output of the inverter output current sensor is largely unaffected, or even completely unaffected, by changes in Tj. Therefore, by comparing the output of the IGBT current mirror sensor with the output of the inverter output current sensor, the IGBT's Tj can be derived.

[0024] Typically, inverter current sensors are used in on-board electric powertrains for output current control and to provide protection against occasional faults caused by unstable control or external short circuits. The IGBT current mirror sensor locally monitors the current. The IGBT current mirror sensor can be used to protect the IGBT from faults caused by external short circuits or internal short circuits. Internal short circuits include short circuits caused by simultaneous conduction of the upper IGBT and the lower IGBT. Typically, the output current sensor uses a Hall effect sensor or a GMR (giant magnetoresistance) sensor with high accuracy. The IGBT current mirror sensor is a mirror IGBT chip that causes part of the current of the main IGBT to flow.

[0025] Figure 1 An electrified vehicle 112, which may be referred to as a plug-in hybrid electric vehicle (PHEV), is depicted. The plug-in hybrid electric vehicle 112 may include one or more electric motors 114 mechanically connected to a hybrid transmission 116. The electric motors 114 can operate as either motors or generators. Furthermore, the hybrid transmission 116 is mechanically connected to an engine 118. The hybrid transmission 116 is also mechanically connected to a drive shaft 120, which is mechanically connected to wheels 122. The electric motors 114 can provide propulsion and deceleration capabilities when the engine 118 is on or off. The electric motors 114 can also function as generators and can provide fuel economy benefits by recovering energy that would normally be lost as heat in the friction braking system. The electric motors 114 can also reduce vehicle emissions by allowing the engine 118 to operate at a more efficient speed and allowing the hybrid electric vehicle 112 to operate in an electric mode with the engine 118 off under certain conditions. The electrified vehicle 112 may also be a battery electric vehicle (BEV). In a BEV configuration, the engine 118 may not be present. In other configurations, the electrified vehicle 112 may be a full hybrid electric vehicle (FHEV) without plug-in capability.

[0026] The traction battery or battery pack 124 stores energy that can be used by the motor 114. The vehicle battery pack 124 can provide a high-voltage direct current (DC) output. The traction battery 124 can be electrically connected to one or more power electronics modules 126. One or more contactors 142 can isolate the traction battery 124 from other components when open, and can connect the traction battery 124 to other components when closed. The power electronics module 126 is also electrically connected to the motor 114 and provides the ability to transfer energy bidirectionally between the traction battery 124 and the motor 114. For example, the traction battery 124 can provide a DC voltage, while the motor 114 can operate using three-phase alternating current (AC). The power electronics module 126 can convert the DC voltage into three-phase AC current to operate the motor 114. In regenerative mode, the power electronics module 126 can convert the three-phase AC current from the motor 114, which is acting as a generator, into a DC voltage compatible with the traction battery 124.

[0027] The vehicle 112 may include a variable voltage converter (VVC) 152 electrically connected between the traction battery 124 and the power electronics module 126. The VVC 152 may be a DC / DC boost converter configured to increase or step up the voltage provided by the traction battery 124. By increasing the voltage, the current demand may be reduced, resulting in a reduction in the wiring size of the power electronics module 126 and the electric motor 114. Additionally, the electric motor 114 may operate with higher efficiency and lower losses.

[0028] In addition to providing energy for propulsion, the traction battery 124 can also provide energy for other vehicle electrical systems. The vehicle 112 may include a DC / DC converter module 128 that converts the high-voltage DC output of the traction battery 124 into a low-voltage DC power source compatible with low-voltage vehicle loads. The output of the DC / DC converter module 128 can be electrically connected to an auxiliary battery 130 (e.g., a 12V battery) for charging the auxiliary battery 130. Low-voltage systems can be electrically connected to the auxiliary battery 130. One or more electrical loads 146 can be connected to the high-voltage bus. The electrical loads 146 can have associated controllers that appropriately operate and control the electrical loads 146. Examples of electrical loads 146 may include fans, electric heating elements, and / or air conditioning compressors.

[0029] The electrified vehicle 112 may be configured to recharge the traction battery 124 via an external power source 136. The external power source 136 may be connected to an electrical outlet. The external power source 136 may be electrically connected to a charger or electric vehicle supply equipment (EVSE) 138. The external power source 136 may be the power distribution grid or the grid provided by a public utility. The EVSE 138 may provide circuitry and controls to regulate and manage energy transfer between the power source 136 and the vehicle 112. The external power source 136 may provide DC power or AC power to the EVSE 138. The EVSE 138 may have a charging connector 140 for plugging into a charging port 134 of the vehicle 112. The charging port 134 may be any type of port configured to transfer power from the EVSE 138 to the vehicle 112. The charging port 134 may be electrically connected to a charger or an onboard power conversion module 132. The power conversion module 132 may condition the power supplied from the EVSE 138 to provide appropriate voltage and current levels to the traction battery 124. The power conversion module 132 may interface with the EVSE 138 to coordinate power delivery to the vehicle 112. The EVSE connector 140 may have pins that mate with corresponding recesses of the charging port 134. Alternatively, the various components described as being electrically coupled or connected may transfer power using wireless inductive coupling.

[0030] One or more wheel brakes 144 may be provided to slow down and prevent vehicle 112 from moving. Wheel brakes 144 may be hydraulically actuated, electrically actuated, or some combination thereof. Wheel brakes 144 may be part of a braking system 150. Braking system 150 may include other components for operating wheel brakes 144. For simplicity, the figures depict a single connection between braking system 150 and one of wheel brakes 144. Connections between braking system 150 and other wheel brakes 144 are implicit. Braking system 150 may include a controller to monitor and coordinate braking system 150. Braking system 150 may monitor the brake components and control wheel brakes 144 to slow the vehicle. Braking system 150 may respond to driver commands and may also operate autonomously to implement functions such as stability control. When requested by another controller or sub-function, the controller of braking system 150 may implement a method for applying the requested braking force.

[0031] The electronic modules in the vehicle 112 may communicate via one or more vehicle networks. The vehicle network may include multiple channels for communication. One channel of the vehicle network may be a serial bus such as a controller area network (CAN). One of the channels of the vehicle network may include Ethernet as defined by the Institute of Electrical and Electronics Engineers (IEEE) 802 family of standards. Other channels of the vehicle network may include discrete connections between modules and may include power signals from the auxiliary battery 130. Different signals may be transmitted over different channels of the vehicle network. For example, a video signal may be transmitted over a high-speed channel (e.g., Ethernet) while control signals may be transmitted over CAN or discrete signals. The vehicle network may include any hardware components and software components that assist in transmitting signals and data between modules. Although the vehicle network is Figure 1 Not shown, but implied, the vehicle network may be connected to any electronic modules present in the vehicle 112. A vehicle system controller (VSC) 148 may be present to coordinate the operation of the various components.

[0032] Figure 2 A diagram of a VVC 152 configured as a boost converter is depicted. The VVC 152 may include input terminals that may be connected to terminals of a traction battery 124 via a contactor 142. The VVC 152 may include output terminals that may be connected to terminals of a power electronics module 126. The VVC 152 may be operated to cause a voltage at the output terminals to be higher than a voltage at the input terminals. The vehicle 112 may include a VVC controller 200 that monitors and controls electrical parameters (e.g., voltage and current) at various locations in the VVC 152. In some configurations, the VVC controller 200 may be included as part of the VVC 152. The VVC controller 200 may determine an output voltage reference. The VVC controller 200 can be based on electrical parameters and voltage references A control signal sufficient to cause the VVC 152 to achieve the desired output voltage is determined. In some configurations, the control signal may be implemented as a pulse width modulated (PWM) signal, wherein the duty cycle of the PWM signal varies. The control signal may operate at a predetermined switching frequency. The VVC controller 200 may use the control signal to command the VVC 152 to provide the desired output voltage. The specific control signal used to operate the VVC 152 may be directly related to the amount of voltage boost provided by the VVC 152.

[0033] The output voltage of the VVC 152 can be controlled to achieve a desired reference voltage. In some configurations, the VVC 152 can be a boost converter. In a boost converter configuration, the VVC controller 200 controls the duty cycle, the input voltage V in , output voltage V outThe ideal relationship between and duty cycle D can be shown using the following equation:

[0034]

[0035] The desired duty cycle D can be determined by measuring the input voltage (e.g., the traction battery voltage) and setting the output voltage to a reference voltage. VVC 152 can be a buck converter that steps down the voltage from input to output. In a buck configuration, different expressions relating the input and output voltages to the duty cycle can be derived. In some configurations, VVC 152 can be a buck-boost converter that can step up or down the input voltage. The control strategy described herein is not limited to a specific variable voltage converter topology.

[0036] Reference Figure 2 The VVC 152 can increase or "step up" the potential of the power provided by the traction battery 124. The traction battery 124 can provide high voltage (HV) DC power. In some configurations, the traction battery 124 can provide a voltage between 150 volts and 400 volts. The contactor 142 can be electrically connected in series between the traction battery 124 and the VVC 152. When the contactor 142 is closed, HV DC power can be transferred from the traction battery 124 to the VVC 152. An input capacitor 202 can be electrically connected in parallel with the traction battery 124. The input capacitor 202 can stabilize the bus voltage and reduce any voltage and current ripple. The VVC 152 can receive the HV DC power and increase or "step up" the potential of the input voltage according to the duty cycle.

[0037] An output capacitor 204 may be electrically connected between the output terminals of the VVC 152. The output capacitor 204 may stabilize the bus voltage and reduce voltage and current ripple at the output of the VVC 152.

[0038] Further references Figure 2The VVC 152 may include a first switching device 206 and a second switching device 208 for stepping up an input voltage to provide a boosted output voltage. Switching devices 206 and 208 may be configured to selectively direct current to an electrical load (e.g., the power electronics module 126 and the motor 114). Each switching device 206 and 208 may be independently controlled by a gate drive circuit (not shown) of the VVC controller 200 and may include any type of controllable switch (e.g., an insulated gate bipolar junction transistor (IGBT) or a field effect transistor (FET)). The gate drive circuit may provide an electrical signal based on a control signal (e.g., the duty cycle of a PWM control signal) to each of the switching devices 206 and 208. A diode may be connected across each of the switching devices 206 and 208. Each switching device 206 and 208 may have associated switching losses. Switching losses are power losses incurred during state changes of the switching device (e.g., on / off transitions and off / on transitions). Switching losses can be quantified by the current flowing through the switching devices 206 and 208 during a transition and the voltage across the switching devices 206 and 208. The switching devices may also have associated conduction losses that occur when the devices are turned on.

[0039] The vehicle system may include sensors for measuring electrical parameters of the VVC 152. The first voltage sensor 210 may be configured to measure an input voltage (eg, the voltage of the battery 124) and provide a corresponding input signal (V bat In one or more embodiments, the first voltage sensor 210 can measure the voltage across the input capacitor 202, which corresponds to the battery voltage. The second voltage sensor 212 can measure the output voltage of the VVC 152 and provide a corresponding input signal (V dc In one or more embodiments, the second voltage sensor 212 can measure the voltage across the output capacitor 204, which corresponds to the DC bus voltage. The first voltage sensor 210 and the second voltage sensor 212 can include circuitry for scaling the voltage to a level suitable for the VVC controller 200. The VVC controller 200 can include circuitry for filtering and digitizing the signals from the first voltage sensor 210 and the second voltage sensor 212.

[0040] An input inductor 214 (often referred to as a boost inductor) may be electrically connected in series between the traction battery 124 and the switching devices 206 and 208. The input inductor 214 may switch between storing energy in the VVC 152 and releasing energy from the VVC 152, thereby providing a variable voltage and current as the output of the VVC 152 and achieving a desired voltage boost. A current sensor 216 may measure the input current flowing through the input inductor 214 and provide a corresponding current signal (I L The input current flowing through the input inductor 214 may be a function of the voltage difference between the input voltage and the output voltage of the VVC 152, the on-time of the switching devices 206 and 208, and the inductance L of the input inductor 214. The VVC controller 200 may include circuitry for scaling, filtering, and digitizing the signal from the current sensor 216.

[0041] The VVC controller 200 may be configured to control the output voltage of the VVC 152. The VVC controller 200 may receive input from the VVC 152 or other controllers via the vehicle network and determine a control signal. The VVC controller 200 may monitor the input signal To determine the control signal. For example, the VVC controller 200 may provide a control signal corresponding to the duty cycle command to the gate drive circuit. The gate drive circuit may then control each switching device 206, 208 based on the duty cycle command.

[0042] The control signals of the VVC 152 can be configured to drive the switching devices 206, 208 at a specific switching frequency. Within each cycle of the switching frequency, the switching devices 206, 208 can be operated at a specified duty cycle. The duty cycle defines the amount of time that the switching devices 206, 208 are in the on state and the off state. For example, a 100% duty cycle can cause the switching devices 206, 208 to operate in a continuous on state with no cutoff. A 0% duty cycle can cause the switching devices 206, 208 to operate in a continuous off state with no cutoff. A 50% duty cycle can cause the switching devices 206, 208 to operate in the on state for half a cycle and in the off state for half a cycle. The control signals for the two switching devices 206, 208 can be complementary. That is, the control signal sent to one of the switching devices (e.g., switching device 206) can be the inverse version of the control signal sent to the other switching device (e.g., switching device 208). Complementary control of the switching devices 206, 208 is utilized to avoid shoot-through conditions where current flows directly through the high-side switching device 206 and the low-side switching device 208. The high-side switching device 206 is also referred to as a pass device 206 and the low-side switching device 208 is also referred to as a charge device 208.

[0043] The current controlled by switching devices 206 and 208 may include a ripple component whose magnitude varies with the magnitude of the current, the duty cycle of switching devices 206 and 208, and the switching frequency. The worst-case ripple current magnitude occurs during relatively high input current conditions relative to the input current. When the duty cycle is fixed, an increase in inductor current causes the ripple current magnitude to increase. The ripple current magnitude is also related to the duty cycle. The highest magnitude ripple current occurs when the duty cycle is equal to 50%. Based on these facts, it is beneficial to implement measures to reduce the ripple current magnitude under high current and mid-range duty cycle conditions.

[0044] When designing VVC 152, the switching frequency and inductance value of inductor 214 may be selected to meet the maximum allowable ripple current amplitude. A ripple component can be a periodic variation that occurs on a DC signal. The ripple component can be defined based on its amplitude and frequency. The ripple component may have harmonics within the audible frequency range, which can increase the vehicle's noise signature. Furthermore, the ripple component can make it difficult to accurately control devices powered by the source. During switching transients, switching devices 206 and 208 may be turned off at the maximum inductor current (DC current plus ripple current), which can generate large voltage spikes across switching devices 206 and 208. Due to size and cost constraints, the inductance value may be selected based on the conduction current. Generally, as the current increases, the inductance may decrease due to saturation.

[0045] The switching frequency can be selected to limit the magnitude of the ripple current component under worst-case conditions (e.g., conditions with the highest input current and / or a duty cycle approaching 50%). The switching frequency of switching devices 206, 208 can be selected to be a frequency (e.g., 10 kHz) greater than the switching frequency (e.g., 5 kHz) of the motor / generator inverter connected to the output of VVC 152. In some applications, the switching frequency of VVC 152 can be selected to be a predetermined fixed frequency. The predetermined fixed frequency is typically selected to meet noise and ripple current specifications. However, the selection of a predetermined fixed frequency may not provide optimal performance across the entire operating range of VVC 152. A predetermined fixed frequency may provide optimal results for a particular set of operating conditions, but may compromise performance for other operating conditions.

[0046] Increasing the switching frequency can reduce the ripple current amplitude and lower the voltage stress on the switching devices 206 and 208, but may result in higher switching losses. When the switching frequency is selected for the worst-case ripple condition, the VVC 152 may only operate under the worst-case ripple condition for a small percentage of the total operating time. This can result in unnecessarily high switching losses that can reduce fuel economy. Furthermore, a fixed switching frequency can concentrate the noise spectrum into a very narrow range. The increased noise density within this narrow range can lead to significant noise, vibration, and harshness (NVH) issues.

[0047] The VVC controller 200 can be configured to vary the switching frequency of the switching devices 206, 208 based on the duty cycle and input current. Variation of the switching frequency can improve fuel economy by reducing switching losses and also reduce NVH issues while maintaining a ripple current target under worst-case operating conditions.

[0048] During relatively high current conditions, switching devices 206 and 208 may experience increased voltage stress. At the maximum operating current of VVC 152, it may be desirable to select a relatively high switching frequency to reduce the ripple component amplitude while maintaining a reasonable level of switching losses. The switching frequency may be selected based on the input current amplitude, such that as the input current amplitude increases, the switching frequency increases. The switching frequency may be increased to a predetermined maximum switching frequency. The predetermined maximum switching frequency may be a level that provides a compromise between lower ripple component amplitude and higher switching losses. The switching frequency may be varied in discrete steps or continuously within the operating current range.

[0049] The VVC controller 200 can be configured to reduce the switching frequency in response to the input current being less than a predetermined maximum current. The predetermined maximum current can be the maximum operating current of the VVC 152. The switching frequency can be varied based on the magnitude of the current input to the switching devices 206 and 208. When the current is greater than the predetermined maximum current, the switching frequency can be set to the predetermined maximum switching frequency. As the current decreases, the magnitude of the ripple component decreases. By operating at a lower switching frequency as the current decreases, switching losses are reduced. The switching frequency can be varied based on the power input to the switching devices. Since the input power is a function of the input current and the battery voltage, the input power and the input current can be utilized in a similar manner.

[0050] Because ripple current is also affected by the duty cycle, the switching frequency can be varied based on the duty cycle. The duty cycle can be determined based on the ratio of the input voltage to the output voltage. Therefore, the switching frequency can also vary based on the ratio between the input voltage and the output voltage. When the duty cycle approaches 50%, the predicted ripple current amplitude is at its maximum, and the switching frequency can be set to a predetermined maximum frequency. The predetermined maximum frequency can be a maximum switching frequency value selected to minimize the ripple current amplitude. The switching frequency can be varied in discrete steps or continuously within the duty cycle range.

[0051] The VVC controller 200 may be configured to reduce the switching frequency from a predetermined maximum frequency in response to the magnitude of the difference between the duty cycle and the duty cycle at which the predicted ripple component amplitude is maximum (e.g., 50%). When the magnitude of the difference is less than a threshold, the switching frequency may be set to the predetermined frequency. As the magnitude of the difference decreases, the switching frequency may be increased toward the predetermined maximum frequency to reduce the ripple component amplitude. When the magnitude of the difference is less than a threshold, the switching frequency may be set to the predetermined maximum frequency.

[0052] The switching frequency may be limited to between a predetermined maximum frequency and a predetermined minimum frequency. The predetermined minimum frequency may be a frequency level greater than a predetermined switching frequency of the power electronics module 126 connected to the output of the variable voltage converter 152. The switching frequency may also be based on parasitic inductance associated with the gate of the IGBT.

[0053] Reference Figure 3 , a system 300 is provided for controlling a power electronics module (PEM) 126 . Figure 3 The PEM 126 is shown as including a plurality of switches 302 (e.g., IGBTs) configured to collectively operate as an inverter having a first phase leg 316, a second phase leg 318, and a third phase leg 320. Although the inverter is shown as a three-phase converter, the inverter may include additional phase legs. For example, the inverter may be a four-phase converter, a five-phase converter, a six-phase converter, etc. Furthermore, the PEM 126 may include multiple converters, each inverter in the PEM 126 including three or more phase legs. For example, the system 300 may control two or more inverters in the PEM 126. The PEM 126 may also include a DC-to-DC converter having high-power switches (e.g., IGBTs) to convert the power electronics module input voltage to the power electronics module output voltage via step-up, step-down, or a combination thereof.

[0054] like Figure 3As shown, the inverter can be a DC to AC converter. In operation, the DC to AC converter receives DC power from the DC power link 306 via the DC bus 304 and converts the DC power into AC power. The AC power is converted to AC power via the phase current i a 、i b and i c transmission to drive an AC motor, also referred to as motor 114 (such as in Figure 3 ). In this example, the DC power link 306 may include a DC battery to provide DC power to the DC bus 304. In another example, the inverter may operate as an AC-to-DC converter that converts AC power from the AC motor 114 (e.g., a generator) into DC power, where the DC bus 304 may provide DC power to the DC power link 306. In addition, the system 300 may control the PEM 126 in other power electronics topologies.

[0055] Continue to refer to Figure 3 Each of the phase bridges 316, 318, and 320 in the inverter includes a power switch 302. The power switch 302 can be implemented by various types of controllable switches. In one embodiment, each power switch 302 can include a diode and a transistor (eg, an IGBT). Figure 3 The diode is marked as D a1 、D a2 、D b1 、D b2 、D c1 and D c2 ,and Figure 3 The IGBTs are marked as S a1 、S a2 、S b1 、S b2 、S c1 and S c2 . Power switch S a1 、S a2 、D a1 and D a2 It is part of the phase bridge A of the three-phase converter. Figure 3 In FIG, it is marked as the first phase bridge A 316. Similarly, the power switch S b1 、S b2 、D b1 and D b2 is part of the phase bridge B 318 of the three-phase converter, and the power switch S c1 、S c2 、D c1 and D c2It is part of the phase bridge C 320 of the three-phase converter. The inverter may include any number of power switches 302 or circuit elements depending on the specific configuration of the inverter. The diode (D xx ) and IGBT(S xx ) are connected in parallel, however, since the polarity is reversed for proper operation, this configuration is often referred to as an anti-parallel connection. The diodes in this anti-parallel configuration are also referred to as freewheeling diodes.

[0056] like Figure 3 As shown, set the current sensor CS a , CS b and CS c To sense the currents in phase bridges 316, 318 and 320 respectively. Figure 3 The current sensor CS is shown separated from the PEM 126. a , CS b and CS c However, depending on the configuration of the PEM 126, the current sensor CS a , CS b and CS c May be integrated as part of the PEM 126. Figure 3 Current sensor CS in a , CS b and CS c are mounted to phase bridges A, B and C (i.e. Figure 3 The phase bridges 316, 318 and 320 in the system are connected in series and provide feedback signals i for the system 300 respectively. as 、i bs and i cs (also Figure 3 Feedback signal i as 、i bs and i cs The current signal may be a raw current signal processed by the logic device (LD) 310, or may be embedded with data or information about the current flowing through the phase bridges 316, 318, and 320, respectively, or may be encoded with the data or information. In addition, the power switch 302 (e.g., IGBT) may include current sensing capability. The current sensing capability may include a device configured to provide a current signal indicating i as 、i bs and i cs The data / signal may indicate the direction, magnitude, or both direction and magnitude of the current flowing through phase bridges A, B, and C, respectively.

[0057] Refer again Figure 3, the system 300 includes a logic device (LD) or controller 310. The controller or LD 310 may be implemented by various types of electronic devices and / or microprocessor-based computers or controllers, or a combination thereof. To implement the method of controlling the PEM 126, the controller 310 may execute a computer program or algorithm that is embedded with or encoded with the method and stored in volatile memory 312 and / or permanent memory 312. Alternatively, the logic may be encoded into discrete logic, a microprocessor, a microcontroller, or a logic array or gate array stored on one or more integrated circuit chips. As Figure 3 As shown in the embodiment, the controller 310 receives and processes the feedback signal i as 、i bs and i cs To control the phase current i a 、i b and i c , so that the phase current i a 、i b and i c Flow through the phase bridges 316, 318, and 320 and into the corresponding windings of the motor 114 according to various current or voltage patterns. For example, the current pattern may include the phase currents i flowing into and out of the DC bus 304 or the DC bus capacitor 308. a 、i b and i c pattern. Figure 3 The DC bus capacitor 308 in FIG. 1 is shown as separate from the PEM 126 . However, the DC bus capacitor 308 may be integrated as part of the PEM 126 .

[0058] like Figure 3 As shown, a storage medium 312 such as a computer readable memory (hereinafter referred to as "memory") can store a computer program or algorithm embedded with or encoded with the method. In addition, the memory 312 can store data or information about various operating conditions or components in the PEM 126. For example, the memory 312 can store data or information about the current flowing through each phase bridge 316, 318, and 320. Figure 3 As shown, the memory 312 may be part of the controller 310. However, the memory 312 may be located in any suitable location accessible to the controller 310.

[0059] like Figure 3As shown, the controller 310 sends at least one control signal 236 to the power converter system 126. The power converter system 126 receives the control signal 236 to control the switching configuration of the inverter, thereby controlling the current flowing through each phase bridge 316, 318, and 320. The switching configuration is a set of switching states of the power switches 302 in the inverter. Generally speaking, the switching configuration of the inverter determines how the inverter converts power between the DC power link 306 and the motor 114.

[0060] To control the switching configuration of the inverter, the inverter changes the switching state of each power switch 302 in the inverter to a closed state or an open state based on the control signal 236. In the illustrated embodiment, to switch the power switches 302 to a closed state or an open state, the controller or LD 310 provides a gate voltage (Vg) to each power switch 302, thereby driving the switching state of each power switch 302. The gate voltage Vg a1 、Vg a2 、Vg b1 、Vg b2 、Vg c1 and Vg c2 (exist Figure 3 ) controls the switching state and characteristics of each power switch 302. Figure 3 302. Although shown as a voltage-driven device, the inverter can be a current-driven device or can be controlled by other strategies that switch the power switch 302 between a closed state and an open state. The controller 310 can change the gate drive of each IGBT based on the speed of the motor 114, the mirror current, or the temperature of the IGBT switch. The gate drive change can be selected based on multiple gate drive currents, in which the gate drive current change is proportional to the change in the IGBT switching speed.

[0061] Also like Figure 3As shown, each of the phase bridges 316, 318 and 320 includes two switches 302. However, only one switch in each of the phase bridges 316, 318 and 320 can be in a closed state without short-circuiting the DC power link 306. Therefore, in each phase bridge, the switching state of the lower switch is generally opposite to the switching state of the corresponding upper switch. The upper switches are generally referred to as high-side switches (i.e., 302A, 302B, 302C) and the lower switches are generally referred to as low-side switches (i.e., 302D, 302E, 302F). Therefore, the high state of the phase bridge refers to the upper switch in the phase bridge being in a closed state and the lower switch being in an open state. Similarly, the low state of the phase bridge refers to the upper switch of the phase bridge being in an open state and the lower switch being in a closed state. As a result, the IGBTs with current mirroring capability can be all IGBTs, a subset of IGBTs (e.g., S a1 、S b1 、S c1 ) or a single IGBT.

[0062] exist Figure 3 During the active state of the three-phase converter example shown in FIG, two situations may occur: (1) two phase bridges are in a high state while the third phase bridge is in a low state, or (2) one phase bridge is in a high state while the other two phase bridges are in a low state. Thus, one phase bridge in the three-phase converter (which can be defined as the "reference" phase for a given active state of the inverter) is in a state opposite to the states of the other two phase bridges (or "non-reference" phases) having the same state. Thus, the non-reference phases are either both in a high state or both in a low state during the active state of the inverter.

[0063] Figure 4 Figure 1 is a schematic diagram of a control circuit with a temperature estimation circuit. The gate driver / control circuit typically receives an input signal from a processor, controller, or other circuit, and its output is connected to the gate of the IGBT. The input signal can be shaped by a buffer stage consisting of switches (e.g., BJTs or MOSFETs). This shaping can include buffering, delaying, amplifying, or adjusting the rise / fall slope or pattern.

[0064] Figure 4 The circuit shown in FIG utilizes a current sense pin (e.g., a current mirror pin) of a solid-state switch (e.g., an IGBT) to operate a hybrid vehicle powertrain (e.g., a DC-DC converter / VVC or a traction inverter). During operation when the load current and the corresponding mirror current exceed a threshold (e.g., during a short circuit), the controller may disable the load switch. In addition, referring to FIG. Figure 3 , current sensors for EV / HEV traction inverters or EV / HEV DC-DC converters (e.g., Figure 3 Current sensor CS in a, CS b , and CS c or Figure 2 The current sensor 216 in the motor (e.g., 114) can be used to provide vector control of the motor (e.g., 114) or to provide direct torque control (e.g., Id and Iq). Here, the circuit is also configured to be based on the current sensor of the EV / HEV traction inverter or EV / HEV DC-DC converter (e.g., Figure 3 Current sensor CS in a , CS b , and CS c or Figure 2 The operation of the inverter and the DC / DC converter is compensated by the difference between the current sensor 216 in the current sensor 216 and the mirror current of the mirror switch.

[0065] Resistor R S The voltage signal at both ends is sent to the microprocessor through the signal processing 406 and the isolation circuit 412. At the same time, the controller 414 receives the signal output from the current sensor. The signal can be analog and received via the ADC, or it can be a digital signal received from the digital input / output pin 416. S The signal associated with the voltage across the terminals is compared with the output current sensor signal, and the controller 414 can obtain or adjust the IGBT junction temperature. Here, filtering 406 and signal isolation 412 can be accomplished by a digital circuit such as a DSP, where the filtering can be IIR, FIR, or other filter types.

[0066] Figure 4FIG2 is a schematic diagram of a powertrain controller and circuit 400 with a temperature estimation circuit. Powertrain control circuit 400 drives the gate of an IGBT 402 with a current mirror. IGBT 402 has a load switch and a mirror switch, with the load switch configured to drive load 408. The load in this diagram is shown connected to the emitter (where IGBT 402 functions as a high-side switch), but the load can also be connected to the collector so that IGBT 402 is configured as a low-side switch. Powertrain control circuit 400 includes a gate driver block 404 that directs current to and from the gate of IGBT 402. Gate driver block 404 can include a solid-state device (SSD) such as a metal oxide semiconductor field effect transistor (MOSFET), a bipolar junction transistor (BJT), an insulated gate bipolar junction transistor (IGBT), or other type of common gate drive switch. Control of the gate driver block 404 is performed by control logic 410 including a controller 414 (e.g., a processor, a controller, a microcontroller, combinational logic, or other circuit / electromechanical circuit), which is driven by an open-loop filtered comparator 406 that provides a signal isolator 412 that feeds the control logic 410. The control logic 410 is also regulated by negative feedback, which can be an analog signal 418 or a digital signal 416. The analog signal 418 or the digital signal 416 can come from a current sensor (e.g., Figure 3 Current sensor CS in a , CS b , and CS c or Figure 2 The current sensor 216 in FIG1 is used to generate the current signal and is received by the controller 414. The controller estimates the IGBT junction temperature based on the current sensor signal and the current mirror output. The controller can take protective action if the junction temperature is high.

[0067] The current mirror structure of IGBT 402 is usually to multiply the IGBT collector current i by a fractional value. cMirror image of the current flow IGBT monolithic integrated device. The current mirror output is typically used to detect overcurrent conditions and / or short circuit conditions. Here, the current mirror output is used to actively control the operation of the powertrain system. The control logic 410 is controlled by an open loop filter comparator 406, which is based on the output of the current mirror sense pin connected to a sense resistor Rs, which is used to convert the current signal into a voltage signal. Since the current mirror signal may have spikes during the switching transients of the IGBT 402, an RC filter can be used to filter the noise. Here, the filter is formed by Rf and Cf and is a low pass filter, wherein the filter rolls off at a frequency based on the frequency response of the capacitor, in other embodiments, the use of a band stop filter can be utilized. The output of the RC filter is compared to a reference voltage V corresponding to the current level threshold. ref The reference voltage can be compared to a constant value or can vary based on operation. For example, the reference voltage can be an output from the controller 410 (e.g., an integrated PWM signal or an output from a D / A converter), or the reference voltage can come from a variable voltage regulator controlled by the controller. When the current exceeds the threshold, the comparator outputs a high signal, which is detected by the control logic 410 after passing through the signal isolator 412. The controller 414 can output a signal to the powertrain and gate control logic 404.

[0068] Typically, the current mirror sensor is a small IGBT that is monolithically integrated (i.e., on-chip or monolithic) with the main IGBT. Therefore, the characteristics of the current mirror sensor are affected by temperature changes of the small IGBT and the main IGBT. When a constant V ref When used as an input to the filtered comparator 406, the mirror current threshold used to switch the comparator in block 406 will vary as temperature changes (eg, will be different at different temperatures).

[0069] Here, a novel IGBT junction temperature estimation method utilizes an output current sensor (e.g., from an inverter or DC / DC converter) and an IGBT current mirror sensor. This system offers higher accuracy than model-based junction temperature estimation methods. Furthermore, it can be combined with model-based methods. This system can also be implemented using low-cost components compared to on-die temperature sensing diodes, thereby reducing IGBT chip / manufacturing costs. Furthermore, this system uses the inverter output current sensor and IGBT current mirror sensor, which are already present in most traction inverters. Therefore, additional circuitry is limited to low-pass filters, comparators, and signal isolators.

[0070] Figure 4 The circuit in the paper implements the proposed output voltage signal V from the IGBT current mirror sensor. Rs With threshold V refComparative conception. Rs First, by R f and C f If the threshold is triggered, the signal is sent to the control logic 410 (e.g., a microprocessor, controller, or circuit). At the same time, the control logic (e.g., via the ADC (analog-to-digital converter) of the controller 414) reads the output current sensor (e.g., Figure 3 Current sensor CS in a , CS b , and CS c or Figure 2 The difference in the current received at this time (for example, the current received by the current mirror sensor and the current sensor) is equivalent to the different IGBT junction temperatures Tj. Figure 6 The Tj and V Rs An exemplary theoretical relationship between .

[0071] Figure 5 5 is a schematic diagram of an insulated gate bipolar junction transistor (IGBT) having a main IGBT 502, a current mirror IGBT 504, and a sense resistor 506. Here, the internal schematic diagram of the IGBT device 500 has the main IGBT 502 and the current mirror IGBT 504 connected in parallel. The emitter pin of the current mirror IGBT 504 is connected to the sense resistor (Rs) 506. When the voltage across Rs (i.e., V Rs ) reaches V ref When the current flowing through the current mirror is mirror It can be expressed by Equation 2 shown below:

[0072] i mirror_threshold =V ref / R s (2)

[0073] Figure 6 is a graphical representation 600 of the collector current 602 of the IGBT versus the voltage 604 between the collector and emitter and of the mirror current 606 of the mirror device monolithically integrated with the IGBT versus the voltage 608 between the collector and emitter. This diagram shows V ce Relative to i c The curve 610 and V ce_mirror +V Rs Relative to i mirror Curve 612 and V at -40°C ce Relative to i c Curve 614 and V ce_mirror +V Rs Relative to i mirrorThese different current thresholds are shown as i threshold_-40C and i threshold_150C IGBTs typically have a positive temperature coefficient. At elevated temperatures, the on-state voltage decreases, thereby increasing the collector-to-emitter current and the corresponding voltage V ce .like Figure 6 As shown, at high current levels, i threshold_-40C Greater than i threshold_150C .

[0074] Applying these to conventional gate drive circuits will disable fast switching at higher current levels when the temperature is low. Since the di / dt (e.g., surge voltage) of the cutoff at higher current levels may be higher, the operation of conventional gate drive circuits is not preferred. In addition, at lower temperatures, the device breakdown voltage is lower. In order to compensate for the effect of temperature on the characteristics of the mirror IGBT, Figure 4 The circuit in the figure adjusts V for different temperatures. ref . Especially reducing V at lower temperatures ref This will result in a lower current threshold at lower temperatures.

[0075] The control logic or functions performed by the controller may be represented by a flowchart or similar diagram in one or more of the accompanying figures. These figures provide representative control strategies and / or logic that can be implemented using one or more processing strategies (such as event-driven, interrupt-driven, multi-tasking, multi-threading, etc.). Therefore, the various steps or functions shown may be performed in the order shown, in parallel, or omitted in some cases. Although not always explicitly shown, those skilled in the art will recognize that one or more of the steps or functions shown may be repeated depending on the specific processing strategy used. Similarly, the order of processing is not necessarily required to achieve the functions and advantages described herein, but is provided for ease of illustration and description. The control logic may be implemented primarily in the form of software executed by a microprocessor-based vehicle, engine, and / or powertrain controller (such as a controller). Of course, the control logic may be implemented in the form of software, hardware, or a combination of software and hardware in one or more controllers, depending on the specific application. When implemented in software, the control logic may be provided in one or more computer-readable storage devices or media that have stored data representing code or instructions executed by a computer to control the vehicle or its subsystems. The computer-readable storage device or medium may include one or more of a number of known physical devices that utilize electronic, magnetic, and / or optical storage to store executable instructions and associated calibration information, operating variables, and the like.

[0076] The processes, methods or algorithms disclosed herein may be transmitted to a processing device, a controller or a computer, or implemented by the processing device, a controller or a computer, wherein the processing device, the controller or the computer may include any existing programmable electronic control unit or a dedicated electronic control unit. Similarly, the processes, methods or algorithms may be stored in a variety of forms as data and instructions that can be executed by a controller or a computer, wherein the various forms include, but are not limited to, information permanently stored in a non-writable storage medium (such as a read-only memory (ROM) device) and information variably stored in a writable storage medium (such as a floppy disk, a magnetic tape, a compact disk (CD), a random access memory (RAM) device, and other magnetic and optical media). The processes, methods or algorithms may also be implemented in a software executable object. Alternatively, the processes, methods or algorithms may be implemented in whole or in part using appropriate hardware components (such as an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a state machine, a controller or other hardware components or devices) or a combination of hardware components, software components and firmware components.

[0077] Although exemplary embodiments have been described above, it is not intended that these embodiments describe all possible forms encompassed by the claims. The words used in the specification are descriptive rather than restrictive, and it should be understood that various changes may be made without departing from the spirit and scope of the present disclosure. As previously mentioned, the features of the various embodiments may be combined to form further embodiments of the present invention that may not be explicitly described or shown. Although each embodiment may have been described as providing advantages or being superior to other embodiments or prior art embodiments for one or more desired characteristics, it will be appreciated by those skilled in the art that, depending on the specific application and embodiment, one or more features or characteristics may be compromised to achieve the desired overall system properties. These properties may include, but are not limited to, cost, strength, durability, life cycle cost, marketability, appearance, packaging, size, maintainability, weight, manufacturability, ease of assembly, etc. Therefore, embodiments described as being inferior to other embodiments or prior art embodiments in one or more characteristics are not outside the scope of the present disclosure and may be expected to be used for specific applications.

Claims

1. A vehicle power transmission system comprising: a power inverter including a load switch and a current sensor, the load switch being monolithically integrated with a mirror switch, the current sensor being configured to measure current flowing through the load switch to provide current feedback for vector control of a motor; The controller is configured to operate the power inverter to reduce the temperature of the load switch based on a difference between a mirror current of the mirror switch generated due to a temperature change of the load switch and an output of the current sensor.

2. The vehicle powertrain system of claim 1, wherein: The controller is further configured to disable the load switch in response to the mirror current exceeding a threshold.

3. The vehicle powertrain system of claim 1, wherein: Operating the power inverter to reduce the temperature of the load switch includes reducing a pulse width modulation duty cycle of the load switch.

4. The vehicle powertrain system of claim 1, wherein: Operating the power inverter to reduce the temperature of the load switch includes reducing a torque demand of a vehicle powertrain.

5. The vehicle powertrain system of claim 1, wherein: Operating the power inverter to reduce the temperature of the load switch includes reducing a switching frequency of the load switch to reduce a switching time of the load switch.

6. The vehicle powertrain system of claim 1, wherein: Operating the power inverter to reduce the temperature of the load switch includes disabling the load switch and providing propulsion via the internal combustion engine.

7. The vehicle powertrain system of claim 1, wherein: The current sensor is a Hall effect sensor or a giant magnetoresistive sensor.

8. The vehicle powertrain system of claim 1, wherein: The difference is based on the mirror switch voltage measured across a resistor that sinks the mirror current.

9. The vehicle powertrain system of claim 1, wherein: The load switch and the mirror switch are insulated gate bipolar junction transistors.

10. A method of controlling a vehicle powertrain system, comprising: disabling an inverter switch in response to the mirror current exceeding a threshold, the inverter switch being monolithically integrated with the mirror switch that enables the mirror current to flow; controlling a vehicle powertrain system based on a signal from an inverter current sensor; Based on a difference between the mirror current generated due to temperature variation of the inverter switch and the signal, the vehicle powertrain is operated to reduce the temperature of the inverter switch.

11. The method according to claim 10, wherein: The control of the vehicle's powertrain is vector control.

12. The method of claim 10, further comprising: Filter the mirror current.

13. The method of claim 10, wherein: Operating the vehicle powertrain to reduce a temperature of an inverter switch includes reducing a pulse width modulation duty cycle of a load switch of the vehicle powertrain.

14. The method of claim 10, wherein: Operating the vehicle powertrain to reduce a temperature of an inverter switch includes reducing a switching frequency of a load switch of the vehicle powertrain to reduce a switching time of the load switch.

15. The method of claim 10, wherein: Operating the vehicle powertrain to reduce a temperature of an inverter switch is based on a model-based junction temperature estimation model that accumulates an error over time, the method further comprising adjusting the model-based junction temperature estimation model based on the difference to minimize the error.

Images