LLC resonant converter and control circuit and control method thereof

Abstract

The application discloses an LLC resonant converter and a control circuit and a control method thereof. The LLC resonant converter comprises a control circuit, a first transformer and a second transformer, the first driving winding and the second driving winding of the second transformer are magnetically coupled with a load winding, and the control circuit is used for controlling the conduction state of a path between the same name end and the opposite name end of the first driving winding, thereby controlling the conduction state of the first bipolar transistor and the second bipolar transistor to adjust the resonant frequency. The control circuit of the LLC resonant converter can realize consistent turn-off time of the two bipolar transistors, and reduce the size of the control switch tube.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of power supply, more particularly, to an LLC resonant converter and a control circuit and a control method thereof. BACKGROUND

[0002] A resonant switching converter is a power converter that uses a controlled switch to obtain a square wave voltage and a resonant tank to achieve energy transfer. An LLC resonant converter is a third-order resonant network composed of inductors and capacitors, which can achieve load regulation from full load to no load in a narrow frequency range. The LLC resonant converter has high power density and fewer electronic components, and has a smooth current waveform, which is beneficial to improve electromagnetic interference and can achieve zero voltage switching (ZVS) and zero current switching (ZCS) of the control switch in the entire operating range, which helps to obtain extremely high efficiency, and is widely used.

[0003] The LLC resonant converter includes first and second power tubes that are alternately turned on and off to convert a direct current input voltage into a square wave voltage. The square wave voltage is input to a resonant tank to generate a resonant current at a resonant frequency. The first and second power tubes are, for example, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) or bipolar transistors. Using MOSFETs as power tubes can achieve excellent switching performance, but the control circuit is complex, resulting in high circuit cost. Using bipolar transistors as power tubes can significantly reduce circuit cost.

[0004] In the LLC resonant converter, a transformer coupled to the resonant tank is used to obtain the drive current of the bipolar transistor, thereby achieving self-oscillating converter (SOC). In order to adjust the switching frequency of the bipolar transistor, a control circuit is used to provide an additional path between the base-emitter of one bipolar transistor, thereby forcing the bipolar transistor to switch from the on state to the off state in advance, so that the LLC resonant converter can adjust the resonant current to obtain the desired direct current output voltage / current.

[0005] However, in the above existing LLC resonant converter, the control circuit only provides a path for one bipolar transistor, and the other bipolar transistor is turned off by the magnetic coupling of the two driving windings of the transformer. Therefore, the turn-off time of the two bipolar transistors is not consistent, causing the problem of asymmetry of the system, and further causing the problem of magnetic bias of the current transformer. In addition, the bipolar transistor works in the saturation region in the on state, and during the control of the turn-off of the bipolar transistor in the short-circuit mode, current needs to be extracted from the base of the bipolar transistor to exit the saturation region. In order to achieve fast turn-off, the on-resistance of the control switch tube needs to be small to obtain a large current, resulting in a large size of the control switch tube.

[0006] It is desirable to further improve the control method of the LLC resonant converter to achieve consistent turn-off time of the two bipolar transistors and reduce the size of the control switch tube. SUMMARY

[0007] In view of the above problems, the present application provides an LLC resonant converter and a control circuit and control method thereof, wherein the control circuit is used to control the on state of the path between the same name end and the opposite name end of the first driving winding to control the on state of the two bipolar transistors, so as to achieve consistent turn-off time of the two bipolar transistors and reduce the size of the control switch tube.

[0008] According to a first aspect of the present application, an LLC resonant converter is provided, comprising: a first transformer comprising a primary winding and a secondary winding; a second transformer comprising a load winding, and a first driving winding and a second driving winding magnetically coupled with the load winding; and a first bipolar transistor and a second bipolar transistor connected in series with each other between a positive power supply terminal and a negative power supply terminal of a direct current input voltage, and a load winding of the second transformer, a resonant element and the primary winding are connected between an intermediate node of the first bipolar transistor and the second bipolar transistor and the negative power supply terminal, the resonant element is connected with the primary winding to form a resonant loop to generate a resonant current, and a resonant output voltage is provided across the primary winding, wherein the LLC resonant converter further comprises a control circuit for controlling the on state of the path between the same name end and the opposite name end of the first driving winding, thereby controlling the on state of the first bipolar transistor and the second bipolar transistor to adjust the resonant frequency.

[0009] Preferably, the first drive winding comprises a same-name end, an opposite-name end and a tap end, the second drive winding comprises a same-name end and an opposite-name end, the base of the first bipolar transistor is connected to the tap end of the first drive winding to receive a first drive current generated according to an induced current of the resonant current, and the base of the second bipolar transistor is connected to the opposite-name end of the second drive winding to receive a second drive current generated according to the induced current of the resonant current.

[0010] Preferably, the first bipolar transistor and the second bipolar transistor are NPN bipolar transistors respectively, the emitter of the first bipolar transistor and the collector of the second bipolar transistor are commonly connected to the intermediate node, the opposite-name end of the first drive winding is connected to the emitter of the first bipolar transistor, and the same-name end of the second drive winding is connected to the emitter of the second bipolar transistor.

[0011] Preferably, the first control switch and the second control switch are reversely connected in series between the same-name end and the opposite-name end of the first drive winding, wherein the control circuit generates a clock signal according to the current sampling signal of the resonant current and / or the voltage feedback signal of the resonant output voltage, and generates a first gate drive signal and a second gate drive signal according to the clock signal, thereby controlling the first control switch and the second control switch to periodically short the same-name end and the opposite-name end of the first drive winding, so that the resonant period of the LLC resonant converter follows the clock signal.

[0012] Preferably, the first control switch and the second control switch are N-type MOSFETs respectively, the drain of the first control switch and the drain of the second control switch are connected to each other, the source of the first control switch is connected to the same-name end of the first drive winding, the source of the second control switch is connected to the opposite-name end of the first drive winding, and the source of the second control switch is grounded.

[0013] Preferably, the gate of the first control switch receives the first gate drive signal, and the gate of the second control switch receives the second gate drive signal.

[0014] Preferably, the first control switch and the second control switch are N-type MOSFETs respectively, the source of the first control switch and the source of the second control switch are connected to each other, the drain of the first control switch is connected to the same-name end of the first drive winding, the drain of the second control switch is connected to the opposite-name end of the first drive winding, and the intermediate node of the first control switch and the second control switch is grounded.

[0015] Preferably, the gate of the first control switch tube receives a second gate drive signal, and the gate of the second control switch tube receives a first gate drive signal.

[0016] Preferably, the control circuit further comprises: a sampling resistor connected in series with the primary winding of the first transformer, and one end of the sampling resistor connected to the intermediate node of the first bipolar transistor and the second bipolar transistor, wherein a current sampling signal of the resonant current is obtained across the sampling resistor.

[0017] Preferably, the control circuit comprises: a clock signal generation module configured to generate the clock signal with a corresponding clock period according to the current sampling signal of the resonant current and / or a voltage feedback signal of the resonant output voltage; a logic module connected to the clock signal generation module and configured to generate the first control signal and the second control signal according to the clock signal; a first drive module connected to the logic module and configured to generate the first gate drive signal according to the first control signal; and a second drive module connected to the logic module and configured to generate the second gate drive signal according to the second control signal.

[0018] Preferably, the control circuit further comprises: a first comparator configured to compare the current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and a second comparator configured to compare the current sampling signal of the resonant current with a second current threshold to obtain a second comparison signal, wherein the logic module is connected to the first comparator to obtain the first comparison signal, and connected to the second comparator to obtain the second comparison signal, the logic module controls the first gate drive signal to be in an active state when the clock signal is invalid or the current sampling signal is less than the first current threshold, and controls the second gate drive signal to be in an active state when the clock signal is active or the current sampling signal is greater than the second current threshold.

[0019] Preferably, the control circuit further comprises: a first current source and a switch tube connected in series between the power supply end and the same-named end of the first drive winding; and a one-shot module having a first input end connected to the first comparator to obtain the first comparison signal, a second input end connected to the second comparator to obtain the second comparison signal, and an output end connected to the switch tube to provide a start signal, wherein during start-up of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the start signal is active, the switch tube is turned on, the first current source injects an excitation current into the first drive winding via the switch tube, and the first control switch tube and the second control switch tube are turned off.

[0020] Preferably, the logic module comprises: a NOT gate for obtaining an inverted signal of the clock signal; a first OR gate, a first input end of which receives the inverted signal of the clock signal, and a second input end of which receives the first comparison signal; a first AND gate, a first input end of which is connected to an output end of the first OR gate, and a second input end of which receives the start signal; a second OR gate, a first input end of which receives the clock signal, and a second input end of which receives the second comparison signal; and a second AND gate, a first input end of which is connected to an output end of the second OR gate, and a second input end of which receives the start signal, wherein the start signal is valid at a low level.

[0021] Preferably, the first drive module is connected to a same-name end of the first drive winding, the first control signal is level-shifted with respect to the same-name end to obtain the first gate drive signal, and the second drive module is connected to an opposite-name end of the first drive winding, the second control signal is level-shifted with respect to the opposite-name end to obtain the second gate drive signal.

[0022] Preferably, the clock signal generation module comprises: a compensation module and a first capacitor connected to an output end of the compensation module, a compensation signal being generated across the first capacitor; an oscillator for generating an oscillation signal of a corresponding frequency according to the compensation signal; and a frequency division module for generating the clock signal with a 50% duty cycle according to the oscillation signal, wherein the compensation module compares the current sampling signal with a current reference signal and / or compares the voltage feedback signal with a voltage reference signal to generate the compensation signal.

[0023] Preferably, the control circuit and the first control switch tube and the second control switch tube are integrated into a single chip.

[0024] According to a second aspect of the present application, there is provided a control circuit for an LLC resonant converter, the LLC resonant converter comprising a first driving winding coupled with a resonant tank, the control circuit comprising: a clock signal generating module configured to generate a clock signal having a corresponding clock period according to a current sampling signal of a resonant current and / or a voltage feedback signal of a resonant output voltage; a logic module connected with the clock signal generating module and configured to generate a first control signal and a second control signal according to the clock signal; a first driving module connected with the logic module and configured to generate a first gate driving signal according to the first control signal; and a second driving module connected with the logic module and configured to generate a second gate driving signal according to the second control signal, wherein the control circuit controls a first control switch and a second control switch to periodically short a same end and an opposite end of the first driving winding by using the first gate driving signal and the second gate driving signal, so as to generate a driving current at a tap end of the first driving winding, and make a resonant period of the LLC resonant converter follow the clock signal.

[0025] Preferably, the control circuit further comprises: a first comparator configured to compare the current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and a second comparator configured to compare the current sampling signal of the resonant current with a second current threshold to obtain a second comparison signal, wherein the logic module is connected with the first comparator to obtain the first comparison signal, and connected with the second comparator to obtain the second comparison signal, and the logic module controls the first gate driving signal to be in an active state when the clock signal is inactive or the current sampling signal is less than the first current threshold, and controls the second gate driving signal to be in the active state when the clock signal is active or the current sampling signal is greater than the second current threshold.

[0026] Preferably, the control circuit further comprises: a first current source and a switch connected in series between a power supply end and the same end of the first driving winding; and a single trigger module having a first input connected with the first comparator to obtain the first comparison signal, a second input connected with the second comparator to obtain the second comparison signal, and an output connected with the switch to provide a start signal, wherein during start-up of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the start signal is active, the switch is turned on, the first current source injects an excitation current into the first driving winding via the switch, and the first control switch and the second control switch are turned off.

[0027] Preferably, the logic module comprises: a NOT gate for obtaining an inverted signal of the clock signal; a first OR gate, a first input end of which receives the inverted signal of the clock signal, and a second input end of which receives the first comparison signal; a first AND gate, a first input end of which is connected to an output end of the first OR gate, and a second input end of which receives the start signal; a second OR gate, a first input end of which receives the clock signal, and a second input end of which receives the second comparison signal; and a second AND gate, a first input end of which is connected to an output end of the second OR gate, and a second input end of which receives the start signal, wherein the start signal is valid at a low level.

[0028] Preferably, the first driving module is connected to a same-name end of the first driving winding, the first control signal is level-shifted with respect to the same-name end to obtain the first gate driving signal, and the second driving module is connected to an opposite-name end of the first driving winding, the second control signal is level-shifted with respect to the opposite-name end to obtain the second gate driving signal.

[0029] Preferably, the clock signal generation module comprises: a compensation module and a first capacitor connected to an output end of the compensation module, a compensation signal being generated across the first capacitor; an oscillator for generating an oscillation signal of a corresponding frequency according to the compensation signal; and a frequency division module for generating the clock signal of 50% duty cycle according to the oscillation signal, wherein the compensation module compares the current sampling signal with a current reference signal and / or compares the voltage feedback signal with a voltage reference signal to generate the compensation signal, so that the frequency of the clock signal is adjusted in relation to the resonant current and / or the resonant output voltage.

[0030] Preferably, the first control switch tube and the second control switch tube are N-type MOSFETs, the drain of the first control switch tube and the drain of the second control switch tube are connected to each other, the source of the first control switch tube is connected to the same-name end of the first driving winding, the source of the second control switch tube is connected to the opposite-name end of the first driving winding, and the source of the second control switch tube is grounded, wherein the gate of the first control switch tube receives a first gate driving signal, and the gate of the second control switch tube receives a second gate driving signal.

[0031] Preferably, the first control switch tube and the second control switch tube are N-type MOSFETs, the source of the first control switch tube and the source of the second control switch tube are connected to each other, the drain of the first control switch tube is connected to the same terminal of the first driving winding, the drain of the second control switch tube is connected to the different terminal of the first driving winding, and the intermediate node of the first control switch tube and the second control switch tube is grounded, wherein the gate of the first control switch tube receives a second gate drive signal, and the gate of the second control switch tube receives a first gate drive signal.

[0032] Preferably, the control circuit and the first control switch tube and the second control switch tube are integrated into a single chip.

[0033] According to a third aspect of the present application, a control method for an LLC resonant converter is provided, the LLC resonant converter comprising a first driving winding coupled with a resonant tank, the control method comprising: generating a clock signal with a corresponding clock period according to a current sampling signal of a resonant current and / or a voltage feedback signal of a resonant output voltage; generating a first control signal and a second control signal according to the clock signal; generating a first gate drive signal and a second gate drive signal according to the first control signal and the second control signal, respectively; and controlling a first control switch tube and a second control switch tube to periodically short the same terminal and the different terminal of the first driving winding using the first gate drive signal and the second gate drive signal, so as to generate a driving current at a tap terminal of the first driving winding, and make a resonant period of the LLC resonant converter follow the clock signal.

[0034] Preferably, the first gate drive signal is in an active state when the clock signal is inactive or the current sampling signal is less than a first current threshold, and the second gate drive signal is in an active state when the clock signal is active or the current sampling signal is greater than a second current threshold.

[0035] Preferably, the control method further comprises: during startup of the LLC resonant converter, injecting an excitation current to the first driving winding via a switch tube, and turning off the first control switch tube and the second control switch tube.

[0036] According to the LLC resonant converter of the embodiments of the present application, the control circuit controls the conduction state of the path between the same terminal and the different terminal of the first driving winding, thereby controlling the conduction state of the first bipolar transistor and the second bipolar transistor to adjust the resonant frequency. The LLC resonant converter can achieve consistent turn-off time of the two bipolar transistors, and reduce the size of the control switch tube.

[0037] In a preferred embodiment, the first bipolar transistor obtains a first drive current generated according to an induced current of the resonant current from a tap end of the first drive winding, and the second bipolar transistor obtains a second drive current generated according to an induced current of the resonant current from a opposite end of the second drive winding, to realize a self-oscillating converter (SOC). Under the control of the self-oscillating drive signal, the first bipolar transistor and the second bipolar transistor are alternately turned on and turned off to convert the direct current input voltage into a square wave voltage. The square wave voltage is input into the resonant circuit to generate the resonant current at the resonant frequency. Therefore, through the resonant circuit, the electric energy is transmitted from the primary side of the first transformer to the secondary side of the first transformer. The LLC resonant converter generates the drive current of the first bipolar transistor and the second bipolar transistor based on the induced current of the first drive winding, which can further improve the consistency of the turn-off time of the two bipolar transistors.

[0038] In a preferred embodiment, the control circuit generates a compensation signal according to the current sampling signal of the resonant current and / or the voltage feedback signal of the resonant output voltage to adjust the clock period of the clock signal, thereby controlling the switching period of the first control switch tube and the second control switch tube, to provide an additional path and excitation current for the first drive winding, and further control the on-off state of the first bipolar transistor and the second bipolar transistor via the magnetic coupling of the first drive winding and the second drive winding, to realize the consistent turn-off time of the two bipolar transistors, thereby adjusting the resonant frequency. Therefore, the LLC resonant converter can adjust the resonant current in the manner of adjusting the clock period to obtain the desired direct current output voltage / current, thereby realizing constant current or constant voltage control, and simplifying the control circuit and reducing the circuit cost. BRIEF DESCRIPTION OF DRAWINGS

[0039] The above and other objects, features and advantages of the present application will become more apparent from the following description of the preferred embodiments of the present application taken in conjunction with the accompanying drawings, in which:

[0040] Figure 1 A schematic circuit diagram of an LLC resonant converter according to the prior art is shown.

[0041] Figure 2 A schematic circuit diagram of an LLC resonant converter according to a first embodiment of the present application is shown.

[0042] Figure 3 A schematic circuit diagram of an LLC resonant converter according to a first embodiment of the present application is shown. Figure 2 A schematic circuit diagram of the control circuit of the LLC resonant converter shown.

[0043] Figure 4 A schematic circuit diagram of the control circuit of the LLC resonant converter shown. Figure 2 An equivalent circuit diagram of the LLC resonant converter shown during startup.

[0044] Figure 5 shows Figure 2 Waveform diagram showing the LLC resonant converter operating in self-oscillation mode.

[0045] Figures 6a to 6j shows Figure 2 Equivalent circuit diagram showing the LLC resonant converter operating in different stages in self-oscillation mode.

[0046] Figure 7 shows

[0047] Figure 8 shows Figure 7 Schematic circuit diagram showing the control circuit in the LLC resonant converter. DETAILED DESCRIPTION

[0048] Various embodiments of the present application will be described in greater detail below with reference to the accompanying drawings. In the various drawings, like elements are denoted by like or similar reference numerals for the sake of clarity. Various parts in the drawings are not drawn to scale.

[0049] Figure 1 shows The LLC resonant converter 10 comprises a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D1 and D2, a resonant capacitor Cr, an output capacitor Co and a resonant inductor Lr.

[0050] At the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr and the resonant inductor Lr form a resonant tank. Between the positive input terminal and the negative input terminal of the LLC resonant converter 10, the bipolar transistors Q1 and Q2 are connected in series, with the middle node connected to the resonant tank. The second transformer T2 comprises three windings, i.e. a load winding W1, a drive winding W2 and W3, around the same core. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Meanwhile, the drive windings W2 and W3 are coupled to the bases of the bipolar transistors Q1 and Q2 respectively, but in opposite directions. That is, the same terminal of the drive winding W2 is connected to the base of the bipolar transistor Q1, and the opposite terminal of the drive winding W3 is connected to the base of the bipolar transistor Q2.

[0051] At the secondary side of the first transformer T1, the diodes D1 and D2 form a rectifier circuit. The two terminals of the secondary winding Ls are connected to the anodes of the diodes D1 and D2 respectively, and the middle tap of the secondary winding Ls is connected to ground. The output capacitor Co is connected between the cathodes of the diodes D1 and D2 and ground, and provides a DC output voltage across its two terminals to the load Rd.

[0052] The LLC resonant converter 10 is a topology of self-oscillating LLC half-bridge drive. The windings of the second transformer T2 are used to provide drive current to the bases of the bipolar transistors Q1 and Q2 to achieve self-oscillating drive (SOC). Under the control of the self-oscillating drive signal, the bipolar transistors Q1 and Q2 are alternately turned on and off to convert the DC input voltage into a square wave voltage. The square wave voltage is input to the resonant tank to generate a resonant current at the resonant frequency. Thus, through the resonant tank, the power is transferred from the primary side of the first transformer T1 to the secondary side of the first transformer T1.

[0053] The operating frequency of the LLC resonant converter 10 according to the prior art is the inherent SOC oscillation frequency, and thus the regulation of the DC output voltage / current cannot be achieved.

[0054] Figure 2 A schematic circuit diagram of an LLC resonant converter according to a first embodiment of the present application is shown. The LLC resonant converter 20 comprises a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D1 and D2, a resonant capacitor Cr, a resonant inductor Lr, control switches M1 and M2, and a control circuit 100.

[0055] On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr and the resonant inductor Lr form a resonant tank. Between the positive input terminal and the negative input terminal of the LLC resonant converter 20, the bipolar transistors Q1 and Q2 are connected in series, with the middle node connected to the resonant tank. In the resonant tank, a sampling resistor Rs is connected in series with the primary winding Lp, so that a sampling signal for characterizing the inductive current flowing through the primary winding Lp can be obtained.

[0056] On the secondary side of the first transformer T1, the diodes D1 and D2 form a rectifier circuit. The anodes of the diodes D1 and D2 are respectively connected to the two ends of the secondary winding, and the middle tap of the secondary winding is connected to ground. The output capacitor Co is connected between the cathodes of the diodes D1 and D2 and ground, and provides a DC output voltage across its two terminals to the load Rd.

[0057] The second transformer T2 comprises three windings around the same core, i.e. a load winding W1, drive windings W2 and W3. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Meanwhile, the drive windings W2 and W3 are respectively coupled to the bases of the bipolar transistors Q1 and Q2, but in opposite directions. For example, the drive winding W2 has a same-name end, an opposite-name end and a tap end between the two, and the drive winding W3 has a same-name end and an opposite-name end. The same-name end DR1 of the drive winding W2 is connected to the current terminal of the control circuit 100 to obtain the excitation current I1.

[0058] The control switch tubes M1 and M2 are connected in reverse series between the same end DR1 and the different end DR2 of the drive winding W2 of the second transformer T2. The control switch tubes M1 and M2 are, for example, N-type MOSFETs (Metal-Oxide-Semiconductor Field Effect Transistor) respectively, and each include a body diode connected in reverse. The drains of the control switch tubes M1 and M2 are connected to each other, the source of the control switch tube M1 is connected to the same end DR1 of the drive winding W2, and the source of the control switch tube M2 is connected to the different end DR2 of the drive winding W2.

[0059] The intermediate nodes of the bipolar transistors Q1 and Q2 are connected to the different end DR2 of the drive winding W2. Specifically, the base and the emitter of the bipolar transistor Q1 are connected to the tap end and the different end DR2 of the drive winding W2 respectively. The base and the emitter of the bipolar transistor Q2 are connected to the different end and the same end of the drive winding W3 respectively.

[0060] The turns ratio of the second transformer T2 is selected according to the electrical characteristics of the bipolar transistors Q1 and Q2. In this embodiment, the turns ratio of the load winding W1 : the drive winding W2 : the drive winding W3 of the second transformer T2 is 1 : 18 : 6, wherein the tap end of the drive winding W2 divides the drive winding W2 into a first sub-winding between the same end DR1 and the tap end and a second sub-winding between the tap end and the different end DR2, and the turns ratio of the first sub-winding : the second sub-winding is 12 : 6. In an alternative embodiment, the turns ratio of the load winding W1 : the drive winding W2 : the drive winding W3 of the second transformer T2 is 1 : 24 : 6 or 2 : 36 : 12. The fixed turns ratio of the first sub-winding and the second sub-winding of the drive winding W2 causes the drive voltage of the bipolar transistor Q1 to be in a fixed proportional relationship with the terminal voltage of the drive winding W2 (i.e. the voltage between the same end DR1 and the different end DR2).

[0061] For example, in a state where only one of the control switch tubes M1 and M2 is on, the terminal voltage of the drive winding W2 is the forward voltage drop of the body diode of the control switch tube, and thus the terminal voltage of the drive winding W2 is 0.7 V, the drive signal of the bipolar transistor Q1 is clamped at 0.7 / 3 V, so that the bipolar transistor Q1 maintains an off state.

[0062] The control circuit 100 obtains a current sampling signal CS of the resonant current from a sampling resistor Rs of the resonant converter 20, obtains a voltage feedback signal FB of the resonant output voltage from an auxiliary winding La of the first transformer Tl of the resonant converter 20, generates a gate drive signal VG1 of the control switch Ml and a gate drive signal VG2 of the control switch M2 according to the current sampling signal CS and the voltage feedback signal FB, and generates the excitation current II. In this embodiment, the ground of the control circuit 100, the intermediate node of the bipolar transistors Ql and Q2, and the same-named end of the auxiliary winding La are grounded together.

[0063] Under the control of the control circuit 100, the control switches Ml and M2 have three working modes, respectively, double-off, double-on, and single-on. In the starting phase of the LLC resonant converter, the control switches Ml and M2 are in the double-off state, and the drive winding W2 of the second transformer T2 of the resonant converter 20 injects the excitation current, thereby starting the self-oscillating working mode. In the self-oscillating working phase of the LLC resonant converter, the control switches Ml and M2 are in the double-on or single-on state, thereby forcing the advance switching of the on and off states of the bipolar transistors Ql and Q2, so that the LLC resonant converter can obtain the desired DC output voltage / current by adjusting the resonant current.

[0064] According to the LLC resonant converter 20 of this embodiment, the drive windings W2 and W3 are used to provide drive currents to drive the bases of the bipolar transistors Ql and Q2 to realize self-oscillating converter (SOC). Under the control of the self-oscillating drive signal, the bipolar transistors Ql and Q2 are alternately turned on and off to convert the DC input voltage into a square wave voltage. This square wave voltage is input into the resonant circuit to generate a resonant current at the resonant frequency. Therefore, through the resonant circuit, the electric energy is transmitted from the primary side of the first transformer Tl to the secondary side of the first transformer Tl.

[0065] The control circuit 100 controls the on state of the control switches Ml and M2, thereby providing an additional path and excitation current for the drive winding W2, and further controls the on state of the bipolar transistors Ql and Q2 via the magnetic coupling of the drive windings W2 and W3 to realize the consistent off time of the two bipolar transistors, thereby adjusting the resonant frequency. Therefore, the LLC resonant converter 20 can obtain the desired DC output voltage / current by adjusting the resonant frequency, thereby realizing constant current control, and simplifying the control circuit and reducing the circuit cost.

[0066] Figure 3 It is shown that Figure 2A schematic circuit diagram of a control circuit in the illustrated LLC resonant converter. The control circuit 100 comprises a clock signal generating module 110, a logic module 120, comparators A12 and A13, a monoflop module 104, drive modules 105 and 106, a current source Is1, a switch tube M3. The clock signal generating module 110 comprises a compensation module 101, an oscillator 111, a frequency dividing module 103, a capacitor Cc. The logic module 120 comprises a NOT gate A14, OR gates A15 and A16, AND gates A17 and A18.

[0067] The compensation module 101 receives a current sampling signal CS and / or a voltage feedback signal FB, and generates a compensation signal COMP across the capacitor Cc according to the comparison result of the current sampling signal CS and a current reference signal, and / or according to the comparison result of the voltage feedback signal FB and a voltage reference signal. In constant current control mode, the compensation signal COMP increases if the DC output current is too small, and decreases if the DC output current is too large. In constant voltage control mode, the compensation signal COMP increases if the DC output voltage is too small, and decreases if the DC output voltage is too large.

[0068] The oscillator 111 further comprises a comparator A11, a current source Is2, a capacitor C11. The current source Is2 and the capacitor C11 are connected in series between a power supply terminal and a ground terminal, and the intermediate node provides a triangular wave signal. The non-inverting input terminal of the comparator A11 receives the triangular wave signal, and the inverting input terminal receives the compensation signal COMP, and the output terminal provides an oscillation signal OSC. The frequency of the oscillation signal OSC is related to the compensation signal COMP, and the frequency of the oscillation signal OSC is low when the compensation signal COMP is high, and the frequency of the oscillation signal OSC is high when the compensation signal COMP is low. The input terminal of the frequency dividing module 103 receives the oscillation signal OSC, and the output terminal provides a clock signal CLK. The frequency dividing module 103 generates the clock signal CLK with 50% duty cycle according to the oscillation signal OSC.

[0069] The comparator A12 compares the current sampling signal CS of the resonant current with a first current threshold (e.g. -100mV) to obtain a first comparison signal, and the comparator A13 compares the current sampling signal CS of the resonant current with a second current threshold (e.g. +100mV) to obtain a second comparison signal. The monoflop module 104 judges whether the current sampling signal CS is located in a predetermined current interval according to the first comparison signal and the second comparison signal, and generates an enable signal Ts.

[0070] During the startup of the LLC resonant converter 20, the current sampling signal CS is located in the predetermined current interval, and the enable signal Ts is in the active state of low level.

[0071] In this embodiment, the control switch tubes Ml and M2 are N-type MOSFETs, and the switch tube M3 is a P-type MOSFET. The start-up signal Ts is active at low level, so that the switch tube M3 is turned on. At the same time, the start-up signal Ts is one of the input signals of the AND gates A17 and A18, so that the gate drive signals VG1 and VG2 provided by the drive modules 105 and 106 are disabled, and the control switch tubes Ml and M2 are turned off. The current source Isl is connected in series with the switch tube M3 between the power supply terminal and the common terminal DRl of the drive winding W2.

[0072] Therefore, during the start-up of the LLC resonant converter 20, the current source Isl injects the excitation current I1 to the common terminal DRl of the drive winding W2 via the switch tube M3, so that the LLC resonant converter 20 starts to operate in the self-oscillation mode. The equivalent circuit diagram of the LLC resonant converter 20 during the start-up is shown in Fig. 4. Figure 4

[0073] Further, after the LLC resonant converter 20 starts to operate in the self-oscillation mode, the current sampling signal CS of the resonant current exceeds the above-mentioned predetermined current range in the fixed time of a plurality of switching periods, so that the start-up signal Ts maintains the inactive state at high level. Since the switch tube M3 is turned off, the current source Isl stops injecting the excitation current I1 to the common terminal DRl of the drive winding W2 via the switch tube M3.

[0074] The control switch tubes Ml and M2 are controlled according to the clock signal CLK to turn on or off. As shown in the figure, the first input terminal of the OR gate A15 is connected to the output terminal of the frequency division module 103 via the inverter A14 to receive the inverted signal of the clock signal CLK, and the second input terminal is connected to the output terminal of the comparator A12 to receive the first comparison signal; the first input terminal of the OR gate A16 is connected to the output terminal of the frequency division module 103 to receive the clock signal CLK, and the second input terminal is connected to the output terminal of the comparator A13 to receive the second comparison signal. The first input terminal of the AND gate A17 is connected to the output terminal of the OR gate A15, and the second input terminal receives the start-up signal Ts, and the output terminal is connected to the drive module 105. The first input terminal of the AND gate A18 is connected to the output terminal of the OR gate A16, and the second input terminal receives the start-up signal Ts, and the output terminal is connected to the drive module 106. Therefore, the control switch tube Ml is turned off only when the clock signal CLK is active and the current sampling signal CS is greater than the first current threshold, and is turned on in other cases. The control switch tube M2 is turned off only when the clock signal CLK is inactive and the current sampling signal CS is less than the second current threshold, and is turned on in other cases.

[0075] ​Drive modules 105 and 106 are used to generate gate drive signals VG1 and VG2 for controlling switch M1 and switch M2, respectively. Drive modules 105 and 106 are, for example, level shifting circuits. Further, drive module 105 is connected to the same-name terminal DR1 of drive winding W2, and performs level shifting on the output signal of AND gate A17 relative to the same-name terminal DR1 to obtain gate drive signal VG1. Drive module 106 is connected to the opposite-name terminal DR2 of drive winding W2, and performs level shifting on the control signal relative to the opposite-name terminal DR2 to obtain gate drive signal VG2. Therefore, even when there is a positive or negative voltage at the same-name terminal DR1 and the opposite-name terminal DR2 of drive winding W2, the level shifting circuit can provide a suitable gate drive signal VG1 between the gate and source of control switch M1, and a suitable gate drive signal VG2 between the gate and source of control switch M2.

[0076] Figure 5 Show Figure 2 The waveform diagram of the LLC resonant converter in self-oscillation mode is shown. The figure shows the relationship between the resonant current CR obtained by the control circuit 100, the excitation current CT of the second transformer T2, and the clock signal CLK over time.

[0077] The resonant current CR is the current flowing through the primary winding Lp of the first transformer T1. In this application, the current flowing to the same-name terminal of the primary winding Lp of the first transformer T1 is defined as a positive current. The magnetizing current CT is the magnetizing current of the load winding W1 of the second transformer T2. In this application, the magnetizing current flowing to the same-name terminal of the load winding W1 of the second transformer T2 is defined as a positive current.

[0078] Furthermore, the current sampling signal CS of the resonant current is a voltage signal obtained on the resistor Rs, and the voltage drop in the direction of the current flowing to the intermediate node of bipolar transistors Q1 and Q2 is defined as a positive voltage. When the resonant current is a positive current, a negative voltage signal of the current sampling signal CS is obtained, and when the resonant current is a negative current, a positive voltage signal of the current sampling signal CS is obtained.

[0079] The resonant current CR and the excitation current CT of the second transformer T2 intersect periodically at points A, B, C, and D. The clock signal CLK has two levels, high and low (1, 0), and the resonant current CR also has two levels, positive and negative (>0, <0). These can be combined in pairs to form four different states, thus generating different circuit stages.

[0080] During the low time period of the clock signal CLK, the resonant current CR is converted from negative current to positive current. During the high time period of the clock signal CLK, the resonant current CR is converted from positive current to negative current. The resonant current CR exists in a predetermined current interval defined by a first current threshold (e.g. -100mV) and a second current threshold (e.g. +100mV) around zero current. During the time period when the clock signal CLK is low (0), the resonant current CR experiences three circuit phases in sequence, i.e. a time period t1 greater than the second current threshold, a time period t2 within the predetermined current interval, and a time period t3 less than the first current threshold. During the time period when the clock signal CLK is high (1), the resonant current CR experiences three circuit phases in sequence, i.e. a time period t4 less than the first current threshold, a time period t5 within the predetermined current interval, and a time period t6 greater than the second current threshold. Therefore, in each switching cycle of the LLC resonant converter, there are six circuit phases as shown in the following table.

[0081] Table 1, Different circuit phases in a switching cycle of the LLC resonant converter

[0082] Stage CLK CS M1 M2 Q1 Q2 Stage 1 0 CS > 100 mV ON ON Off then reverse amplification Desaturate then off Stage 2 0 0 < CS < 100 mV ON Off Reverse saturation Off Stage 2 0 - 100 mV < CS < 0 ON Off Forward on Off Stage 3 0 CS <- 100 mV ON Off Forward on Off Stage 4 1 CS <- 100 mV ON ON Desaturate then off Off then reverse amplification Stage 5 1 - 100 mV < CS < 0 Off ON Off Reverse saturation Stage 5 1 0 < CS < 100 mV Off ON Off Forward on Stage 6 1 CS > 100 mV Off ON Off Forward on

[0083] Figures 6a to 6c It is shown that Figure 2 The equivalent circuit diagram of the LLC resonant converter in the first phase t1 is shown.

[0084] In the first phase t1, the clock signal CLK is low, the resonant current CR is negative current, and the current sampling signal CS of the resonant current is greater than 100mV. In combination with Figure 3 As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is invalid, the inverted signal generated by the clock signal CLK via the inverter A14 is valid, the gate drive signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is valid, so that the control switch tube M1 is in the conducting state. At the same time, the second comparison signal generated by the comparator A13 is valid, the clock signal CLK is invalid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the control switch tube M2 is in the conducting state. Therefore, the control switch tubes M1 and M2 are in the double conducting state, and the driving winding W2 of the second transformer T2 forms a bidirectional path between the same name end and the opposite name end via the control switch tubes M1 and M2.

[0085] As Figure 6aAs shown, before point A, the resonant current CR and the magnetizing current CT are both negative currents, and the absolute value of the resonant current CR is greater than the absolute value of the magnetizing current CT. The net current is the difference between the resonant current CR and the magnetizing current CT, and the net current direction is still negative during this stage. Bipolar transistor Q1 is in the off state under the base-emitter bias voltage provided by the drive winding W2 of the second transformer T2, while bipolar transistor Q2 enters the off state from the saturation state under the base-emitter bias voltage provided by the drive winding W3 of the second transformer T2. The induced current in the drive winding W2 of the second transformer T2 is a positive current, flowing from the opposite-named terminal to the same-named terminal via control switches M1 and M2. Since bipolar transistor Q2 is off, no induced current is generated in the drive winding W3 of the second transformer T2. Because both bipolar transistors Q1 and Q2 are in the off state, the resonant current charges capacitor Cm, and the node voltage Vmid gradually increases.

[0086] like Figure 6b As shown, as the node voltage Vmid increases, the PN junction of the base-collector of bipolar transistor Q1 turns on, thus transitioning from the cutoff state to the reverse amplification state. Bipolar transistor Q2 remains in the cutoff state under the base-emitter bias voltage provided by the drive winding W3 of the second transformer T2. The induced current in the drive winding W2 of the second transformer T2 is a forward current, flowing from the opposite-named terminal to the same-named terminal via control switches M1 and M2. Since bipolar transistor Q2 is cut off, no induced current is generated in the drive winding W3 of the second transformer T2.

[0087] like Figure 6c As shown, after point A, both the resonant current CR and the magnetizing current CT are negative currents, and the absolute value of the resonant current CR is less than the absolute value of the magnetizing current CT. Therefore, the net current direction of the resonant current CR and the magnetizing current CT becomes positive. At this time, bipolar transistor Q1 remains in reverse amplification mode, while bipolar transistor Q2 remains in cutoff mode. The induced current in the drive winding W2 of the second transformer T2 becomes a reverse current, that is, it flows from the same-name terminal to the opposite-name terminal through the control switches M1 and M2. Since bipolar transistor Q2 is cut off, no induced current is generated in the drive winding W3 of the second transformer T2.

[0088] Figure 6d and 6e It shows Figure 2 The equivalent circuit diagram of the LLC resonant converter shown is shown in the second stage t2.

[0089] In the second stage t2, the clock signal CLK remains low, the resonant current CR changes from negative to positive, and the current sampling signal CS of the resonant current is greater than -100mV and less than 100mV. Combined with...Figure 3 As shown in the diagram, the first comparison signal generated by comparator A12 is invalid, while the inverted signal generated by inverter A14 from the clock signal CLK is valid. The gate drive signal VG1, generated based on the first comparison signal and the inverted signal of clock CLK, is valid, causing control switch M1 to be in the ON state. Simultaneously, the second comparison signal generated by comparator A13 is invalid, the clock signal CLK is invalid, and the gate drive signal VG2, generated based on the second comparison signal and clock CLK, is invalid, causing control switch M2 to be in the OFF state. Therefore, control switches M1 and M2 are in a single ON state, forming a unidirectional path for the drive winding W2 of the second transformer T2 from the opposite-named terminal to the same-named terminal via control switches M1 and M2.

[0090] like Figure 6d As shown, when the resonant current CR is negative (i.e., the current sampling signal CS of the resonant current is greater than 0 and less than 100mV), both the resonant current CR and the excitation current CT are negative, and the absolute value of the resonant current CR is less than the absolute value of the excitation current CT. Therefore, the net current direction of the resonant current CR and the excitation current CT remains positive. At this time, the bipolar transistor Q1 is in reverse saturation, and the bipolar transistor Q2 remains in the off state. When the drive winding W2 of the second transformer T2 allows bidirectional current flow, the induced current is a reverse current, that is, it flows from the same-name terminal to the opposite-name terminal via the control switches M1 and M2. However, the drive winding W2 of the second transformer T2 only has a unidirectional path from the opposite-name terminal to the same-name terminal. Therefore, the induced current generated by the drive winding W2 of the second transformer T2 provides the drive current for the bipolar transistor Q1. Since the bipolar transistor Q2 is off, no induced current is generated in the drive winding W3 of the second transformer T2.

[0091] like Figure 6e As shown, when the resonant current CR changes from negative to positive (i.e., the current sampling signal CS of the resonant current is less than 0 and greater than -100mV), the resonant current CR is positive, and the excitation current CT is negative. Therefore, the net current direction of the resonant current CR and the excitation current CT remains positive. At this time, bipolar transistor Q1 is in the forward conducting state, and bipolar transistor Q2 remains in the off state. The PN junction of the base-emitter of bipolar transistor Q1 is conducting, forming an additional path from the tap end to the opposite end of the drive winding W2 of the second transformer T2. Therefore, the induced current in the drive winding W2 of the second transformer T2 is a reverse current, that is, it flows from the tap end to the opposite end through the PN junction of the base-emitter of bipolar transistor Q1. Since bipolar transistor Q2 is off, no induced current is generated in the drive winding W3 of the second transformer T2.

[0092] Figure 6e It showsFigure 2 Equivalent circuit diagram of the LLC resonant converter shown in the third phase t3.

[0093] In the third phase t3, the clock signal CLK is maintained as low, the resonant current CR is maintained as positive current, and the current sampling signal CS of the resonant current is less than -100mV. In combination with Figure 3 As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is valid, the inverted signal of the clock signal CLK generated via the inverter A14 is valid, and the gate drive signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is valid, so that the control switch tube M1 is in the on state. At the same time, the second comparison signal generated by the comparator A13 is invalid, the clock signal CLK is invalid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is invalid, so that the control switch tube M2 is in the off state. Therefore, the control switch tubes M1 and M2 are in the single conduction state, and the driving winding W2 of the second transformer T2 forms a unidirectional path from the opposite end to the same end via the control switch tubes M1 and M2.

[0094] In the third phase t3, the state of the induced current in the driving windings W2 and W3 of the second transformer T2 remains unchanged, as Figure 6e shown.

[0095] Figures 6f to 6h It is shown Figure 2 Equivalent circuit diagram of the LLC resonant converter shown in the fourth phase t4.

[0096] In the fourth phase t4, the clock signal CLK is high, the resonant current CR is positive current, and the current sampling signal CS of the resonant current is less than -100mV. In combination with Figure 3 As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is valid, the inverted signal of the clock signal CLK generated via the inverter A14 is invalid, and the gate drive signal VG1 generated according to the first comparison signal and the inverted signal of the clock signal CLK is valid, so that the switch tube M1 is in the on state. At the same time, the second comparison signal generated by the comparator A13 is invalid, the clock signal CLK is valid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch tube M2 is in the on state. Therefore, the switch tubes M1 and M2 are in the double conduction state, and the driving winding W2 of the second transformer T2 forms a bidirectional path between the same end and the opposite end via the control switch tubes M1 and M2.

[0097] As Figure 6fAs shown, before point B, the resonant current CR and the magnetizing current CT are positive currents, and the absolute value of the resonant current CR is greater than that of the magnetizing current CT. The net current is the difference between the resonant current CR and the magnetizing current CT, and the direction of the net current is still positive. The bipolar transistor Q2 is in the cut-off state under the action of the base-emitter bias voltage provided by the drive winding W3 of the second transformer T2, and the bipolar transistor Q1 enters the cut-off state from the saturation state under the action of the base-emitter bias voltage provided by the drive winding W2 of the second transformer T2. The induced current of the drive winding W2 of the second transformer T2 is a reverse current, that is, flows from the same end to the opposite end via the control switch tubes M1 and M2. Since the bipolar transistor Q2 is cut off, no induced current is generated in the drive winding W3 of the second transformer T2. Since the bipolar transistors Q1 and Q1 are both cut off, the resonant current CR discharges the capacitor Cm, and the node voltage Vmid gradually decreases.

[0098] As shown in FIG. 4, when the node voltage Vmid is lower than the ground potential, the base-collector PN junction of the bipolar transistor Q2 is turned on, and thus the bipolar transistor Q2 enters the reverse amplification state from the cut-off state. The bipolar transistor Q1 maintains the cut-off state under the action of the base-emitter bias voltage provided by the drive winding W2 of the second transformer T2. The induced current of the drive winding W2 of the second transformer T2 is a reverse current, that is, flows from the same end to the opposite end via the control switch tubes M1 and M2. Since the bipolar transistor Q2 is in the reverse amplification state, no induced current is generated in the drive winding W3 of the second transformer T2. Figure 6g As shown in FIG. 4, when the node voltage Vmid is lower than the ground potential, the base-collector PN junction of the bipolar transistor Q2 is turned on, and thus the bipolar transistor Q2 enters the reverse amplification state from the cut-off state. The bipolar transistor Q1 maintains the cut-off state under the action of the base-emitter bias voltage provided by the drive winding W2 of the second transformer T2. The induced current of the drive winding W2 of the second transformer T2 is a reverse current, that is, flows from the same end to the opposite end via the control switch tubes M1 and M2. Since the bipolar transistor Q2 is in the reverse amplification state, no induced current is generated in the drive winding W3 of the second transformer T2.

[0099] Figure 6h As shown in FIG. 4, when the node voltage Vmid is lower than the ground potential, the base-collector PN junction of the bipolar transistor Q2 is turned on, and thus the bipolar transistor Q2 enters the reverse amplification state from the cut-off state. The bipolar transistor Q1 maintains the cut-off state under the action of the base-emitter bias voltage provided by the drive winding W2 of the second transformer T2. The induced current of the drive winding W2 of the second transformer T2 is a reverse current, that is, flows from the same end to the opposite end via the control switch tubes M1 and M2. Since the bipolar transistor Q2 is in the reverse amplification state, no induced current is generated in the drive winding W3 of the second transformer T2.

[0100] Figure 6i and 6j shows Figure 2 shows the equivalent circuit diagram of the LLC resonant converter in the fifth stage t5.

[0101] ​In the fifth stage t5, the clock signal CLK remains high, the resonant current CR changes from positive current to negative current, and the current sampling signal CS of the resonant current is greater than -100 mV and less than 100 mV. In combination with Figure 3 As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the clock signal CLK via the inverter A14 is invalid, and the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch tube M1 is in the off state. At the same time, the second comparison signal generated by the comparator A13 is invalid, the clock signal CLK is valid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch tube M2 is in the on state. Therefore, the switch tubes M1 and M2 are in the single on state, and the driving winding W2 of the second transformer T2 forms a unidirectional path from the same end to the different end via the control switch tubes M1 and M2.

[0102] As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the clock signal CLK via the inverter A14 is invalid, and the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch tube M1 is in the off state. At the same time, the second comparison signal generated by the comparator A13 is invalid, the clock signal CLK is valid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch tube M2 is in the on state. Therefore, the switch tubes M1 and M2 are in the single on state, and the driving winding W2 of the second transformer T2 forms a unidirectional path from the same end to the different end via the control switch tubes M1 and M2. Figure 6i As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the clock signal CLK via the inverter A14 is invalid, and the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch tube M1 is in the off state. At the same time, the second comparison signal generated by the comparator A13 is invalid, the clock signal CLK is valid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch tube M2 is in the on state. Therefore, the switch tubes M1 and M2 are in the single on state, and the driving winding W2 of the second transformer T2 forms a unidirectional path from the same end to the different end via the control switch tubes M1 and M2.

[0103] As shown in the working mode of the control circuit, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the clock signal CLK via the inverter A14 is invalid, and the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch tube M1 is in the off state. At the same time, the second comparison signal generated by the comparator A13 is invalid, the clock signal CLK is valid, and the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch tube M2 is in the on state. Therefore, the switch tubes M1 and M2 are in the single on state, and the driving winding W2 of the second transformer T2 forms a unidirectional path from the same end to the different end via the control switch tubes M1 and M2. Figure 6jAs shown, when the resonant current CR changes from positive current to negative current, i.e. the current sampling signal CS of the resonant current is greater than 0 and less than 100mV, the resonant current CR is negative current and the excitation current CT is positive current. Therefore, the net current direction of the resonant current CR and the excitation current CT is maintained as negative current. At this time, the bipolar transistor Q2 is in forward conducting state and the bipolar transistor Q1 is maintained in cut-off state. In the case that the driving winding W2 of the second transformer T2 allows bidirectional current to flow, the induced current is forward current, i.e. flows from the opposite end to the same end via the control switches M1 and M2. However, there is only a unidirectional path from the same end to the opposite end in the driving winding W2 of the second transformer T2, therefore, the driving winding W2 of the second transformer T2 actually does not generate induced current. Since the bipolar transistor Q2 is in forward conducting state, the induced current generated in the driving winding W3 of the second transformer T2 is forward current, i.e. flows from the opposite end to the same end via the PN junction of the base-emitter of the bipolar transistor Q2.

[0104] Figure 6j The working manner of the control circuit shown is known, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the inverter A14 via the clock signal CLK is invalid, the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch M1 is in cut-off state. At the same time, the second comparison signal generated by the comparator A13 is valid, the clock signal CLK is valid, the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch M2 is in conducting state. Therefore, the control switches M1 and M2 are in single conducting state, forming a unidirectional path from the same end to the opposite end of the driving winding W2 of the second transformer T2 via the control switches M1 and M2. Figure 2 The equivalent circuit diagram of the LLC resonant converter shown in the sixth stage t6.

[0105] In the sixth stage t6, the clock signal CLK is maintained as high level, the resonant current CR is maintained as negative current, and the current sampling signal CS of the resonant current is greater than 100mV. In combination with the working manner of the control circuit shown, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the inverter A14 via the clock signal CLK is invalid, the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch M1 is in cut-off state. At the same time, the second comparison signal generated by the comparator A13 is valid, the clock signal CLK is valid, the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch M2 is in conducting state. Therefore, the control switches M1 and M2 are in single conducting state, forming a unidirectional path from the same end to the opposite end of the driving winding W2 of the second transformer T2 via the control switches M1 and M2. Figure 3 The working manner of the control circuit shown is known, the first comparison signal generated by the comparator A12 is invalid, the inverse signal generated by the inverter A14 via the clock signal CLK is invalid, the gate drive signal VG1 generated according to the first comparison signal and the inverse signal of the clock signal CLK is invalid, so that the switch M1 is in cut-off state. At the same time, the second comparison signal generated by the comparator A13 is valid, the clock signal CLK is valid, the gate drive signal VG2 generated according to the second comparison signal and the clock signal CLK is valid, so that the switch M2 is in conducting state. Therefore, the control switches M1 and M2 are in single conducting state, forming a unidirectional path from the same end to the opposite end of the driving winding W2 of the second transformer T2 via the control switches M1 and M2.

[0106] In the sixth stage t6, the induced current states in the driving windings W2 and W3 of the second transformer T2 are maintained unchanged, as shown. Figure 6j

[0107] Figure 7 The schematic circuit diagram of the LLC resonant converter according to the second embodiment of the present application is shown. The LLC resonant converter 30 comprises a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D1 and D2, a resonant capacitor Cr, a resonant inductor Lr, control switches M1 and M2, and a control circuit 200.​

[0108] On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonance capacitor Cr and the resonance inductor Lr form a resonance circuit. Between the positive input terminal and the negative input terminal of the LLC resonant converter 30, the bipolar transistors Q1 and Q2 are connected in series, and the middle node of the two is connected to the resonance circuit. In the resonance circuit, the sampling resistor Rs is connected in series with the primary winding Lp, so that a sampling signal for characterizing the inductive current flowing through the primary winding Lp can be obtained.

[0109] On the secondary side of the first transformer T1, the diodes D1 and D2 form a rectifier circuit. The anodes of the diodes D1 and D2 are respectively connected to the two ends of the secondary winding, and the middle tap of the secondary winding is connected to ground. The output capacitor Co is connected between the cathodes of the diodes D1 and D2 and the ground, and provides a DC output voltage across the two terminals to the load Rd.

[0110] The second transformer T2 includes three windings, i.e. the load winding W1, the drive windings W2 and W3, around the same core. In the resonance circuit, the load winding W1 is connected in series with the primary winding Lp. Meanwhile, the drive windings W2 and W3 are respectively coupled to the bases of the bipolar transistors Q1 and Q2, but in opposite directions. For example, the drive winding W2 has a same-name end, an opposite-name end and a tap end between the two, and the drive winding W3 has a same-name end and an opposite-name end. The same-name end DR1 of the drive winding W2 is connected to the current terminal of the control circuit 200 to obtain the excitation current I1.

[0111] The control switching tubes M1 and M2 are connected in reverse series between the same-name end DR1 and the opposite-name end DR2 of the drive winding W2 of the second transformer T2. The control switching tubes M1 and M2 are, for example, N-type MOSFETs, and each includes a body diode connected in reverse. Unlike the first embodiment, the sources of the control switching tubes M1 and M2 are connected to each other, the drain of the control switching tube M1 is connected to the same-name end DR1 of the drive winding W2, and the drain of the control switching tube M2 is connected to the opposite-name end DR2 of the drive winding W2. Further, the middle nodes of the control switching tubes M1 and M2 are connected to ground.

[0112] The middle nodes of the bipolar transistors Q1 and Q2 are connected to the opposite-name end DR2 of the drive winding W2. Specifically, the base and the emitter of the bipolar transistor Q1 are respectively connected to the tap end and the opposite-name end of the drive winding W2. The base and the emitter of the bipolar transistor Q2 are respectively connected to the opposite-name end and the same-name end of the drive winding W3.

[0113] The turns ratio of the second transformer T2 is selected according to the electrical characteristics of the bipolar transistors Q1 and Q2. In this embodiment, the turns ratio of the load winding W1 : the drive winding W2 : the drive winding W3 of the second transformer T2 is 1 : 18 : 6, wherein the tap end of the drive winding W2 divides the drive winding W2 into a first sub-winding between the same end DR1 and the tap end, and a second sub-winding between the tap end and the opposite end DR2, and the turns ratio of the first sub-winding : the second sub-winding is 12 : 6. In alternative embodiments, the turns ratio of the load winding W1 : the drive winding W2 : the drive winding W3 of the second transformer T2 is 1 : 24 : 6 or 2 : 36 : 12. The fixed turns ratio of the first sub-winding and the second sub-winding of the drive winding W2 makes the drive voltage of the bipolar transistor Q1 in a fixed proportional relationship with the terminal voltage of the drive winding W2 (i.e. the voltage between the same end DR1 and the opposite end DR2). For example, in the state that only one of the control switch tubes M1 and M2 is single-conducting, the terminal voltage of the drive winding W2 is the forward voltage drop of the body diode of the control switch tube, thus the terminal voltage of the drive winding W2 is 0.7 V, and the drive signal of the bipolar transistor Q1 is clamped at 0.7 / 3 V, so that the bipolar transistor Q1 maintains the off state.

[0114] The control circuit 200 obtains a current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 30, and obtains a voltage feedback signal FB of the resonant output voltage from the auxiliary winding La of the first transformer T1 of the resonant converter 30, and generates the gate drive signal VG1 of the control switch tube M1 and the gate drive signal VG2 of the control switch tube M2 according to the current sampling signal CS and the voltage feedback signal FB, and excites the current II. In this embodiment, the ground end of the control circuit 200, the intermediate node of the control switch tubes M1 and M2, and the same end of the auxiliary winding La are grounded together.

[0115] Under the control of the control circuit 200, the control switch tubes M1 and M2 have three working modes, which are double-off, double-conducting, and single-conducting. In the starting phase of the LLC resonant converter, the control switch tubes M1 and M2 are in the double-off state, and the drive winding W2 of the second transformer T2 of the resonant converter 30 injects the excitation current, thereby starting the self-excited oscillation mode. In the self-excited oscillation working phase of the LLC resonant converter, the control switch tubes M1 and M2 are in the double-conducting or single-conducting state, thereby forcing the advance switching of the on and off states of the bipolar transistors Q1 and Q2, so that the LLC resonant converter can adopt the mode of adjusting the resonant current to obtain the desired DC output voltage / current.

[0116] According to the LLC resonant converter 30 of the embodiment, the drive windings W2 and W3 are used to provide drive current to drive the bases of the bipolar transistors Q1 and Q2 to achieve self-oscillation driving. Under the control of the self-oscillation driving signal, the bipolar transistors Q1 and Q2 are alternately turned on and turned off to convert the direct current input voltage into a square wave voltage. The square wave voltage is input to the resonant circuit to generate a resonant current at a resonant frequency. Therefore, through the resonant circuit, the electric energy is transferred from the primary side of the first transformer T1 to the secondary side of the first transformer T1.

[0117] The control circuit 200 controls the conduction states of the switching transistors M1 and M2 to provide an additional path and excitation current for the drive winding W2, and further controls the conduction states of the bipolar transistors Q1 and Q2 via the magnetic coupling of the drive windings W2 and W3 to achieve consistent turn-off time of the two bipolar transistors, thereby adjusting the resonant frequency. Therefore, the LLC resonant converter 30 can obtain the desired direct current output voltage / current in the manner of adjusting the resonant frequency, thereby achieving constant current control, and simplifying the control circuit and reducing the circuit cost.

[0118] In the first embodiment and the second embodiment, the control switching transistors M1 and M2 are N-type MOSFETs, for example, and are connected in anti-parallel between the same terminals DR1 and the different terminals DR2 of the drive winding W2 of the second transformer T2. In the first embodiment, the gate drive voltages of the control switching transistors M1 and M2 are negative voltages, for example, and the anodes of the parasitic diodes of the control switching transistors M1 and M2 are connected to each other to form an internal parasitic PNP-type transistor. The parasitic PNP-type transistor can also be turned on in the case of single conduction of the control transistors M1 and M2. For this reason, a special silicon-on-insulator (SOI) process is required in the integrated circuit process, and level shifting is required in the control circuit. In the second embodiment, the intermediate nodes of the control switching transistors M1 and M2 are grounded, and therefore the gate drive voltages of the control switching transistors M1 and M2 are positive voltages, thereby simplifying the control circuit. The cathodes of the parasitic diodes of the control switching transistors M1 and M2 are connected to each other to form an internal parasitic NPN-type transistor. The parasitic NPN-type transistor will not be turned on in the case of single conduction of the control transistors M1 and M2. Therefore, in the second embodiment, a special silicon-on-insulator (SOI) process is not required in the integrated circuit process, and level shifting is not required in the control circuit.

[0119] Figure 8 It is shown Figure 7A schematic circuit diagram of a control circuit in the LLC resonant converter shown. The control circuit 200 comprises a clock signal generating module 110, a logic module 120, comparators A12 and A13, a monoflop module 104, drive modules 105 and 106, a current source Is1, a switch tube M3. The clock signal generating module 110 comprises a subtraction module 201, a compensation module 202, an oscillator 111, a frequency division module 103, a capacitor Cc. The logic module 120 comprises a NOT gate A14, OR gates A15 and A16, AND gates A17 and A18.

[0120] The two input terminals of the subtraction module 201 are connected to Figure 7 The two terminals of the sampling resistor RS shown. In this embodiment, one terminal of the sampling resistor RS is connected to the same terminal of the load winding W1 of the second transformer T2, and the other terminal is connected to the different terminal DR2 of the drive winding W2 of the second transformer T2, i.e. the middle node of the bipolar transistor Q1 and Q2. Therefore, the two input terminals of the subtraction module 201 obtain the current sampling signal CS and the node voltage Vmid respectively, and perform subtraction operation to obtain the voltage drop on the sampling resistor RS, thereby obtaining the corrected current sampling signal CS1 proportional to the resonant current.

[0121] The compensation module 202 receives the corrected current sampling signal CS1 and / or the voltage feedback signal FB, and generates a compensation signal COMP across the capacitor Cc according to the comparison result of the corrected current sampling signal CS1 with the current reference signal, and / or according to the comparison result of the voltage feedback signal FB with the voltage reference signal. In constant current control mode, if the direct current output is too small, the compensation signal COMP increases, and if the direct current output is too large, the compensation signal COMP decreases. In constant voltage control mode, if the direct voltage output is too small, the compensation signal COMP increases, and if the direct voltage output is too large, the compensation signal COMP decreases.

[0122] The oscillator 111 further comprises a comparator A11, a current source Is2, a capacitor C11. The current source Is2 and the capacitor C11 are connected in series between the power supply terminal and the internal ground, and the middle node thereof provides a triangular wave signal. The non-inverting input terminal of the comparator A11 receives the triangular wave signal, and the inverting input terminal receives the compensation signal COMP, and the output terminal provides an oscillation signal OSC. The frequency of the oscillation signal OSC is related to the compensation signal COMP, and in the case that the compensation signal COMP is high, the frequency of the oscillation signal OSC is low, and in the case that the compensation signal COMP is low, the frequency of the oscillation signal OSC is high. The input terminal of the frequency division module 103 receives the oscillation signal OSC, and the output terminal provides the clock signal CLK. The frequency division module 103 generates the clock signal CLK with 50% duty cycle according to the oscillation signal OSC.

[0123] The comparator A12 compares the modified current sampling signal CS1 of the resonant current with a first current threshold (for example, -100 mV) to obtain a first comparison signal, and the comparator A13 compares the modified current sampling signal CS1 of the resonant current with a second current threshold (for example, +100 mV) to obtain a second comparison signal. The one-shot module 104 judges whether the modified current sampling signal CS1 is located in a predetermined current interval according to the first comparison signal and the second comparison signal, and generates an enabling signal Ts.

[0124] During the starting of the LLC resonant converter 30, the modified current sampling signal CS1 is located in the predetermined current interval, and the enabling signal Ts is in an active state of low level.

[0125] In this embodiment, the control switch tubes M1 and M2 are N-type MOSFETs, and the switch tube M3 is a P-type MOSFET. The enabling signal Ts is active at low level, so that the switch tube M3 is turned on. At the same time, the enabling signal Ts is one of the input signals of the AND gates A17 and A18, so that the gate drive signal VG1 provided by the drive module 106 and the gate drive signal VG2 provided by the drive module 105 are invalid, so that the control switch tubes M1 and M2 are turned off. The current source Is1 is connected in series with the switch tube M3 between the power supply end and the same end DR1 of the drive winding W2.

[0126] Therefore, during the starting of the LLC resonant converter 30, the current source Is1 injects the exciting current I1 into the same end DR1 of the drive winding W2 through the switch tube M3, so that the LLC resonant converter 30 starts to work in a self-oscillation manner.

[0127] Further, after the LLC resonant converter 30 starts to work in the self-oscillation manner, the modified current sampling signal CS1 of the resonant current exceeds the above-mentioned predetermined current interval in a fixed time of a plurality of switching periods, so that the enabling signal Ts maintains an inactive state of high level. Since the switch tube M3 is turned off, the current source Is1 stops injecting the exciting current I1 into the same end DR1 of the drive winding W2 through the switch tube M3.

[0128] The control switch tubes M1 and M2 control their conduction states according to the clock signal CLK. As shown in the figure, the first input terminal of the OR gate A15 is connected to the output terminal of the frequency dividing module 103 via the inverter A14 to receive the inverted signal of the clock signal CLK, and the second input terminal is connected to the output terminal of the comparator A12 to receive the first comparison signal; the first input terminal of the OR gate A16 is connected to the output terminal of the frequency dividing module 103 to receive the clock signal CLK, and the second input terminal is connected to the output terminal of the comparator A13 to receive the second comparison signal. The first input terminal of the AND gate A17 is connected to the output terminal of the OR gate A15, the second input terminal receives the start signal Ts, and the output terminal is connected to the drive module 105. The first input terminal of the AND gate A18 is connected to the output terminal of the OR gate A16, the second input terminal receives the start signal Ts, and the output terminal is connected to the drive module 106. Therefore, in the case that the clock signal CLK is effective and the corrected current sampling signal CS1 is greater than the first current threshold value, the control switch tube M1 is turned off, and in other cases, it is in the conduction state. In the case that the clock signal CLK is not effective and the corrected current sampling signal CS1 is less than the second current threshold value, the control switch tube M2 is turned off, and in other cases, it is in the conduction state.

[0129] The drive module 106 is used to generate the gate drive signal VG1 of the control switch tube M1, and the drive module 105 is used to generate the gate drive signal VG2 of the control switch tube M2. The drive modules 105 and 106 are, for example, a plurality of inverters connected in series, respectively.

[0130] In the above embodiment, it is described that in the LLC resonant converter, the short circuit of the drive winding of the upper bipolar transistor is controlled, and the short circuit state is released at the appropriate time, so as to control the alternating conduction of the bipolar transistor, so that the switching period of the bipolar transistor follows the period of the internal switching control signal of the control circuit, and further, the frequency of the switching control signal is controlled according to the negative feedback of the resonant current, so as to realize the constant of the output current or voltage. However, the present application is not limited to this. It can be understood that based on the similar working principle, the current path control of the drive winding of the lower bipolar transistor of the LLC resonant converter can also achieve the same technical effect.

[0131] In the above first embodiment, it is described that in the LLC resonant converter, the intermediate node of the two bipolar transistors is grounded, and the compensation module in the control circuit directly obtains the voltage drop of the sampling resistor as the current sampling signal, the first drive module is connected to the same name terminal of the first drive winding, and the first control signal is level shifted with respect to the same name terminal to obtain the first gate drive signal; the second drive module is connected to the opposite name terminal of the second drive winding, and the second control signal is level shifted with respect to the opposite name terminal to obtain the second gate drive signal.

[0132] In the second embodiment described above, the intermediate node of two control switch tubes in the LLC resonant converter is grounded, and the two input terminals of the subtraction module in the control circuit are connected to the two ends of the sampling resistor to obtain the voltage drop of the sampling resistor as the corrected current sampling signal, the first driving module generates the first gate drive signal relative to the intermediate node of the control switch tubes M1 and M2; the second driving module generates the second gate drive signal relative to the intermediate node of the control switch tubes M1 and M2.

[0133] In accordance with the embodiments of the present application as described above, these embodiments do not describe all the details, nor limit the application to only the specific embodiments described. Obviously, many modifications and variations are possible in light of the above teachings. The embodiments were chosen and described in order to explain the principles of the application and its practical application, to thereby enable others skilled in the art to best utilize the application and various embodiments with various modifications as are suited to the particular use contemplated. The application is only limited by the scope of the appended claims, and the full scope of equivalents thereof.

Claims

1. An LLC resonant converter, comprising: a first transformer comprising a primary winding and a secondary winding; a second transformer comprising a load winding, and a first driving winding and a second driving winding magnetically coupled to the load winding; and a first bipolar transistor and a second bipolar transistor connected in series to each other between a positive supply terminal and a negative supply terminal of a direct current input voltage, a load winding of the second transformer, a resonant element and the primary winding being connected between an intermediate node of the first bipolar transistor and the second bipolar transistor and the negative supply terminal, the resonant element being connected to the primary winding to form a resonant circuit to generate a resonant current, and a resonant output voltage being provided across the primary winding, wherein the LLC resonant converter further comprises a control circuit for controlling a conduction state of a path between a same-name terminal and an opposite-name terminal of the first driving winding, thereby controlling a conduction state of the first bipolar transistor and the second bipolar transistor to adjust a resonant frequency, the first driving winding comprising the same-name terminal, the opposite-name terminal and a tap terminal, and the second driving winding comprising the same-name terminal and the opposite-name terminal, a base of the first bipolar transistor being connected to the tap terminal of the first driving winding to receive a first driving current generated according to an induced current of the resonant current, a base of the second bipolar transistor being connected to the opposite-name terminal of the second driving winding to receive a second driving current generated according to the induced current of the resonant current, a first control switch and a second control switch connected in anti-series between the same-name terminal and the opposite-name terminal of the first driving winding, the control circuit generating a clock signal according to a current sampling signal of the resonant current and / or a voltage feedback signal of the resonant output voltage, and generating a first gate drive signal and a second gate drive signal according to the clock signal, thereby controlling the first control switch and the second control switch to periodically short the same-name terminal and the opposite-name terminal of the first driving winding, so that a resonant period of the LLC resonant converter follows the clock signal. the first bipolar transistor and the second bipolar transistor are NPN bipolar transistors, respectively, an emitter of the first bipolar transistor and a collector of the second bipolar transistor being commonly connected to the intermediate node, the opposite-name terminal of the first driving winding being connected to the emitter of the first bipolar transistor, and the same-name terminal of the second driving winding being connected to the emitter of the second bipolar transistor.

2. The LLC resonant converter of claim 1, wherein, the first control switch and the second control switch are N-type MOSFETs, respectively, a drain of the first control switch and a drain of the second control switch being connected to each other, a source of the first control switch being connected to the same-name terminal of the first driving winding, a source of the second control switch being connected to the opposite-name terminal of the first driving winding, and the source of the second control switch being grounded.

3. The LLC resonant converter of claim 1, wherein, a gate of the first control switch receives the first gate drive signal, and a gate of the second control switch receives the second gate drive signal.

4. The LLC resonant converter of claim 3, wherein, ​ 5. The LLC resonant converter of claim 1, wherein, The first control switch tube and the second control switch tube are N-type MOSFETs, the source of the first control switch tube and the source of the second control switch tube are connected to each other, the drain of the first control switch tube is connected to the same end of the first drive winding, the drain of the second control switch tube is connected to the opposite end of the first drive winding, and the intermediate node of the first control switch tube and the second control switch tube is grounded.

6. The LLC resonant converter of claim 5, wherein, The gate of the first control switch tube receives a second gate drive signal, and the gate of the second control switch tube receives a first gate drive signal.

7. The LLC resonant converter of claim 1, further comprising: A sampling resistor is connected in series with the primary winding of the first transformer, and one end of the sampling resistor is connected to the intermediate node of the first bipolar transistor and the second bipolar transistor, wherein a current sampling signal of the resonant current is obtained across the sampling resistor.

8. The LLC resonant converter of claim 1, wherein, The control circuit comprises: a clock signal generation module configured to generate the clock signal with a corresponding clock period according to the current sampling signal of the resonant current and / or a voltage feedback signal of the resonant output voltage; a logic module connected to the clock signal generation module and configured to generate a first control signal and a second control signal according to the clock signal; a first drive module connected to the logic module and configured to generate the first gate drive signal according to the first control signal; and a second drive module connected to the logic module and configured to generate the second gate drive signal according to the second control signal.

9. The LLC resonant converter of claim 8, wherein, The control circuit further comprises: a first comparator configured to compare the current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and a second comparator configured to compare the current sampling signal of the resonant current with a second current threshold to obtain a second comparison signal, wherein the logic module is connected to the first comparator to obtain the first comparison signal and connected to the second comparator to obtain the second comparison signal, in a case where the clock signal is invalid or the current sampling signal is less than the first current threshold, the logic module controls the first gate drive signal to be in an effective state, in a case where the clock signal is valid or the current sampling signal is greater than the second current threshold, the logic module controls the second gate drive signal to be in an effective state.

10. The LLC resonant converter of claim 9, wherein, The control circuit further comprises: a first current source and a switch tube connected in series between a power supply end and the same end of the first drive winding; and a single-trigger module having a first input end connected to the first comparator to obtain the first comparison signal, a second input end connected to the second comparator to obtain the second comparison signal, and an output end connected to the switch tube to provide a start signal, wherein during startup of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the start signal is effective, the switch tube is turned on, the first current source injects an excitation current into the first drive winding via the switch tube, and the first control switch tube and the second control switch tube are turned off.

11. The LLC resonant converter of claim 10, wherein, The logic module comprises: a NOT gate for obtaining an inverted signal of the clock signal; a first OR gate, a first input end of which receives the inverted signal of the clock signal, and a second input end of which receives the first comparison signal; a first AND gate, a first input end of which is connected to an output end of the first OR gate, and a second input end of which receives the start signal; a second OR gate, a first input end of which receives the clock signal, and a second input end of which receives the second comparison signal; and a second AND gate, a first input end of which is connected to an output end of the second OR gate, and a second input end of which receives the start signal, wherein the start signal is valid at a low level.

12. The LLC resonant converter of claim 8, wherein, The first drive module is connected to a same-name end of the first drive winding, the first control signal is level-shifted relative to the same-name end to obtain the first gate drive signal, and the second drive module is connected to an opposite-name end of the first drive winding, the second control signal is level-shifted relative to the opposite-name end to obtain the second gate drive signal.

13. The LLC resonant converter of claim 8, wherein, The clock signal generation module comprises: a compensation module and a first capacitor connected at an output end of the compensation module, the compensation module comparing the current sampling signal with a current reference signal and / or comparing the voltage feedback signal with a voltage reference signal to generate a compensation signal, and the first capacitor generating the compensation signal across two ends thereof; an oscillator for generating an oscillation signal of a corresponding frequency according to the compensation signal; and a frequency division module for generating the clock signal with a 50% duty cycle according to the oscillation signal. The control circuit and the first control switch tube and the second control switch tube are integrated into a single chip.

14. The LLC resonant converter of claim 1, wherein, 15. A control circuit for an LLC resonant converter as claimed in claim 1, the LLC resonant converter comprising a first drive winding coupled with a resonant tank, the control circuit comprising: a clock signal generation module for generating a clock signal with a corresponding clock period according to a current sampling signal of a resonant current and / or a voltage feedback signal of a resonant output voltage; a logic module connected with the clock signal generation module, for generating a first control signal and a second control signal according to the clock signal; a first drive module connected with the logic module, for generating a first gate drive signal according to the first control signal; and a second drive module connected with the logic module, for generating a second gate drive signal according to the second control signal, wherein the control circuit controls a first control switch tube and a second control switch tube to periodically short a same-name end and an opposite-name end of the first drive winding by using the first gate drive signal and the second gate drive signal, so as to generate a drive current at a tap end of the first drive winding, and the resonant period of the LLC resonant converter follows the clock signal.

16. The control circuit of claim 15, further comprising: a first comparator for comparing the current sampling signal of the resonant current with a first current threshold to obtain a first comparison signal; and a second comparator for comparing the current sampling signal of the resonant current with a second current threshold to obtain a second comparison signal. ​ ​ ​ The logic module is connected with the first comparator to obtain the first comparison signal, and is connected with the second comparator to obtain the second comparison signal, In the case that the clock signal is invalid, or the current sampling signal is less than the first current threshold, the logic module controls the first gate drive signal to be in an effective state, In the case that the clock signal is valid, or the current sampling signal is greater than the second current threshold, the logic module controls the second gate drive signal to be in an effective state.

17. The control circuit of claim 16, further comprising: a first current source and a switch tube connected in series between a power supply end and a same-named end of the first drive winding; and a single trigger module, a first input end of which is connected with the first comparator to obtain the first comparison signal, a second input end of which is connected with the second comparator to obtain the second comparison signal, and an output end of which is connected with the switch tube to provide a start signal, wherein, during starting of the LLC resonant converter, the current sampling signal is greater than the first current threshold and less than the second current threshold, the start signal is effective, the switch tube is turned on, the first current source injects an excitation current to the first drive winding via the switch tube, and the first control switch tube and the second control switch tube are turned off. The logic module comprises:

18. The control circuit of claim 17, wherein, a NOT gate for obtaining an inverted signal of the clock signal; a first OR gate, a first input end of which receives the inverted signal of the clock signal, and a second input end of which receives the first comparison signal; a first AND gate, a first input end of which is connected to an output end of the first OR gate, and a second input end of which receives the start signal; a second OR gate, a first input end of which receives the clock signal, and a second input end of which receives the second comparison signal; and a second AND gate, a first input end of which is connected to an output end of the second OR gate, and a second input end of which receives the start signal, wherein, the start signal is effective when at a low level. The first drive module is connected to the same-named end of the first drive winding, and the first control signal is level-shifted with respect to the same-named end to obtain the first gate drive signal, and 19. The control circuit of claim 15, wherein, the second drive module is connected to the different-named end of the first drive winding, and the second control signal is level-shifted with respect to the different-named end to obtain the second gate drive signal. The clock signal generation module comprises:

20. The control circuit of claim 15, wherein, a compensation module and a first capacitor connected at an output end of the compensation module, an inverted signal being generated at both ends of the first capacitor; an oscillator for generating an oscillation signal of a corresponding frequency according to the inverted signal; and a frequency division module for generating the clock signal of 50% duty cycle according to the oscillation signal, wherein, the compensation module compares the current sampling signal with a current reference signal, and / or compares the voltage feedback signal with a voltage reference signal, to generate the inverted signal, so that the frequency of the clock signal is adjusted in relation to the resonant current and / or the resonant output voltage. The clock signal generation module comprises: a compensation module and a first capacitor connected at an output end of the compensation module, an inverted signal being generated at both ends of the first capacitor; an oscillator for generating an oscillation signal of a corresponding frequency according to the inverted signal; and a frequency division module for generating the clock signal of 50% duty cycle according to the oscillation signal, wherein, the compensation module compares the current sampling signal with a current reference signal, and / or compares the voltage feedback signal with a voltage reference signal, to generate the inverted signal, so that the frequency of the clock signal is adjusted in relation to the resonant current and / or the resonant output voltage.

21. The control circuit of claim 15, wherein, The first control switch and the second control switch are N-type MOSFETs, the drain of the first control switch and the drain of the second control switch are connected to each other, the source of the first control switch is connected to the same terminal of the first driving winding, the source of the second control switch is connected to the different terminal of the first driving winding, and the source of the second control switch is grounded, wherein the gate of the first control switch receives a first gate drive signal, and the gate of the second control switch receives a second gate drive signal.

22. The control circuit of claim 15, wherein, The first control switch and the second control switch are N-type MOSFETs, the source of the first control switch and the source of the second control switch are connected to each other, the drain of the first control switch is connected to the same terminal of the first driving winding, the drain of the second control switch is connected to the different terminal of the first driving winding, and the intermediate node of the first control switch and the second control switch is grounded, wherein the gate of the first control switch receives a second gate drive signal, and the gate of the second control switch receives a first gate drive signal.

23. The control circuit of claim 15, wherein, The control circuit, the first control switch and the second control switch are integrated into a single chip.

24. A control method for the LLC resonant converter as claimed in claim 1, the LLC resonant converter comprising a first driving winding coupled with a resonant tank, the control method comprising: generating a clock signal with a corresponding clock period according to a current sampling signal of a resonant current and / or a voltage feedback signal of a resonant output voltage; generating a first control signal and a second control signal according to the clock signal; generating a first gate drive signal and a second gate drive signal according to the first control signal and the second control signal, respectively; and controlling a first control switch and a second control switch to periodically short the same terminal and the different terminal of the first driving winding with the first gate drive signal and the second gate drive signal, generating a driving current at a tap terminal of the first driving winding, so that the resonant period of the LLC resonant converter follows the clock signal.

25. The control method according to claim 24, wherein In the case that the clock signal is invalid, or the current sampling signal is less than a first current threshold, the first gate drive signal is in an effective state, In the case that the clock signal is valid, or the current sampling signal is greater than a second current threshold, the second gate drive signal is in an effective state.

26. The control method of claim 24, further comprising: During the startup of the LLC resonant converter, an excitation current is injected into the first driving winding via a switch, and the first control switch and the second control switch are turned off.

Images