High-Order Miller N-Path Filter

Abstract

An N-path filter (600) having one or more branches selectively coupled to a shared circuit node (606), the N-path filter comprising a first branch having a first feedback path and a second feedback path. The first feedback path comprises a Miller amplifier (620) having an input coupled to an input voltage; and a first capacitor (C1) coupled to both the input voltage and the output of the Miller amplifier. The second feedback path comprises a node common to the first feedback path. The second feedback path also comprises a first high pass filter (R HP1 , C HP1 ), which is coupled to the output of the Miller amplifier; and a second capacitor (C2), which is coupled to both the first capacitor and the first high pass filter.

Description

[0001] Priority claim

[0002] This patent application claims priority to non-provisional application No. 15 / 963,056, filed on April 25, 2018, entitled “HIGH ORDER MILLER N-PATH FILTER,” which provisional application has been assigned to the present assignee and is expressly incorporated herein by reference. Technical Field

[0003] The present disclosure relates generally to wireless receivers and more particularly to an application of the Miller effect on an N-path filter configured as a bandpass filter to implement a high-order Miller N-path filter. Background Art

[0004] Advances in technology have resulted in computing devices that are smaller in size and more powerful. For example, there are currently a variety of portable personal computing devices or user equipment (UE), which include wireless computing devices such as portable wireless telephones, personal digital assistants (PDAs), and paging devices that are small, lightweight, and easy for users to carry. More specifically, portable wireless telephones such as cellular telephones and Internet Protocol (IP) telephones can communicate voice and data packets over wireless networks.

[0005] In some types of wireless networks, the UE communicates with one or more network base stations. In some scenarios, different base stations can use different radio access technologies (RATs). The term RAT refers to the physical connection of a radio-based communication network. Examples of different RATs include, but are not limited to, third generation partnership project (3GPP) technology (e.g., third generation technology (3G), fourth generation technology (4G), and fifth generation technology (5G)), millimeter wave (mmW) technology (ultra-high frequency (EHF)), Bluetooth technology, and Wi-Fi technology. In millimeter wave (mmW) systems, multiple antennas are used for beamforming (e.g., within 30 gigahertz (GHz), 60 GHz, etc.).

[0006] Different RATs may have different capabilities. For example, a UE may have the ability to access a Long Term Evolution (LTE) network and a millimeter wave wavelength (mmW) network. The downlink / uplink (DL / UL) access link between an LTE base station and a UE is typically more reliable than the access link between an mmW base station and a UE. However, the capacity of an LTE link is typically lower than the capacity of an mmW link.

[0007] In a UE that can simultaneously transmit and receive wireless communications according to a wireless network, transmit (TX) leakage can cause performance limitations to receive (RX) circuitry. TX leakage and other interference in the RX circuitry can be modulated and down-converted to baseband along with the received wireless signal. TX leakage and interference can have relatively larger voltage swings than the received signal and can saturate the output of a receiver that converts the received signal from radio frequency (RF) to baseband.

[0008] In carrier aggregation (CA) architectures, blocking (TX leakage and interference) is a performance limitation of the RX circuitry. For operation in CA with a low noise amplifier (LNA) followed by a cascode device or a transconductance stage, the LNA output is a high impedance node. The high impedance causes large blocking swings and linearity issues. For concurrent CA operation, if the signal in one CA receive path is large and acts as an interferer, the noise figure in the other CA receive path may be degraded. Summary of the invention

[0009] An N-path filter having one or more branches selectively coupled to a shared circuit node, the N-path filter comprising a first branch having a first feedback path and a second feedback path. The first feedback path comprises an amplifier (e.g., a Miller amplifier or an amplifier based on the Miller effect) whose input is coupled to an input voltage; and a first capacitor coupled to both the input voltage and the output of the Miller amplifier. The second feedback path comprises a node common to the first feedback path. The second feedback path also comprises a first high-pass filter coupled to the output of the Miller amplifier; and a second capacitor coupled to both the first capacitor and the first high-pass filter.

[0010] An N-path filter having one or more branches selectively coupled to a shared circuit node, the N-path filter comprising a first branch having a first feedback path and a second feedback path. The first feedback path comprises a Miller amplifier having an input coupled to an input voltage; and a first device for generating an impedance coupled to both the input voltage and the output of the Miller amplifier. The second feedback path comprises a node common to the first feedback path. The second feedback path also comprises a first high pass filter coupled to the output of the Miller amplifier; and a second device for generating an impedance coupled to both the first impedance generating device and the first high pass filter.

[0011] A method for filtering a wireless signal at a receiver includes: receiving a radio frequency signal at a shared circuit node, the shared circuit node selectively coupled to each of a plurality of branches of an N-path filter. Each branch of the N-path filter includes a first feedback path and a second feedback path, the first feedback path includes a Miller amplifier, and the second feedback path has a common node with the first feedback path. The method also includes: when the radio frequency signal is received at a first frequency, generating a high impedance at the common node to prevent the radio frequency signal from traversing the first feedback path and the second feedback path. The method also includes: when the radio frequency signal is received at a second frequency, generating a low impedance at the common node to allow the radio frequency signal to traverse the first feedback path and / or the second feedback path. The second frequency is higher than the first frequency.

[0012] This has been a fairly broad overview of the features and technical advantages of the present disclosure so that the following specific embodiments may be better understood. Additional features and advantages of the present disclosure are described below. It will be appreciated by those skilled in the art that the present disclosure may be easily used as a basis for modifying or designing other structures for achieving the same purpose of the present disclosure. It will also be appreciated by those skilled in the art that such equivalent constructions do not deviate from the teachings of the present disclosure as set forth in the appended claims. When considered in conjunction with the accompanying drawings, the novel features that are considered to be the characteristics of the present disclosure in terms of its organization and method of operation, as well as other purposes and advantages, may be better understood from the following description. However, it is to be clearly understood that each of the figures in the accompanying drawings is provided for the purpose of illustration and description only and is not intended to be a definition of limitations to the present disclosure. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] For a more complete understanding of the present disclosure, reference is now made to the following description taken in conjunction with the accompanying drawings.

[0014] Figure 1 A wireless device in communication with a wireless communication system is shown.

[0015] Figure 2 According to one aspect of the present disclosure Figure 1 Block diagram of a wireless device in FIG.

[0016] FIG. 3A to FIG. 3D Four examples of carrier aggregation (CA) are shown.

[0017] Figure 4A An example receive chain of a radio frequency (RF) front end including an N-path filter in accordance with certain aspects of the present disclosure is shown.

[0018] Figure 4B is a diagram showing a method for controlling a Figure 4A An example timing diagram of the signals for the switching operation of the transistors in an N-path filter.

[0019] Figure 5A Example frequency responses of an N-path filter according to certain aspects of the present disclosure are shown.

[0020] Figure 5B Example frequency responses of N-path filters according to various aspects of the present disclosure are shown.

[0021] Figure 6 An N-path filter including a single high pass filtering path is shown in accordance with various aspects of the present disclosure.

[0022] Figure 7 An N-path filter including two high-pass filtering paths according to various aspects of the present disclosure is shown.

[0023] Figure 8 is a graph showing a first frequency response of a low-order bandpass filter and a second frequency response of a high-order bandpass filter.

[0024] Fig. 9 Represents a method including two high pass filtering paths according to various aspects of the present disclosure Figure 7 The Thévenin equivalent of an N-path filter.

[0025] Fig.10 An N-path filter having multiple branches is shown, wherein each branch includes two high-pass filtering paths.

[0026] Fig.11 An N-path filter having multiple branches is shown, wherein each branch includes two high-pass filter paths.

[0027] Fig.12 A simplified flow chart of a method for filtering an interfering signal according to various aspects of the present disclosure is depicted.

[0028] Fig.13 is a block diagram illustrating an exemplary wireless communication system in which configurations of the present disclosure may be advantageously employed. DETAILED DESCRIPTION

[0029] The specific embodiments described below in conjunction with the accompanying drawings are intended as descriptions of various configurations and are not intended to represent the only configurations in which the concepts described herein may be practiced. In order to provide a thorough understanding of the various concepts, the specific embodiments include specific details. However, it is apparent to those skilled in the art that these concepts may be practiced without these specific details. In some instances, well-known structures and components are shown in block diagram form to avoid obscuring these concepts. As described herein, the use of the term "and / or" is intended to represent "inclusive or", while the use of the term "or" is intended to represent "exclusive or".

[0030] In a user equipment (UE) that can simultaneously transmit and receive wireless signals according to a wireless network, transmit (TX) leakage can cause performance limitations to receive (RX) circuitry. TX leakage and other interference (or jammers) in the RX circuitry can be modulated and down-converted to baseband along with the received wireless signal. TX leakage and interference can have relatively larger voltage swings than the received signal and can saturate the output of a receiver that converts the received signal from radio frequency (RF) to baseband.

[0031] In a carrier aggregation (CA) architecture, blocking (transmit (TX) leakage and interference) is a performance limitation of the RX circuit system. TX leakage is the leakage from the TX chain (e.g., the transmit path including the transmitter) to the receive (RX) chain (e.g., the receive path including the receiver) in the transceiver or transceiver front end. The signal coupled into the RX chain along with the TX leakage can be modulated at the local oscillator (LO) frequency and its multiples (e.g., two and three times the LO frequency) and down-converted to the baseband (BB). Larger voltage swings can saturate the BB output. Although this may reduce the signal-to-noise ratio (SNR), this can be mitigated by reducing the BB gain.

[0032] In non-CA applications, TX leakage or blockers and jammers are limiting factors for improving noise figure (NF) and linearity. Large blockers and TX leakage may prevent the use of some architectures that are well suited to achieve NF increase. For example, TX jammers can cause TX second-order intercept point (IIP2) issues and limit the receiver's signal-to-noise and distortion ratio (SNDR) performance.

[0033] Therefore, circuits for rejecting TX leakage or blocking to maintain sensitivity during concurrent CA or non-CA operation are important. Some techniques for reducing the effects of interference include large TX capacitors, which are used in front of the transimpedance amplifier (TIA) to absorb the TX interference current. However, this approach specifies a large TX capacitor to draw the current of the transmission interference into the TX capacitor to reduce the voltage swing (since the voltage swing is inversely proportional to the capacitance of the TX capacitor). In addition, if the TX interference is too close to the in-band, the TX rejection is small, which results in specifying a lower gain mode, thereby reducing the noise figure (NF). Even with a large TX capacitor, this small amount of TX rejection still holds. Therefore, the circuit cannot achieve the best signal-to-noise ratio.

[0034] Furthermore, the bandwidth of the desired signal is becoming larger (e.g., on the order of 80 MHz). Therefore, it is becoming increasingly challenging to implement a high quality factor (high Q) bandpass filter at RF to reject out-of-band interference and TX leakage while avoiding attenuation of the desired signal in the RX band.

[0035] N-path filters can be used to provide a high-Q bandpass filter at radio frequencies. An N-path filter can include N branches. For example, an N-path filter can consist of N identical linear time-invariant (LTI) networks and 2N mixers driven by time / phase shifted versions of a clock signal. If the LTI network exhibits a low-pass characteristic near direct current (DC), mixing through the mixers produces a bandpass filter response with the passband centered near the mixing frequency. That is, the input signal is downconverted to baseband, filtered through the LTI network, and then upconverted again to the original frequency band of the input signal. The center frequency is determined by the mixing frequency and is insensitive to the filter component values. A high mixing frequency combined with a narrow low-pass filter bandwidth provides a very high Q filter.

[0036] Some techniques use the Miller effect to reduce the size of capacitors used in N-path filters to improve the performance of the RX circuit system. For example, amplification (using a Miller amplifier) ​​can be applied to increase the voltage swing across the capacitor. Thus, the size of the capacitor used for the N-path filter is reduced. However, these techniques may not be able to achieve high-order filtering when the Miller effect is applied to the capacitor. Therefore, it is necessary to design a high-order high-Q bandpass filter to alleviate these problems.

[0037] Various aspects of the present disclosure include an N-path filter having one or more branches that are selectively coupled to a shared circuit node. The N-path filter is coupled to a low noise amplifier (LNA). For example, the N-path filter can be coupled to the input of the LNA, the output of the LNA, or between two LNAs of a receiver. The N-path filter can be configured as a bandpass filter. Each branch includes a first feedback path and a second feedback path (or a high-pass filter path). When one or more branches include multiple branches, each of the multiple branches can be selectively coupled to a shared circuit node. The synthetic impedance can be achieved by coupling multiple branches in parallel.

[0038] In one aspect of the present disclosure, a first feedback path of a first branch of an N-path filter includes an amplifier (e.g., a Miller amplifier or an amplifier based on the Miller effect) and a first capacitor. The Miller amplifier includes a first terminal (e.g., an input) coupled to an input voltage. The first terminal may correspond to a shared circuit node. For example, the N-path filter may be connected or coupled to a circuit (e.g., a Thévenin equivalent circuit) via an input impedance (e.g., a series resistor) that includes an input signal from a voltage source Vin. The Thévenin equivalent circuit may represent an equivalent of a signal received by an antenna and amplified by an LNA. The voltage source Vin may be considered an input voltage.

[0039] The first capacitor is coupled to both the input voltage and the second terminal (e.g., output) of the Miller amplifier. For example, the input of the Miller amplifier is coupled to a shared circuit node, which is coupled to a voltage source or input voltage Vin. In one aspect of the present disclosure, the first capacitor is coupled to the input voltage through a first switch or transistor and is coupled to the output of the Miller amplifier through a second transistor.

[0040] The second feedback path has a first node in common with the first feedback path. The second feedback path includes a first high pass filter and a second capacitor. The first node in common with the first feedback path merges the second feedback path into the first feedback path. For example, the second capacitor is coupled to the input of the Miller amplifier via the first transistor. The first high pass filter is coupled to the output of the Miller amplifier. For example, the first high pass filter can be coupled to the output of the Miller amplifier via the second transistor.

[0041] In one aspect of the present disclosure, the second feedback path may further include a first buffer. The first buffer may be located between the first high pass filter and the second capacitor. The first buffer may be implemented according to a bias source follower configuration.

[0042] In some aspects of the present disclosure, the N-path filter may include a third feedback path and a third capacitor. The third feedback path may be merged with the second feedback path. The third feedback path includes a second high-pass filter, which is coupled to the output of the first high-pass filter. The third capacitor may be coupled to both the first capacitor and the second capacitor and the second high-pass filter. The first buffer is also coupled between the output of the first high-pass filter and the input of the second high-pass filter. Depending on the shunt configuration, the N-path filter can be integrated into a receive (RX) chain before or after a low noise amplifier (LNA).

[0043] Various aspects of the present disclosure may be Figure 1 systems and Fig.13 More specifically, various aspects of the present disclosure may be implemented in Figure 2 is implemented in the wireless device 200.

[0044] Figure 1 A wireless device 110 is shown that may include the disclosed N-path filter in communication with a wireless communication system 120. The wireless communication system 120 may be a 5G system, a long term evolution (LTE) system, a code division multiple access (CDMA) system, a global mobile communication (GSM) system, a wireless local area network (WLAN) system, or some other wireless system. The CDMA system may implement Wideband CDMA (WCDMA), Time Division Synchronous CDMA (TD-SCDMA), CDMA2000, or other versions of CDMA. For simplicity, Figure 1A wireless communication system 120 is shown that includes two base stations 130 and 132 and one system controller 140. In general, a wireless system may include any number of base stations and any number of network entities.

[0045] The wireless device 110 may be referred to as a user equipment (UE), a mobile station, a terminal, an access terminal, a subscriber unit, a station, etc. The wireless device 110 may also be a cellular phone, a smart phone, a tablet, a wireless modem, a personal digital assistant (PDA), a handheld device, a laptop, a smartbook, a netbook, a cordless phone, a wireless local loop (WLL) station, a Bluetooth device, etc. The wireless device 110 may be capable of communicating with the wireless communication system 120. The wireless device 110 may also be capable of receiving signals from a broadcast station (e.g., broadcast station 134), receiving signals from satellites in one or more global navigation satellite systems (GNSS) (e.g., satellite 150), etc. The wireless device 110 may support one or more radio technologies for wireless communication, such as 5G, LTE, CDMA2000, WCDMA, TD-SCDMA, GSM, 802.11, etc.

[0046] The wireless device 110 may support carrier aggregation, which is an operation on multiple carriers. Carrier aggregation may also be referred to as multi-carrier operation. According to one aspect of the present disclosure, the wireless device 110 may be able to operate in a low frequency band of 698 to 960 megahertz (MHz), a mid-frequency band of 1475 to 2170 MHz, and / or a high frequency band of 2300 to 2690 MHz, an ultra-high frequency band of 3400 to 3800 MHz, and a long term evolution (LTE) in an LTE unlicensed band (LTE-U / LAA) of 5150 to 5950 MHz. Low frequency band, mid-frequency band, high frequency band, ultra-high frequency band, and LTE-U refer to five groups of frequency bands (or frequency band groups), each of which includes a plurality of frequency bands (or simply referred to as "bands"). For example, in some systems, each frequency band may cover up to 200 MHz and may include one or more carriers. For example, in LTE, each carrier may cover up to 40 MHz. Of course, the range of each of the frequency bands is exemplary and non-limiting, and other frequency ranges may be used. LTE Release 11 supports 35 frequency bands, which are referred to as LTE / UMTS bands and are listed in 3GPP TS 36.101. Wireless device 110 may be configured with up to 5 carriers in one or both frequency bands in LTE Release 11.

[0047] Figure 2 Shows such as Figure 1 A block diagram of an exemplary design of a wireless device 200, such as the wireless device 110, is shown. Figure 2An example of a transceiver 220 is shown, which may be a wireless transceiver (WTR). In general, conditioning of the signals in the transmitter 230 and the receiver 250 may be performed by one or more stages of one or more amplifiers, one or more filters, upconverters, downconverters, etc. The arrangement of these circuit blocks may be similar to Figure 2 The arrangement of the configuration shown is different. Furthermore, Figure 2 Other circuit blocks not shown in the figure may also be used to condition the signals in the transmitter 230 and the receiver 250. Unless otherwise specified, Figure 2 Any signal in or any other illustration in the drawings may be single-ended or differential. Figure 2 Some circuit blocks in .

[0048] exist Figure 2 In the example shown, the wireless device 200 generally includes a transceiver 220 and a data processor 210. The data processor 210 may include a memory (not shown) for storing data and program codes, and may generally include analog and digital processing elements. The transceiver 220 may include a transmitter 230 and a receiver 250 that support two-way communication. In general, the wireless device 200 may include any number of transmitters and / or receivers for any number of communication systems and frequency bands. All or a portion of the transceiver 220 may be implemented on one or more analog integrated circuits (ICs), radio frequency (RF) integrated circuits (RFICs), mixed signal ICs, etc.

[0049] A transmitter or receiver may be implemented using a superheterodyne architecture or a direct conversion architecture. In a superheterodyne architecture, the signal is frequency converted between RF and baseband in multiple stages of the receiver, for example, from RF to an intermediate frequency (IF) in one stage and from IF to baseband in another stage. In a direct conversion architecture, the signal is frequency converted between RF and baseband in one stage. Superheterodyne and direct conversion architectures may use different circuit blocks and / or have different requirements. Figure 2 In the example shown, transmitter 230 and receiver 250 are implemented using a direct conversion architecture.

[0050] In the transmit path, the data processor 210 processes the data to be transmitted. The data processor 210 also provides in-phase (I) and quadrature (Q) analog output signals to the transmitter 230 in the transmit path. In an exemplary aspect, the data processor 210 includes digital-to-analog converters (DACs) 214a and 214b for converting the digital signals generated by the data processor 210 into in-phase (I) and quadrature (Q) analog output signals (e.g., I and Q output currents) for further processing.

[0051] In the transmitter 230, low pass filters 232a and 232b filter the in-phase (I) and quadrature (Q) analog transmission signals, respectively, to reduce the undesired images caused by the existing digital-to-analog conversion. Amplifiers (Amp) 234a and 234b amplify the signals from the low pass filters 232a and 232b, respectively, and provide in-phase (I) and quadrature (Q) baseband signals. An up-converter 240 including up-converter mixers 241a and 241b up-converts the in-phase (I) and quadrature (Q) baseband signals together with the in-phase (I) and quadrature (Q) transmission (TX) local oscillator (LO) signals from the TX LO signal generator 290 to provide an up-converted signal. Filter 242 filters the up-converted signal to reduce the undesired images caused by the frequency up-conversion and the interference in the receiving band. A power amplifier (PA) 244 amplifies the signal from filter 242 to obtain a desired output power level and provides a transmit RF signal. The transmit RF signal is routed through a duplexer / switch 246 and transmitted via an antenna 248.

[0052] In the receive path, antenna 248 receives a communication or wireless signal and provides a received radio frequency (RF) signal that is routed through duplexer / switch 246 and provided to a low noise amplifier (LNA) 252. Duplexer / switch 246 is designed to operate at a specific receive (RX) to transmit (TX) (RX to TX) duplexer frequency spacing so that the RX signal is isolated from the TX signal. The received RF signal is amplified by LNA 252 and filtered by filter 254 to obtain the desired RF input signal. Down-conversion mixers 261a and 261b mix the output of filter 254 with the in-phase (I) and quadrature (Q) receive (RX) LO signals (i.e., LO_I and LO_Q) from RX LO signal generator 280 to generate in-phase (I) and quadrature (Q) baseband signals. The in-phase (I) and quadrature (Q) baseband signals are amplified by amplifiers 262a and 262b and filtered by low-pass filters 264a and 264b to obtain in-phase (I) and quadrature (Q) analog input signals, which are provided to the data processor 210. In the exemplary configuration shown, the data processor 210 includes analog-to-digital converters (ADCs) 216a and 216b for converting the analog input signals into digital signals for further processing by the data processor 210.

[0053] exist Figure 22, a transmit local oscillator (TX LO) signal generator 290 generates an in-phase (I) and quadrature (Q) TX LO signal for frequency up-conversion, while a receive local oscillator (RX LO) signal generator 280 generates an in-phase (I) and quadrature (Q) RX LO signal for frequency down-conversion. Each LO signal is a periodic signal having a specific base frequency. A phase-locked loop (PLL) 292 receives timing information from the data processor 210 and generates a control signal for adjusting the frequency and / or phase of the TX LO signal from the TX LO signal generator 290. Similarly, the PLL 282 receives timing information from the data processor 210 and generates a control signal for adjusting the frequency and / or phase of the RX LO signal from the RX LO signal generator 280.

[0054] The wireless device 200 may support carrier aggregation and may (i) receive multiple downlink signals transmitted by one or more cells on multiple downlink carriers of different frequencies, and / or (ii) transmit multiple uplink signals to one or more cells on multiple uplink carriers. For intra-band carrier aggregation, transmissions are sent on different carriers in the same frequency band. For inter-band carrier aggregation, transmissions are sent on multiple carriers of different frequency bands. However, it should be understood by those skilled in the art that the various aspects described herein may be implemented in systems, devices, and / or architectures that do not support carrier aggregation.

[0055] The wireless device 300 may support carrier aggregation, which is operation on multiple carriers. Carrier aggregation may also be referred to as multi-carrier operation.

[0056] Generally speaking, carrier aggregation (CA) can be divided into two types: intra-band CA and inter-band CA. Intra-band CA refers to operation on multiple carriers within the same frequency band, and inter-band CA refers to operation on multiple carriers in different frequency bands.

[0057] Figure 3A An example of continuous intra-band CA is shown. Figure 3A In the example shown, a wireless device (eg, wireless device 110) is configured with four contiguous carriers in the same frequency band, which is a frequency band in the low frequency band. A wireless device may send and / or receive transmissions on multiple contiguous carriers within the same frequency band.

[0058] Figure 3B An example of non-continuous intra-band CA is shown. Figure 3B In the example shown, a wireless device (e.g., wireless device 110) is configured with four non-contiguous carriers in the same frequency band, which is a frequency band in the low frequency band. The carriers may be separated by 5 MHz, 10 MHz, or some other amount. The wireless device may send and / or receive transmissions on multiple non-contiguous carriers within the same frequency band.

[0059] Figure 3C An example of inter-band CA in the same band group is shown. Figure 3C In the example shown, a wireless device (e.g., wireless device 110) is configured with four carriers in two frequency bands of the same frequency band group, which is a low frequency band. The wireless device can be configured with four carriers in different frequency bands in the same frequency band group (e.g., Figure 3C Transmissions are sent and / or received on multiple carriers in low frequency band 1 (LB1) and low frequency band 2 (LB2).

[0060] Figure 3D An example of inter-band CA in different band groups is shown. Figure 3D In the example shown, a wireless device (e.g., wireless device 110) is configured with four carriers in two frequency bands in different frequency band groups, the four carriers including two carriers in one frequency band in the low frequency band and two additional carriers in another frequency band in the high frequency band. The wireless device may be configured with four carriers in two frequency bands in different frequency band groups (e.g., Figure 3D Transmissions may be sent and / or received on multiple carriers in low frequency bands and high frequency bands in the same frequency band. FIG. 3A to FIG. 3D Four examples of carrier aggregation are shown. Other combinations of frequency bands and frequency band groups may also support carrier aggregation. For example, carrier aggregation may be supported for low and high bands, mid and high bands, high and high bands, and other frequency band combinations in unlicensed spectrum with ultra-high bands and long term evolution (LTE-U).

[0061] Figure 4A An example receive chain (e.g., including receiver 250) of an RF front end including an N-path filter 402 according to certain aspects of the present disclosure is illustrated. In certain aspects, Figure 2 The LNA 252 may include two separate LNAs (e.g., a first LNA 452A and a second LNA 452B). For some aspects, the first LNA 452A may be external to an RF integrated circuit (RFIC), while the second LNA 452B may be included in the RFIC along with other circuits (e.g., mixers 261a and 261b and low pass filters 264a and 264b). The low pass filters 264a and 264b may be baseband filters (BBF). For other aspects, the first LNA 452A and the second LNA 452B may be included in the RFIC along with other circuits. The N-path filter 402 may be connected to a node 406 between the LNAs 452A and 452B. In this manner, the N-path filter 402 may be used as a shunt filter having a frequency response 410, dedicated to passing signals in a desired receive (RX) band and rejecting signals having frequencies outside the band (including TX leakage and interference).

[0062] N-path filter 402 has N=4 parallel branches that are selectively connected to node 406, which is the common node of the branches. Those skilled in the art of N-path filters will appreciate that there may be more or less than N=4 branches in any of the various aspects of the present disclosure.

[0063] N-path filter 402 may include a number of switches 404 (eg, N switches, one switch in each filter branch), which may be used in Figure 4A The switches 404 in the N-path filter are implemented using n-channel metal oxide semiconductor (NMOS) transistors, which are labeled as transistors M1, M2, M3, and M4, respectively. For other aspects, the switches 404 in the N-path filter can be implemented using p-channel metal oxide semiconductor (PMOS) transistors or a combination of PMOS and NMOS transistors.

[0064] The four transistors M1, M2, M3 and M4 can be controlled using four 25% duty cycle signals P1, P2, P3 and P4, respectively. Figure 4B As shown in the timing diagram 412 of . In this way, one switch 404 can be opened before or when the next switch in the control signal sequence is closed. That is, each of the transistors M1, M2, M3 and M4 can be driven so that the transistors are activated in sequence and the periods during which each transistor is activated (e.g., each switch 404 is closed) ideally do not overlap, although a small amount of overlap can be tolerated for practical implementations. The duty cycle of the control signal can be a function of the number of filter branches N (e.g., equal to 1 / N). If any, the amount of overlap in the control signals P1, P2, P3, etc. can be a small fraction of the duty cycle (e.g., 1 / 10).

[0065] When closed, each switch 404 can connect a corresponding impedance ZA, ZB, ZC, or ZD to a node 406. Impedances ZA, ZB, ZC, and ZD may all have the same impedance value. One end of each impedance ZA, ZB, ZC, or ZD can be connected to a corresponding switch 404, and the other end of each impedance can be connected to a reference potential (e.g., electrical ground, a power supply voltage, or a bias voltage) of the N-path filter 402.

[0066] In this configuration, the center frequency of the frequency response 410 of the N-path filter 402 can be approximately equal to the switching frequency of the control signals P1, P2, P3, and P4 of the transistors M1, M2, M3, and M4, respectively. For example, the switching frequency can be considered to be the inverse of the period between the rising edges of the control signal P1, as shown by the vertical dashed line in the timing diagram 412. The control signals P1, P2, P3, and P4 can have the same frequency (e.g., the switching frequency), but have different phases. Moreover, the bandwidth of the frequency response 410 can be twice the bandwidth of the pole of the branch impedance (ZA, ZB, ZC, or ZD).

[0067] Figure 5A An example frequency response of an N-path filter is illustrated. Frequency response 502 may correspond to an N-path filter configured such that each of the branch impedances (e.g., ZA, ZB, ZC, and ZD) has only one pole (e.g., implemented using a resistor-capacitor (RC) load). An N-path filter with only one pole may provide a narrowband high-Q bandpass filter, but it may not provide sufficient out-of-band rejection when used as a filter for a wide RX band 503. Various aspects of the present disclosure provide an N-path filter with a wideband frequency response 504. For example, an N-path filter according to the present disclosure may provide a high-Q bandpass filter at a TX leakage frequency (f TX ) 506 provides rejection similar to a narrowband N-path filter but with a wide and flat passband shape.

[0068] Figure 5B Example frequency responses of an N-path filter according to various aspects of the present disclosure are illustrated. Frequency response 508 may correspond to an N-path filter without a frequency-dependent capacitor, high-pass filter, and / or buffer configured. Frequency response 508 includes transmission interference that may degrade the quality of a received RF signal. Frequency response 510 corresponds to an N-path filter configured with a frequency-dependent capacitor, high-pass filter, and / or buffer. Frequency-dependent capacitors, high-pass filters, and buffers may be implemented in the feedback path to provide high-pass feedback to create a sharper frequency response or a sharper filter. The sharper frequency response 510 eliminates transmission interference.

[0069] Some N-path filters may include only switches and capacitors. For example, an N-path filter may include only switches such as M1a and M1b (eg Figure 6 as shown) and capacitor C1 (as Figure 6 ). However, such an N-path filter may not achieve a flat frequency response or a wider bandwidth within the transmission frequency or transmission interference and high rejection. Thus, various aspects of the present disclosure include additional circuit systems that achieve a flat frequency response or a wider bandwidth within the transmission interference and high rejection.

[0070] Figure 6 An N-path filter 600 including a single high-pass filter path according to various aspects of the present disclosure is illustrated. For illustrative purposes, only a single branch of the N-path filter 600 is shown. However, the N-path filter 600 may include multiple branches that operate according to different phases (e.g., zero degrees, ninety degrees, one hundred and eighty degrees, two hundred and seventy degrees, etc.). For example, the N-path filter 600 includes a single branch (at a phase of zero degrees) that can be used to provide an impedance (e.g., Z A , Z B , Z C or Z D As illustrated, the N-path filter 600 includes a first transistor M1a, a second transistor M1b, a third transistor M1c, a fourth transistor M1d, a first capacitor C1, a second capacitor C2, a first high-pass capacitor CHP1, and a first resistor RHP1.

[0071] The N-path filter 600 is illustrated as being connected to a Thévenin equivalent circuit 602 (via node 606) having an input signal (e.g., voltage source Vin) and an input impedance (e.g., series resistor RS), which may represent an equivalent of a signal received by an antenna (e.g., antenna 248) and amplified by an LNA (e.g., first LNA 452A). In some aspects, node 606 may be located between two LNAs of a receiver (e.g., between LNAs 452A and 452B), before an LNA of a receiver (e.g., before LNA 452A), or after one or more LNAs of a receiver (e.g., after LNA 452B). Alternatively, node 606 may be coupled to a mixer (e.g., mixer 261a or mixer 261b).

[0072] The first transistor M1a, the second transistor M1b and the first capacitor C1 are connected in series with the Miller amplifier 620 to form a first feedback path. The second feedback path (or high-pass filter path) of the first branch includes a third transistor M1c, a fourth transistor M1d, a second capacitor C2, a first high-pass capacitor CHP1 and a first resistor RHP1. The first high-pass capacitor CHP1 and the first resistor RHP1 form a high-pass filter. In this case, the high-pass filter path includes only one zero (e.g., zero at 5.3MHz). The third transistor M1c and the fourth transistor M1d form a buffer coupled to the high-pass filter. In one aspect of the present disclosure, the buffer is a source follower (e.g., a voltage buffer or a bias source follower device). The source follower isolates the second capacitor C2 from the first high capacitor CHP1 and the first resistor RHP1. Otherwise, the second capacitor C2 may change the high-pass filter characteristics, which makes it difficult to program the zero frequency.

[0073] The second feedback path is merged with the first feedback path so that the current at a common node (e.g., one or more nodes 622 and / or 624) is shared between the first feedback path and the second feedback path. For example, the second feedback path is coupled to the first feedback path via nodes 622 and 624. Node 622 is located at the source (or drain) of the first transistor M1a, and node 624 is located at the drain (or source) of the second transistor M1b. The second capacitor C2 is coupled between node 622 and node 626, and the first capacitor C1 is coupled between node 622 and node 624. The first high-pass capacitor CHP1 is located between node 616 and node 624. Node 606 is coupled to a first terminal (e.g., input) of the Miller amplifier 620 and the drain (or source) of the first transistor M1a. The source (or drain) of the second transistor M1b is coupled to a second terminal (e.g., output) of the Miller amplifier 620.

[0074] The voltage at node 624 passes through a high-pass filter created by the first resistor RHP1 and the first high-pass capacitor CHP1. The third transistor M1c senses the high-pass signal voltage at node 616, and the source follower provides the same voltage at node 626, which drives the second capacitor C2. The first capacitor C1 and the second capacitor C2 are coupled to node 622, which means that the current flowing through the first capacitor C1 and the second capacitor C2 converges at node 622. The voltage at node 626 is a high-pass version of the voltage at node 624, which is the voltage after the Miller amplifier. Therefore, the current flowing through the second capacitor C2 is different from the current flowing through the first capacitor C1. For example, the current flowing through the second capacitor C2 only includes current with high-frequency components. As a result, there are more high-frequency components at the node 622 through which the second capacitor C2 and the corresponding high-path filter path are coupled, thereby obtaining stronger feedback at high frequencies. Therefore, a low impedance is generated at the transmission (TX) interference frequency, which causes a low TX interference voltage swing at nodes 606 and 622.

[0075] The source of the third transistor M1c is coupled to the drain of the fourth transistor M1d at node 626. The power supply 612 is coupled to the drain of the third transistor M1c, and the source of the fourth transistor M1d is coupled to ground 614 (or a reference potential). The buffer is coupled to the high pass filter via the gate of the third transistor M1c. For example, the gate of the third transistor M1c is coupled to a node 616 between the first high pass capacitor CHP1 and the first resistor RHP1. The third transistor M1c is biased based on a bias voltage VB provided at a node 618 of the high pass filter. The high pass filter and the buffer are configured so that a frequency dependent capacitor (e.g., the second capacitor C2) can be implemented in the N path filter 600. As the frequency increases, the impedance of the second capacitor C2 decreases, which can result in sharper filtering.

[0076] The first transistor M1a and the second transistor M1b can be used as a mixer to down-convert or up-convert the signal received by the N-path filter 600. In order to be used as a mixer, the gate of each transistor in the first transistor M1a and the second transistor M1b is coupled to the same LO (or different LO) to operate at the same or different LO frequencies. The first transistor M1a and the second transistor M1b can be driven by the phase and frequency of the same local oscillator (LO). For example, the N-path filter 600 can operate at a phase of zero (0) degrees, so that the first transistor M1a and the second transistor M1b are in phase with each other (for example, at a phase of zero (0) degrees). However, the first transistor M1a and the second transistor M1b can be driven by different LO frequencies. For example, the first transistor M1a and the second transistor M1b can be driven by different LO frequencies so that the transistors operate at different frequencies to achieve down-conversion or up-conversion of the radio frequency or baseband signal received by the N-path filter. Whether up-conversion occurs in the first transistor M1a or the second transistor M1b depends on the signal received at the node 606.

[0077] For example, nodes 622 and 624 may include or be characterized by baseband voltage, while nodes 606 and 628 include or are characterized by radio frequency signals. At in-band frequencies close to DC, the impedance associated with the first capacitor C1 and the second capacitor C2 is large. In this case, the received signal (which is close to DC) cannot pass through the first capacitor C1 and the second capacitor C2, so there is almost no feedback or no feedback (for example, high impedance plays an open circuit role). Thus, the first transistor M1a down-converts the signal received at node 606 (local oscillator frequency plus a low frequency close to DC), so that the signal at node 622 is a down-converted version of the signal at node 606. However, when there is an out-of-band frequency (for example, including a transmitter (TX) interference frequency of a local oscillator frequency plus a high frequency (for example, 100MHz)), the impedance C2 of the first capacitor C1 and the second capacitor is small, so feedback becomes stronger. In this case, the signal flows through node 628 to node 624 (down conversion), to the first capacitor C1 and the second capacitor C2, to node 622, and then to node 606 (up conversion). The signal at node 606 feeds the Miller amplifier (feedback loop).

[0078] Various aspects of the present disclosure use the Miller effect to achieve high impedance (e.g., capacitance) with reduced impedance (e.g., capacitor of reduced size) rather than large impedance (e.g., large capacitor). The Miller effect explains the increase in equivalent input impedance (e.g., capacitance) of an inverting voltage amplifier due to the amplification of the impedance effect between the input terminal and the output terminal. For example, a Miller amplifier is configured to amplify the Miller capacitance formed at the input node of the Miller amplifier.

[0079] Miller amplifier 620 is included in N-path filter 600 to reduce filter bandwidth without increasing the value of the capacitor. Miller amplifier 620 increases the loop gain associated with N-path filter 600. In one aspect, the Miller amplifier applies amplification (e.g., a gain of -4) to produce more voltage swing across the first capacitor C1 and / or the second capacitor C2. For example, for a total volt of 5 volts across capacitors C1 and / or C2, the capacitors (C1 and / or C2) experience 1 (positive) and 4 volts (negative). Therefore, when 1 volt is applied at node 606, node 628 swings 4 volts consistent with the Miller effect. As a result, the current flowing through capacitors C1 and / or C2 is five times more than in conventional implementations because the voltage across one or more capacitors is five times greater. Therefore, because no more current flows through one or more capacitors, more capacitance can be seen from Vin without adding large capacitors.

[0080] In one aspect, one or more notches are established in a feedback path (e.g., a first feedback path and / or a second feedback path) based on one or more capacitors (e.g., C1) to form a bandpass filter. At higher frequencies, more capacitance is provided, which results in a sharper frequency response curve and creates a sharper filter, such as Figure 5B As shown. The sharpening of the frequency response occurs due to the Miller effect applied to the second capacitor C2 while creating high-order filtering. The value of capacitor C1 depends on R S and the desired bandwidth, the R S is the series resistance of the Thévenin equivalent circuit 602. For example, if the desired bandwidth is ten (10) MHz and the series resistance R S is one hundred (100) ohms, then the capacitor C1 is eight (8) pF, where the Miller gain is -4. The value of capacitor C2 is determined by the high pass filter zero frequency, the series resistor R S and the desired bandwidth. For example, for a zero at 5.3 MHz, the capacitance of capacitor C2 may be fifteen (15) picofarads.

[0081] exist Figure 6 In this aspect shown, the high pass filter path includes a filter with one zero. The high pass filter path is used in conjunction with the low pass filter to form a band pass filter. High pass filter feedback is provided to the first feedback path to create a sharper frequency response. For example, the high pass filter feedback is provided to the second capacitor C2, which is only visible at higher frequencies.

[0082] Figure 7An N-path filter 700 including two high-pass filter paths according to various aspects of the present disclosure is illustrated. For illustrative purposes, only a single branch of the N-path filter 700 is shown. However, the N-path filter 700 may include multiple branches operating according to different phases (e.g., zero degrees, ninety degrees, one hundred and eighty degrees, two hundred and seven degrees, etc.).

[0083] For illustration purposes, Figure 7 Some of the markings and numbers of the equipment and features of Figure 6 The markings and numbers are similar. Figure 6 In addition to the characteristics of Figure 7 Another high pass filter path (or third feedback path) is included. The third feedback path includes a third capacitor C3 connected in series with the second buffer and the second high pass filter. The second buffer includes a fifth transistor M1e and a sixth transistor M1f. The second high pass filter includes a second high pass capacitor CHP2 and a second resistor RHP2.

[0084] A terminal of the third capacitor C3 is coupled to the same node 622 as a terminal of the second capacitor C2. Another terminal of the third capacitor C3 is coupled to a node 726, which is common to the transistors M1e and M1f. For example, the source of the fifth transistor M1e and the drain of the sixth transistor M1f are coupled to each other via the node 726. The high pass filter is coupled between the gate of the fifth transistor M1e and the node 626. A node 716 between the second high pass capacitor CHP2 and the second resistor RHP2 is also coupled to the gate of the fifth transistor M1e.

[0085] Similar to the second capacitor C2, the third capacitor C3 is a frequency dependent capacitor. However, the third capacitor C3 is only visible at frequencies above the frequency where the second capacitor C2 is visible. The additional high pass filter path introduces another zero (two zeros total), which allows the path through the third capacitor C3 to provide more out-of-band rejection, such as Figure 8 shown.

[0086] Figure 8 8 is a graph 800 showing a first frequency response 802 of a low-order bandpass filter and a second frequency response 804 of a high-order bandpass filter. For example, a low-order bandpass filter may be Figure 6 The N-path filter 600 corresponds to a single zero associated with the high pass filter path. The high order band pass filter can be connected to Figure 7800, wherein two zeros are associated with the high pass filter path. The RF bandwidth 803 of the two filters is the same. For example, in this case, both the lower band pass filter and the upper band pass filter have the same RF bandwidth 803 (e.g., twenty megahertz (MHz)). For an offset of 40 megahertz (40 MHz), the rejection of the high order band pass filter is four decibels (4 dB) more than the rejection of the low order band pass filter.

[0087] Fig. 9 The Thévenin equivalent circuit 900 of the N-path filter 700 including two high-pass filter paths according to various aspects of the present disclosure is represented. The Thévenin equivalent circuit 900 illustrates a baseband equivalent model of the N-path filter 700 as seen from the antenna represented by the Thévenin equivalent circuit 602. The first high-pass filter path and the second high-pass filter path have transfer functions in the S-domain represented by H(s). The s-plane is a complex plane in which Laplace transforms are plotted. It is a mathematical domain in which processes modeled using time-based functions in the time domain are treated as equations in the frequency domain rather than viewing them. For example, the transfer function H(s) can be represented by the following equation:

[0088]

[0089] Where S is the complex frequency parameter for Laplace transform, R B represents a high-pass resistor (eg, the first resistor RHP1 or the second resistor RHP2), and C B represents a high-pass capacitor (eg, a first high-pass capacitor CHP1 or a second high-pass capacitor CHP2).

[0090] The gain of the N-path filter 700 can be calculated as follows:

[0091]

[0092] Where V1(s) is the output voltage of the N-path filter 700, R SW is the switch resistance of the branch switch (for example, the branch resistance of transistor M1a), R S is the series resistance of the Thévenin equivalent circuit 602, and A V is the gain of the Miller amplifier. It should be noted that with the series resistor R S It gets bigger, and the transmission rejection increases.

[0093] Fig.10 An N-path filter 1000 having multiple branches is illustrated, wherein each branch includes two high-pass filter paths. For illustrative purposes, Fig.10 Some of the markings and numbers of the equipment and features of Figure 7 However, the N-path filter 1000 includes multiple branches, rather than Figure 7 For example, the plurality of branches may include a first branch 1010 and a second branch 1030. Each of the first branch 1010 and the second branch 1030 is connected to Figure 7 1010 is similar to the single branch shown in . However, for the purpose of illustration, only a portion of the second branch 1030 is shown. For example, the second branch 1030 shows transistors M2a and M2b and a first capacitor C1. The first branch 1010 can operate according to a first phase (e.g., zero degrees), while the second branch 1030 can operate according to a second phase (e.g., ninety degrees).

[0094] In this aspect, a first terminal of the first capacitor C1 is coupled to both a node 606 (associated with an input voltage) and an input of a RF Miller amplifier 620 via a first transistor or switch M1a. A second terminal of the first capacitor C1 is coupled to an output of the RF Miller amplifier 620 via a second transistor M1b. The first transistor M1a and the second transistor M1b are configured as a mixer to up-convert or down-convert a RF signal through the N-path filter 1000.

[0095] A local oscillator (not shown) is coupled to the gate of the first transistor M1a and the gate of the second transistor M1b to configure each of the transistors to function as a mixer. The first branch 1010 and the second branch 1030 are selectively coupled to a shared circuit node (e.g., node 606) that receives an input voltage Vin. However, in this aspect, the Miller amplifier 620 is selectively shared across the first branch 1010 and the second branch 1030. Therefore, only one Miller amplifier supports both branches.

[0096] Fig.11 An N-path filter 1100 having multiple branches is illustrated, wherein each branch includes two high-pass filter paths. For illustration purposes, Fig.11 Some of the markings and numbers of the equipment and features of Figure 7 The markings and numbers are similar.

[0097] The N-path filter 1100 includes multiple branches, rather than Figure 7 A single branch is shown. Additionally, Fig.11 Use baseband Miller amplifiers for baseband amplification instead of Figure 7 For example, the plurality of branches may include a first branch 1110 and a second branch 1130. Each of the first branch 1110 and the second branch 1130 is connected to Figure 7The single branch shown is similar. A first terminal of the Miller amplifier 620 is coupled or connected to the source of the transistor M1a and a node 622 at the first terminal of the first capacitor C1 of the first branch 1110. A second terminal of the Miller amplifier 620 is coupled or connected to a node 624 at the second terminal of the first capacitor C1 of the first branch 1110. However, after the Miller amplifier 620, the out-of-band signal (e.g., TX interference) does not see the mixer (e.g., transistor M1a).

[0098] For illustration purposes, only a portion of the second branch 1130 is shown. For example, the second branch 1130 shows transistor M2a (without transistor M2b) and a first capacitor C1. Similar to the first branch 1110, a first terminal of the Miller amplifier 1120 of the second branch 1130 is coupled or connected to a source of the transistor M2a and a node 1122 at a first terminal of the first capacitor C1 of the second branch 1130. A second terminal of the Miller amplifier 1120 is coupled or connected to a node 1124 at a second terminal of the first capacitor C1 of the second branch 1130. The first branch 1110 can operate according to a first phase (e.g., zero degrees), while the second branch 1130 can operate according to a second phase (e.g., ninety degrees).

[0099] A first terminal of the first capacitor C1 of the first branch 1110 is coupled to a node 606 (associated with the input voltage Vin) via a first transistor or switch M1a. A second terminal of the first capacitor C1 is coupled to the output of the RF Miller amplifier 620.

[0100] The first transistor M1a is configured as a mixer. A local oscillator (not shown) is coupled to the gate of the first transistor M1a to configure the transistor to function as a mixer. The first branch 1110 and the second branch 1130 are selectively coupled to a shared circuit node (e.g., node 606) associated with the input voltage Vin. However, in this aspect, the Miller amplifier 620 and the Miller amplifier 1120 are respectively assigned to the first branch 1110 and the second branch 1130. Thus, a Miller amplifier is assigned to each of the branches. Although not shown, it should also be envisioned that, as shown in FIG. Fig.11 as well as Figure 6 and Figure 7 As shown, a Miller amplifier is provided. In this case, the Miller amplifier is selectively enabled depending on which mode is selected.

[0101] Fig.12A simplified flow chart of a method 1200 for filtering a wireless signal at a receiver (e.g., a user device) is depicted. At block 1202, a radio frequency signal is received at a shared circuit node that is selectively coupled to each of a plurality of branches of an N-path filter. Each branch of the N-path filter includes a first feedback path that includes a Miller amplifier; and a second feedback path that has a node in common with the first feedback path. At block 1204, when a radio frequency signal is received at a first frequency, a high impedance is generated at the common node to prevent the radio frequency signal from traversing the first feedback path and the second feedback path. At block 1206, when a second radio frequency signal is received at a second frequency that is higher than the first frequency, a low impedance is generated at the common node to allow the radio frequency signal to traverse the first feedback path and / or the second feedback path.

[0102] According to one aspect of the present disclosure, an N-path filter is described. The filter includes a first device, a second device, and a third device for generating impedance. The first impedance generating device may be, for example, a first capacitor C1. The second impedance generating device may be, for example, a second capacitor C2. The third impedance generating device may be, for example, a third capacitor C3. In another aspect, the aforementioned device may be any module or any device or material configured to perform the functions listed by the aforementioned device.

[0103] Fig.13 is a block diagram illustrating an exemplary wireless communication system in which the configuration of the present disclosure may be advantageously employed. For purposes of illustration, Fig.13 Three remote units 1320, 1330, and 1350 and two base stations 1340 are shown. It should be appreciated that the wireless communication system can have more remote units and base stations. Remote units 1320, 1330, and 1350 include IC devices 1325A, 1325B, and 1325C, which include the disclosed N-path filters. It should be appreciated that other devices may also include the disclosed N-path filters, such as base stations, switching devices, and network devices. Fig.13 Forward link signals 1380 from base station 1340 to remote units 1320, 1330, and 1350 and reverse link signals 1390 from remote units 1320, 1330, and 1350 to base station 1340 are shown.

[0104] exist Fig.13, remote unit 1320 is shown as a mobile phone, remote unit 1330 is shown as a portable computer, and remote unit 1350 is shown as a fixed location remote unit in a wireless local loop system. For example, the remote unit may be a mobile phone, a handheld personal communication system (PCS) unit, a portable data unit such as a personal digital assistant (PDA), a GPS-enabled device, a navigation device, a set-top box, a music player, a video player, an entertainment unit, a fixed location data unit (such as a meter reading device), or other communication devices that store or retrieve data or computer instructions, or a combination thereof. Although Fig.13 Remote units according to various aspects of the present disclosure are illustrated, but the present disclosure is not limited to these exemplary illustrated units. Various aspects of the present disclosure may be suitably used in many devices including N-path filters.

[0105] For firmware and / or software implementations, the methods can be implemented using modules (e.g., procedures, functions, etc.) that perform the functions described herein. A machine-readable medium embodying instructions in a tangible manner can be used to implement the methods described herein. For example, software code can be stored in a memory and executed by a processor unit. The memory can be implemented within a processor unit or outside a processor unit. As used herein, the term "memory" refers to a type of long-term, short-term, volatile, non-volatile, or other memory, and is not limited to a specific type of memory or the number of memories, or the type of medium on which the memory is stored.

[0106] If implemented in firmware and / or software, the function may be stored as one or more instructions or codes on a computer-readable medium. Examples include computer-readable media encoded using data structures and computer-readable media encoded using computer programs. Computer-readable media include physical computer storage media. Storage media may be available media that can be accessed by a computer. As an example and not limitation, such computer-readable media may include RAM, ROM, EEPROM, CD-ROM or other optical disk storage devices, magnetic disk storage devices or other magnetic storage devices, or other media that can be used to store desired program codes in the form of instructions or data structures and that can be accessed by a computer; as used herein, disks and discs include compact disks (CDs), laser disks, optical disks, digital versatile disks (DVDs), floppy disks, and Blu-ray disks, wherein disks generally reproduce data magnetically, and discs reproduce data optically using lasers. The above combinations should also be included within the scope of computer-readable media.

[0107] In addition to being stored on a computer-readable medium, instructions and / or data may also be provided as signals on a transmission medium included in a communication device. For example, a communication device may include a transceiver with signals indicating instructions and data. The instructions and data are configured to cause one or more processors to implement the functions outlined in the claims.

[0108] The various illustrative logical blocks, modules, and circuits described in conjunction with the disclosure herein may be implemented or executed using a general purpose processor, a digital signal processor (DSP), an application specific integrated circuit (ASIC), a field programmable gate array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general purpose processor may be a microprocessor, but alternatively, the processor may be any conventional processor, controller, microcontroller, or state machine. The processor may also be implemented as a combination of computing devices, for example, a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in combination with a DSP core, or any other such configuration.

[0109] Although the present disclosure and its advantages have been described in detail, it should be understood that various changes, substitutions and modifications can be made herein without departing from the technology of the present disclosure defined by the appended claims. For example, relational terms such as "above" and "below" are used relative to a substrate or an electronic device. Of course, if the substrate or electronic device is reversed, the top becomes the bottom, and vice versa. Additionally, if it is lateral orientation, the top and the bottom can refer to the side of the substrate or electronic device. Moreover, the scope of the present application is not intended to be limited to the specific configurations of the processes, machines, manufactures, and material compositions, means, methods and steps described in the specification. It should be easily understood by a person of ordinary skill in the art based on the disclosure that according to the present disclosure, processes, machines, manufactures, material compositions, means, methods or steps that currently exist or are to be developed later to perform functions substantially the same as the corresponding configurations described herein or achieve results substantially the same as the corresponding configurations described herein can be utilized. Therefore, the appended claims are intended to include such processes, machines, manufactures, material compositions, means, methods or steps within their scope.

Claims

1. An N-path filter, comprising: The first branch includes: The first feedback path includes: amplifier having an input coupled to an input voltage, and a first capacitor coupled to both the input voltage and the output of the amplifier; and A second feedback path having a common node with the first feedback path, the second feedback path comprising: a first high pass filter, coupled to the output of the amplifier, a second capacitor coupled to both the first capacitor and the first high pass filter, and A buffer is coupled to the output of the first high pass filter and to the second capacitor, the buffer comprising a biased source follower device.

2. The N-path filter according to claim 1, further comprising: The third feedback path includes: a second high pass filter coupled to the output of the first high pass filter, and A third capacitor is coupled to both the first capacitor and the second capacitor and to the second high pass filter.

3. The N-path filter of claim 2, wherein the buffer is coupled to an input of the second high pass filter.

4. The N-path filter of claim 1 , integrated into a receive (RX) chain before or after a low noise amplifier (LNA) according to a split configuration.

5. The N-path filter of claim 1, wherein the amplifier comprises a Miller amplifier or an amplifier based at least in part on the Miller effect.

6. The N-path filter of claim 5, wherein the Miller amplifier comprises a radio frequency Miller amplifier, and wherein, A first terminal of the first capacitor is coupled to both the input voltage and an input of the Miller amplifier via a first switch; and A second terminal of the first capacitor is coupled to the output of the Miller amplifier via a second switch.

7. The N-path filter of claim 5, wherein the Miller amplifier comprises a baseband Miller amplifier, and wherein, a first terminal of the first capacitor coupled to a first terminal of a first switch and to the input of the Miller amplifier; as well as A second terminal of the first capacitor is coupled to the output of the Miller amplifier, and a second terminal of the first switch is coupled to the input voltage.

8. The N-path filter of claim 1 , further comprising a second branch, wherein the first branch and the second branch are selectively coupled to a shared circuit node that receives the input voltage, wherein the first branch and the second branch operate according to a first phase and a different second phase, respectively, and wherein the amplifier is selectively shared across the first branch and the second branch.

9. The N-path filter of claim 1 , further comprising a second branch, wherein the first branch and the second branch are selectively coupled to a shared circuit node that receives the input voltage, wherein the first branch and the second branch operate according to a first phase and a different second phase, respectively, and wherein the amplifier is assigned to the first branch and another amplifier is assigned to the second branch.

10. A method for filtering a wireless signal at a receiver, comprising: receiving a radio frequency signal at a shared circuit node, the shared circuit node selectively coupled to each of a plurality of branches of an N-path filter, wherein each branch of the N-path filter includes a first feedback path and a second feedback path, the first feedback path including an amplifier and the second feedback path having a common node with the first feedback path; generating a high impedance at the common node when the RF signal is received at a first frequency to prevent the RF signal from traversing the first feedback path and / or the second feedback path; as well as When the RF signal is received at a second frequency higher than the first frequency, a low impedance is generated at the common node to allow the RF signal to traverse the first feedback path and / or the second feedback path. The method of claim 10 , wherein the first frequency comprises an in-band frequency and the second frequency comprises an out-of-band frequency.

12. The method according to claim 10, further comprising: Down-converting the radio frequency signal of the first frequency.

13. The method according to claim 10, further comprising: amplifying the radio frequency signal received at the second frequency using the amplifier; Down-converting the amplified radio frequency signal; providing the radio frequency signal, the radio frequency signal being down-converted to an up-converter via the low impedance generated by first and second capacitors associated with the first and second feedback paths; as well as The radio frequency signal is up-converted and provided, the radio frequency signal being up-converted to the shared circuit node.

14. The method of claim 10, wherein the amplifier comprises a Miller amplifier or an amplifier based at least in part on the Miller effect.

15. An N-path filter, comprising: The first branch includes: The first feedback path includes: amplifier having an input coupled to an input voltage, and a first device for generating an impedance, the first impedance generating device being coupled to both the input voltage and the output of the amplifier; and A second feedback path having a common node with the first feedback path, the second feedback path comprising: a first high pass filter, coupled to the output of the amplifier, a second device for generating an impedance, the second impedance generating device being coupled to both the first impedance generating device and the first high pass filter, and A buffer is coupled to the output of the first high pass filter and to the second impedance generating device, the buffer comprising a biased source follower device.

16. The N-path filter according to claim 15, further comprising: The third feedback path includes: a second high pass filter coupled to the output of the first high pass filter, and A third device is used to generate impedance, the third impedance generating device is coupled to both the first impedance generating device and the second impedance generating device, and is coupled to the second high pass filter. 17 . The N-path filter of claim 16 , further comprising a second buffer coupled to an output of the second high pass filter and to the third impedance generating device.

18. The N-path filter of claim 15, integrated into a receive (RX) chain before or after a low noise amplifier (LNA) according to a split configuration.

19. The N-path filter of claim 15, wherein the amplifier comprises a Miller amplifier or an amplifier based at least in part on the Miller effect.

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