Apparatus and method for analog electronic fiber dispersion and bandwidth precompensation (EDPC) for use in 50GBPS and greater PAMN optical transceivers

Abstract

The present invention provides a cost-effective high data rate optical data transceiver that includes an electronic analog transversal filter that simultaneously provides one or more of bandwidth compensation and forward impairment compensation for a transmitted optical signal. These devices may include an analog transversal filter circuit configured to condition an analog modulator signal from a PAM processor. The conditioning may involve approximate Hilbert transform, dispersion precompensation, or both.

Description

Technical Field

[0001] The technical field of the present invention relates to optical data transceivers, and more particularly to fiber-optic transceiver modules that provide greater than or equal to 50 Gigabits per second ("Gbps") per wavelength in high throughput short to medium distance fiber-optic communication links (e.g., in access networks, data centers, data center interconnects, and campus networks). Background Art

[0002] Traditional fiber-optical communication links transmit binary data directly as binary, for example, sending a "power on" burst to represent a binary "1", or sending a "power off" burst to represent a binary "0". Various alternatives refer to this type of encoding in various ways, and in most cases, taking certain liberties, can be collectively described as on-off keying ("OOK"). "Bits" are transmitted at a bit rate of 1 / T bits per second, or "bps". For contemporary high-speed fiber-optic links, bit rates are typically expressed as "Gigabits" (10 9 ) per second or "Gbps", and accordingly, T (bit period) will be expressed in picoseconds or "ps". For example, 10Gbps links with a 100ps bit period are almost ubiquitous. Following terminology developed for communications and control theory, the physical communications link and medium from transmitter to receiver or vice versa is often referred to as a communications "device".

[0003] As other technologies advance and the demand for transmitting more data increases, a huge challenge arises because available communication equipment cannot reliably support OOK for data rates significantly greater than 10Gbps per optical wavelength. Complex optical link systems that utilize coherent optical technology with high-performance digital signal processing (DSP) control have emerged to provide 100Gbps / wavelength and higher bit rates, which are suitable for high-end applications and have high costs compared to existing communication equipment. However, there are many valuable applications that require more than 10Gbps but do not have such "high-end applications", but trying to use coherent systems for such less demanding applications does not alleviate the "high cost" associated with coherent technology. Therefore, it is necessary to establish a more cost-effective method to provide increased data capacity in important applications using communication equipment that does not meet high-end requirements. This situation is common, for example, when many independent interconnections need to be made and the length of the optical fiber equipment is in the medium range, such as from a few hundred meters to about 40-80km (sometimes referred to as "short haul") . This situation is encountered in many valuable applications such as access networks, data centers, data center interconnects, and campus networks.

[0004] One key approach that has been particularly successful is the switch from OOK to multi-level signaling, transmitting more than 1 bit of binary data per period T. In this case, the optical transmission per period T is a more densely valued "symbol" rather than just a bit. In that case, the terminology for the physical transmission rate changes from "bits per second" to the symbol rate, often called the baud ("Bd") rate. For example, 25*10 9 Symbols per second (approximately) is referred to as "25GBd", which has a corresponding symbol period T = 40ps. Such links are also often referred to in terms of effective bit rate, so a 25GBd system can support, for example, 50Gbps (2 bits per symbol) or 100Gbps (4 bits per symbol), but in any case the symbol period T remains 40ps.

[0005] Coherent systems take this dense symbol approach to practical extremes, such as demonstrating 600Gbps on a single wavelength by using 64GBd and 12 bits per symbol (with a portion of the data siphoned off to improve quality). The approach described in this article takes a less extreme but generally more cost-effective approach. Perhaps the most fundamental and widely sought-after improvement is to replace OOK with 4-level pulse amplitude modulation (PAM4) to provide 2 bits per symbol. The same principles could easily be applied to consider PAM8 (3 bits per symbol), PAM16 (4 bits per symbol), or any corresponding quadrature amplitude modulation (QAMn), but PAM4 has many supporting technologies, so the application is particularly interesting. Summary of the invention

[0006] For this invention, we propose a novel approach that exploits a technical discovery that adding an analog linear equalizer chip in a transceiver, for example, using a short-haul 100Gbps PAM4 DSP chip (to convert 4x25Gbps or 2x50Gbps on the host side to 2x50Gbps on the line side), can improve performance in edge or inaccessible applications for that data rate. The added analog chip can provide electronic dispersion pre-compensation (EDPC), while the PAM4 DSP can provide electronic dispersion post-compensation for up to 40km or more distances, as well as other filtering enhancements, without requiring any optical dispersion compensation. The enhancements provided in this manner are significantly more efficient in size, cost, and power consumption than a corresponding upgrade in DSP performance or the addition of fixed or adjustable optical dispersion compensators. The EDPC chip can provide any of the following enhancements: (a) approximation of the Hilbert transform of the PAM4 signal to transform it into a single sideband (PAM4-SSB) signal, together with additional fiber dispersion precompensation; and (b) dispersion precompensation for conventional double sideband (PAM4-DSB) PAM4 signals. Conventional commercial digital PAM4 chips for transceivers are responsible for the host side interface, FEC, and a powerful line side post-compensation equalizer. This operating principle is applicable to 50Gbps per wavelength and 100Gbps per wavelength supported by current transceiver standards, and can be equally applicable to higher data rates that will appear in future transceivers. In order to increase the transmission distance to SSMF beyond 40km, the number of equalizer taps in the EDPC can be increased (limited by the power consumption of the pluggable module) and / or a simple passive optical dispersion compensator can be further added to keep the remaining fiber dispersion range within the 40km window. Passive (fixed) dispersion compensation elements are known in the art and can be purchased from Proximion AB. In this case, since the distance will be greater than 40 km, there will necessarily be one or two optical amplifiers, which can additionally be used to compensate for the losses caused by the passive optical dispersion compensator.

[0007] Many OSSB schemes have been proposed for binary data transmission to use analog Hilbert transformers (note that digital Hilbert transformers can also be used, but require new DSP chips), for example, [8, 9] for baseband 10Gbps NRZ signals and

[10] for microwave subcarrier 2.5Gbps NRZ signals.

[0008] In a first aspect, the present invention relates to an optical transceiver module for n-level pulse amplitude modulation (PAMn) optical symbols, thereby providing an interface between electronic data signals on a host and optical symbols transmitted at a baud rate through an optical communication device, where n≥2, and the transceiver includes: a receiver portion, a PAMn digital signal processing (DSP) circuit, a transmitter portion, and an analog transversal filter circuit. The receiver portion may include at least one optical receiver to convert a received optical signal into an analog electrical signal. The PAMn DSP circuit may generally provide at least a logical interface with electronic host data, a forward error correction (FEC) function, analog-to-digital conversion of the electrical signal from the optical receiver, digital adaptive filtering of the converted received signal, and reconstruction of the filtered received signal into data. The transmitter portion generally includes at least one laser and at least one interference modulator. The analog transversal filter circuit is configured as an electronic dispersion precompensator (EDPC) to filter a multi-level transmission signal, such as a PAMn signal provided by a PAMn DSP, wherein the filtered signal emitted from the EDPC is connected to the signal input of the interference modulator. In variations of this aspect, the current required to drive the modulator electrodes according to the filtered signal voltage may be provided by an electronic amplifier external to the EDPC, or the current may be provided directly by an amplifier integrated within the EDPC.

[0009] On the other hand, the present invention relates to a method for extending a transmission distance using readable symbols of an optical transmitter operating with PAMn modulation, wherein the method comprises: using an analog transversal filter circuit to condition an analog modulator signal from a PAMn processor to perform an approximate Hilbert transform, dispersion precompensation, or both to form an conditioned modulator signal; and modulating an optical laser using an optical interferometer modulator based on the conditioned modulator signal.

[0010] In another aspect, the present invention relates to a method for determining tap weights of an analog transversal filter circuit, wherein the analog transversal filter circuit is configured to condition an analog modulator signal from a PAMn processor to perform an approximate Hilbert transform, dispersion precompensation, or both to form an conditioned modulator signal, wherein the method includes iteratively correcting the tap weights to improve dispersion for a composite range of lengths of multiple optical fiber devices. BRIEF DESCRIPTION OF THE DRAWINGS

[0011] Figure 1A Depicted is a schematic diagram of an analog transversal filter that approximates the Hilbert transform using delay elements for one symbol period.

[0012] Figure 1B Depicted is a schematic diagram of an analog transversal filter that approximates the Hilbert transform using delay elements with two-thirds of the sampling period.

[0013] Figure 2A Depicted is an optical Mach-Zehnder modulator configuration for providing SSB modulation using dual drive capability, where d1 and d2 are independent.

[0014] Figure 2B Depicted is an optical Mach-Zehnder modulator configuration for providing SSB modulation using IQ drive capability, where d 1a =-d 1b , d 2a =-d 2b , and d1 and d2 are independent.

[0015] Figure 3A Depicted is a block diagram of an optical transceiver module that provides a 100Gbps link using PAM4 with two 50Gbps wavelengths.

[0016] Figure 3B Depicted is a block diagram of an optical transceiver module that provides a 100 Gbps link using PAM4 at one 100 Gbps wavelength.

[0017] Figure 4A Shown is the optical configuration of a dual wavelength transmitter whose light outputs are combined on a single optical fiber.

[0018] Figure 4B Shown is the optical configuration of a dual wavelength transmitter whose light outputs are kept separate on separate output fibers.

[0019] Figure 5A The optical configuration of a dual wavelength receiver is shown with light inputs presented combined on a single optical fiber.

[0020] Figure 5B Shown is the optical configuration of a dual wavelength receiver having light inputs presented separately on separate input optical fibers.

[0021] Fig. 6A Depicted is a block diagram of an optical transceiver module that provides a 400Gbps link using PAM4 with eight 50Gbps wavelengths.

[0022] Figure 6B Depicted is a block diagram of an optical transceiver module that provides a 400Gbps link using PAM4 with four 100Gbps wavelengths.

[0023] Figure 7 A communications device for determining optimal tap weighting parameters for an optical data link is depicted.

[0024] Figure 8 Presented for Figure 7 Flow chart of the preferred optimization process of communication equipment.

[0025] Fig.9A Depicted is a simulated optical eye diagram transmitted by a transmitter with EDPC tap weights obtained by computational convolution of the Hilbert transform and dispersion precompensation of 10 km of optical fiber.

[0026] Fig. 9B Describes the Fig.9A The received signal level of the transmitted signal.

[0027] Fig. 9C Describes the simulation of Figure 8 The algorithm optimizes the EDPC tap weights of the transmitter to emit the optical eye diagram.

[0028] Fig.9D Depicts the image before DSP processing Fig. 9C The received signal level of the transmitted signal.

[0029] Fig. 10A yes Figure 3A A graph showing experimental results for one of the 50 Gbps wavelengths of the illustrated embodiment.

[0030] Fig. 10B yes Figure 3A A graph showing experimental results of optical signal-to-noise ratio as a function of one of the wavelengths at 50 Gbps for the illustrated embodiment. DETAILED DESCRIPTION

[0031] Optical data transceivers are described herein, and also include electronic analog transverse filters to provide key signal conditioning functions in converting digital data formatted by PAMn (n≥2) into appropriately modulated optical signals emitted from the optical transmitter of the transceiver. Herein, the electronic analog transverse filter circuit will be referred to as an electronic dispersion precompensator (EDPC). The following signal conditioning method can be equally applicable to C, L and O optical bands and any other practically useful optical bands in optical fiber links. The transceiver with EDPC described herein can provide a very efficient method to extend the transmission distance, while also providing a smaller footprint for incorporating into standard module formats, which should be followed to be incorporated into conventional systems. The description and examples herein will distinguish parallel optical data streams according to "per wavelength". For transceivers using parallel optical fibers, even if the wavelengths between the optical fibers are not different, the "per wavelength" here can be equally understood as "per optical fiber". The present invention adapts to parallel wavelengths / parallel optical fibers through simple integration and / or discrete replication. It should be assumed that the description herein is applicable to any reasonable number of parallel wavelengths and / or parallel optical fibers supported by optical fiber equipment.

[0032] In general, the transceivers in this article are applicable to various PAMn modulation formats and other directly detected multi-bit symbol formats, but since PAM4 is widely adopted in existing deployments, the discussion focuses on PAM4 for a more specific description. PAM4 modulation has been widely used in short-haul transmission systems, as shown in Table 1 below. Table 1 references IEEE standards or multi-source agreement (MSA) standards related to PAM4 modulation. All entries except the last one use a 1.3μm wavelength to avoid fiber dispersion in standard single-mode fiber (SSMF). However, operating at a 1.3μm wavelength brings certain other issues, such as (a) higher optical fiber transmission loss (≤0.35dB / km) than at 1.55μm (≤0.20dB / km), and (b) the technology for 1.3μm does not provide DWDM (dense wavelength division multiplexing) wavelengths to increase the number of supported wavelengths, and therefore does not increase the total transmission per fiber.

[0033] Table 1

[0034]

[0035]

[0036]

[0037] (“DC” = Data Center; “CO” = Central Office, e.g., access network; EML = Externally Modulated Laser; DML = Directly Modulated Laser; SiPho = Silicon Photonics)

[0038] As a result, a commercial 100Gbps optical transceiver module [6] has emerged, which uses two 1.55μm wavelengths to carry 100Gbps data for data center interconnect (DCI). Each of the two wavelengths in the transceiver module carries approximately 50Gbps at 25GBd based on PAM4 modulation. However, the 50Gbps PAM4 wavelength is very sensitive to the fiber dispersion inherent in the fiber equipment. It cannot tolerate the dispersion of even a few kilometers of SSMF, so a tunable optical dispersion compensator (ODC) is required in the transmission link. This active tunable ODC is a complex and management-intensive item of optical hardware, which makes the operation of optical networks very complex and cumbersome. To solve this problem, people have conducted intensive research [7] and proposed to eliminate the fiber dispersion compensation for transmitting 50Gbps or 100Gbps PAM4 per wavelength. However, almost all proposals are related to new digital signal processing (DSP) algorithms, which will require a large investment in manufacturing, acquiring and operating custom DSP ASICs.

[0039] Another application that requires short-haul chromatic dispersion (CD) compensation is 100Gbps / wavelength over 10km at 1.3μm CWDM wavelengths. The nominal wavelengths defined in 400G-LR4 [1] are 1271, 1291, 1311 and 1331nm, each with a drift / tolerance of ±6.5nm. Many typical optical transmitters exhibit “linear frequency modulation”, where the optical wavelength is slightly deflected when the transmitter level is changed to a new symbol. When an optical transmitter has this linear frequency modulation, it will bring about greater dispersion losses. The dispersion losses are also not uniform for all channels, and if the linear frequency modulation is positive, the dispersion losses will be greater for longer wavelengths (such as 1331nm), and if the linear frequency modulation is negative, the dispersion losses will be greater for shorter wavelengths (such as 1271nm). There are proposals to increase the number of CWDM wavelengths to 8, which will result in even greater parallax.

[0040] Another application that requires CD compensation for 50Gbps / wavelength or 100Gbps / wavelength within 10-20km is the forward and backhaul links feeding wireless / mobile networks. For service areas where the amount of fiber is insufficient, CWDM or DWDM is required to support the required data bandwidth, and fiber CD may cause significant system performance loss. In this case, it is actually necessary to identify electronic CD compensation methods.

[0041] PAM4 (or PAMn) OSSB

[0042] 4-level pulse amplitude modulation (PAM4) data symbols may be transmitted over an optical single sideband (OSSB) optical carrier using known data formats and optical modulation techniques. As is well known, this compresses the optical bandwidth of the transmission, thereby reducing the dispersion impairments accumulated as the signal propagates along the length of the optical fiber. A general description of a PAM4-based transceiver may be found in U.S. Patent 7,380,993, entitled "Optical Transceiver for 100 Gigabits / Second Transmission," which is incorporated herein by reference. Embodiments of the transceivers described herein may configure an EDPC to approximate a Hilbert transform of the electrical PAM4 data to condition the electrical signal applied to the optical modulator for PAM4 OSSB transmission. The corresponding processing may be performed on the PAMn data symbols by summarizing the discussion of this section to illustrate the corresponding data symbols.

[0043] The optical SSB signal can be mathematically represented as

[0044]

[0045] Where m(t) is the modulating signal (which can be a broadband digital signal or a narrowband microwave signal), is its Hilbert transform, and ω cis the optical carrier frequency. In this embodiment, m(t) can be a PAM4 signal of 50Gbps or 100Gbps, or can easily be at a higher data rate, such as PAMn (n≥4), where other elements of the optical link can support such higher rates. The Hilbert transformer is an all-pass filter that provides a phase shift of -90 degrees at all positive frequencies and a phase shift of +90 degrees at all negative frequencies. Compared with traditional optical double-sideband (ODSB) signals, the core motivation of OSSB is to remove half of the spectral components, that is, to suppress the spectral components on the positive or negative frequency side of the optical carrier as much as possible. This in turn can:

[0046] a) reduce the effects of cumulative fiber dispersion, including signal fading and signal-to-signal beat frequency interference after square-law detection, and

[0047] b) Support post-detection compensation for accumulated dispersion in the electrical domain.

[0048] The analog Hilbert transformer can be approximated by a finite impulse response filter (FIR) based on a tapped delay line [9, 11], which has a finite number of taps. The finite impulse filter based on a tapped delay line is an alternative name for the analog transversal filter. Due to the finite number of taps, the sideband suppression is not ideal. The fewer taps, the more residual power is retained in the suppressed sidebands. According to

[11] , the tap weights in the analog Hilbert transformer are as follows:

[0049] For integer sample delays:

[0050] For arbitrary delays:

[0051] Where n corresponds to the nth tap relative to the center tap (the value of n from the center tap to the input is negative), and τ is the tap delay in samples. Typically, the number of taps in the analog circuit is at least three, in other embodiments at least five, and in other embodiments 5 to 25. As further described below, the weight of the center tap is +1 or -1. In simpler symmetrical circuits, the number of taps is an odd number, although conceivable structures include structures with an even number of taps. Those of ordinary skill in the art will understand that other ranges of tap numbers within the above explicit ranges are contemplated and included in the present disclosure.

[0052] Figure 1A and Figure 1B Depicted is the EDPC circuit configuration to approximate the Hilbert transform and generate a filtered signal for the optical modulator The adder section of the EDPC actually provides two coordination outputs, one for (center tap is +1.0 coefficient), and the other is for (Center tap is -1.0 coefficient). Figure 1A A finite impulse response filter 101 having a unit delay period of T is utilized, wherein the taps 103 have tap weights obtained from formula (2). T is the symbol rate. The outputs from the taps 103 go to an amplifier / attenuator 105, which sums at an output unit 107 to send the summed transformed signal (formula (4) below) to an output 109. An alternative center amplifier / attenuator may be provided to another output 113, with the negative value of the original signal added to the Hilbert transform (formula 5 below) for driving a modulator, as described below. An alternative center amplifier / attenuator may be provided by having a single tap weighted amplifier / attenuator providing positive and negative outputs. The amplifier / attenuator 105 determines the tap weights, shown from left to right, -2 / (3π) (n=--3), -2 / π (n=-1), 1 or -1 (n=0), 2 / π (n=1) and 2 / (3π) (n=3). Since equation (2) indicates that even if the tap weights are set to zero, in this example, the external two tap weight amplifiers / attenuators are each assigned a delay of ±3T to maintain five non-negligible tap weights. n = 0 taps are set to provide an untransformed input, which is added to the transformed signal of equations 2 and 3. Similarly, Figure 1B An EDPC circuit configuration 121 based on a unit delay of (2 / 3)T and a tap 123 with tap weights derived from equation (3) to generate a corresponding filtered signal is depicted. The output from tap 123 goes to amplifier 125 where it is summed with the transformed signal at 127 (equation (4) below) for output 129. For this embodiment, the tap weights are determined by equation (3) using a delay unit of 2T / 3. Similarly, an alternative center amplifier / attenuator can be provided to another output 131, the Hilbert transform is summed with the negative of the original signal (equation 5 below) for driving a modulator as described below. For reasons and consequences described later herein, the preferred balance between signal bandwidth, tap delay spacing, and number of taps varies depending on the nature of the application. For example, by making further modifications to the tap weights (as described below), Figure 1B The configuration shown has been used to demonstrate excellent transmission performance for 50 Gbps / wavelength at DWDM wavelengths and distances up to 40 km, as will be described below.

[0053] The filtered signal is only part of the transformation of the PAM4 data into a PAM4 OSSB optical signal. It is further desirable to provide an optical carrier with in-phase and quadrature phase modulation according to formula (1). Preferably, the modulation is provided by driving an interferometric optical modulator, in particular a Mach-Zehnder type optical modulator (MZM), using a signal from the EDPC.

[0054] This can be achieved by using a dual drive base MZM (DDMZM) 201 or an IQ nested MZM 203, respectively. Figure 2A (See [9]) and Figure 2B The IQ nested MZM 203 comprises an I MZM 205 and a QMZM 207, which themselves are used to nest the respective arms 209, 211 of their MZMs. Figure 2A The basic MZM 201 is generally preferred in appropriate applications due to its simplicity. Figure 2B The IQ nested MZM 203 can provide better signal integrity range for applications that might otherwise be barely achievable, but such an embodiment will typically involve an additional inverting amplifier for the EDPC signal, resulting in greater power requirements. The MZM modulator includes a semiconductor optical waveguide that interacts with an RF electrode. Although lithium niobate-based modulators are commercially available and other semiconductor / electro-optic modulator types can be used, indium phosphide (InP)-based MZMs are particularly ideal due to size, efficiency, and performance considerations. InP-based IQ MZMs are available from NeoPnics and can be found in their High Bandwidth Coherent Driver Modulator (HB-CDM) product line. Additionally, note that the electrical signal amplitude required to drive the passive electrode 213 of the MZM is substantially less than that of a conventional OOK or PAM4 signal. This is because OSSB signals require a higher ratio of the optical carrier to signal beat frequency than the signal to signal beat frequency. As a result, the amplified output signal from the EDPC (such as Figure 1A and Figure 1B ) directly drives the MZM without the need for an additional driver amplifier.

[0055] The output of the EDPC chip may include a basic amplifier suitable for directly driving the MZM electrodes. In an alternative embodiment, an analog electrical amplifier is used to amplify the output of the EDPC chip to provide a drive current to the MZM electrodes. The modulator signal from the EDPC is reflected in the following formula, where equations (4) and (5) are used for Figure 2A , and formulas (6) and (7) are used for Figure 2B :

[0056]

[0057]

[0058]

[0059]

[0060] The configuration of the EDPC chip relative to the MZM eliminates the need for an otherwise required transmitter driver amplifier, further reducing the associated cost, size, and significant power consumption. Figure 1A and Figure 1B As shown, although multiple FIR elements can be used in a single chip or multiple chips, a single FIR can be configured to produce both outputs of d1 and d2. With respect to equations (6) and (7), the EDPC chip can further include a buffer amplifier for receiving the output of the summing unit to direct the signal to the output. The buffer amplifier can be easily configured to output a positive output or a negative output or both, or can be used as an integrated driver. The output unit ( Figure 1A 107 and Figure 1B 127) in may include a summing unit and a buffer amplifier, and for appropriate embodiments, the buffer amplifier may output a positive output and a negative output, such as Figure 1A and Figure 1B As shown by the dotted line in .

[0061] Chromatic dispersion (CD) precompensation

[0062] To further improve the transmission distance, we describe an alternative embodiment that includes partial fiber dispersion pre-compensation in the simulated EDPC. Since the actual length of the transmission link is usually not available to the transceiver, the CD at the receiver end must be estimated or adaptively compensated. Since making the estimate involves additional power consumption, this estimate is unlikely to be used in practice. The power-constrained number of taps in the receiver adaptive equalizer provided by the short-reach PAM4 chip limits the maximum dispersion that can be adaptively compensated. For the purposes of illustration, it can be assumed for the moment that the receiver's adaptive equalizer is able to compensate for the CD corresponding to the maximum transmission length L. Due to the symmetry of the adaptive equalization, this means that the range of CD that can be compensated by the receiver's adaptive equalizer is between -CD L and+CD L We find that by pre-compensating for the dispersion corresponding to a fixed equivalent transmission distance L at the transmitter side, the meaningful optical transmission range can be effectively extended, thereby doubling the transmission distance.

[0063] To illustrate, the proposed precompensation method is considered in two limiting cases:

[0064] 1) Transmission distance is zero: On the transmitter side, the dispersion with opposite sign (-CD L) will be applied to the signal. Since this value is within its compensation range, it can be processed by the receiver's adaptive equalizer.

[0065] 2) The transmission distance is 2L. On the transmitter side, the dispersion (-CD) corresponding to the transmission length L has the opposite sign. L ) will be applied to the signal again, thus precompensating it by half the transmission length. After propagating 2L in the fiber, at the receiver, the residual CD will be equal to CD L Since this value is still within its compensation range, it can again be processed by the receiver's adaptive equalizer.

[0066] Therefore, in these embodiments, the analog EDPC is designed to provide a combined Hilbert transform and dispersion precompensation simultaneously. This is achieved by convolving the tap weights of the Hilbert transformer with the tap weights of the dispersion precompensation FIR. The tap weights of the CD precompensation FIR can be obtained by calculating the inverse Fourier transform of the conjugate of the dispersion fiber transfer function H(f) of the fiber device length L [8].

[0067]

[0068] Where D is the fiber dispersion parameter, L is the length of the fiber, λ is the wavelength of the signal, f is the low-pass equivalent frequency, and c is the speed of light.

[0069] Figure 3A (Module 301) and Figure 3B (Module 303) shows a method based on the Hilbert transform and fiber CD precompensation (hereinafter referred to as "HT+CD -1 ”) EDPC optical transceiver modules are suitable for 2λx 50Gbps and 1λx 100Gbps respectively. Figure 3A and Figure 3B All are envisioned as 100G QSFP28 modules, but the optical modules can be implemented in pluggable form factors such as QSFP28, DD-QSFP, OSFP, CFP2, or CFP8 under the industry MSA standards. Figure 3A The dual wavelength 2x50G optical transceiver 301 in the embodiment is simplified to a single wavelength 50Gbps QSFP or SFP optical transceiver module for wireless fronthaul or midhaul or passive optical network (PON) applications.

[0070] refer to Figure 3A, the optical transceiver module 301 includes a 2x50Gbps PAM4 DSP 311, which is connected to a 2-wavelength receiver optical subassembly (ROSA) 313 on the receiving side, and the 2-wavelength ROSA receives two wavelength-separated optical signals from a demultiplexer 315, which is connected to an optical input 317 entering the module. The ROSA 313 typically includes a photodetector (such as a PIN diode or APD (avalanche photodiode)) and a transimpedance amplifier (TIA) for each wavelength, and converts the optical signal into an electrical signal. The demultiplexer 315 may include a grating, a filter, and / or a demultiplexer based on a 1:2 Mach-Zehnder interferometer. The transmission signal from the PAM4 DSP 311 is directed to a 2x analog EDPC 321, which may be two analog circuits, such as Figure 1A and Figure 1B The pre-compensated signal from the 2x analog EDPC to the 2x MZM 323, the 2x MZM 323 receives light input from the λ1 laser 325 and the λ2 laser 327, which is modulated and directed to the output 329.

[0071] refer to Figure 3B , the optical transceiver module 303 includes a 1×100Gbps PAM4 DSP 331, which is connected to a 1-wavelength ROSA 333 on the receiving side, which receives a one-wavelength optical signal from an optical input 335 entering the module. The ROSA 333 typically includes a photodetector (such as a PIN or APD) and a TIA for converting the one-wavelength optical signal into an electrical signal. The transmission signal from the PAM4 DSP 331 is directed to a 1x analog EDPC 341, which can be an analog circuit such as Figure 1A or Figure 1B The pre-compensated signal from the 1x analog EDPC 341 to the 1x MZM 343, the 1x MZM 343 receives the optical input from the λ1 laser 345, which is modulated and directed to the output 349.

[0072] To apply CD precompensation with the Hilbert transform, the following equations 9 to 12 replace Figure 2A and Figure 2B The formula is based only on the Hilbert transform. Figure 3A Two λ embodiments, using d1, d2, d3 and d4, for Figure 3B An embodiment of λ, using only d1 and d2.

[0073]

[0074]

[0075]

[0076]

[0077]

[0078] F -1 = Inverse Fourier Transform

[0079] By adjusting the attenuated output of each tap from two FIR elements within the analog chip (one FIR for equations (9) and (10) and one FIR for equations (11) and (12)), the analog circuit can be designed to approximately perform the desired transformation from equation 9 to equation 12. In a transceiver embodiment that supports multiple simultaneous emission wavelengths and / or transmission fibers, preferably, the analog EDPC chip may include multiple FIR groups for filtering multiple signal streams simultaneously. For example, in the above example, the analog EDPC may have two parallel FIR groups, one for solving equations (9) and (10) relative to signal m1, and the other for solving equations (11) and (12) relative to signal m2. In particular, the response of the FIR can be written as:

[0080]

[0081] where c(t) is set equal to d1, d2, d3, and d4, respectively, and δ represents the Dirac delta function, where c n is the tap weight, and τ is the delay period of the analog filter, such as T( Figure 1A ) or 2T / 3( Figure 1B ) (or other appropriate delay), T is the symbol rate. 2N+1 equals the number of taps. This relationship can be evaluated by simulation for a set number of taps, or can be corrected empirically. The general theory of correcting communication distortion with electronic filters is described in Rudin, "Automatic Equalization Using Transversal Filters," IEEE Spectrum, January 1967, pp. 53-59, which is incorporated herein by reference.

[0082] like Figure 4A and Figure 4B As shown, in the optical transceiver module 301 ( Figure 3A ) can contain one of two alternative structures 403, 405 of a dual wavelength emitter. The first structure ( Figure 4A ) has an integrated optical combiner 407 so the output is connected to a single optical fiber 409 which in turn can be connected to 40 with 100 GHz spacing. + Channel DWDM multiplexer, or connected to a remote receiver via dark fiber. The second structure 405 ( Figure 4B) has two transmitter output fibers 411, 413, which can be connected to 80 with 50 GHz spacing. + The two input ports of the channel DWDM multiplexer. 80 + The channel DWDM multiplexer usually consists of two 40-channel even and odd (ITU-T) channels. + The channels consist of a DWDM multiplexer and can be combined by an interleaver.

[0083] Similarly, if Figure 5A and Figure 5B As shown, the optical transceiver module 301 ( Figure 3A )'s dual wavelength optical receiver can have one of two different alternative structures corresponding to the two transmitter structures. Figure 5A The structure corresponds to Figure 3A For example, from 40 + The optical input fiber 505 of the channel DEMUX is connected to Figure 3A The demultiplexer 315 receives the two-channel DEMUX 507. The λ1 output of the two-channel DEMUX 507 is directed to the ROSA 509, and the λ2 output of the two-channel DEMUX 507 is directed to the ROSA 511. Figure 5B , separate optical fibers 521, 523 carrying inputs at λ1 and λ2, respectively, direct their respective inputs to ROSA 525 and ROSA 527. The input optical fibers 521, 523 may be, for example, from 80 + channel DEMUX, which usually consists of two 40 + Channel DWDM demultiplexer, two 40 + The channel DWDM demultiplexer has even and odd ITU-T channels respectively, which can be combined by an interleaver.

[0084] By quadrupling the number of wavelengths, PAM4 channels, and analog Hilbert transformers, Figure 3A and Figure 3B The module concept in is extended to 400Gbps DD-QSFP (or OSFP, CFP2, CFP8) transceiver modules, such as Fig. 6A and Figure 6B shown. Specifically, Figure 3A and Figure 3B Based on the Hilbert transform and fiber CD precompensation (hereinafter referred to as "HT+CD -1 ”) EDPC optical transceiver modules 301, 303 can be expanded to 400Gbps, respectively for 8λx 50Gbps and 4λx100Gbps. Fig. 6A and Figure 6BThey are specifically intended for use as 400G QSFP28 modules, but the optical modules can be implemented in pluggable form factors of the QSFP28, DD-QSFP, OSFP, CFP2 or CFP8 standards under the industry MSA.

[0085] refer to Fig. 6A , the optical transceiver module 601 includes an 8x50Gbps PAM4 DSP 611, which is connected on the receive side to an 8-wavelength receiver optical subassembly (ROSA) 613, which receives 8-wavelength separated optical signals from a demultiplexer 615, which is connected to an optical input 617 entering the module. The ROSA 613 typically includes a photodetector for each wavelength and converts the optical signal into an electrical signal. The demultiplexer 615 may include a grating, an interference filter, and / or an AWG-based demultiplexer. The transmission signal from the PAM4 DSP 611 is directed to an 8x analog EDPC 621, which may be eight analog circuits, such as Figure 1A and Figure 1B Precompensated signal from 8x analog EDPC to 8x MZM 623, 8x MZM 623 receives optical input from laser bank 625 consisting of eight (λ1 ... λ8) lasers, which are modulated by eight channel multiplexer 627 and directed to output 629. Similar multiplexing elements can be used as described for the demultiplexing function.

[0086] refer to Figure 6B , the optical transceiver module 603 includes a 4x100Gbps PAM4 DSP 631, which is connected to a 4-wavelength ROSA 633 on the receiving side, which receives a one-wavelength optical signal from an optical input 635 entering the module. The ROSA 633 typically includes a photodetector (such as a PIN or APD) and a TIA for converting the one-wavelength optical signal into an electrical signal. The transmission signal from the PAM4 DSP 631 is directed to a 4x analog EDPC 641, which can be an analog circuit, such as Figure 1A and Figure 1B The pre-compensated signal from the 4x analog EDPC 641 to the 4x MZM 643 receives an optical input from a laser bank consisting of four (λ1 . . . λ4) lasers 645 which are modulated by a four channel multiplexer 647 and directed to an output 649 .

[0087] The N:1 multiplexer and 1:N demultiplexer in the figure are just examples of many possible configurations. Likewise, there are passive (fixed) dispersion compensating elements in the optical path after the multiplexer. For the purposes of the drawings, the passive dispersion compensating elements may be considered within the frame of the multiplexer, although in actual devices they may be packaged separately. Moreover, for embodiments in which a multiplexer is not shown, there may be a passive dispersion element whose position in the optical path is essentially at the location of the multiplexer without the multiplexing function. A combination of an Mx1 coupler and an L:1 multiplexer / demultiplexer may be used to perform the same function. In the case of DWDM applications, multiple TX fibers and RX fibers may also be used, as well as MPO (multi-fiber push-on) connectors.

[0088] In the module representation of the attached figure, all PAM4 DSP chips can be initially designed for short distance (≤10km) and low power consumption, and typically include host side SERDES (serializer / deserializer), line side forward error correction (FEC) encoder / decoder, line side equalizer (e.g., CTLE, feedforward and decision feedback equalizer), MSB / LSB amplitude and skew adjustment, digital-to-analog converter (DAC) and analog-to-digital converter (ADC) and many other functions. It is conceivable that these PAM4 DSP chips can be further replaced by simpler analog-based implementations that do not require DAC and ADC. However, the number of feedforward equalizer taps in analog PAM4 chips may not be sufficient to obtain sufficient post-compensation for dispersion for the required transmission distance. Similarly, analog PAM4 chips typically do not include FEC. Although we know that the higher the FEC coding gain, the stronger the equalizer (e.g., the more taps), the longer the transmission distance. Given the maturity of existing DSP-based PAM4 chips, it seems generally more suitable to use them for the transceiver modules described in this article.

[0089] Iterative tap weight optimization

[0090] As shown in formulas (5)-(8), use HT+CD -1 The driving signal of the tap weight is based on theoretical calculation. In practice, the bandwidth of the transmitter and receiver will also affect the performance. Therefore, based on the configurable FIR filter, we can convert the calculated HT+CD -1The tap weights are used as initial conditions and improved tap weights are sought that can also compensate for both transmitter bandwidth limitations and receiver bandwidth limitations through a convergence process. Adjustments to the amplifier / attenuator outputs are used to adjust the tap weights. The performance of tap weights optimized for a certain target distance can degrade rapidly when the actual distance differs from the optimization target. In order to obtain a set of tap weights that work over a wide range of transmission distances, a process of performing joint optimization for multiple transmission distances (or fiber lengths) within the desired range is proposed herein. Note that this process can also be used to optimize a single transmission distance.

[0091] The search algorithm uses a test signal sent through a length of L1, ..L N The weighted sum of the EVM (error vector magnitude) obtained after a set of optical fibers is used as the optimization metric J.

[0092]

[0093] where w n are arbitrary weights that can be adjusted to achieve the desired EVM vs. distance distribution. Figure 7 A schematic diagram of the architecture of a system 701 for implementing iterative tap weight optimization is depicted in . Transmitter 703 can be one of the transmitter components of the transceiver described herein. Transmitter 703 then outputs light from fiber group 705 into individual optical fibers and then transmits the light to receiver 707. Receiver 707 uses a suitable processor to calculate the EVM and transmits the EVM to a controller 709 connected to both transmitter 703 and receiver 707. For convenience, controller 709 can be used in the manufacturing / configuration process of preparing the transceiver and can be an analog circuit, a digital processor, or a combination of both. Controller 709 can be used to configure the FIR and is not typically used by the end user, although controller 709 can be used to reconfigure the FIR during a subsequent time after a period of use.

[0094] Note that in a production line, a programmable dispersion compensator can be used to simulate CD, replacing the lengths L1..L NA set of physical fiber links. For a description of programmable dispersion compensators, see, for example, U.S. Patent 6,879,426 to Weiner, entitled "System and Method for Programmable Polarization-Independent Phase Compensation of Optical Signals," which is incorporated herein by reference. For an EDPC with n taps, the search for the best taps is an n-dimensional optimization problem that can usually be solved by a known search method such as steepest descent. In practical devices, the number of taps can be small, n ≤ 5, and their values ​​must be specified as integers in a limited range (in the experimental devices described later, their values ​​are integers between -100 and 100). For this case, it is likely to use Figure 8 A simplified search process is shown in the flowchart.

[0095] It is convenient to view the tap weights as a vector h, which is formed by cascading the tap weights required to generate signals d1 (Formula (9)) and d2 (Formula (10)), respectively. For example, consider an EDPC with five analog taps. Then, the tap weights that generate signal d1 are [I1 I2 I3 I4 I5], and the tap weights that generate signal d2 are [Q1 Q2 Q3 Q4 Q5]. Then, the vector h is [I1 I3 I2 I4 I5 Q1 Q2 Q3 Q4 Q5]. It is observed that changes in the middle tap weights have a greater impact on the performance of the EDPC than changes in the outer tap weights. Therefore, it is preferred to adjust the tap weights in order of their importance. To this end, it is convenient to sort the vector h in descending order of importance, h = [I3 Q3 I2 Q2 I4 Q4 I1 Q1I5 Q5], and then adjust the elements of h from beginning to end. The best tap is searched iteratively. Initially, the controller loads the EDPC with the tap weights that are analytically calculated as the convolution of the Hilbert transform and the CD precompensation weights corresponding to one of the target fiber lengths L. Then, the signal is subsequently transmitted to each fiber in the optimization target fiber set, and the EVM value is obtained from the receiver by the controller. The optimization cost function is calculated by the controller according to formula (14).

[0096] A similar iterative process can be used whether simulating J from a programmable dispersion compensator or measuring using a set of fiber lengths connected to an optical receiver. Figure 8 The iterative process 801 in FIG. 803 specifies (803) a set of fiber lengths (L1, ..., L2) within a desired range. N ) and their importance (w1, ..., w N), as described above in equation (14). The initial tap weights are then specified (805) in the order of importance indicated in the previous paragraph with a step size s (which can be selected by the user as a reasonable initial value). Like most stochastic convergence optimization procedures, the initial value of the step size affects the balance between convergence speed and convergence quality towards obtaining the best results. For a given setting, it is often necessary to evaluate different initial step sizes based on the expected quality to determine a preferred value or range. The vector h0 is loaded (807) into the EDPC. For each L, a signal is transmitted (809) to obtain EVM1, ..., EVM N Using these values, the initial value of the cost function J0(h0) is calculated (811).

[0097] At each iteration k( Figure 8 813), the controller generates two test tap weight sets h by respectively increasing and decreasing one of the tap weights by a value specified as the step size s. + and h - Then, for each fiber length L, the tap weight vector is loaded into the EDPC. 1..N The test signal is transmitted and the obtained EVM is summed according to formula (14) to obtain the vector h, h + and h - The corresponding three cost functions J, J + and J - Next, the cost function values ​​are compared and the minimum function value with the corresponding h is selected for the next iteration. The iterations will continue until the cost function stops decreasing. The iterations go through each tap separately, where the parameter "i" counts the taps. In this case, the step size s will be halved and the iterations will continue. This cycle continues until the step size equals 1.

[0098] The proposed search procedure has been verified by simulating the final EDPC in an optical link.

[0099] Fig.9A shows a simulated transmit optical eye diagram produced by a transmitter with EDPC tap weights obtained by computational convolution of the Hilbert transform and dispersion precompensation for 10km of fiber, with a target transmission distance of 20km (i.e., no iterative optimization). Since the deterministic filter is calculated for 10km, an open eye diagram can be seen after 10km transmission even without the Rx adaptive equalizer. Therefore, Fig. 9BThe eye diagram shows the received signal level before DSP processing, and it can be seen that the eye is almost gone, but still open. Note that if the same pre-compensated optical signal was transmitted over 20 km of fiber, no open eye would be seen before DSP, because the transmitted signal is only pre-compensated for half that distance, so the remaining 10 km of dispersion washes out the open eye. Fig. 9C and Fig.9D shows that when using the optimized tap (by Figure 8 We can see that the eye diagram is relative to the transmitted and received eye diagrams. Fig.9A and Fig. 9B There has been a noticeable improvement.

[0100] Based on the above, a set of tap weights suitable for any distance within 40km can be found using a short-haul PAM4 DSP chip (when only 5 EDPC taps are used). By setting L = 20km and based on the Hilbert transform and CD pre-compensation (HT + CD -1 ) and a set of optimized weights subsequently found by the iterative search. If the use of this set of tap weights results in poor link performance, we may alternatively choose to use two sets of tap weights and select the best tap weights as part of link initialization. In this case, the first tap weights are found by, for example, setting L=10km (refining the actual tap weights as previously described) to cover distances between 0 and 20km; and similarly determining the second set of tap weights by refining L=around 30km to cover distances between 20 and 40km. These two sets of tap weights can be saved in the transceiver configuration memory, and when the transceiver is enabled on the actual fiber optic equipment, the appropriate settings for optimal performance can be selected. Depending on power consumption requirements, more taps can be used to further improve link BER performance or increase transmission distance. For example, Figure 1A and Figure 1B The 5 taps shown can be increased to 7 or 9 taps.

[0101] PAM4 ODSB

[0102] Above, the transceiver embodiments described are all based on OSSB transmission. For some considerations, it is preferred to use conventional optical double side band (ODSB) transmission. For these embodiments, no Hilbert transform is performed. In another embodiment of the transceiver, the EDPC chip can also be advantageously used in ODSB signal transmission. The drive signal to the two MZI electrodes is:

[0103]

[0104]

[0105] in

[0106]

[0107] The same principle can be applied to line side optics based on 1x100G, 8x50G or 4x100G optical transceivers etc. The optimization process for tap weights can still follow Figure 8 In general, at a specific distance, an ODSB-PAM4 transceiver can achieve a lower raw error rate than OSSB-PAM4. However, the optimized tap weights for ODSB-PAM4 cannot cover as wide a transmission distance range as OSSB-PAM4. As a result, more sets of tap weights for different distance ranges will need to be stored in the transceiver memory, which may be operationally inconvenient for service providers.

[0108] Experimental Results

[0109] follow Figure 3A The optical transceiver configuration shown in Figure 1 was used to configure a field experiment, although only one of the two wavelengths was activated to demonstrate the invention. A commercial short-range PAM4 DSP was used in the experiment (it converts 4x25Gbps on the host side to 2x50Gbps on the line side). The EDP C chip is an analog transverse filter manufactured in the IBM 90nm 9HP SiGe BiCMOS process with a cutoff frequency of 300GHz

[12] . The MZM structure is shown in Figure 1. Figure 2B shown.

[0110] After a direct 40 km fiber link without optical dispersion compensation, we obtain the bit error rate (BER) as a function of the received optical power, as Fig. 10A As shown in this figure, we can see that when the pre-FEC BER threshold is set to 1e-3 (determined by the PAM4 DSP chip used in the experiment), a receiver sensitivity of <-10.5dBm after 40km and <-11dBm after 20 / 30km can be achieved (the minimum received optical power is kept below 10 -3 Receiver sensitivity can be further improved by replacing the older TIA (spectral noise density of 17pA / √Hz) in our experiments with a more recent TIA (spectral noise density of 12pA / √Hz).

[0111] For DWDM systems with DWDM booster amplifiers and preamplifiers, we also measured the BER performance versus the optical signal-to-noise ratio (OSNR), as Fig. 10BWe can see in this figure that when the pre-FEC BER threshold is set to 1e-3 (shown by the black dashed line), the required OSNR of <33dB after 40km and <32dB after 20 / 30km can be achieved.

[0112] The references cited above in square brackets are incorporated herein by reference:

[0113] 【1】http: / / 100glambda.com / specifications

[0114] 【2】IEEE802.3bsTM / D3.5

[0115] [3] N. Eiselt et al., “Evaluation of Real-Time 8×56.25Gbps(400G)PAM-4 forInter-Data Center Application Over 80km of SSMF at 1550nm”, J. Lightwave Tech., 35(4), 955-962, 2017.

[0116] [4] S. Yin et al., “100-km DWDM Transmission of 56-Gbps PAM4 per λ via Tunable Laser and 10-Gbps InP MZM”, Photon. Tech. Lett., 27(24), 2531-2534, 2015.

[0117] 【5】U.S. Patent 9,722,722 B2, August 1, 2017.

[0118] 【6】U.S. Patent 9,553,670 B2, January 24, 2017.

[0119] 【7】M. Morsy and D. V. Plant, “A comparative study of technology options for next generation intra-and inter-datacenter interconnects,” in W4E.1, 2018 Optical Fiber Communications Conference.

[0120] [8] M. Sieben et al., “Optical Single Sideband Transmission at 10 Gbps Using Only Electrical Dispersion Compensation”, J. Lightwave Tech., Vol. 17, No. 10, pp. 1742-1749, October 1999.

[0121] 【9】U.S. Patent 5,880,870, March 9, 1999.

[0122]

[10] U.S. Patent 7,206,520 B2, April 17, 2007.

[0123]

[11] L. R. Rabiner and R. W. Schafer, “On the behavior of minimax FIR digital Hilbert transformers”, Bell System Tech. J., vol. 53, no. 2, pp. 363–390, February 1974.

[0124]

[12] Edem Ibragimov et al., “Coherent Analog Low Power, Small Size 400 / 200 / 100GB / s Receiver Based on Bipolar SiGe Technology,” Th1A.5, 2018 Optical Fiber Communications Conference.

[0125] The above embodiments are intended to be illustrative and not restrictive. Additional embodiments are within the claims. In addition, although the invention has been described with reference to specific embodiments, those skilled in the art will recognize that changes can be made in form and detail without departing from the spirit and scope of the invention. Limitations are made by reference to any combination of the above documents so that no subject matter contrary to the explicit disclosure herein is combined. Unless otherwise specifically noted, to the extent that components, elements, ingredients, or other separators are used herein to describe specific structures, compositions, and / or methods, it should be understood that the disclosure herein covers specific embodiments, embodiments containing specific components, elements, ingredients, other separators, or combinations thereof, and embodiments consisting essentially of such specific components, ingredients, other separators, or combinations thereof, which embodiments may include other features that do not change the basic nature of the subject matter, as suggested in the discussion.

Claims

1. An optical transceiver module for n-level pulse amplitude modulation (PAMn) optical symbols, wherein n≥2, the optical transceiver module provides an interface between electronic data signals on a host and optical symbols transmitted at a baud rate through an optical communication device, the optical transceiver module comprising: a receiver section including at least one optoelectronic receiver to convert a received optical signal into an analog electrical signal; PAMn digital signal processing DSP circuit, providing at least a logical interface with electronic host data, forward error correction (FEC) function, analog-to-digital conversion of electrical signals from optical receivers, digital adaptive filtering of converted received signals and reconstruction of filtered received signals into data; a transmitter section including at least one laser and at least one interference modulator; as well as an analog transversal filter circuit connected to the output of the PAMn DSP and configured as an electronic dispersion precompensator EDPC for filtering the PAMn transmission signal provided by the PAMn DSP, wherein a filtered signal transmitted from said EDPC is connected to a signal input of an interference modulator; The EDPC filter circuit comprises: a finite impulse response filter including at least three taps, an amplifier / attenuator for adjusting the tap weights, and an electrical synthesizer for combining signals from the taps.

2. The optical transceiver module according to claim 1, wherein: The at least three taps are 5 taps, 7 taps or 9 taps.

3. The optical transceiver module according to claim 1 or 2, wherein: The amplifier / attenuator is field programmable.

4. The optical transceiver module according to any one of claims 1 to 3, wherein: The EDPC also includes an integrated driver for an output from the electrical synthesizer, wherein the driver output power rating is sufficient to directly drive the interference modulator.

5. The optical transceiver module according to any one of claims 1 to 4, wherein: The EDPC performs an approximate Hilbert transform on the analog electrical signal.

6. The optical transceiver module according to any one of claims 1 to 5, wherein: The EDPC implements distortion pre-correction based on a predicted distortion corresponding to a multiple of half a maximum correction length provided by the receiver.

7. The optical transceiver module according to claim 6, wherein: The distortion precorrection is selected from a set of pre-stored tap weights, wherein members of the set represent different multiples of half a maximum correction length.

8. The optical transceiver module according to claim 6 or claim 7, wherein: The EDPC also implements an approximate Hilbert transform.

9. The optical transceiver module according to any one of claims 1 to 8, wherein: The PAMn DSP is a PAM4 chip designed for the MSA standard in Table 1.

10. The optical transceiver module according to any one of claims 1 to 9, wherein: The optical transceiver module provides a data flow greater than or equal to 50 Gbps / wavelength×channel, and the optical transceiver module is functionally accommodated in a standard QSFP28, DD-QSFP, OSFP, CFP2 or CFP8 pluggable package.

11. An optical transceiver module according to any one of claims 1 to 10, wherein the interferometric modulator is a DDMZM, and wherein two modulator signals are provided from the EDPC to two DDMZM arms.

12. The optical transceiver module according to any one of claims 1 to 10, wherein the interfering modulator is an IQ nested MZM, and wherein four modulator signals are provided from the EDPC to four MZM arms of the IQ nested MZM.

13. The optical transceiver module according to claim 1, further comprising: A fixed dispersion compensation module is arranged on an optical path after the multiplexer in the optical transceiver module.

14. A method for extending transmission distance using readable symbols of an optical transmitter operating with PAMn modulation, the method comprising: conditioning an analog modulator signal from a PAMn processor using an analog transversal filter circuit to perform an approximate Hilbert transform, dispersion precompensation, or both an approximate Hilbert transform and dispersion precompensation to form a conditioned modulator signal; as well as modulating the optical laser using an optical interferometer modulator based on the adjusted modulator signal; The analog transversal filter circuit comprises: a finite impulse response filter including at least three taps, an amplifier / attenuator for adjusting the tap weights, and an electrical synthesizer for combining signals from the taps.

15. The method according to claim 14, wherein: The PAMn processor includes a PAM4 DSP configured within an optical transceiver module.

16. The method according to claim 14 or claim 15, wherein: The analog transversal filter circuit uses at least three taps to approximate the Hilbert transform.

17. A method according to any one of claims 14 or 15, wherein: The analog transversal filter circuit implements dispersion pre-compensation by setting tap weights.

18. The method according to claim 17, wherein: The analog transversal filter also implements an approximate Hilbert transform.

19. The method according to claim 17, wherein: The tap weights are operatively selected from a set of predetermined tap weights, wherein members of the set provide compensation for different dispersion accumulations.

20. The method according to any one of claims 14 to 19, wherein: The optical interferometric modulator performing the modulation is an IQ nested MZM, and wherein four modulator signals are provided to four MZM arms of the IQ nested MZM.

21. The method according to any one of claims 14 to 19, wherein: The optical interferometric modulator performing the modulation is a DDMZM, and wherein two modulator signals are provided to two arms of the DDMZM.

22. A method for determining tap weights of an analog transversal filter circuit, the analog transversal filter circuit being configured to condition an analog modulator signal from a PAMn processor to perform an approximate Hilbert transform, dispersion precompensation, or both an approximate Hilbert transform and dispersion precompensation to form a conditioned modulator signal, the method comprising: Iterative correction of tap weights to improve dispersion over a composite range of multiple fiber lengths; The analog transversal filter circuit comprises: a finite impulse response filter including at least three taps, an amplifier / attenuator for adjusting the tap weights, and an electrical synthesizer for combining signals from the taps.

23. The method according to claim 22, wherein: The iterative correction is performed through error vector magnitude measurements of optical receivers located at the target distance for each fiber length.

24. A method according to claim 22 or claim 23, wherein: Tap weight settings are determined for a plurality of different composite ranges of optical fiber lengths, and the settings for each composite range of lengths are stored in a digital memory of the optical transceiver as a member of a set for operational selection.

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